HCNR200 and HCNR201 High-Linearity Analog Optocouplers Data Sheet Lead (Pb) Free RoHS 6 fully compliant RoHS 6 fully compliant options available; -xxxE denotes a lead-free product Description Features The HCNR200/201 high-linearity analog optocoupler consists of a high-performance AlGaAs LED that illumi‑ nates two closely matched photodiodes. The input pho‑ todiode can be used to monitor, and therefore stabilize, the light output of the LED. As a result, the non-linearity and drift characteristics of the LED can be virtually elimi‑ nated. The output photodiode produces a photocurrent that is linearly related to the light output of the LED. The close matching of the photo-diodes and advanced de‑ sign of the package ensure the high linearity and stable gain characteristics of the optocoupler. • Low nonlinearity: 0.01% • K3 (IPD2/IPD1) transfer gain HCNR200: ±15% HCNR201: ±5% • Low gain temperature coefficient: ‑65 ppm/°C • Wide bandwidth – DC to >1 MHz • Worldwide safety approval – UL 1577 recognized (5 kV rms/1 min rating) – CSA approved – IEC/EN/DIN EN 60747-5-2 approved VIORM = 1414 V peak (option #050) • Surface mount option available (Option #300) • 8-Pin DIP package - 0.400” spacing • Allows flexible circuit design The HCNR200/201 can be used to isolate analog signals in a wide variety of applications that require good stabil‑ ity, linearity, bandwidth and low cost. The HCNR200/201 is very flexible and, by appropriate design of the appli‑ cation circuit, is capable of operating in many different modes, including: unipolar/bipolar, ac/dc and inverting/ non-inverting. The HCNR200/201 is an excellent solution for many analog isolation problems. Schematic LED CATHODE 1 8 - NC VF LED ANODE PD1 CATHODE PD1 ANODE + IF 2 7 3 6 IPD1 4 I PD2 5 NC Applications • Low cost analog isolation • Telecom: Modem, PBX • Industrial process control: Transducer isolator Isolator for thermocouples 4 mA to 20 mA loop isola‑ tion • SMPS feedback loop, SMPS feedforward • Monitor motor supply voltage • Medical PD2 CATHODE PD2 ANODE CAUTION: It is advised that normal static precautions be taken in handling and assembly of this component to prevent damage and/or degradation which may be induced by ESD. Ordering Information HCNR200/HCNR201 is UL Recognized with 5000 Vrms for 1 minute per UL1577. Option IEC/EN/DIN EN Part RoHS non RoHS Surface Gull Tape UL 5000 Vrms/ 60747-5-2 Number Compliant Compliant Package Mount Wing & Reel 1 Minute rating VIORM = 1414 Vpeak Quantity -000E no option 400 mil X 42 per tube -300E #300 Widebody X X X 42 per tube HCNR200-500E #500 DIP-8 X X X 750 per reel HCNR201-050E #050 X X 42 per tube -350E #350 X X X X 42 per tube -550E #550 X X X X 750 per reel X X To order, choose a part number from the part number column and combine with the desired option from the option column to form an order entry. Example 1: HCNR200-550E to order product of Gull Wing Surface Mount package in Tape and Reel packaging with IEC/EN/ DIN EN 60747-5-2 VIORM = 1414 Vpeak Safety Approval and UL 5000 Vrms for 1 minute rating and RoHS compliant. Example 2: HCNR201 to order product of 8-Pin Widebody DIP package in Tube packaging with UL 5000 Vrms for 1 minute rating and non RoHS compliant. Option datasheets are available. Contact your Avago sales representative or authorized distributor for information. Remarks: The notation ‘#XXX’ is used for existing products, while (new) products launched since July 15, 2001 and RoHS compliant will use ‘–XXXE.’ 2 Package Outline Drawings 0.20 (0.008) 0.30 (0.012) 11.30 (0.445) MAX. 8 7 6 5 MARKING A HCNR200 yyww EEE * PIN ONE 1 2 3 LOT ID 9.00 (0.354) TYP. 10.16 (0.400) TYP. 11.00 (0.433) MAX. 0° 15° 4 1.50 (0.059) MAX. 1 5.10 (0.201) MAX. LED 2 3 1.70 (0.067) 1.80 (0.071) 3.10 (0.122) 3.90 (0.154) 0.40 (0.016) 0.56 (0.022) 2.54 (0.100) TYP. 4 8 NC 7 K2 K1 0.51 (0.021) MIN. NC 6 PD1 PD2 5 DIMENSIONS IN MILLIMETERS AND (INCHES). MARKING : yy - Year ww - Work Week Marked with black dot - Designates Lead Free option E XXX = 050 ONLY if option #050,#350,#550 (or -050,-350,-550) ordered (otherwise blank) * - Designates pin 1 NOTE: FLOATING LEAD PROTRUSION IS 0.25 mm (10 mils) MAX. Figure 1a. 8 PIN DIP 3 Gull Wing Surface Mount Option #300 11.15 ± 0.15 (0.442 ± 0.006) 8 7 6 LAND PATTERN RECOMMENDATION 5 9.00 ± 0.15 (0.354 ± 0.006) 1 2 3 13.56 (0.534) 4 1.3 (0.051) 2.29 (0.09) 12.30 ± 0.30 (0.484 ± 0.012) 1.55 (0.061) MAX. 11.00 MAX. (0.433) 4.00 MAX. (0.158) 1.78 ± 0.15 (0.070 ± 0.006) 2.54 (0.100) BSC 0.75 ± 0.25 (0.030 ± 0.010) DIMENSIONS IN MILLIMETERS (INCHES). LEAD COPLANARITY = 0.10 mm (0.004 INCHES). NOTE: FLOATING LEAD PROTRUSION IS 0.25 mm (10 mils) MAX. Figure 1b. 8 PIN Gull Wing Surface Mount Option #300 4 1.00 ± 0.15 (0.039 ± 0.006) + 0.076 0.254 - 0.0051 + 0.003) (0.010 - 0.002) 7° NOM. Solder Reflow Temperature Profile 300 PREHEATING RATE 3 °C + 1 °C/–0.5 °C/SEC. REFLOW HEATING RATE 2.5 °C ± 0.5 °C/SEC. 200 PEAK TEMP. 245 °C PEAK TEMP. 240 °C TEMPERATURE (°C) 2.5 C ± 0.5 °C/SEC. 30 SEC. 160 °C 150 °C 140 °C PEAK TEMP. 230 °C SOLDERING TIME 200 °C 30 SEC. 3 °C + 1 °C/–0.5 °C 100 PREHEATING TIME 150 °C, 90 + 30 SEC. 50 SEC. TIGHT TYPICAL LOOSE ROOM TEMPERATURE 0 0 50 100 150 200 250 TIME (SECONDS) NOTE: NON-HALIDE FLUX SHOULD BE USED. Recommended Pb-Free IR Profile TIME WITHIN 5 °C of ACTUAL PEAK TEMPERATURE tp Tp TEMPERATURE TL Tsmax * 245 +0/-5 °C 15 SEC. 217 °C 150 - 200 °C RAMP-UP 3 °C/SEC. MAX. RAMP-DOWN 6 °C/SEC. MAX. Tsmin ts PREHEAT 60 to 180 SEC. 25 tL 60 to 150 SEC. NOTES: THE TIME FROM 25 °C to PEAK TEMPERATURE = 8 MINUTES MAX. Tsmax = 200 °C, Tsmin = 150 °C NOTE: NON-HALIDE FLUX SHOULD BE USED. t 25 °C to PEAK TIME Regulatory Information The HCNR200/201 optocoupler features a 0.400” wide, eight pin DIP package. This package was specifically designed to meet worldwide regulatory requirements. The HCNR200/201 has been approved by the following organizations: UL IEC/EN/DIN EN 60747-5-2 Recognized under UL 1577, Component Recognition Program, FILE E55361 Approved under IEC 60747-5-2:1997 + A1:2002 EN 60747-5-2:2001 + A1:2002 DIN EN 60747-5-2 (VDE 0884 Teil 2):2003-01 (Option 050 only) CSA Approved under CSA Component Acceptance Notice #5, File CA 88324 5 Insulation and Safety Related Specifications Parameter Symbol Value Units Conditions Min. External Clearance L(IO1) 9.6 mm (External Air Gap) Measured from input terminals to output terminals, shortest distance through air Min. External Creepage L(IO2) 10.0 mm (External Tracking Path) Measured from input terminals to output terminals, shortest distance path along body Min. Internal Clearance 1.0 mm (Internal Plastic Gap) Through insulation distance conductor to conductor, usually the direct distance between the photoemitter and photodetector inside the optocoupler cavity Min. Internal Creepage 4.0 mm (Internal Tracking Path) The shortest distance around the border between two different insulating materials measured between the emitter and detector Comparative Tracking Index CTI Isolation Group 200 V IIIa DIN IEC 112/VDE 0303 PART 1 Material group (DIN VDE 0110) Option 300 – surface mount classification is Class A in accordance with CECC 00802. IEC/EN/DIN EN 60747-5-2 Insulation Characteristics (Option #050 Only) Description Symbol Installation classification per DIN VDE 0110/1.89, Table 1 For rated mains voltage ≤600 V rms For rated mains voltage ≤1000 V rms Characteristic I-IV I-III Climatic Classification (DIN IEC 68 part 1) 55/100/21 Pollution Degree (DIN VDE 0110 Part 1/1.89) 2 Maximum Working Insulation Voltage Unit VIORM 1414 V peak Input to Output Test Voltage, Method b* VPR = 1.875 x VIORM, 100% Production Test with tm = 1 sec, Partial Discharge < 5 pC VPR 2651 V peak Input to Output Test Voltage, Method a* VPR = 1.5 x VIORM, Type and sample test, tm = 60 sec, Partial Discharge < 5 pC VPR 2121 V peak VIOTM 8000 V peak TS IS PS,OUTPUT 150 400 700 °C mA mW RS >109 Ω Highest Allowable Overvoltage* (Transient Overvoltage, tini = 10 sec) Safety-Limiting Values (Maximum values allowed in the event of a failure, also see Figure 11) Case Temperature Current (Input Current IF, PS = 0) Output Power Insulation Resistance at TS, VIO = 500 V *Refer to the front of the Optocoupler section of the current catalog for a more detailed description of IEC/EN/DIN EN 60747-5-2 and other prod‑ uct safety regulations. Note: Optocouplers providing safe electrical separation per IEC/EN/DIN EN 60747-5-2 do so only within the safety-limiting values to which they are qualified. Protective cut-out switches must be used to ensure that the safety limits are not exceeded. 6 Absolute Maximum Ratings Storage Temperature...............................................................................................-55°C to +125°C Operating Temperature (TA).................................................................................. -55°C to +100°C Junction Temperature (TJ).......................................................................................................... 125°C Reflow Temperature Profile...............................................See Package Outline Drawings Section Lead Solder Temperature.............................................................................................260°C for 10s (up to seating plane) Average Input Current - IF ......................................................................................................... 25 mA Peak Input Current - IF ................................................................................................................ 40 mA (50 ns maximum pulse width) Reverse Input Voltage - VR .............................................................................................................2.5 V (IR = 100 µA, Pin 1-2) Input Power Dissipation..................................................................................... 60 mW @ TA = 85°C (Derate at 2.2 mW/°C for operating temperatures above 85°C) Reverse Output Photodiode Voltage.........................................................................................30 V (Pin 6-5) Reverse Input Photodiode Voltage.............................................................................................30 V (Pin 3-4) Recommended Operating Conditions Storage Temperature..................................................................................................-40°C to +85°C Operating Temperature.............................................................................................-40°C to +85°C Average Input Current - IF ................................................................................................... 1 - 20 mA Peak Input Current - IF ................................................................................................................ 35 mA (50% duty cycle, 1 ms pulse width) Reverse Output Photodiode Voltage...................................................................................0 - 15 V (Pin 6-5) Reverse Input Photodiode Voltage.......................................................................................0 - 15 V (Pin 3-4) 7 Electrical Specifications TA = 25°C unless otherwise specified. Parameter Symbol Device Min. Typ. Max. Units Test Conditions Fig. Note Transfer Gain K3 HCNR200 0.85 1.00 1.15 5 nA < IPD < 50 µA, 0 V < VPD < 15 V 2,3 1 HCNR201 0.95 1.00 1.05 5 nA < IPD < 50 µA, 0 V < VPD < 15 V 1 HCNR201 0.93 1.00 1.07-40°C < TA < 85°C, 5 nA < IPD < 50 µA, 0 V < VPD < 15 V 1 Temperature ∆K3 /∆TA -65 ppm/°C -40°C < TA < 85°C, 2,3 Coefficient of 5 nA < IPD < 50 µA, Transfer Gain 0 V < VPD < 15 V DC NonLinearity NLBF HCNR200 0.01 0.25 % (Best Fit) HCNR201 0.01 0.05 5 nA < IPD < 50 µA, 0 V < VPD < 15 V 4,5, 6 2 5 nA < IPD < 50 µA, 0 V < VPD < 15 V 2 HCNR2010.01 0.07-40°C < TA < 85°C, 5 nA < IPD < 50 µA, 0 V < VPD < 15 V 2 DC Nonlinearity NLEF 0.016 (Ends Fit) % 5 nA < IPD < 50 µA, 0 V < VPD < 15 V Input Photo- K1 HCNR200 0.25 0.50 0.75 % diode Current Transfer Ratio HCNR201 0.36 0.48 0.72 (IPD1/IF) IF = 10 mA, 0 V < VPD1 < 15 V Temperature ∆K1/∆TA-0.3 %/°C-40°C < TA < 85°C, Coefficient IF = 10 mA of K1 0 V < VPD1 < 15 V 7 Photodiode Leakage Current 8 Photodiode Reverse Breakdown Voltage Photodiode Capacitance ILK 0.5 25 nA BVRPD 30 150 V CPD 22 pF IF = 0 mA, 0 V < VPD < 15 V IR = 100 µA VPD = 0 V LED Forward VF 1.3 1.6 1.85 V IF = 10 mA Voltage 1.2 1.6 1.95 IF = 10 mA, -40°C < TA < 85°C LED Reverse Breakdown Voltage Temperature Coefficient of Forward Voltage BVR 2.5 9 V ∆VF /∆TA-1.7 mV/°C LED Junction CLED 80 pF Capacitance 8 7 IF = 100 µA IF = 10 mA f = 1 MHz, VF = 0 V 9, 10 3 AC Electrical Specifications TA = 25°C unless otherwise specified. Test Parameter Symbol Device Min. Typ. Max. Units Conditions LED Bandwidth f -3dB 9 MHz Application Circuit Bandwidth: High Speed High Precision 1.5 10 Application Circuit: IMRR High Speed 95 Note 16 17 6 6 16 6, 7 IF = 10 mA MHz kHz dB Fig. freq = 60 Hz Package Characteristics TA = 25°C unless otherwise specified. Test Parameter Symbol Device Min. Typ. Max. Units Conditions Fig. Note Input-Output VISO 5000 V rms Momentary-Withstand Voltage* RH ≤50%, t = 1 min. 4, 5 Resistance (Input-Output) VO = 500 VDC 4 1011TA = 100°C, VIO = 500 VDC 4 Capacitance (Input-Output) RI-O1012 CI-O 1013 0.4 0.6 Ω pF f = 1 MHz 4 *The Input-Output Momentary Withstand Voltage is a dielectric voltage rating that should not be interpreted as an input-output continuous voltage rating. For the continuous voltage rating refer to the VDE 0884 Insulation Characteristics Table (if applicable), your equipment level safety specification, or Application Note 1074, “Optocoupler Input-Output Endurance Voltage.” Notes: 1. K3 is calculated from the slope of the best fit line of IPD2 vs. IPD1 with eleven equally distributed data points from 5 nA to 50 µA. This is approxi‑ mately equal to IPD2/IPD1 at IF = 10 mA. 2. BEST FIT DC NONLINEARITY (NLBF) is the maximum deviation expressed as a percentage of the full scale output of a “best fit” straight line from a graph of IPD2 vs. IPD1 with eleven equally distributed data points from 5 nA to 50 µA. IPD2 error to best fit line is the deviation below and above the best fit line, expressed as a percentage of the full scale output. 3. ENDS FIT DC NONLINEARITY (NLEF) is the maximum deviation expressed as a percentage of full scale output of a straight line from the 5 nA to the 50 µA data point on the graph of IPD2 vs. IPD1. 4. Device considered a two-terminal device: Pins 1, 2, 3, and 4 shorted together and pins 5, 6, 7, and 8 shorted together. 5. In accordance with UL 1577, each optocoupler is proof tested by applying an insulation test voltage of ≥6000 V rms for ≥1 second (leakage detection current limit, II-O of 5 µA max.). This test is performed before the 100% production test for partial discharge (method b) shown in the IEC/EN/DIN EN 60747-5-2 Insulation Characteris-tics Table (for Option #050 only). 6. Specific performance will depend on circuit topology and components. 7. IMRR is defined as the ratio of the signal gain (with signal applied to VIN of Figure 16) to the isolation mode gain (with VIN connected to input common and the signal applied between the input and output commons) at 60 Hz, expressed in dB. 9 1.04 1.02 1.00 0.98 10.0 20.0 30.0 40.0 50.0 0.005 0.0 -0.005 -0.01 = DELTA K3 MEAN = DELTA K3 MEAN ± 2 • STD DEV -0.015 -0.02 -55 60.0 -25 5 Figure 2. Normalized K3 vs. input IPD. 0.02 0.015 0.01 0 V < VPD < 15 V 5 nA < IPD < 50 µA -25 5 35 65 95 125 DELTA NLBF – DRIFT OF BEST-FIT NL – % PTS NLBF – BEST-FIT NON-LINEARITY – % 0.025 0.00 -55 0.01 0.005 0.0 -0.005 -0.01 -0.015 -0.02 -55 = DELTA NLBF MEAN = DELTA NLBF MEAN ± 2 • STD DEV -25 5 65 95 125 TA = 25 °C, 0 V < VPD < 15 V -0.03 0.0 10.0 20.0 6.0 4.0 2.0 65 95 Figure 8. Typical photodiode leakage vs. temperature. CNR200 fig 8 125 50.0 60.0 1.2 -55°C 1.1 -40°C 1.0 25°C 0.9 85°C 100°C 0.8 0.7 0.6 0.5 NORMALIZED TO K1 CTR AT IF = 10 mA, TA = 25°C 0 V < VPD1 < 15 V 0.4 0.3 0.2 0.0 2.0 4.0 6.0 8.0 10.0 12.0 14.0 16.0 HCNR200 fig 7 1.8 10 1 0.1 0.01 0.001 0.0001 1.20 40.0 Figure 7. Input photodiode CTR vs. LED input current. TA = 25°C 8.0 30.0 IF – LED INPUT CURRENT – mA 100 VPD = 15 V IF – FORWARD CURRENT – mA ILK – PHOTODIODE LEAKAGE – nA 35 HCNR200 fig 6 TA – TEMPERATURE – °C 10 -0.02 HCNR200 fig 4 Figure 6. NLBF drift vs. temperature. 10.0 35 -0.01 Figure 4. IPD2 error vs. input IPD (see note 4). 0 V < VPD < 15 V 5 nA < IPD < 50 µA HCNR200 fig 5 5 0.00 TA – TEMPERATURE – °C Figure 5. NLBF vs. temperature. -25 0.01 IPD1 – INPUT PHOTODIODE CURRENT – µA 0.02 0.015 TA – TEMPERATURE – °C 0.0 -55 125 = ERROR MEAN = ERROR MEAN ± 2 • STD DEV 0.02 HCNR200 fig 3 = NLBF 50TH PERCENTILE = NLBF 90TH PERCENTILE 0.005 95 Figure 3. K3 drift vs. temperature. HCNR200 fig 2 0.03 65 0.03 TA – TEMPERATURE – °C IPD1 – INPUT PHOTODIODE CURRENT – µA 0.035 35 NORMALIZED K1 – INPUT PHOTODIODE CTR 0.94 0.0 NORMALIZED TO BEST-FIT K3 AT TA = 25°C, 0 V < VPD < 15 V 0 V < VPD < 15 V 0.01 VF – LED FORWARD VOLTAGE – V 0.96 0.015 IPD2 ERROR FROM BEST-FIT LINE (% OF FS) 0.02 = NORM K3 MEAN = NORM K3 MEAN ± 2 • STD DEV DELTA K3 – DRIFT OF K3 TRANSFER GAIN NORMALIZED K3 – TRANSFER GAIN 1.06 1.30 1.40 1.50 1.60 VF – FORWARD VOLTAGE – VOLTS Figure 9. LED input current vs. forward voltage. CNR200 fig 9 IF = 10 mA 1.7 1.6 1.5 1.4 1.3 1.2 -55 -25 5 35 65 95 125 TA – TEMPERATURE – °C Figure 10. LED forward voltage vs. temperature. HCNR200 fig 10 1000 PS OUTPUT POWER – mV IS INPUT CURRENT – mA 900 800 700 600 500 400 300 200 100 0 0 25 50 75 100 125 150 175 TS – CASE TEMPERATURE – °C Figure 11. Thermal derating curve dependence of safety limiting value with case temperature per IEC/EN/DIN EN 60747-5-2. CNR200 fig 11 R2 VIN R1 IPD1 PD1 + A1 LED IPD2 A2 + PD2 VOUT IF A) BASIC TOPOLOGY VCC VIN C1 R1 - A1 + PD1 R2 C2 LED R3 PD2 PD2 A2 + VOUT B) PRACTICAL CIRCUIT Figure 12. Basic isolation amplifier. CNR200 fig 12 VCC VIN - - + + A) POSITIVE INPUT VOUT B) POSITIVE OUTPUT VIN - - + + C) NEGATIVE INPUT D) NEGATIVE OUTPUT Figure 13. Unipolar circuit topologies. CNR200 fig 13 11 VOUT VCC1 VCC2 VCC1 IOS1 IOS2 VIN - - + + VOUT A) SINGLE OPTOCOUPLER VCC + VIN VOUT + + B) DUAL OPTOCOUPLER Figure 14. Bipolar circuit topologies. CNR200 fig 14 R2 +IIN R1 D1 - VOUT PD2 PD1 + + LED R3 -IIN A) RECEIVER VCC VIN R1 LED +IOUT R2 PD1 + D1 Q1 + PD2 R3 B) TRANSMITTER Figure 15. Loop-powered 4-20 mA current loop circuits. CNR200 fig 15 12 -IOUT VCC2 +5 V VCC1 +5 V LED R3 10 K VIN R5 10 K R2 68 K R7 470 VOUT Q2 2N3904 R1 68 K Q1 2N3906 Q4 2N3904 Q3 2N3906 R4 10 PD1 R6 10 PD2 Figure 16. High-speed low-cost analog isolator. VCC1 +15 V VCC2 +15 V CNR200 fig 16 C3 0.1µ C5 0.1µ R4 2.2 K R5 270 Q1 2N3906 INPUT BNC R6 6.8 K C1 47 P R1 200 K 1% C2 33 P 7 C4 0.1µ 6 LT1097 50 K - 2 A2 3 + 4 PD2 C6 0.1µ R3 33 K VEE1 -15 V R2 1% 7 2 6 3 A1 LT1097 + 4 PD1 174 K D1 1N4150 LED VEE2 -15 V Figure 17. Precision analog isolation amplifier. CNR200 fig 17 C3 C1 R6 180 K R2 180 K OC1 PD1 VIN R1 50 K BALANCE R3 180 K VCC1 VEE1 +15 V -15 V Figure 18. Bipolar isolation amplifier. 13 D1 + R4 680 OC1 LED OC2 LED + D2 C2 10 pf R5 680 R 50 K GAIN OC1 PD2 + OC2 PD1 10 pf 10 pf OC2 PD2 VMAG OUTPUT BNC C3 C1 R5 180 K D1 GAIN + R1 220 K R6 50 K D3 - VIN 10 pf 10 pf OC1 PD1 R2 10 K R4 680 R3 4.7 K VMAG + OC1 LED + D2 OC1 PD2 D4 + C2 VCC 10 pf + - R7 6.8 K R8 2.2 K V IGN VCC1 VEE1 +15 V -15 V Figure 19. Magnitude/sign isolation amplifier. Figure 20. SPICE model listing. 14 OC2 6N13 0.001 µF +ILOOP HCNR200 LED R1 10 kΩ Z1 5.1 V R4 100 Ω HCNR200 PD1 + - 2N3906 + VOUT LM158 HCNR200 PD2 0.001 µF -ILOOP VCC 5.5 V 0.1 µF LM158 - R2 10 kΩ R5 80 kΩ R3 25 Ω 2 Design Equations: VOUT / ILOOP = K3 (R5 R3) / R1 + R3) K3 = K2 / K1 = Constant = 1 HCNR200 fig 21 Note: The two OP-AMPS shown are two separate LM158, and not two channels in a single dual package, otherwise the loop side and output side will not be properly isolated. Figure 21. 4 to 20 mA HCNR200 receiver circuit. Q3 2N3904 Vcc 5.5V R2 150 Ω Q2 IC3 + LM158 C3 100nF C1 1nF Q4 Vin 0.8V~4V “0” @ 2200Hz “1” @ 1200Hz PD1/IC1 + LM158 IC2 Q1 2N3906 R4 10k Ω 1nF C2 PD2/IC1 R3 10k Ω Design Equations: (ILOOP/Vin)=K3(R5+R3)/(R5R1) K3 = K2/K1 = Constant ≈ 1 Note: The two OP-AMPS shown are two separate LM158 IC’s, and NOT dual channels in a single package, otherwise, the LOOP side and input side will not be properly isolated; The 5V1 Zener should be properly selected to ensure that it conducts at 187µA; Figure 22. 4 to 20 mA HCNR200 transmitter circuit. 15 2N3904 5V1 R7 3k2Ω 2N3904 LED/IC1 HCNR200 R1 80k Ω +I LOOP 12V~40V 4 ~ 20mA 4mA (Vin=0.8V) 20mA(Vin=4V) R8 100kΩ R6 150 Ω R5 25 Ω - I LOOP Theory of Operation Figure 1 illustrates how the HCNR200/201 high-linearity optocoupler is configured. The basic optocoupler con‑ sists of an LED and two photodiodes. The LED and one of the photodiodes (PD1) is on the input leadframe and the other photodiode (PD2) is on the output leadframe. The package of the optocoupler is constructed so that each photodiode receives approximately the same amount of light from the LED. Notice that IPD1 depends ONLY on the input voltage and the value of R1 and is independent of the light output characteristics of the LED. As the light output of the LED changes with temperature, amplifier A1 adjusts IF to compensate and maintain a constant current in PD1. Also notice that IPD1 is exactly proportional to VIN, giving a very linear relationship between the input voltage and the photodiode current. An external feedback amplifier can be used with PD1 to monitor the light output of the LED and automatically adjust the LED current to compensate for any non-linear‑ ities or changes in light output of the LED. The feedback amplifier acts to stabilize and linearize the light output of the LED. The output photodiode then converts the stable, linear light output of the LED into a current, which can then be converted back into a voltage by another amplifier. The relationship between the input optical power and the output current of a photodiode is very linear. There‑ fore, by stabilizing and linearizing IPD1, the light output of the LED is also stabilized and linearized. And since light from the LED falls on both of the photodiodes, IPD2 will be stabilized as well. Figure 12a illustrates the basic circuit topology for implement ing a simple isolation amplifier using the HCNR200/201 optocoupler. Besides the optocoupler, two external op-amps and two resistors are required. This simple circuit is actually a bit too simple to function properly in an actual circuit, but it is quite useful for ex‑ plaining how the basic isolation amplifier circuit works (a few more components and a circuit change are required to make a practical circuit, like the one shown in Figure 12b). The operation of the basic circuit may not be immedi‑ ately obvious just from inspecting Figure 12a, particu‑ larly the input part of the circuit. Stated briefly, amplifier A1 adjusts the LED current (IF), and therefore the current in PD1 (IPD1), to maintain its “+” input terminal at 0 V. For example, increasing the input voltage would tend to in‑ crease the voltage of the “+” input terminal of A1 above 0 V. A1 amplifies that increase, causing IF to increase, as well as IPD1. Because of the way that PD1 is connected, IPD1 will pull the “+” terminal of the op-amp back toward ground. A1 will continue to increase IF until its “+” termi‑ nal is back at 0 V. Assuming that A1 is a perfect op-amp, no current flows into the inputs of A1; therefore, all of the current flowing through R1 will flow through PD1. Since the “+” input of A1 is at 0 V, the current through R1, and therefore IPD1 as well, is equal to VIN/R1. Essentially, amplifier A1 adjusts IF so that IPD1 = VIN/R1. 16 The physical construction of the package determines the relative amounts of light that fall on the two photodiodes and, therefore, the ratio of the photodiode currents. This results in very stable operation over time and tempera‑ ture. The photodiode current ratio can be expressed as a constant, K, where K = IPD2/IPD1. Amplifier A2 and resistor R2 form a trans-resistance am‑ plifier that converts IPD2 back into a voltage, VOUT, where VOUT = IPD2*R2. Combining the above three equations yields an overall expression relating the output voltage to the input volt‑ age, VOUT /VIN = K*(R2/R1). Therefore the relationship between VIN and VOUT is con stant, linear, and independent of the light output characteristics of the LED. The gain of the basic isolation amplifier circuit can be adjusted simply by adjusting the ratio of R2 to R1. The parameter K (called K3 in the electri‑ cal specifications) can be thought of as the gain of the optocoupler and is specified in the data sheet. Remember, the circuit in Figure 12a is simplified in order to explain the basic circuit operation. A practical circuit, more like Figure 12b, will require a few additional compo nents to stabilize the input part of the circuit, to limit the LED current, or to optimize circuit performance. Example application circuits will be discussed later in the data sheet. Circuit Design Flexibility Circuit design with the HCNR200/201 is very flexible because the LED and both photodiodes are accessible to the designer. This allows the designer to make perf ormance trade-offs that would otherwise be difficult to make with commercially available isolation amplifiers (e.g., bandwidth vs. accuracy vs. cost). Analog isolation circuits can be designed for applications that have either unipolar (e.g., 0-10 V) or bipolar (e.g., ±10 V) signals, with positive or negative input or output voltages. Several simplified circuit topologies illustrating the design flex‑ ibility of the HCNR200/201 are discussed below. to worry about. However, the second circuit requires two optocouplers, separate gain adjustments for the posi‑ tive and negative portions of the signal, and can exhibit crossover distortion near zero volts. The correct circuit to choose for an application would depend on the require‑ ments of that particular application. As with the basic isolation amplifier circuit in Figure 12a, the circuits in Fig‑ ure 14 are simplified and would require a few additional components to function properly. Two example circuits that operate with bipolar input signals are discussed in the next section. The circuit in Figure 12a is configured to be non-invert‑ ing with positive input and output voltages. By simply changing the polarity of one or both of the photodiodes, the LED, or the op-amp inputs, it is possible to implement other circuit configurations as well. Figure 13 illustrates how to change the basic circuit to accommodate both positive and negative input and output voltages. The in‑ put and output circuits can be matched to achieve any combination of positive and negative voltages, allowing for both inverting and non-inverting circuits. As a final example of circuit design flexibility, the simpli‑ fied schematics in Figure 15 illustrate how to implement 4-20 mA analog current-loop transmitter and receiver circuits using the HCNR200/201 optocoupler. An impor‑ tant feature of these circuits is that the loop side of the circuit is powered entirely by the loop current, eliminat‑ ing the need for an isolated power supply. All of the configurations described above are unipolar (single polarity); the circuits cannot accommodate a sig‑ nal that might swing both positive and negative. It is pos‑ sible, however, to use the HCNR200/201 optocoupler to implement a bipolar isolation amplifier. Two topologies that allow for bipolar operation are shown in Figure 14. The circuit in Figure 14a uses two current sources to offset the signal so that it appears to be unipolar to the optocoupler. Current source IOS1 provides enough offset to ensure that IPD1 is always positive. The second current source, IOS2, provides an offset of opposite polarity to ob‑ tain a net circuit offset of zero. Current sources IOS1 and IOS2 can be implemented simply as resistors connected to suitable voltage sources. The circuit in Figure 14b uses two optocouplers to obtain bipolar operation. The first optocoupler handles the pos‑ itive voltage excursions, while the second optocoupler handles the negative ones. The output photodiodes are connected in an antiparallel configuration so that they produce output signals of opposite polarity. The first circuit has the obvious advantage of requiring only one optocoupler; however, the offset performance of the circuit is dependent on the matching of IOS1 and IOS2 and is also dependent on the gain of the optocoupler. Changes in the gain of the optocoupler will directly af‑ fect the offset of the circuit. The offset performance of the second circuit, on the other hand, is much more stable; it is independent of optocoupler gain and has no matched current sources 17 The input and output circuits in Figure 15a are the same as the negative input and positive output circuits shown in Figures 13c and 13b, except for the addition of R3 and zener diode D1 on the input side of the circuit. D1 regu‑ lates the supply voltage for the input amplifier, while R3 forms a current divider with R1 to scale the loop current down from 20 mA to an appropriate level for the input circuit (<50 µA). As in the simpler circuits, the input amplifier adjusts the LED current so that both of its input terminals are at the same voltage. The loop current is then divided between R1 and R3. IPD1 is equal to the current in R1 and is given by the following equation: IPD1 = ILOOP*R3/(R1+R3). Combining the above equation with the equations used for Figure 12a yields an overall expression relating the output voltage to the loop current, VOUT/ILOOP = K*(R2*R3)/(R1+R3). Again, you can see that the relationship is constant, lin‑ ear, and independent of the characteristics of the LED. The 4-20 mA transmitter circuit in Figure 15b is a little dif‑ ferent from the previous circuits, particularly the output circuit. The output circuit does not directly generate an output voltage which is sensed by R2, it instead uses Q1 to generate an output current which flows through R3. This output current generates a voltage across R3, which is then sensed by R2. An analysis similar to the one above yields the following expression relating output current to input voltage: ILOOP /VIN = K*(R2+R3)/(R1*R3). The preceding circuits were presented to illustrate the flexibility in designing analog isolation circuits using the HCNR200/201. The next section presents several com‑ plete schematics to illustrate practical applications of the HCNR200/201. Example Application Circuits The circuit shown in Figure 16 is a high-speed low-cost circuit designed for use in the feedback path of switchmode power supplies. This application requires good bandwidth, low cost and stable gain, but does not re‑ quire very high accuracy. This circuit is a good example of how a designer can trade off accuracy to achieve improvements in bandwidth and cost. The circuit has a bandwidth of about 1.5 MHz with stable gain character‑ istics and requires few external components. Although it may not appear so at first glance, the circuit in Figure 16 is essentially the same as the circuit in Fig‑ ure 12a. Amplifier A1 is comprised of Q1, Q2, R3 and R4, while amplifier A2 is comprised of Q3, Q4, R5, R6 and R7. The circuit operates in the same manner as well; the only difference is the performance of amplifiers A1 and A2. The lower gains, higher input currents and higher offset voltages affect the accuracy of the circuit, but not the way it operates. Because the basic circuit operation has not changed, the circuit still has good gain stability. The use of discrete transistors instead of op-amps allowed the design to trade off accuracy to achieve good band‑ width and gain stability at low cost. To get into a little more detail about the circuit, R1 is se‑ lected to achieve an LED current of about 7-10 mA at the nominal input operating voltage according to the fol‑ lowing equation: IF = (VIN/R1)/K1, where K1 (i.e., IPD1/IF) of the optocoupler is typically about 0.5%. R2 is then selected to achieve the desired output voltage according to the equation, VOUT/VIN = R2/R1. The purpose of R4 and R6 is to improve the dynamic re‑ sponse (i.e., stability) of the input and output circuits by lowering the local loop gains. R3 and R5 are selected to provide enough current to drive the bases of Q2 and Q4. And R7 is selected so that Q4 operates at about the same collector current as Q2. The next circuit, shown in Figure 17, is designed to achieve the highest possible accuracy at a reasonable cost. The high accuracy and wide dynamic range of the circuit is achieved by using low-cost precision op-amps with very low input bias currents and offset voltages and is limited by the performance of the optocoupler. The circuit is de‑ signed to operate with input and output voltages from 1 mV to 10 V. 18 The circuit operates in the same way as the others. The only major differences are the two compensation capaci‑ tors and additional LED drive circuitry. In the high-speed circuit discussed above, the input and output circuits are stabilized by reducing the local loop gains of the input and output circuits. Because reducing the loop gains would decrease the accuracy of the circuit, two compen‑ sation capacitors, C1 and C2, are instead used to improve circuit stability. These capacitors also limit the bandwidth of the circuit to about 10 kHz and can be used to reduce the output noise of the circuit by reducing its bandwidth even further. The additional LED drive circuitry (Q1 and R3 through R6) helps to maintain the accuracy and bandwidth of the circuit over the entire range of input voltages. Without these components, the transconductance of the LED driver would decrease at low input voltages and LED currents. This would reduce the loop gain of the input circuit, reducing circuit accuracy and bandwidth. D1 pre‑ vents excessive reverse voltage from being applied to the LED when the LED turns off completely. No offset adjustment of the circuit is necessary; the gain can be adjusted to unity by simply adjusting the 50 kohm potentiometer that is part of R2. Any OP-97 type of opamp can be used in the circuit, such as the LT1097 from Linear Technology or the AD705 from Analog Devices, both of which offer pA bias currents, µV offset voltages and are low cost. The input terminals of the op-amps and the photodiodes are connected in the circuit using Kelvin connections to help ensure the accuracy of the circuit. The next two circuits illustrate how the HCNR200/201 can be used with bipolar input signals. The isolation amplifier in Figure 18 is a practical implementation of the circuit shown in Figure 14b. It uses two optocouplers, OC1 and OC2; OC1 handles the positive portions of the input sig‑ nal and OC2 handles the negative portions. Diodes D1 and D2 help reduce crossover distortion by keeping both amplifiers active during both positive and negative portions of the input signal. For example, when the input signal positive, optocoupler OC1 is active while OC2 is turned off. However, the amplifier controlling OC2 is kept active by D2, allowing it to turn on OC2 more rap‑ idly when the input signal goes negative, thereby reduc‑ ing crossover distortion. Balance control R1 adjusts the relative gain for the posi‑ tive and negative portions of the input signal, gain con‑ trol R7 adjusts the overall gain of the isolation amplifier, and capacitors C1-C3 provide compensation to stabilize the amplifiers. The final circuit shown in Figure 19 isolates a bipolar analog signal using only one optocoupler and generates two output signals: an analog signal proportional to the magnitude of the input signal and a digital signal cor‑ responding to the sign of the input signal. This circuit is especially useful for applications where the output of the circuit is going to be applied to an analog-to-digital converter. The primary advantages of this circuit are very good linearity and offset, with only a single gain adjust‑ ment and no offset or balance adjustments. HCNR200/201 SPICE Model Figure 20 is the net list of a SPICE macro-model for the HCNR200/201 high-linearity optocoupler. The macromodel accurately reflects the primary characteristics of the HCNR200/201 and should facilitate the design and understanding of circuits using the HCNR200/201 opto‑ coupler. To achieve very high linearity for bipolar signals, the gain should be exactly the same for both positive and negative input polarities. This circuit achieves excellent linearity by using a single optocoupler and a single input resistor, which guarantees identical gain for both posi‑ tive and negative polarities of the input signal. This pre‑ cise matching of gain for both polarities is much more difficult to obtain when separate components are used for the different input polarities, such as is the previous circuit. The circuit in Figure 19 is actually very similar to the pre‑ vious circuit. As mentioned above, only one optocoupler is used. Because a photodiode can conduct current in only one direction, two diodes (D1 and D2) are used to steer the input current to the appropriate terminal of input photodiode PD1 to allow bipolar input currents. Normally the forward voltage drops of the diodes would cause a serious linearity or accuracy problem. However, an additional amplifier is used to provide an appropriate offset voltage to the other amplifiers that exactly cancels the diode voltage drops to maintain circuit accuracy. Diodes D3 and D4 perform two different functions; the diodes keep their respective amplifiers active indepen‑ dent of the input signal polarity (as in the previous cir‑ cuit), and they also provide the feedback signal to PD1 that cancels the voltage drops of diodes D1 and D2. Either a comparator or an extra op-amp can be used to sense the polarity of the input signal and drive an inex‑ pensive digital optocoupler, like a 6N139. It is also possible to convert this circuit into a fully bipolar circuit (with a bipolar output signal) by using the output of the 6N139 to drive some CMOS switches to switch the polarity of PD2 depending on the polarity of the input signal, obtaining a bipolar output voltage swing. For product information and a complete list of distributors, please go to our website: www.avagotech.com Avago, Avago Technologies, and the A logo are trademarks of Avago Technologies in the United States and other countries. Data subject to change. Copyright © 2005-2014 Avago Technologies. All rights reserved. Obsoletes AV01-0567EN AV02-0886EN - July 1, 2014