LINER LTC3850IGN-2-PBF Dual, 2-phase synchronous step-down switching controller Datasheet

LTC3850-2
Dual, 2-Phase
Synchronous Step-Down
Switching Controller
DESCRIPTION
FEATURES
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The LTC®3850-2 is a high performance dual synchronous
step-down switching regulator controller that drives all
N-channel power MOSFET stages. A constant-frequency
current mode architecture allows a phase-lockable
frequency of up to 780kHz. Power loss and supply noise
are minimized by operating the two controller output
stages out of phase.
Dual, 180° Phased Controllers Reduce Required
Input Capacitance and Power Supply Induced Noise
High Efficiency: Up to 95%
RSENSE or DCR Current Sensing
±1% 0.8V Output Voltage Accuracy
Phase-Lockable Fixed Frequency 250kHz to 780kHz
Supports Pre-Biased Output
Dual N-Channel MOSFET Synchronous Drive
Wide VIN Range: 4V to 30V Operation
Adjustable Soft-Start Current Ramping or Tracking
Foldback Output Current Limiting
Output Overvoltage Protection
Power Good Output Voltage Monitor
28-Pin Narrow SSOP Package
OPTI-LOOP® compensation allows the transient response
to be optimized over a wide range of output capacitance
and ESR values. The LTC3850-2 features a precision 0.8V
reference and a power good output indicator. A wide 4V
to 30V input supply range encompasses most battery
chemistries and intermediate bus voltages.
Independent TK/SS pins for each controller ramp the
output voltages during start-up. Current foldback limits
MOSFET heat dissipation during short-circuit conditions. The MODE/PLLIN pin selects among Burst Mode®
operation, pulse-skipping mode, or continuous inductor
current mode and allows the IC to be synchronized to an
external clock.
APPLICATIONS
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Notebook and Palmtop Computers
Portable Instruments
Battery-Operated Digital Devices
DC Power Distribution Systems
L, LT, LTC, LTM, OPTI-LOOP and Burst Mode are registered trademarks of Linear Technology
Corporation. All other trademarks are the property of their respective owners. Protected by U.S.
Patents, including 5481178, 5705919, 5929620, 6100678, 6144194, 6177787, 6304066, 6580258.
The LTC3850-2 is identical to the LTC3850-1, except they
have different pin assignments.
TYPICAL APPLICATION
High Efficiency Dual 3.3V/2.5V Step-Down Converter
Efficiency
22μF
50V
4.7μF
VIN PGOOD INTVCC
500kHz
MODE/PLLIN
EXTVCC
0.1μF
63.4k
20k
15k
TK/SS1
0.1μF
2.2μH
2.2k
PGND
TK/SS2
0.1μF
1000
80
75
70
100
65
POWER LOSS
55
0.1μF
43.2k
ITH2
SGND
85
60
SENSE2+
RUN2
SENSE2–
VFB2
ITH1
220pF
100μF
6V
90
FREQ/PLLFLTR
SENSE1+
RUN1
SENSE1–
VFB1
10000
EFFICIENCY
POWER LOSS (mW)
2.2k
VIN = 12V
95 VOUT = 3.3V
0.1μF
BOOST1
BOOST2
SW1
SW2
LTC3850-2
BG2
BG1
2.2μH
VOUT1
3.3V
5A
TG2
100
EFFICIENCY (%)
TG1
0.1μF
VIN
7V TO
26V
10nF
10k
VOUT2
2.5V
5A
220pF
15k
50
10
100
1000
LOAD CURRENT (mA)
10
10000
38502 TA01b
100μF
6V
20k
38502 TA01
38502f
1
LTC3850-2
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
Input Supply Voltage (VIN) ......................... 30V to –0.3V
Input Supply Transient Voltage (VIN) < 500ms, with
INTVCC ≥ 5V ........................................... 34V to –0.3V
Top Side Driver Voltages
BOOST1, BOOST2.................................. 34V to –0.3V
Switch Voltage (SW1, SW2) ......................... 30V to –5V
INTVCC , RUN1, RUN2, PGOOD, EXTVCC,
(BOOST1-SW1), (BOOST2-SW2) ................. 6V to –0.3V
SENSE1+, SENSE2+, SENSE1–,
SENSE2– Voltages..................................... 5.5V to –0.3V
MODE/PLLIN, TK/SS1,TK/SS2, FREQ/PLLFLTR
Voltages ................................................ INTVCC to –0.3V
ITH1 , ITH2 , VFB1 , VFB2 Voltages .................. 2.7V to –0.3V
INTVCC Peak Output Current ................................100mA
Operating Temperature Range (Note 2)....–40°C to 85°C
Junction Temperature (Note 3) ............................. 125°C
Storage Temperature Range...................–65°C to 125°C
Lead Temperature (Soldering, 10 sec)
(GN Package) .................................................... 300°C
TOP VIEW
RUN1
1
28 FREQ/PLLFLTR
SENSE1+
2
27 MODE/PLLIN
SENSE1–
3
26 SW1
VFB1
4
25 TG1
TK/SS1
5
24 BOOST1
ITH1
6
23 BG1
SGND
7
22 VIN
ITH2
8
21 INTVCC
TK/SS2
9
20 BG2
VFB2 10
19 PGND
SENSE2–
11
18 BOOST2
SENSE2+
12
17 TG2
RUN2 13
16 SW2
EXTVCC 14
15 PGOOD
GN PACKAGE
28-LEAD NARROW PLASTIC SSOP
TJMAX = 125°C, θJA = 95°C/W
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3850IGN-2#PBF
LTC3850IGN-2#TRPBF
LTC3850GN-2
28-Lead Narrow Plastic SSOP
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
38502f
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LTC3850-2
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C, VIN = 15V, VRUN1,2 = 5V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0.792
0.800
0.808
V
–10
–50
nA
0.002
0.02
%/V
0.01
–0.01
0.1
–0.1
%
%
Main Control Loops
VFB1,2
Regulated Feedback Voltage
ITH1,2 Voltage = 1.2V; (Note 4)
IFB1,2
Feedback Current
(Note 4)
VREFLNREG
Reference Voltage Line Regulation
VIN = 6V to 24V (Note 4)
VLOADREG
Output Voltage Load Regulation
(Note 4)
Measured in Servo Loop; ΔITH Voltage = 1.2V to 0.7V
Measured in Servo Loop; ΔITH Voltage = 1.2V to 1.6V
l
l
l
gm1,2
Transconductance Amplifier gm
ITH1,2 = 1.2V; Sink/Source 5μA; (Note 4)
2.2
IQ
Input DC Supply Current
Normal Mode
Shutdown
(Note 5)
VIN = 15V; EXTVCC Tied to VOUT1; VOUT1 = 5V
VRUN1,2 = 0V
850
30
UVLO
Undervoltage Lockout on INTVCC
VINTVCC Ramping Down
UVLOHYS
UVLO Hysteresis
DFMAX
Maximum Duty Factor
In Dropout
VOVL
Feedback Overvoltage Lockout
Measured at VFB1,2
ISENSE
Sense Pin Bias Current
(Each Channel) VSENSE1,2 = 3.3V
l
mmho
50
μA
μA
3
V
0.5
V
96
97.2
%
0.84
0.86
0.88
V
±1
±2
μA
ITK/SS1,2
Soft-Start Charge Current
VTK/SS1,2 = 0V
VRUN1,2
RUN Pin ON Threshold
VRUN1, VRUN2 Rising
VRUN1,2HYS
RUN Pin ON Hysteresis
VSENSE(MAX)
Maximum Current Sense Threshold
VFB1,2 = 0.7V, VSENSE1,2 = 3.3V
TG RUP
TG Driver Pull-Up On-Resistance
TG High
TG RDOWN
TG Driver Pulldown On-Resistance
TG Low
1.5
Ω
BG RUP
BG Driver Pull-Up On-Resistance
BG High
2.4
Ω
BG RDOWN
BG Driver Pulldown On-Resistance
BG Low
1.1
Ω
TG1,2 tr
TG1,2 tf
TG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
BG1,2 tr
BG1,2 tf
BG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
TG/BG t1D
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
BG/TG t2D
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
tON(MIN)
Minimum On-Time
(Note 7)
90
ns
l
0.9
1.3
1.7
μA
1.1
1.22
1.35
V
80
l
40
50
mV
60
2.6
mV
Ω
INTVCC Linear Regulator
VINTVCC
Internal VCC Voltage
7V < VIN < 24V
VLDO INT
INTVCC Load Regulation
ICC = 0mA to 50mA
VEXTVCC
EXTVCC Switchover Voltage
EXTVCC Ramping Positive
VLDO EXT
EXTVCC Voltage Drop
ICC = 20mA, VEXTVCC = 5V
VLDOHYS
EXTVCC Hysteresis
4.8
l
4.5
5
5.2
V
0.5
2
%
100
mV
4.7
50
200
V
mV
38502f
3
LTC3850-2
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C, VIN = 15V, VRUN1,2 = 5V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Oscillator and Phase-Locked Loop
fNOM
Nominal Frequency
VFREQ = 1.2V
450
500
550
kHz
fLOW
Lowest Frequency
VFREQ = 0V
210
250
290
kHz
fHIGH
Highest Frequency
VFREQ ≥ 2.4V
700
780
860
kHz
RMODE/PLLIN
MODE/PLLIN Input Resistance
IFREQ
Phase Detector Output Current
Sinking Capability
Sourcing Capability
250
kΩ
fMODE < fOSC
fMODE > fOSC
–13
13
μA
μA
0.1
PGOOD Output
VPGL
PGOOD Voltage Low
IPGOOD = 2mA
IPGOOD
PGOOD Leakage Current
VPGOOD = 5V
VPG
PGOOD Trip Level
VFB with Respect to Set Regulated Voltage
VFB Ramping Negative
VFB Ramping Positive
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3850I-2 is guaranteed to meet performance specifications
over the –40°C to 85°C operating temperature range.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC3850IGN-2: TJ = TA + (PD • 95°C/W)
–5
5
– 7.5
7.5
0.3
V
±2
μA
–10
10
%
%
Note 4: The LTC3850I-2 is tested in a feedback loop that servos VITH1,2 to
a specified voltage and measures the resultant VFB1,2.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 7: The minimum on-time condition is specified for an inductor
peak-to-peak ripple current ≥40% of IMAX (see Minimum On-Time
Considerations in the Applications Information section).
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Output Current
and Mode
Efficiency vs Output Current
and Mode
100
90
BURST
DCM
50
40
30
DCM
60
50
40
30
CCM
20
20
10
10
0
0
10
95
70
100
1000
LOAD CURRENT (mA)
CIRCUIT OF FIGURE 14
10000
38502 G01
CCM
EFFICIENCY
100
1000
LOAD CURRENT (mA)
CIRCUIT OF FIGURE 14
10000
38502 G02
1500
1000
90
POWER LOSS
85
VIN = 12V
VOUT = 3.3V
10
2000
VOUT = 3.3V
IOUT = 2A
BURST
80
EFFICIENCY (%)
70
60
100
POWER LOSS (mW)
EFFICIENCY (%)
80
100
EFFICIENCY (%)
VIN = 12V
90 VOUT = 1.8V
Efficiency and Power Loss
vs Input Voltage
500
80
5
10
15
20
INPUT VOLTAGE (V)
0
25
38502 G03
CIRCUIT OF FIGURE 14
38502f
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LTC3850-2
TYPICAL PERFORMANCE CHARACTERISTICS
Load Step
(Burst Mode Operation)
Load Step
(Forced Continuous Mode)
ILOAD
2A/DIV
200mA TO 2.5A
ILOAD
2A/DIV
200mA TO 2.5A
IL
2A/DIV
IL
2A/DIV
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
40μs/DIV
38502 G04
CIRCUIT OF FIGURE 14
VIN = 12V, VOUT = 1.8V
Load Step
(Pulse-Skipping Mode)
Inductor Current at Light Load
FORCED
CONTINUOUS
MODE
2A/DIV
ILOAD
2A/DIV
200mA TO 2.5A
IL
2A/DIV
Burst Mode
OPERATION
2A/DIV
VOUT
100mV/DIV
AC COUPLED
PULSE-SKIPPING
MODE
2A/DIV
40μs/DIV
VOUT
2V/DIV
38502 G05
40μs/DIV
CIRCUIT OF FIGURE 14
VIN = 12V, VOUT = 1.8V
38502 G06
1μs/DIV
CIRCUIT OF FIGURE 14
VIN = 12V, VOUT = 1.8V
CIRCUIT OF FIGURE 14
VIN = 12V, VOUT = 1.8V
ILOAD = 100μA
Prebiased Output at 2V
Coincident Tracking
RUN1
2V/DIV
VTK/SS
500mV/DIV
38502 G07
VOUT1, 3.3V
3Ω LOAD, 1V/DIV
VOUT2, 1.8V
1.5Ω LOAD
1V/DIV
VFB
500mV/DIV
2.5ms/DIV
38502 G08
1ms/DIV
38502 G09
38502f
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LTC3850-2
TYPICAL PERFORMANCE CHARACTERISTICS
Tracking Up and Down
with External Ramp
Quiescent Current
vs Input Voltage without EXTVCC
INTVCC Line Regulation
5.25
5
TK/SS1
TK/SS2
2V/DIV
5.00
VOUT1
3.3V
3Ω LOAD
1V/DIV
VOUT2
1.8V
1.5Ω LOAD
1V/DIV
3
2
4.75
4.50
4.25
4.00
1
38502 G10
10ms/DIV
INTVCC VOLTAGE (V)
SUPPLY CURRENT (mA)
4
3.75
0
3.50
15
10
5
20
0
25
5
10
INPUT VOLTAGE (V)
15
20
38502 G11
VSENSE (mV)
60
40
20
0
–20
–40
0.5
1
VITH (V)
1.5
2
80
100
70
90
60
50
40
30
20
10
0
80
70
60
50
40
30
20
10
0
1
2
4
3
VSENSE COMMON MODE VOLTAGE (V)
38502 G13
5
0
0
20
40
60
DUTY CYCLE (%)
38502 G14
80
100
38502 G15
TK/SS Pull-Up Current
vs Temperature
Maximum Current Sense Voltage vs
Feedback Voltage (Current Foldback)
2.00
80
70
60
TK/SS CURRENT (μA)
MAXIMUM CURRENT SENSE VOLTAGE (mV)
Maximum Current Sense
Threshold vs Duty Cycle
CURRENT SENSE THRESHOLD (mV)
CURRENT SENSE THRESHOLD (mV)
80
0
38502 G12
Maximum Current Sense Threshold
vs Common Mode Voltage
Current Sense Threshold
vs ITH Voltage
25
INPUT VOLTAGE (V)
50
40
30
1.75
1.50
1.25
20
10
0
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
FEEDBACK VOLTAGE (V)
38502 G16
1.00
–50
–25
50
25
0
TEMPERATURE (°C)
75
100
38502 G17
38502f
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LTC3850-2
TYPICAL PERFORMANCE CHARACTERISTICS
Regulated Feedback Voltage
vs Temperature
Shutdown (RUN) Threshold
vs Temperature
RUN PIN VOLTAGE (V)
1.4
1.3
ON
1.2
OFF
1.1
1.0
–50
–25
50
25
0
TEMPERATURE (°C)
900
804
800
802
800
798
796
–25
50
25
0
TEMPERATURE (°C)
4
75
100
Shutdown Current
vs Input Voltage
50
40
INPUT CURRENT (μA)
FREQUENCY (kHz)
2
50
25
0
TEMPERATURE (°C)
–25
38502 G20
410
FALLING
VFREQ = 0V
200
–50
100
75
420
3
VFREQ = 1.2V
500
Oscillator Frequency
vs Input Voltage
5
400
390
30
20
10
1
380
50
25
0
TEMPERATURE (°C)
75
100
0
10
5
15
20
15
20
25
38502 G23
Quiescent Current
vs Temperature without EXTVCC
5
VIN = 15V
40
30
20
10
–25
10
38502 G22
Shutdown Current
vs Temperature
0
–50
5
INPUT VOLTAGE (V)
38502 G21
50
25
INPUT VOLTAGE (V)
QUIESCENT CURRENT (mA)
–25
SHUTDOWN CURRENT (μA)
INTVCC VOLTAGE (V)
600
38502 G19
Undervoltage Lockout Threshold
(INTVCC) vs Temperature
0
–50
700
300
38502 G18
RISING
VFREQ = INTVCC
400
794
–50
100
75
806
FREQUENCY (kHz)
REGULATED FEEDBACK VOLTAGE (mV)
1.5
Oscillator Frequency
vs Temperature
50
25
0
TEMPERATURE (°C)
75
100
38502 G24
4
3
2
1
0
–50
–25
50
25
0
TEMPERATURE (°C)
75
100
38502 G25
38502f
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LTC3850-2
PIN FUNCTIONS
RUN1, RUN2 (Pin 1, Pin 13): Run Control Inputs. A voltage
above 1.2V on either pin turns on the IC. However, forcing
either of these pins below 1.2V causes the IC to shut down
that particular channel. There are 0.5μA pull-up currents
for these pins. Once the RUN pin rises above 1.2V, an
additional 4.5μA pull-up current is added to the pin.
SENSE1+, SENSE2+ (Pin 2, Pin 12): Current Sense
Comparator Inputs. The (+) inputs to the current
comparators are normally connected to DCR sensing
networks or current sensing resistors.
SENSE1–, SENSE2– (Pin 3, Pin 11): Current Sense
Comparator Inputs. The (–) inputs to the current
comparators are connected to the outputs.
TK/SS1, TK/SS2 (Pin 5, Pin 9): Output Voltage Tracking
and Soft-Start Inputs. When one channel is configured to
be master of the two channels, a capacitor to ground at
this pin sets the ramp rate for the master channel’s output
voltage. When the channel is configured to be the slave
of two channels, the VFB voltage of the master channel is
reproduced by a resistor divider and applied to this pin.
Internal soft-start currents of 1.3μA charge the soft-start
capacitors.
ITH1, ITH2 (Pin 6, Pin 8): Current Control Thresholds and
Error Amplifier Compensation Points. Each associated
channels’ current comparator tripping threshold increases
with its ITH control voltage.
VFB1, VFB2 (Pin 4, Pin 10): Error Amplifier Feedback
Inputs. These pins receive the remotely sensed feedback
voltages for each channel from external resistive dividers
across the outputs.
SGND (Pin 7): Signal Ground. All small-signal components
and compensation components should connect to this
ground, which in turn connects to PGND at one point.
EXTVCC (Pin 14): External Power Input to an Internal Switch
Connected to INTVCC. This switch closes and supplies the
IC power, bypassing the internal low dropout regulator,
whenever EXTVCC is higher than 4.7V. Do not exceed 6V
on this pin and ensure VIN > VEXTVCC at all times.
PGOOD (Pin 15): Power Good Indicator Output. Open-drain
logic out that is pulled to ground when either channel
output exceeds the ±7.5% regulation window, after the
internal 17μs power bad mask timer expires.
PGND (Pin 19): Power Ground Pin. Connect this pin closely
to the sources of the bottom N-channel MOSFETs, the (–)
terminal of CVCC and the (–) terminal of CIN.
INTVCC (Pin 21): Internal 5V Regulator Output. The control circuits are powered from this voltage. Decouple this
pin to PGND with a 4.7μF low ESR tantalum or ceramic
capacitor.
VIN (Pin 22): Main Input Supply. Decouple this pin to
PGND with a capacitor (0.1μF to 1μF). For applications
where the main input power is 5V, tie the VIN and INTVCC
pins together.
BG1, BG2 (Pins 23, 20): Bottom Gate Driver Outputs. These
pins drive the gates of the bottom N-Channel MOSFETs
and swings between PGND and INTVCC.
BOOST1, BOOST2 (Pins 24, 18): Boosted Floating Driver
Supplies. The (+) terminal of the boost-strap capacitors
connect to these pins. These pins swing from a diode
voltage drop below INTVCC up to VIN + INTVCC.
TG1, TG2 (Pins 25, 17): Top Gate Driver Outputs. These are
the outputs of floating drivers with a voltage swing equal
to INTVCC superimposed on the switch nodes voltages.
SW1, SW2 (Pins 26, 16): Switch Node Connections to
Inductors. Voltage swing at these pins are from a body
diode voltage drop below ground to VIN.
MODE/PLLIN (Pin 27): Force Continuous Mode, Burst
Mode, or Pulse-Skipping Mode Selection Pin and
External Synchronization Input to Phase Detector Pin.
Connect this pin to SGND to force both channels into
the continuous mode of operation. Connect to INTVCC
to enable pulse-skipping mode of operation. Leaving the
pin floating will enable Burst Mode operation. A clock on
the pin will force the controller into continuous mode of
operation and synchronize the internal oscillator.
FREQ/PLLFLTR (Pin 28): The Phase-Locked Loop’s
Low-Pass Filter is Tied to This Pin. Alternatively, this pin
can be driven with a DC voltage to vary the frequency of
the internal oscillator.
38502f
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LTC3850-2
FUNCTIONAL DIAGRAM
FREQ/PLLFLTR
MODE/PLLIN
EXTVCC
VIN
VIN
4.7V
F
0.8V
5V
REG
+
–
PLL-SYNC
CIN
+
–
MODE/SYNC
DETECT
+
INTVCC
INTVCC
F
BOOST
OSC
BURSTEN
S
R
3k
+
ON
–
ICMP
IREV
+
–
CB
TG
FCNT
Q
M1
SW
SWITCH
LOGIC
AND
ANTISHOOT
THROUGH
L1
VOUT
SENSE+
DB
SENSE–
+
COUT
RUN
BG
OV
M2
CVCC
SLOPE COMPENSATION
PGND
PGOOD
INTVCC
UVLO
+
1
51k
ITHB
SLOPE RECOVERY
ACTIVE CLAMP
UV
R2
–
+
SLEEP
VIN
0.74V
VFB
R1
OV
–
–
–
+
SS
+
–
RUN
+
0.86V
SGND
1.3μA
EA
– + +
0.8V
REF
0.64V
1.2V
0.5μA
0.55V
ITH
RC
CC1
RUN
TK/SS
CSS
38502 FD
38502f
9
LTC3850-2
OPERATION
Main Control Loop
The LTC3850-2 is a constant-frequency, current mode
step-down controller with two channels operating 180
degrees out-of-phase. During normal operation, each
top MOSFET is turned on when the clock for that channel
sets the RS latch, and turned off when the main current
comparator, ICMP, resets the RS latch. The peak inductor
current at which ICMP resets the RS latch is controlled by
the voltage on the ITH pin, which is the output of each error
amplifier EA. The VFB pin receives the voltage feedback
signal, which is compared to the internal reference voltage
by the EA. When the load current increases, it causes a
slight decrease in VFB relative to the 0.8V reference, which
in turn causes the ITH voltage to increase until the average
inductor current matches the new load current. After the
top MOSFET has turned off, the bottom MOSFET is turned
on until either the inductor current starts to reverse, as
indicated by the reverse current comparator IREV, or the
beginning of the next cycle.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin. When
the EXTVCC pin is left open or tied to a voltage less than
4.7V, an internal 5V linear regulator supplies INTVCC power
from VIN. If EXTVCC is taken above 4.7V, the 5V regulator is
turned off and an internal switch is turned on connecting
EXTVCC. Using the EXTVCC pin allows the INTVCC power
to be derived from a high efficiency external source such
as one of the LTC3850-2 switching regulator outputs.
Each top MOSFET driver is biased from the floating bootstrap capacitor CB, which normally recharges during each
off cycle through an external diode when the top MOSFET
turns off. If the input voltage VIN decreases to a voltage
close to VOUT, the loop may enter dropout and attempt
to turn on the top MOSFET continuously. The dropout
detector detects this and forces the top MOSFET off for
about one-twelfth of the clock period every third cycle to
allow CB to recharge. However, it is recommended that a
load be present during the drop-out transition to ensure
CB is recharged.
Shutdown and Start-Up (RUN1, RUN2 and TK/SS1,
TK/SS2 Pins)
The two channels of the LTC3850-2 can be independently
shut down using the RUN1 and RUN2 pins. Pulling either
of these pins below 1.2V shuts down the main control
loop for that controller. Pulling both pins low disables both
controllers and most internal circuits, including the INTVCC
regulator. Releasing either RUN pin allows an internal
0.5μA current to pull up the pin and enable that controller. Alternatively, the RUN pin may be externally pulled up
or driven directly by logic. Be careful not to exceed the
Absolute Maximum Rating of 6V on this pin.
The start-up of each controller’s output voltage VOUT is
controlled by the voltage on the TK/SS1 and TK/SS2 pins.
When the voltage on the TK/SS pin is less than the 0.8V
internal reference, the LTC3850-2 regulates the VFB voltage
to the TK/SS pin voltage instead of the 0.8V reference. This
allows the TK/SS pin to be used to program a soft-start
by connecting an external capacitor from the TK/SS pin
to SGND. An internal 1.3μA pull-up current charges this
capacitor, creating a voltage ramp on the TK/SS pin. As the
TK/SS voltage rises linearly from 0V to 0.8V (and beyond),
the output voltage VOUT rises smoothly from zero to its final
value. Alternatively the TK/SS pin can be used to cause the
start-up of VOUT to “track” that of another supply. Typically,
this requires connecting to the TK/SS pin an external resistor
divider from the other supply to ground (see the Applications Information section). When the corresponding RUN
pin is pulled low to disable a controller, or when INTVCC
drops below its undervoltage lockout threshold of 3V, the
TK/SS pin is pulled low by an internal MOSFET. When in
undervoltage lockout, both controllers are disabled and
the external MOSFETs are held off.
Light Load Current Operation (Burst Mode Operation,
Pulse-Skipping, or Continuous Conduction)
The LTC3850-2 can be enabled to enter high efficiency Burst
Mode operation, constant-frequency pulse-skipping mode,
or forced continuous conduction mode. To select forced
continuous operation, tie the MODE/PLLIN pin to a DC
voltage below 0.8V (e.g., SGND). To select pulse-skipping
38502f
10
LTC3850-2
OPERATION
mode of operation, tie the MODE/PLLIN pin to INTVCC. To
select Burst Mode operation, float the MODE/PLLIN pin.
When a controller is enabled for Burst Mode operation,
the peak current in the inductor is set to approximately
one-third of the maximum sense voltage even though
the voltage on the ITH pin indicates a lower value. If the
average inductor current is higher than the load current,
the error amplifier EA will decrease the voltage on the ITH
pin. When the ITH voltage drops below 0.5V, the internal
sleep signal goes high (enabling “sleep” mode) and both
external MOSFETs are turned off.
In sleep mode, the load current is supplied by the output
capacitor. As the output voltage decreases, the EA’s output
begins to rise. When the output voltage drops enough, the
sleep signal goes low, and the controller resumes normal
operation by turning on the top external MOSFET on the
next cycle of the internal oscillator. When a controller is
enabled for Burst Mode operation, the inductor current is
not allowed to reverse. The reverse current comparator
(IREV) turns off the bottom external MOSFET just before the
inductor current reaches zero, preventing it from reversing and going negative. Thus, the controller operates in
discontinuous operation. In forced continuous operation,
the inductor current is allowed to reverse at light loads or
under large transient conditions. The peak inductor current
is determined by the voltage on the ITH pin, just as in normal operation. In this mode, the efficiency at light loads is
lower than in Burst Mode operation. However, continuous
mode has the advantages of lower output ripple and less
interference with audio circuitry.
When the MODE/PLLIN pin is connected to INTVCC, the
LTC3850-2 operates in PWM pulse-skipping mode at
light loads. At very light loads, the current comparator
ICMP may remain tripped for several cycles and force the
external top MOSFET to stay off for the same number of
cycles (i.e., skipping pulses). The inductor current is not
allowed to reverse (discontinuous operation). This mode,
like forced continuous operation, exhibits low output ripple
as well as low audio noise and reduced RF interference
as compared to Burst Mode operation. It provides higher
low current efficiency than forced continuous mode, but
not nearly as high as Burst Mode operation.
Frequency Selection and Phase-Locked Loop
(FREQ/PLLFLTR and MODE/PLLIN Pins)
The selection of switching frequency is a trade-off between
efficiency and component size. Low frequency operation
increases efficiency by reducing MOSFET switching losses,
but requires larger inductance and/or capacitance to maintain low output ripple voltage. The switching frequency
of the LTC3850-2’s controllers can be selected using the
FREQ/PLLFLTR pin. If the MODE/PLLIN pin is not being
driven by an external clock source, the FREQ/PLLFLTR
pin can be used to program the controller’s operating
frequency from 250kHz to 780kHz.
A phase-locked loop (PLL) is available on the LTC3850-2
to synchronize the internal oscillator to an external clock
source that is connected to the MODE/PLLIN pin. The
controller is operating in forced continuous mode when
it is synchronized. A series R-C should be connected
between the FREQ/PLLFLTR pin and SGND to serve as
the PLL’s loop filter.
Power Good (PGOOD Pin)
The PGOOD pin is connected to an open drain of an internal
N-channel MOSFET. The MOSFET turns on and pulls the
PGOOD pin low when either VFB pin voltage is not within
±7.5% of the 0.8V reference voltage. The PGOOD pin is
also pulled low when either RUN pin is below 1.2V or when
the LTC3850-2 is in the soft-start or tracking phase. When
the VFB pin voltage is within the ±7.5% requirement, the
MOSFET is turned off and the pin is allowed to be pulled
up by an external resistor to a source of up to 6V. The
PGOOD pin will flag power good immediately when both
VFB pins are within the ±7.5% window. However, there is
an internal 17μs power bad mask when either VFB goes
out of the ±7.5% window.
Output Overvoltage Protection
An overvoltage comparator, OV, guards against transient
overshoots (> 7.5%) as well as other more serious conditions that may overvoltage the output. In such cases,
the top MOSFET is turned off and the bottom MOSFET is
turned on until the overvoltage condition is cleared.
38502f
11
LTC3850-2
APPLICATIONS INFORMATION
The Typical Application on the first page is a basic LTC38502 application circuit. LTC3850-2 can be configured to use
either DCR (inductor resistance) sensing or low value resistor sensing. The choice between the two current sensing
schemes is largely a design trade-off between cost, power
consumption, and accuracy. DCR sensing is becoming
popular because it saves expensive current sensing resistors and is more power efficient, especially in high current
applications. However, current sensing resistors provide
the most accurate current limits for the controller. Other
external component selection is driven by the load requirement, and begins with the selection of RSENSE (if RSENSE is
used) and inductor value. Next, the power MOSFETs are selected. Finally, input and output capacitors are selected.
SENSE+ and SENSE– Pins
The SENSE+ and SENSE– pins are the inputs to the current
comparators. The common mode input voltage range of
the current comparators is 0V to 5V. Both SENSE pins
are high impedance inputs with small base currents of
less than 1μA. When the SENSE pins ramp up from 0V
to 1.4V, the small base currents flow out of the SENSE
pins. When the SENSE pins ramp down from 5V to 1.1V,
the small base currents flow into the SENSE pins. The
high impedance inputs to the current comparators allow
accurate DCR sensing. However, care must be taken not
to float these pins during normal operation.
VIN
INTVCC
BOOST
TG
LTC3850-2
Low Value Resistors Current Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 2a. RSENSE is chosen based on the required
output current.
The current comparator has a maximum threshold
VSENSE(MAX). The input common mode range of the current
comparator is 0V to 5V. The current comparator threshold
TO SENSE FILTER,
NEXT TO THE CONTROLLER
COUT
INDUCTOR OR RSENSE
VIN
INTVCC
SENSE RESISTOR
PLUS PARASITIC
INDUCTANCE
BOOST
ESL
VIN
INDUCTOR
TG
VOUT
38502 F01
Figure 1. Sense Lines Placement
with Inductor or Sense Resistor
VIN
RS
SW
Filter components mutual to the sense lines should be placed
close to the LTC3850-2, and the sense lines should run close
together to a Kelvin connection underneath the current sense
element (shown in Figure 1). Sensing current elsewhere can
effectively add parasitic inductance and capacitance to the
current sense element, degrading the information at the sense
terminals and making the programmed current limit unpredictable. If DCR sensing is used (Figure 2b), sense resistor
R1 should be placed close to the switching node, to prevent
noise from coupling into sensitive small-signal nodes. The
capacitor C1 should be placed close to the IC pins.
LTC3850-2
L
SW
DCR
VOUT
BG
BG
PGND
RF
SENSE+
SENSE–
SGND
CF • 2RF ≤ ESL/RS
POLE-ZERO
CANCELLATION
PGND
R1
SENSE+
C1*
CF
R2
SENSE–
SGND
RF
38502 F02b
38502 F02a
*PLACE C1 NEAR SENSE+,
SENSE– PINS
FILTER COMPONENTS
PLACED NEAR SENSE PINS
(2a) Using a Resistor to Sense Current
R1||R2 × C1 =
L
DCR
RSENSE(EQ) = DCR
R2
R1 + R2
(2b) Using the Inductor DCR to Sense Current
Figure 2. Two Different Methods of Sensing Current
38502f
12
LTC3850-2
APPLICATIONS INFORMATION
sets the peak of the inductor current, yielding a maximum
average output current IMAX equal to the peak value less
half the peak-to-peak ripple current, ΔIL. To calculate the
sense resistor value, use the equation:
VSENSE(MAX)
ΔI
I(MAX) + L
2
Because of possible PCB noise in the current sensing loop,
the AC current sensing ripple of ΔVSENSE = ΔIL • RSENSE
also needs to be checked in the design to get a good
signal-to-noise ratio. In general, for a reasonably good
PCB layout, a 15mV ΔVSENSE voltage is recommended as
a conservative number to start with, either for RSENSE or
DCR sensing applications.
RSENSE =
For previous generation current mode controllers, the
maximum sense voltage was high enough (e.g., 75mV for
the LTC1628 / LTC3728 family) that the voltage drop across
the parasitic inductance of the sense resistor represented
a relatively small error. For today’s highest current density
solutions, however, the value of the sense resistor can
be less than 1mΩ and the peak sense voltage can be as
low as 20mV. In addition, inductor ripple currents greater
than 50% with operation up to 1MHz are becoming more
common. Under these conditions the voltage drop across
the sense resistor’s parasitic inductance is no longer negligible. A typical sensing circuit using a discrete resistor is
shown in Figure 2a. In previous generations of controllers,
a small RC filter placed near the IC was commonly used to
reduce the effects of capacitive and inductive noise coupled
inthe sense traces on the PCB. A typical filter consists of
two series 10Ω resistors connected to a parallel 1000pF
capacitor, resulting in a time constant of 20ns.
This same RC filter, with minor modifications, can be used to
extract the resistive component of the current sense signal
in the presence of parasitic inductance. For example, Figure
3 illustrates the voltage waveform across a 2mΩ sense
resistor with a 2010 footprint for the 1.2V/15A converter
shown in Figure 18 operating at 100% load. The waveform
is the superposition of a purely resistive component and a
purely inductive component. It was measured using two
scope probes and waveform math to obtain a differential
measurement. Based on additional measurements of the
inductor ripple current and the on-time and off-time of
the top switch, the value of the parasitic inductance was
determined to be 0.5nH using the equation:
ESL =
VESL(STEP) tON • tOFF
ΔIL
tON + tOFF
If the RC time constant is chosen to be close to the parasitic
inductance divided by the sense resistor (L/R), the resulting waveform looks resistive again, as shown in Figure
4. For applications using low maximum sense voltages,
check the sense resistor manufacturer’s data sheet for
information about parasitic inductance. In the absence of
data, measure the voltage drop directly across the sense
resistor to extract the magnitude of the ESL step and use
the equation above to determine the ESL. However, do not
over-filter. Keep the RC time constant less than or equal
to the inductor time constant to maintain a high enough
ripple voltage on VRSENSE.
The above generally applies to high density / high current
applications where I(MAX) > 10A and low values of inductors
are used. For applications where I(MAX) < 10A, set RF to 10Ω
and CF to 1000pF. This will provide a good starting point.
The filter components need to be placed close to the IC. The
positive and negative sense traces need to be routed as a
differential pair and Kelvin connected to the sense resistor.
Inductor DCR Sensing
For applications requiring the highest possible efficiency at
high load currents, the LTC3850-2 is capable of sensing the
voltage drop across the inductor DCR, as shown in Figure
2b. The DCR of the inductor represents the small amount
of DC winding resistance of the copper, which can be less
than 1mΩ for today’s low value, high current inductors.
In a high current application requiring such an inductor,
conduction loss through a sense resistor would cost several
points of efficiency compared to DCR sensing.
38502f
13
LTC3850-2
APPLICATIONS INFORMATION
VESL(STEP)
VSENSE
20mV/DIV
500ns/DIV
38502 F03
Figure 3. Voltage Waveform Measured
Directly Across the Sense Resistor.
To ensure that the application will deliver full load current over the full operating temperature range, choose
the minimum value for the Maximum Current Sense
Threshold (VSENSE(MAX)) in the Electrical Characteristics
table (40mV).
Next, determine the DCR of the inductor. Where provided,
use the manufacturer’s maximum value, usually given
at 20°C. Increase this value to account for the temperature coefficient of resistance, which is approximately
0.4%/°C. A conservative value for TL(MAX) is 100°C.
To scale the maximum inductor DCR to the desired sense
resistor value, use the divider ratio:
RD =
VSENSE
20mV/DIV
500ns/DIV
38502 F04
Figure 4. Voltage Waveform Measured After the
Sense Resistor Filter. CF = 1000pF, RF = 100Ω.
If the external R1|| R2 • C1 time constant is chosen to be
exactly equal to the L/DCR time constant, the voltage drop
across the external capacitor is equal to the drop across
the inductor DCR multiplied by R2/(R1 + R2). R2 scales the
voltage across the sense terminals for applications where
the DCR is greater than the target sense resistor value.
To properly dimension the external filter components, the
DCR of the inductor must be known. It can be measured
using a good RLC meter, but the DCR tolerance is not
always the same and varies with temperature; consult the
manufacturers’ datasheets for detailed information.
Using the inductor ripple current value from the Inductor
Value Calculation section, the target sense resistor value is:
RSENSE(EQUIV) =
VSENSE(MAX)
ΔI
I(MAX) + L
2
RSENSE(EQUIV)
DCR(MAX) at TL(MAX)
C1 is usually selected to be in the range of 0.047μF to
0.47μF. This forces R1|| R2 to around 2kΩ, reducing
error that might have been caused by the SENSE pins’
±1μA current.
The equivalent resistance R1|| R2 is scaled to the room
temperature inductance and maximum DCR:
R1||R2 =
L
(DCR at 20°C) • C1
The sense resistor values are:
R1=
R1|| R2
R1 • RD
; R2 =
RD
1− RD
The maximum power loss in R1 is related to duty cycle,
and will occur in continuous mode at the maximum input
voltage:
PLOSS R1=
(V
IN(MAX) − VOUT
)• V
OUT
R1
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
38502f
14
LTC3850-2
APPLICATIONS INFORMATION
sense resistors. Light load power loss can be modestly
higher with a DCR network than with a sense resistor, due
to the extra switching losses incurred through R1. However,
DCR sensing eliminates a sense resistor, reduces conduction losses and provides higher efficiency at heavy loads.
Peak efficiency is about the same with either method.
To maintain a good signal to noise ratio for the current
sense signal, use a minimum ΔVSENSE of 10mV to 15mV.
For a DCR sensing application, the actual ripple voltage
will be determined by the equation:
V −V
VOUT
ΔVSENSE = IN OUT
R1• C1 VIN • fOSC
Slope Compensation and Inductor Peak Current
Slope compensation provides stability in constantfrequency architectures by preventing subharmonic
oscillations at high duty cycles. It is accomplished internally by adding a compensating ramp to the inductor
current signal at duty cycles in excess of 40%. Normally,
this results in a reduction of maximum inductor peak current for duty cycles > 40%. However, the LTC3850-2 uses
a patented scheme that counteracts this compensating
ramp, which allows the maximum inductor peak current
to remain unaffected throughout all duty cycles.
Inductor Value Calculation
Given the desired input and output voltages, the inductor
value and operating frequency fOSC directly determine the
inductor’s peak-to-peak ripple current:
IRIPPLE =
VOUT ⎛ VIN – VOUT ⎞
VIN ⎜⎝ fOSC •L ⎟⎠
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors, and output voltage
ripple. Thus, highest efficiency operation is obtained at
low frequency with a small ripple current. Achieving this,
however, requires a large inductor.
A reasonable starting point is to choose a ripple current
that is about 40% of IOUT(MAX). Note that the largest ripple
current occurs at the highest input voltage. To guarantee
that ripple current does not exceed a specified maximum,
the inductor should be chosen according to:
L≥
VIN – VOUT VOUT
•
fOSC •IRIPPLE VIN
Inductor Core Selection
Once the inductance value is determined, the type of inductor must be selected. Core loss is independent of core
size for a fixed inductor value, but it is very dependent
on inductance selected. As inductance increases, core
losses go down. Unfortunately, increased inductance
requires more turns of wire and therefore copper losses
will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Power MOSFET and Schottky Diode
(Optional) Selection
Two external power MOSFETs must be selected for each
controller in the LTC3850-2: one N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
The peak-to-peak drive levels are set by the INTVCC voltage.
This voltage is typically 5V during start-up (see EXTVCC Pin
Connection). Consequently, logic-level threshold MOSFETs
must be used in most applications. The only exception is if
low input voltage is expected (VIN < 5V); then, sub-logic level
threshold MOSFETs (VGS(TH) < 3V) should be used. Pay close
attention to the BVDSS specification for the MOSFETs as well;
most of the logic level MOSFETs are limited to 30V or less.
38502f
15
LTC3850-2
APPLICATIONS INFORMATION
Selection criteria for the power MOSFETs include the
on-resistance RDS(ON), Miller capacitance CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result is
then multiplied by the ratio of the application applied VDS
to the gate charge curve specified VDS. When the IC is
operating in continuous mode the duty cycles for the top
and bottom MOSFETs are given by:
Main Switch Duty Cycle =
VOUT
VIN
Synchronous Switch Duty Cycle =
VIN – VOUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
PMAIN =
VOUT
2
IMAX ) (1+ δ )RDS(ON) +
(
VIN
⎞
( VIN)2 ⎛⎜⎝ IMAX
(R )(C
)•
2 ⎟⎠ DR MILLER
⎡
1
1 ⎤
+
⎥ • fOSC
⎢
⎢⎣ VINTVCC – VTH(MIN) VTH(MIN) ⎥⎦
V –V
2
PSYNC = IN OUT (IMAX ) (1+ δ )RDS(ON)
VIN
where δ is the temperature dependency of RDS(ON) and
RDR (approximately 2Ω) is the effective driver resistance
at the MOSFET’s Miller threshold voltage. VTH(MIN) is the
typical MOSFET minimum threshold voltage.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short-circuit when the synchronous switch is on close
to 100% of the period.
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
The optional Schottky diodes conduct during the dead time
between the conduction of the two power MOSFETs. These
prevent the body diodes of the bottom MOSFETs from turning on, storing charge during the dead time and requiring
a reverse recovery period that could cost as much as 3%
in efficiency at high VIN. A 1A to 3A Schottky is generally
a good compromise for both regions of operation due to
the relatively small average current. Larger diodes result
in additional transition losses due to their larger junction
capacitance.
Soft-Start and Tracking
The LTC3850-2 has the ability to either soft-start by itself
with a capacitor or track the output of another channel or
external supply. When one particular channel is configured
to soft-start by itself, a capacitor should be connected to
its TK/SS pin. This channel is in the shutdown state if its
RUN pin voltage is below 1.2V. Its TK/SS pin is actively
pulled to ground in this shutdown state.
Once the RUN pin voltage is above 1.2V, the channel powers up. A soft-start current of 1.3μA then starts to charge
its soft-start capacitor. Note that soft-start or tracking is
achieved not by limiting the maximum output current of
the controller but by controlling the output ramp voltage
according to the ramp rate on the TK/SS pin. Current
foldback is disabled during this phase to ensure smooth
soft-start or tracking. The soft-start or tracking range is
defined to be the voltage range from 0V to 0.8V on the
TK/SS pin. The total soft-start time can be calculated as:
CSS
1.3µA
Regardless of the mode selected by the MODE/PLLIN pin,
the regulator will always start in pulse-skipping mode up
t SOFTSTART = 0.8 •
38502f
16
LTC3850-2
APPLICATIONS INFORMATION
to TK/SS = 0.64V. Between TK/SS = 0.64V and 0.74V, it
will operate in forced continuous mode and revert to the
selected mode once TK/SS > 0.74V. The output ripple is
minimized during the 100mV forced continuous mode
window ensuring a clean PGOOD signal.
When the channel is configured to track another supply,
the feedback voltage of the other supply is duplicated by
a resistor divider and applied to the TK/SS pin. Therefore,
the voltage ramp rate on this pin is determined by the
ramp rate of the other supply’s voltage. Note that the
small soft-start capacitor charging current is always
flowing, producing a small offset error. To minimize this
error, select the tracking resistive divider value to be small
enough to make this error negligible.
In order to track down another channel or supply after
the soft-start phase expires, the LTC3850-2 is forced into
continuous mode of operation as soon as VFB is below the
undervoltage threshold of 0.74V regardless of the setting
on the MODE/PLLIN pin. However, the LTC3850-2 should
always be set in force continuous mode tracking down
when there is no load. After TK/SS drops below 0.1V, its
channel will operate in discontinuous mode.
Output Voltage Tracking
The LTC3850-2 allows the user to program how its output ramps up and down by means of the TK/SS pins.
Through these pins, the output can be set up to either coincidentally or ratiometrically track another
supply’s output, as shown in Figure 5. In the following
discussions, VOUT1 refers to the LTC3850-2’s output 1 as a
master channel and VOUT2 refers to the LTC3850-2’s output
2 as a slave channel. In practice, though, either phase can
be used as the master. To implement the coincident tracking in Figure 5a, connect an additional resistive divider to
VOUT1 and connect its midpoint to the TK/SS pin of the
slave channel. The ratio of this divider should be the same
as that of the slave channel’s feedback divider shown in
Figure 6a. In this tracking mode, VOUT1 must be set higher
than VOUT2. To implement the ratiometric tracking, the ratio
of the VOUT2 divider should be exactly the same as the
master channel’s feedback divider. By selecting different
resistors, the LTC3850-2 can achieve different modes of
tracking including the two in Figure 5.
So which mode should be programmed? While either
mode in Figure 5 satisfies most practical applications,
some tradeoffs exist. The ratiometric mode saves a pair
of resistors, but the coincident mode offers better output
regulation. This can be better understood with the help
of Figure 7. At the input stage of the slave channel’s error
amplifier, two common anode diodes are used to clamp
the equivalent reference voltage and an additional diode
is used to match the shifted common mode voltage. The
top two current sources are of the same amplitude. In the
coincident mode, the TK/SS voltage is substantially higher
than 0.8V at steady state and effectively turns off D1. D2
and D3 will therefore conduct the same current and offer
tight matching between VFB2 and the internal precision
0.8V reference. In the ratiometric mode, however, TK/SS
equals 0.8V at steady state. D1 will divert part of the bias
current to make VFB2 slightly lower than 0.8V.
Although this error is minimized by the exponential I-V
characteristic of the diode, it does impose a finite amount
of output voltage deviation. Furthermore, when the master
channel’s output experiences dynamic excursion (under
load transient, for example), the slave channel output will
be affected as well. For better output regulation, use the
coincident tracking mode instead of ratiometric.
INTVCC Regulators and EXTVCC
The LTC3850-2 features an NPN linear regulator that supplies power to INTVCC from the VIN supply. INTVCC powers
the gate drivers and much of the LTC3850-2’s internal
circuitry. The linear regulator regulates the voltage at the
INTVCC pin to 5V when VIN is greater than 6.5V. EXTVCC
connects to INTVCC through a P-channel MOSFET and can
supply the needed power when its voltage is higher than
4.7V. Each of these can supply a peak current of 100mA
and must be bypassed to ground with a minimum of 1μF
ceramic capacitor or low ESR electrolytic capacitor. No matter what type of bulk capacitor is used, an additional 0.1μF
ceramic capacitor placed directly adjacent to the INTVCC
and PGND pins is highly recommended. Good bypassing
is needed to supply the high transient currents required
by the MOSFET gate drivers and to prevent interaction
between the channels.
38502f
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LTC3850-2
APPLICATIONS INFORMATION
VOUT1
OUTPUT VOLTAGE
OUTPUT VOLTAGE
VOUT1
VOUT2
TIME
VOUT2
TIME
38502 F03a
(5a) Coincident Tracking
38502 F03b
(5b) Ratiometric Tracking
Figure 5. Two Different Modes of Output Voltage Tracking
VOUT1
VOUT1
VOUT2
R3
R1
R3
TO
VFB1
PIN
TO
TK/SS2
PIN
R4
TO
TK/SS2
PIN
TO
VFB2
PIN
R2
VOUT2
R1
R4
R2
R3
TO
VFB1
PIN
TO
VFB2
PIN
R4
38502 F06
(6a) Coincident Tracking Setup
(6b) Ratiometric Tracking Setup
Figure 6. Setup for Coincident and Ratiometric Tracking
I
I
+
D1
D2
EA2
TK/SS2
–
0.8V
VFB2
D3
38502 F07
Figure 7. Equivalent Input Circuit of Error Amplifier
38502f
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LTC3850-2
APPLICATIONS INFORMATION
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3850-2 to
be exceeded. The INTVCC current, which is dominated
by the gate charge current, may be supplied by either
the 5V linear regulator or EXTVCC. When the voltage on
the EXTVCC pin is less than 4.7V, the linear regulator is
enabled. Power dissipation for the IC in this case is highest and is equal to VIN • IINTVCC. The gate charge current
is dependent on operating frequency as discussed in the
Efficiency Considerations section. The junction temperature can be estimated by using the equations given in
Note 3 of the Electrical Characteristics. For example, the
LTC3850-2 INTVCC current is limited to less than 24mA
from a 24V supply in the GN package and not using the
EXTVCC supply:
TJ = 70°C + (24mA)(24V)(95°C/W) = 125°C
To prevent the maximum junction temperature from being
exceeded, the input supply current must be checked while
operating in continuous conduction mode (MODE/PLLIN
= SGND) at maximum VIN. When the voltage applied to
EXTVCC rises above 4.7V, the INTVCC linear regulator is
turned off and the EXTVCC is connected to the INTVCC.
The EXTVCC remains on as long as the voltage applied
to EXTVCC remains above 4.5V. Using the EXTVCC allows
the MOSFET driver and control power to be derived from
one of the LTC3850-2’s switching regulator outputs during
normal operation and from the INTVCC when the output
is out of regulation(e.g., start-up, short-circuit). If more
current is required through the EXTVCC than is specified,
an external Schottky diode can be added between the
EXTVCC and INTVCC pins. Do not apply more than 6V to
the EXTVCC pin and make sure that EXTVCC < VIN.
Significant efficiency and thermal gains can be realized by
powering INTVCC from the output, since the VIN current
resulting from the driver and control currents will be scaled
by a factor of (Duty Cycle)/(Switcher Efficiency).
Tying the EXTVCC pin to a 5V supply reduces the junction
temperature in the previous example from 125°C to:
TJ = 70°C + (24mA)(5V)(95°C/W) = 81°C
However, for 3.3V and other low voltage outputs, additional circuitry is required to derive INTVCC power from
the output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause
INTVCC to be powered from the internal 5V regulator
resulting in an efficiency penalty of up to 10% at high
input voltages.
2. EXTVCC connected directly to VOUT. This is the
normal connection for a 5V regulator and provides
the highest efficiency.
3. EXTVCC connected to an external supply. If a 5V
external supply is available, it may be used to power
EXTVCC providing it is compatible with the MOSFET
gate drive requirements.
4. EXTVCC connected to an output-derived boost network. For 3.3V and other low voltage regulators,
efficiency gains can still be realized by connecting
EXTVCC to an output-derived voltage that has been
boosted to greater than 4.7V.
For applications where the main input power is 5V, tie
the VIN and INTVCC pins together and tie the combined
pins to the 5V input with a 1Ω or 2.2Ω resistor as shown
in Figure 8 to minimize the voltage drop caused by the
gate charge current. This will override the INTVCC linear
regulator and will prevent INTVCC from dropping too low
due to the dropout voltage. Make sure the INTVCC voltage
is at or exceeds the RDS(ON) test voltage for the MOSFET
which is typically 4.5V for logic level devices.
LTC3850-2
VIN
RVIN
INTVCC
5V
CINTVCC
4.7μF
1Ω
+
CIN
38502 F08
Figure 8. Setup for a 5V Input
38502f
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LTC3850-2
APPLICATIONS INFORMATION
Topside MOSFET Driver Supply (CB, DB)
CIN and COUT Selection
External bootstrap capacitors CB connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs.
Capacitor CB in the Functional Diagram is charged though
external diode DB from INTVCC when the SW pin is low.
When one of the topside MOSFETs is to be turned on,
the driver places the CB voltage across the gate source
of the desired MOSFET. This enhances the MOSFET and
turns on the topside switch. The switch node voltage, SW,
rises to VIN and the BOOST pin follows. With the topside
MOSFET on, the boost voltage is above the input supply:
VBOOST = VIN + VINTVCC. The value of the boost capacitor
CB needs to be 100 times that of the total input capacitance of the topside MOSFET(s). The reverse breakdown of the external Schottky diode must be greater
than VIN(MAX). When adjusting the gate drive level, the
final arbiter is the total input current for the regulator. If
a change is made and the input current decreases, then
the efficiency has improved. If there is no change in input
current, then there is no change in efficiency.
The selection of CIN is simplified by the 2-phase architecture and its impact on the worst-case RMS current drawn
through the input network (battery/fuse/capacitor). It can be
shown that the worst-case capacitor RMS current occurs
when only one controller is operating. The controller with
the highest (VOUT)(IOUT) product needs to be used in the
formula below to determine the maximum RMS capacitor
current requirement. Increasing the output current drawn
from the other controller will actually decrease the input
RMS ripple current from its maximum value. The out-ofphase technique typically reduces the input capacitor’s RMS
ripple current by a factor of 30% to 70% when compared
to a single phase power supply solution.
Undervoltage Lockout
The LTC3850-2 has two functions that help protect the
controller in case of undervoltage conditions. A precision
UVLO comparator constantly monitors the INTVCC voltage
to ensure that an adequate gate-drive voltage is present.
It locks out the switching action when INTVCC is below
3V. To prevent oscillation when there is a disturbance on
the INTVCC, the UVLO comparator has 500mV of precision hysteresis.
Another way to detect an undervoltage condition is to
monitor the VIN supply. Because the RUN pins have a
precision turn-on reference of 1.2V, one can use a resistor
divider to VIN to turn on the IC when VIN is high enough.
An extra 4.5μA of current flows out of the RUN pin once
the RUN pin voltage passes 1.2V. One can program the
hysteresis of the run comparator by adjusting the values
of the resistive divider. For accurate VIN undervoltage
detection, VIN needs to be higher than 4V.
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle (VOUT)/(VIN). To prevent
large voltage transients, a low ESR capacitor sized for the
maximum RMS current of one channel must be used. The
maximum RMS capacitor current is given by:
CIN Required IRMS ≈
1/2
IMAX
⎡⎣( VOUT ) ( VIN – VOUT ) ⎤⎦
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT/2. This simple worst-case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that capacitor manufacturers’ ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet
size or height requirements in the design. Due to the high
operating frequency of the LTC3850-2, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
The benefit of the LTC3850-2 2-phase operation can be
calculated by using the equation above for the higher
power controller and then calculating the loss that would
have resulted if both controller channels switched on at
the same time. The total RMS power lost is lower when
38502f
20
LTC3850-2
APPLICATIONS INFORMATION
both controllers are operating due to the reduced overlap of
current pulses required through the input capacitor’s ESR.
This is why the input capacitor’s requirement calculated
above for the worst-case controller is adequate for the dual
controller design. Also, the input protection fuse resistance,
battery resistance, and PC board trace resistance losses
are also reduced due to the reduced peak currents in a 2phase system. The overall benefit of a multiphase design
will only be fully realized when the source impedance of the
power supply/battery is included in the efficiency testing.
The sources of the top MOSFETs should be placed within
1cm of each other and share a common CIN(s). Separating
the sources and CIN may produce undesirable voltage and
current resonances at VIN.
A small (0.1μF to 1μF) bypass capacitor between the chip
VIN pin and ground, placed close to the LTC3850-2, is also
suggested. A 2.2Ω – 10Ω resistor placed between CIN
(C1) and the VIN pin provides further isolation between
the two channels.
The selection of COUT is driven by the effective series
resistance (ESR). Typically, once the ESR requirement
is satisfied, the capacitance is adequate for filtering. The
output ripple (ΔVOUT) is approximated by:
⎛
1 ⎞
ΔVOUT ≈IRIPPLE ⎜ ESR +
8fCOUT ⎟⎠
⎝
where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage
since IRIPPLE increases with input voltage.
VOUT
RB
1/2 LTC3850-2
CFF
VFB
RA
38502 F09
Figure 9. Setting Output Voltage
To improve the frequency response, a feed-forward capacitor, CFF , may be used. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
Fault Conditions: Current Limit and Current Foldback
The LTC3850-2 includes current foldback to help limit
load current when the output is shorted to ground. If the
output falls below 50% of its nominal output level, then
the maximum sense voltage is progressively lowered from
its maximum programmed value to one-third of the maximum value. Foldback current limiting is disabled during
the soft-start or tracking up. Under short-circuit conditions with very low duty cycles, the LTC3850-2 will begin
cycle skipping in order to limit the short-circuit current.
In this situation the bottom MOSFET will be dissipating
most of the power but less than in normal operation. The
short-circuit ripple current is determined by the minimum
on-time tON(MIN) of the LTC3850-2 (≈ 90ns), the input
voltage and inductor value:
ΔIL(SC) = tON(MIN) •
VIN
L
The resulting short-circuit current is:
Setting Output Voltage
The LTC3850-2 output voltages are each set by an external feedback resistive divider carefully placed across the
output, as shown in Figure 9. The regulated output voltage
is determined by:
⎛ R ⎞
VOUT = 0.8V • ⎜ 1+ B ⎟
⎝ R ⎠
A
ISC =
1/3 VSENSE(MAX)
RSENSE
1
– ΔIL(SC)
2
Phase-Locked Loop and Frequency Synchronization
The LTC3850-2 has a phase-locked loop (PLL) comprised of
an internal voltage-controlled oscillator (VCO) and a phase
detector. This allows the turn-on of the top MOSFET of
controller 1 to be locked to the rising edge of an external
clock signal applied to the MODE/PLLIN pin. The turn-on
38502f
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LTC3850-2
APPLICATIONS INFORMATION
of controller 2’s top MOSFET is thus 180 degrees outof-phase with the external clock. The phase detector is
an edge sensitive digital type that provides zero degrees
phase shift between the external and internal oscillators.
This type of phase detector does not exhibit false lock to
harmonics of the external clock.
The output of the phase detector is a pair of complementary current sources that charge or discharge the external
filter network connected to the FREQ/PLLFLTR pin. The
relationship between the voltage on the FREQ/PLLFLTR
pin and operating frequency is shown in Figure 10 and
specified in the Electrical Characteristics table. Note that the
LTC3850-2 can only be synchronized to an external clock
whose frequency is within range of the LTC3850-2’s internal
VCO. This is guaranteed to be between 250kHz and 780kHz.
A simplified block diagram is shown in Figure 11.
900
800
FREQUENCY (kHz)
700
600
If no clock is applied to MODE/PLLIN pin, the FREQ/
PLLFLTR pin will be high impedance.
If the external clock frequency is greater than the internal
oscillator’s frequency, fOSC , then current is sourced continuously from the phase detector output, pulling up the
FREQ/PLLFLTR pin. When the external clock frequency is
less than fOSC , current is sunk continuously, pulling down
the FREQ/PLLFLTR pin. If the external and internal frequencies are the same but exhibit a phase difference, the current
sources turn on for an amount of time corresponding to the
phase difference. The voltage on the FREQ/PLLFLTR pin is
adjusted until the phase and frequency of the internal and
external oscillators are identical. At the stable operating
point, the phase detector output is high impedance and
the filter capacitor CLP holds the voltage.
The loop filter components, CLP and RLP, smooth out the
current pulses from the phase detector and provide a stable
input to the voltage-controlled oscillator. The filter components CLP and RLP determine how fast the loop acquires
lock. Typically RLP = 10k and CLP is 2200pF to 0.01μF.
Typically, the external clock (on MODE/PLLIN pin) input high
threshold is 1.6V, while the input low thres-hold is 1V.
500
400
300
Minimum On-Time Considerations
200
Minimum on-time tON(MIN) is the smallest time duration that
the LTC3850-2 is capable of turning on the top MOSFET.
It is determined by internal timing delays and the gate
charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that
100
0
0
0.5
1
1.5
2
FREQ/PLLFLTR PIN VOLTAGE (V)
2.5
38502 F10
Figure 10. Relationship Between Oscillator
Frequency and Voltage at the FREQ/PLLFLTR Pin
2.4V
tON(MIN) <
RLP
VOUT
VIN (f)
CLP
MODE/
PLLIN
EXTERNAL
OSCILLATOR
FREQ/
PLLFLTR
DIGITAL
PHASE/
FREQUENCY
DETECTOR
VCO
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
The minimum on-time for the LTC3850-2 is approximately
90ns, with reasonably good PCB layout, minimum 30%
inductor current ripple and at least 10mV – 15mV ripple
38502 F11
Figure 11. Phase-Locked Loop Block Diagram
38502f
22
LTC3850-2
APPLICATIONS INFORMATION
on the current sense signal. The minimum on-time can be
affected by PCB switching noise in the voltage and current
loop. As the peak sense voltage decreases the minimum
on-time gradually increases to 130ns. This is of particular
concern in forced continuous applications with low ripple
current at light loads. If the duty cycle drops below the
minimum on-time limit in this situation, a significant
amount of cycle skipping can occur with correspondingly
larger current and voltage ripple.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3850-2 circuits: 1) IC VIN current, 2) INTVCC
regulator current, 3) I2R losses, 4) Topside MOSFET
transition losses.
1. The VIN current is the DC supply current given in
the Electrical Characteristics table, which excludes
MOSFET driver and control currents. VIN current typically results in a small (<0.1%) loss.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the
control circuit current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges of
the topside and bottom side MOSFETs.
Supplying INTVCC power through EXTVCC from an output-derived source will scale the VIN current required
for the driver and control circuits by a factor of (Duty
Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately
2.5mA of VIN current. This reduces the mid-current loss
from 10% or more (if the driver was powered directly
from VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor.
In continuous mode, the average output current flows
through L and RSENSE, but is “chopped” between the
topside MOSFET and the synchronous MOSFET. If the
two MOSFETs have approximately the same RDS(ON),
then the resistance of one MOSFET can simply be
summed with the resistances of L and RSENSE to obtain
I2R losses. For example, if each RDS(ON) = 10mΩ, RL
= 10mΩ, RSENSE = 5mΩ, then the total resistance is
25mΩ. This results in losses ranging from 2% to 8%
as the output current increases from 3A to 15A for
a 5V output, or a 3% to 12% loss for a 3.3V output.
Efficiency varies as the inverse square of VOUT for the
same external components and output power level. The
combined effects of increasingly lower output voltages
and higher currents required by high performance digital
systems is not doubling but quadrupling the importance
of loss terms in the switching regulator system!
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high
input voltages (typically 15V or greater). Transition
losses can be estimated from:
Transition Loss = (1.7) VIN2 IO(MAX) CRSS f
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is
very important to include these “system” level losses
during the design phase. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at the
switching frequency. A 25W supply will typically require
38502f
23
LTC3850-2
APPLICATIONS INFORMATION
a minimum of 20μF to 40μF of capacitance having a
maximum of 20mΩ to 50mΩ of ESR. The LTC3850-2
2-phase architecture typically halves this input capacitance
requirement over competing solutions. Other losses
including Schottky conduction losses during dead time
and inductor core losses generally account for less than
2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to ΔILOAD (ESR), where ESR is the effective
series resistance of COUT. ΔILOAD also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. The
availability of the ITH pin not only allows optimization of
control loop behavior but also provides a DC coupled and
AC filtered closed loop response test point. The DC step,
rise time and settling at this test point truly reflects the
closed loop response. Assuming a predominantly second
order system, phase margin and/or damping factor can be
estimated using the percentage of overshoot seen at this
pin. The bandwidth can also be estimated by examining the
rise time at the pin. The ITH external components shown
in the Typical Application circuit will provide an adequate
starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1μs to 10μs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking the feedback loop. Placing a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This
is why it is better to look at the ITH pin signal which is in
the feedback loop and is the filtered and compensated
control loop response. The gain of the loop will be increased by increasing RC and the bandwidth of the loop
will be increased by decreasing CC. If RC is increased by
the same factor that CC is decreased, the zero frequency
will be kept the same, thereby keeping the phase shift the
same in the most critical frequency range of the feedback
loop. The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT , causing a rapid drop in VOUT . No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • CLOAD. Thus a 10μF capacitor would
require a 250μs rise time, limiting the charging current
to about 200mA.
38502f
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LTC3850-2
APPLICATIONS INFORMATION
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the IC. These items are also illustrated graphically in the
layout diagram of Figure 12. Figure 13 illustrates the current
waveforms present in the various branches of the 2-phase
synchronous regulators operating in the continuous mode.
Check the following in your layout:
1. Are the top N-channel MOSFETs M1 and M3 located within
1 cm of each other with a common drain connection at
CIN? Do not attempt to split the input decoupling for the
two channels as it can cause a large resonant loop.
2. Are the signal and power grounds kept separate? The
combined IC signal ground pin and the ground return
of CINTVCC must return to the combined COUT (–)
terminals. The VFB and ITH traces should be as short
as possible. The path formed by the top N-channel
MOSFET, Schottky diode and the CIN capacitor should
have short leads and PC trace lengths. The output
capacitor (–) terminals should be connected as close
as possible to the (–) terminals of the input capacitor
by placing the capacitors next to each other and away
from the Schottky loop described above.
3. Do the LTC3850-2 VFB pins’ resistive dividers connect to
the (+) terminals of COUT? The resistive divider must be
connected between the (+) terminal of COUT and signal
ground. The feedback resistor connections should not
be along the high current input feeds from the input
capacitor(s).
4. Are the SENSE+ and SENSE– leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE+ and SENSE– should be as close as possible
to the IC. Ensure accurate current sensing with Kelvin
connections at the sense resistor or inductor, whichever
is used for current sensing.
5. Is the INTVCC decoupling capacitor connected close to
the IC, between the INTVCC and the power ground pins?
This capacitor carries the MOSFET drivers current peaks.
An additional 1μF ceramic capacitor placed immediately
next to the INTVCC and PGND pins can help improve
noise performance substantially.
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), and boost nodes (BOOST1, BOOST2) away
from sensitive small-signal nodes, especially from the
opposite channel’s voltage and current sensing feedback
pins. All of these nodes have very large and fast moving
signals and therefore should be kept on the “output
side” of the LTC3850-2 and occupy minimum PC trace
area. If DCR sensing is used, place the top resistor
(Figure 2b, R1) close to the switching node.
7. Use a modified “star ground” technique: a low impedance, large copper area central grounding point on
the same side of the PC board as the input and output
capacitors with tie-ins for the bottom of the INTVCC
decoupling capacitor, the bottom of the voltage feedback
resistive divider and the SGND pin of the IC.
PC Board Layout Debugging
Start with one controller at a time. It is helpful to use a
DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output
switching node (SW pin) to synchronize the oscilloscope
to the internal oscillator and probe the actual output voltage
as well. Check for proper performance over the operating
voltage and current range expected in the application. The
frequency of operation should be maintained over the input
voltage range down to dropout and until the output load
drops below the low current operation threshold—typically
10% of the maximum designed current level in Burst
Mode operation.
38502f
25
LTC3850-2
APPLICATIONS INFORMATION
TK/SS1
RPU2
PGOOD
PGOOD
VPULL-UP
ITH1
VFB1
L1
SENSE1+
TG1
SENSE1–
SW1
VOUT1
CB1
LTC3850-2
fIN
M1
BOOST1
PLLLPF
RSENSE
M2
BG1
D1
1μF
CERAMIC
MODE/PLLIN
COUT1
+
VIN
RIN
CVIN
RUN1
PGND
RUN2
INTVCC
SENSE2+
BG2
VFB2
+
SENSE2–
CIN
CINTVCC
COUT2
1μF
CERAMIC
M3
BOOST2
GND
+
EXTVCC
+
VIN
SGND
M4
D2
CB2
ITH2
TK/SS2
SW2
RSENSE
TG2
VOUT2
L2
38502 F12
Figure 12. Recommended Printed Circuit Layout Diagram
SW1
L1
D1
RSENSE1
VOUT1
COUT1
RL1
VIN
RIN
CIN
SW2
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
D2
L2
RSENSE2
VOUT2
COUT2
RL2
38502 F13
Figure 13. Branch Current Waveforms
38502f
26
LTC3850-2
APPLICATIONS INFORMATION
4.7μF
D3
M1
0.1μF
L1
3.3μH
22μF
50V
1μF
2.2Ω
VIN PGOOD EXTVCC INTVCC
TG1
TG2
BOOST1
SW1
D4
M2
0.1μF
L2
2.2μH
BOOST2
SW2
LTC3850-2
BG1
6.19k
1%
BG2
4.12k
1%
10k, 1%
MODE/PLLIN
VIN
7V TO
20V
PGND
FREQ/PLLFLTR
1.33k
1%
SENSE2+
–
SENSE2–
0.1μF
0.1μF
SENSE1
33pF
VOUT1
3.3V
5A
SENSE1+
RUN1
COUT1
100μF
X2
1800pF
20k
1%
4.75k
1%
100pF
TK/SS1
33pF
RUN2
VFB1
ITH1
63.4k
1%
1.5k
1%
VFB2
ITH2
SGND
0.1μF
0.1μF
25.5k
1%
2200pF
TK/SS2
3.16k
1%
5.49k
1%
100pF
VOUT2
1.8V
5A
COUT2
100μF
X2
20k
1%
38502 F14
L1, L2: COILTRONICS HCP0703
M1, M2: VISHAY SILICONIX Si4816BDY
COUT1, COUT2: TAIYO YUDEN JMK325BJ107MM
Figure 14. High Efficiency Dual 500kHz 3.3V/1.8V Step-Down Converter
The duty cycle percentage should be maintained from cycle
to cycle in a well-designed, low noise PCB implementation.
Variation in the duty cycle at a subharmonic rate can suggest
noise pickup at the current or voltage sensing inputs or
inadequate loop compensation. Overcompensation of the
loop can be used to tame a poor PC layout if regulator
bandwidth optimization is not required. Only after each
controller is checked for its individual performance should
both controllers be turned on at the same time. A particularly
difficult region of operation is when one controller channel
is nearing its current comparator trip point when the other
channel is turning on its top MOSFET. This occurs around
50% duty cycle on either channel due to the phasing of the
internal clocks and may cause minor duty cycle jitter.
Reduce VIN from its nominal level to verify operation
of the regulator in dropout. Check the operation of the
undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, Schottky and the top
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
SGND pin of the IC.
38502f
27
LTC3850-2
APPLICATIONS INFORMATION
Design Example
As a design example for a two channel medium current regulator, assume VIN = 12V(nominal), VIN = 20V(maximum),
VOUT1 = 3.3V, VOUT2 = 1.8V, IMAX1,2 = 5A, and f = 500kHz
(see Figure 14).
The regulated output voltages are determined by:
⎛ R ⎞
VOUT = 0.8V • ⎜ 1+ B ⎟
⎝ RA ⎠
Using 20k 1% resistors from both VFB nodes to ground,
the top feedback resistors are (to the nearest 1% standard
value) 63.4k and 25.5k.
The frequency is set by biasing the FREQ/PLLFLTR pin to
1.2V (see Figure 10), using a divider from INTVCC. This
voltage will decrease as VIN approaches 5V, lowering the
switching frequency. If a separate 5V supply is connected to
EXTVCC, INTVCC will remain at 5V even if VIN decreases.
The inductance values are based on a 35% maximum
ripple current assumption (1.75A for each channel). The
highest value of ripple current occurs at the maximum
input voltage:
⎛
VOUT
VOUT ⎞
L=
⎟
⎜ 1−
ƒ • ΔIL(MAX) ⎝
VIN(MAX) ⎠
Channel 1 will require 3.2μH, and Channel 2 will require
1.9μH. The next highest standard values are 3.3μH and 2.2μH.
At the nominal input voltage (12V), the ripple will be:
ΔIL(NOM) =
VOUT ⎛
VOUT ⎞
⎟
⎜ 1−
ƒ•L ⎝
VIN(NOM) ⎠
Channel 1 will have 1.45A (29%) ripple, and Channel 2 will
have 1.4A (28%) ripple. The peak inductor current will be
the maximum DC value plus one-half the ripple current,
or 5.725A for Channel 1 and 5.7A for Channel 2.
The minimum on-time occurs on Channel 1 at the maximum
VIN, and should not be less than 90ns:
tON(MIN) =
VOUT
VIN(MAX) ƒ
=
1.8V
= 180ns
20V(500kHz)
The equivalent RSENSE resistor value can be calculated by
using the minimum value for the maximum current sense
threshold (40mV).
RSENSE(EQUIV) =
VSENSE(MIN)
=
ΔIL(NOM)
ILOAD(MAX) +
2
40mV
≅ 7mΩ
1.5A
5A +
2
The equivalent RSENSE is the same for Channel 2.
The Coiltronics (Cooper) HCP0703-2R2 (20mΩ DCRMAX
at 20°C) and HCP0703-3R3 (30mΩ DCRMAX at 20°C) are
chosen. At 100°C, the estimated maximum DCR values are
26.4mΩ and 39.6mΩ. The divider ratios are:
RD =
and
RSENSE(EQUIV)
DCRMAX at TL(MAX)
=
7mΩ
= 0.26;
26.4mΩ
7mΩ
≅ 0.18
39.6mΩ
For each channel, 0.1μF is selected for C1.
L
2.2µH
=
(DCRMAX at 20°C) • C1 20mΩ • 0.1µF
3.3µH
= 1.1k ; and
= 1.1k
30mΩ • 0.1µF
R1||R2 =
For channel 1, the DCRSENSE filter/divider values are:
R1||R2 1.1k
=
≅ 6.19k;
RD
0.18
R1 • RD 6.19k • 0.18
R2 =
=
≅ 1.33k
1− RD
1− 0.18
R1=
38502f
28
LTC3850-2
APPLICATIONS INFORMATION
The power loss in R1 at the maximum input voltage is:
PLOSS R1=
(VIN(MAX) − VOUT ) • VOUT
R1
=
(20V − 3.3V) • 3.3V
= 9mW
6.19k
The power dissipation on the topside MOSFET can be easily
estimated. Choosing a Siliconix Si4816BDY dual MOSFET
results in: RDS(ON) = 0.023Ω/0.016Ω, CMILLER ≅ 100pF.
At maximum input voltage with T(estimated) = 50°C:
3.3V 2
(5) [1+(0.005)(50°C – 25°C)] •
20V
⎞
(0.023Ω) + (20V )2 ⎛⎜⎝ 5A
(2Ω)(100pF ) •
2 ⎟⎠
PMAIN =
The respective values for Channel 2 are R1 = 4.12k, R2 =
1.5k; and PLOSS R1 = 8mW.
Burst Mode operation is chosen for high light load efficiency
(Figure 15) by floating the MODE/PLLIN pin. Power loss
due to the DCR sensing network is slightly higher at light
loads than would have been the case with a suitable sense
resistor (7mΩ). At heavier loads, DCR sensing provides
higher efficiency.
1 ⎤
⎡ 1
⎢ 5 – 2.3 + 2.3 ⎥ ( 500kHz ) = 186mW
⎣
⎦
A short-circuit to ground will result in a folded back current of:
I SC =
100
10
(1/ 3) 50mV – 1 ⎛ 90ns(20V) ⎞ = 2.1A
0.007Ω
2 ⎜⎝
3.3µH ⎟⎠
DCR
80
7mΩ
1
DCR
70
0.1
60
POWER LOSS (mW)
EFFICIENCY (%)
90
50
40
0.01
EFFICIENCY
POWER LOSS
0.1
1
LOAD CURRENT (mA)
with a typical value of RDS(ON) and δ = (0.005/°C)(20)
= 0.1. The resulting power dissipated in the bottom
MOSFET is:
20V – 3.3V
2
2.1A ) (1.125) ( 0.016Ω )
(
20V
= 66mW
PSYNC =
which is less than under full-load conditions.
0.01
10
38502 F15
Figure 15. Design Example Efficiency vs Load
CIN is chosen for an RMS current rating of at least 2A at
temperature assuming only channel 1 or 2 is on. COUT is
chosen with an ESR of 0.02Ω for low output ripple. The
output ripple in continuous mode will be highest at the
maximum input voltage. The output voltage ripple due to
ESR is approximately:
VORIPPLE = RESR (ΔIL) = 0.02Ω(1.5A) = 30mVP–P
38502f
29
LTC3850-2
TYPICAL APPLICATIONS
VIN
7V TO
24V
22μF
50V
2.2Ω
1μF
4.7μF
D3
M1
0.1μF
L2
2.2μH
TG1
TG2
BOOST1
BOOST2
SW1
SW2
LTC3850-2
BG2
BG1
MODE/PLLIN
10Ω
15pF
10Ω
COUT1
220μF
SENSE1–
SENSE2–
L2
3.3μH
10k
1%
10Ω
1000pF
RUN1
20k
1%
1000pF
100pF
10k
1%
TK/SS1
0.1μF
6mΩ
10pF
10Ω
RUN2
EXTVCC
VFB2
ITH2
VFB1
ITH1
63.4k
1%
+
SENSE1+
PGND
FREQ/PLLFLTR
SENSE2+
M2
0.1μF
1000pF
6mΩ
VOUT1
3.3V
5A
D4
VIN PGOOD INTVCC
SGND
0.1μF
105k
1%
1000pF
TK/SS2
3.16k
1%
15k
1%
100pF
20k
1%
VOUT2
5V
5A
+
COUT2
150μF
38502 F16
L1: TDK RLF 7030T-2R2M5R4
L2: TDK ULF10045T-3R3N6R9
COUT1: SANYO 4TPE220MF
COUT2: SANYO 6TPE150MI
Figure 16. 3.3V/5A, 5V/5A Converter Using Sense Resistors
38502f
30
25.5k
20k
C12
100pF
C11
1000pF
C7
1000pF
C6
100pF
CSS
0.1μF
R18
4.99k
R12
7.5k
2.10k
C15
47pF
C10
33pF
C5
0.1μF
SENSE2+ SGND
SENSE2–
TK/SS2
ITH2
VFB2
EXTVCC
RPG
100k
PGOOD
PGOOD SW2 TG2
CVCC
4.7μF
CB2
0.1μF
D4
CMDSH-3
D3
CMDSH-3
CB1
0.1μF
CVIN
1μF
M4
RJK0301DPB
+
38502 F17
L2
0.68μH
R30
4.02k
M3
HAT2168H
PGND
GND
M2
RJK0301DPB
R27
L1
4.02k 0.68μH
M1
HAT2168H
Figure 17. 2.5V/15A, 1.8V/15A Converter with DCR Sensing and Coincident Rail Tracking
FSW = 350kHz
RUN2
BOOST2
PGND
BG2
INTVCC
VIN
VFB1
LTC3850-2
BG1
BOOST1
ITH1
TK/SS1
SENSE1– SENSE1+ RUN1 FREQ/ MODE/ SW1 TG1
PLLFLTR PLLIN
L1, L2: VISHAY IHLP5050EZ-01 0.68μH
COUT1, COUT2: SANYO 4TPD330M
R4
25.5k
R3
20k
R2
20k
R1
43.2k
C4
0.1μF
10k
+
RVIN
2.2Ω
COUT2
330μF
4V
2X
COUT1
330μF
4V
2X
10μF
2x +
VOUT2
1.8V/
15A
VOUT1
2.5V/
15A
CIN
180μF
VIN
7V TO 14V
LTC3850-2
TYPICAL APPLICATIONS
38502f
31
32
CSS2
0.1μF
C12
100pF
C11
1000pF
C7
1000pF
C6
100pF
CSS1
0.1μF
R18
5.9k
R12
5.9k
R20
100Ω
R10
100Ω
R5
10k
PLLIN
400kHz
C5
1000pF
RUN2
EXTVCC
RPG
100k
PGOOD
PGOOD SW2 TG2
BOOST2
PGND
BG2
INTVCC
CVCC
4.7μF
CB2
0.1μF
D4
CMDSH-3
D3
CMDSH-3
CB1
0.1μF
RVIN
2.2Ω
CVIN
1μF
M4
RJK0301DPB
M3
RJK0305DPB
PGND
GND
M2
RJK0301DPB
M1
RJK0305DPB
L2
0.4μH
L1
0.4μH
Figure 18. 1.5V/15A, 1.2V/15A Core-I/O Converter with Sense Resistor Synchronized at 400kHz
R22
100Ω
SENSE2+ SGND
SENSE2–
TK/SS2
ITH2
VFB2
VIN
VFB1
LTC3850-2
BG1
BOOST1
ITH1
TK/SS1
SENSE1– SENSE1+ RUN1 FREQ/ MODE/ SW1 TG1
PLLFLTR PLLIN
L1, L2: VITEC 59PR9875
COUT1, COUT2: 2R5TPE330M9
R4
10k
R3
20k
R2
20k
R1
17.8k
C4
1000pF
R9
100Ω
C2
0.01μF
38502 F18
RSENSE2
0.002Ω
RSENSE1
0.002Ω
+
COUT2
330μF
2.5V
2X
COUT1
330μF
2.5V
2X
+
10μF
2x
+
C1
1000pF
VOUT2
1.2V/15A
VOUT1
1.5V/15A
CIN
180μF
VIN
7V TO 14V
LTC3850-2
TYPICAL APPLICATIONS
38502f
LTC3850-2
TYPICAL APPLICATIONS
5V ± 0.5V
4.7μF
6.3V
2x
1Ω
4.7μF
D3
M1
TG1
0.1μF
L1
0.75μH
VIN PGOOD EXTVCC INTVCC
BG1
PLLIN
750kHz
M2
TG2
BOOST1
SW1
1.2k
1%
D4
0.1μF
L2
0.75μH
BOOST2
SW2
LTC3850-2
MODE/PLLIN
BG2
1.2k
1%
PGND
FREQ/PLLFLTR
SENSE1+
2.94k
1%
0.047μF
SENSE1–
47pF
VOUT1
1.8V
5A
SENSE2+
0.047μF
SENSE2–
RUN1
COUT1
100μF
X2
2200pF
20k
1%
14k
1%
100pF
0.1μF
TK/SS1
100pF
RUN2
VFB1
ITH1
25.5k
1%
4.99k
1%
VFB2
ITH2
SGND
TK/SS2
0.1μF
1nF
10nF
10k
1%
2200pF
14k
1%
100pF
10k
1%
COUT2
100μF
X2
20k
1%
L1, L2: TOKO FDV0630 0.75μH
M1, M2: VISHAY SILICONIX Si4816BDY
COUT1, COUT2: TAIYO YUDEN JMK325BJ107MM
VOUT2
1.2V
5A
38502 F19
Figure 19. 1.8V/5A, 1.2V/5A Core-I/O Converter with a 5V Input Synchronized at 750kHz
38502f
33
LTC3850-2
TYPICAL APPLICATIONS
2.2Ω
VIN1
12V
VIN2
3.3V
1μF
4.7μF
4.7μF
D3
M1
VIN PGOOD EXTVCC INTVCC
TG1
0.1μF
L1
2.2μH
4.7μF
2x
13.0k
TG2
BOOST1
SW1
M2
0.1μF
L2
0.75μH
BOOST2
SW2
LTC3850-2
BG1
3.74k
1%
10k
D4
MODE/PLLIN
BG2
PGND
1.2k
1%
10k
1%
FREQ/PLLFLTR
1.40k
1%
SENSE2+
–
SENSE2–
0.1μF
SENSE1
47pF
VOUT1
2.5V
5A
SENSE1+
0.1μF
RUN1
COUT1
100μF
X2
2200pF
20k
1%
10k
1%
100pF
TK/SS1
0.1μF
100pF
RUN2
VFB1
ITH1
43.2k
1%
4.32k
1%
VFB2
ITH2
SGND
2200pF
TK/SS2
0.1μF
3.16k
1%
6.04k
1%
L1: TOKO FDV0630 2.2μH
L2: TOKO FDV0630 0.75μH
M1, M2: VISHAY SILICONIX Si4816BDY
COUT1, COUT2: TAIYO YUDEN JMK325BJ107MM
10k
1%
100pF
VOUT2
1.2V
5A
COUT2
100μF
X2
20k
1%
38502 F20
Figure 20. 2.5V/5A, 1.2V/5A Core-I/O Converter with Dual Inputs
38502f
34
LTC3850-2
PACKAGE DESCRIPTION
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.386 – .393*
(9.804 – 9.982)
.045 p.005
28 27 26 25 24 23 22 21 20 19 18 17 1615
.254 MIN
.033
(0.838)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 p.0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
1
RECOMMENDED SOLDER PAD LAYOUT
.015 p .004
s 45o
(0.38 p 0.10)
.0075 – .0098
(0.19 – 0.25)
2 3
4
5 6
7
8
9 10 11 12 13 14
.0532 – .0688
(1.35 – 1.75)
.004 – .0098
(0.102 – 0.249)
0o – 8o TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN28 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
38502f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
35
LTC3850-2
TYPICAL APPLICATION
20k
2.55k
10k
RUN
1nF
RJK0305DPB
0.1μF
7.5k
0.1μF
SENSE1– SENSE1+ RUN1 FREQ
MODE SW1 TG1
0.1μF
2.21k
L1
0.56μH
10μF
2x
VIN
7V TO 14V
+
180μF
RJK0301DPB
CMDSH-3
220pF
TK/SS1
2.2nF
BOOST1
ITH1
BG1
VFB1
VIN
2.74k
20k
LTC3850-2
VFB2
2.2Ω
INTVCC
ITH2
BG2
4.7μF
1μF
VOUT
1.1V/30A
PGND
TK/SS2
100μF
2x
CMDSH-3
SENSE2–
BOOST2
RJK0305DPB
SENSE2+
+
SGND
RUN2
EXTVCC PGOOD SW2 TG2
0.1μF
0.1μF
COUT1
330μF
2.5V
4x
L2
0.56μH
RJK0301DPB
2.21k
PGOOD
20k
100k
38502 TA02
RUN
L1, L2: VISHAY IHLP4040DZ-01 0.56μH
COUT: SANYO 2R5TPE330M9
FOR SINGLE OUTPUT, DUAL PHASE OPERATION, TIE THE FOLLOWING PINS TOGETHER:
TK/SS1 TO TK/SS2
VFB1 TO VFB1
RUN1 TO RUN2
ITH1 TO ITH2
Figure 21. 1.1V/30A Dual Phase Core Converter, FSW = 400kHz
RELATED PARTS
PART NUMBER DESCRIPTION
COMMENTS
LTC1625/
LTC1775
No RSENSE™ Current Mode Synchronous Step-Down Controllers
97% Efficiency, No Sense Resistor, 16-Pin SSOP
LTC1735
Synchronous Step-Down Switching Regulator Controller
Programmable Fixed Frequency from 200kHz to 550kHz
LTC1778
No RSENSE Wide Input Range Synchronous Step-Down Controller
Up to 97% Efficiency, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9)(VIN),
IOUT Up to 20A, Extremely Fast Transient Response
LTC3727A-1
Dual, 2-Phase Synchronous Controller
Very Low Dropout; VOUT ≤ 14V, 4V ≤ VIN ≤ 36V
LTC3728
2-Phase 550kHz, Dual Synchronous Step-Down Controller
20A to 200A PolyPhase® Synchronous Controller
QFN and SSOP Packages
LTC3729L-6
Expandable from 2-Phase to 12-Phase,
4V ≤ VIN ≤ 30V, 0.6V ≤ VOUT ≤ 7V
LTC3731
3-Phase, Single Output From 250kHz to 600kHz Synchronous
Step-Down Controller
0.6V ≤ VOUT ≤ 6V, 4.5V ≤ VIN ≤ 32V,
Expandable PolyPhase from 3-Phase to 12-Phase
LTC3810
100V Current Mode Synchronous Step-Down Switching Controller
0.8V ≤ VOUT ≤ 0.93VIN, 6.2V ≤ VIN ≤ 100V, No RSENSE
LTC3826
Low IQ, Dual, 2-Phase Synchronous Step-Down Controller
30μA IQ, 0.8V ≤ VOUT ≤ 10V, 4V ≤ VIN ≤ 36V
LTC3828
Dual, 2-Phase Synchronous Step-Down Controller with Tracking
Up to Six Phases, 0.8V ≤ VOUT ≤ 7V, 4.5V ≤ VIN ≤ 28V
LTC3834/
LTC3834-1
Low IQ, Synchronous Step-Down Controller
30μA IQ, 0.8V ≤ VOUT ≤ 10V, 4V ≤ VIN ≤ 36V
LT3845
Low IQ, High Voltage Single Output Synchronous Step-Down
DC/DC Controller
1.23V ≤ VOUT ≤ 36V, 4V ≤ VIN ≤ 60V, 120μA IQ
LTC3851
High Efficiency Synchronous Step-Down Switching
Regulator Controller
Single Output Version of LTC3850-2
4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 5.5V
PolyPhase is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation.
36 Linear Technology Corporation
38502f
LT 0908 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2007
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