LINER LTC3865EUH Dual, 2-phase synchronous dc/dc controller with pin selectable output Datasheet

LTC3865/LTC3865-1
Dual, 2-Phase Synchronous
DC/DC Controller with
Pin Selectable Outputs
Description
Features
n
n
n
n
n
n
n
n
n
n
n
n
Dual, 180° Phased Controllers
2-Pin VID Output Voltage Programming from 0.6V
to 5V
High Efficiency: Up to 95%
RSENSE or DCR Current Sensing
Phase-Lockable Fixed Frequency 250kHz to 770kHz
Adjustable Current Limit
Dual N-Channel MOSFET Synchronous Drive
Wide VIN Range: 4.5V to 38V Operation
Adjustable Soft-Start Current Ramping or Tracking
Output OV Protection With Reverse Current Limit
Power Good Output Voltage Monitor
32-Pin 5mm × 5mm QFN and 38-Lead TSSOP
Packages
The LTC®3865/LTC3865-1 are high performance dual
synchronous step-down DC/DC switching regulator
controllers that drive all N‑channel synchronous power
MOSFET stages. A constant frequency current mode
architecture allows a phase-lockable frequency of up to
770kHz. Power loss and noise are minimized by operating
the two controller output stages out of phase.
OPTI-LOOP® compensation allows the transient response
to be optimized over a wide range of output capacitance
and ESR values. Independent track/soft-start pins for
each controller ramp the output voltage during start-up.
Current foldback limits MOSFET heat dissipation during
short-circuit conditions. The MODE/PLLIN pin selects
among Burst Mode® operation, pulse-skipping mode,
and continuous inductor current mode. The output voltages can be precisely programmed by pin strapping or
external resistors.
Applications
n
DC Power Distribution Systems
L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, µModule, Linear Technology and the Linear logo
are registered trademarks and No RSENSE is a trademark of Linear Technology Corporation. All
other trademarks are the property of their respective owners. U.S. Patents, including 5481178,
5705919, 5929620, 6100678, 6144194, 6177787, 6304066, 6580258.
The LTC3865/LTC3865-1 are available in low profile (5mm
× 5mm) 32-pin QFN and 38-Lead thermally enhanced
TSSOP packages.
Typical Application
High Efficiency 1.5V/15A, 1.2V/15A Step-Down Converter
+
2.2Ω
22µF
50V
VIN
4.5V TO 24V
Efficiency and Power Loss
vs Load Current
1µF
100
4.7µF
TG2
BOOST1
SW1
BG1
100Ω
SENSE1+
BOOST2
SW2
0.47µH
BG2
PGND
FREQ
SENSE2+
100Ω
SENSE1–
SENSE2–
VID21
VID11
VID12
VID22
VOSENSE2
VOSENSE1
ITH2
ITH1
TK/SS1
SGND TK/SS2
100µF
COUT2,3
1000pF
10k
100pF
0.1µF
0.1µF
2mΩ
162k
15k
60
1
EFFICIENCY
POWER LOSS
50
40
0.1
30
10
0
0.01
100Ω
1000pF
70
20
1000pF
100Ω
+
80
0.1µF
1000pF
2mΩ
VOUT1
1.5V
15A
LTC3865
10
VIN = 12V
VOUT = 1.5V
POWER LOSS (W)
0.47µH
INTVCC
EFFICIENCY (%)
0.1µF
VIN
TG1
90
+
VOUT2
1.2V
15A
0.1
1
10
LOAD CURRENT (A)
0.01
100
3865 TA01b
COUT5,6
100µF
100pF
3865 TA01a
3865f
LTC3865/LTC3865-1
Absolute Maximum Ratings (Note 1)
Input Supply Voltage (VIN).......................... –0.3V to 40V
Topside Driver Voltages
BOOST1, BOOST2................................... –0.3V to 46V
Switch Voltage (SW1, SW2).......................... –5V to 40V
INTVCC, RUN1, RUN2, PGOOD(s), EXTVCC,
(BOOST1-SW1), (BOOST2-SW2).................. –0.3V to 6V
SENSE1+, SENSE2+, SENSE1–, SENSE2–
VOSENSE1, VOSENSE2 Voltages..................... –0.3V to 5.8V
MODE/PLLIN, ILIM, TK/SS1, TK/SS2, VID11,
VID12, VID21, VID22, FREQ Voltages.... –0.3V to INTVCC
ITH1, ITH2 Voltages..................................... –0.3V to 2.7V
INTVCC DC Output Current......................................80mA
Operating Junction Temperature Range (Notes 2, 3)
LTC3865/LTC3865-1........................... –40°C to 125°C
Storage Temperature Range.................... –65°C to 125°C
Lead Temperature (Soldering, 10 sec)
FE Package........................................................ 300°C
Pin Configuration
LTC3865
TOP VIEW
RUN1
1
38 FREQ
SENSE1+
2
37 VID22
SENSE–
3
36 MODE/PLLIN
NC
4
35 SW1
TG1
LTC3865
VOSENSE1
5
34 TG1
32 31 30 29 28 27 26 25
TK/SS1
6
33 BOOST1
SW1
MODE/PLLIN
VID22
FREQ
RUN1
SENSE1+
SENSE1–
TOP VIEW
ITH1
7
32 PGND1
23 BG1
SGND
8
31 BG1
22 VIN
VID11
9
21 INTVCC
VID12 10
24 BOOST1
VOSENSE1 1
TK/SS1 2
ITH1 3
VID11 4
33
SGND
VID12 5
20 EXTVCC
19 BG2
ITH2 6
TK/SS2 7
18 PGND
VOSENSE2 8
17 BOOST2
TG2
SW2
PGOOD
ILIM
RUN2
VID21
SENSE2+
SENSE2–
9 10 11 12 13 14 15 16
UH PACKAGE
32-LEAD (5mm s 5mm) PLASTIC QFN
TJMAX = 125°C, θJA = 34°C/W, θJC = 3°C/W
EXPOSED PAD (PIN 33) IS SGND, MUST BE SOLDERED TO PCB
ITH2 11
TK/SS2 12
VOSENSE2 13
NC 14
SENSE2–
15
39
SGND
30 VIN
29 INTVCC
28 EXTVCC
27 BG2
26 PGND2
25 BOOST2
24 NC
SENSE2+ 16
23 TG2
RUN2 17
22 SW2
VID21 18
21 PGOOD1
ILIM 19
20 PGOOD2
FE PACKAGE
38-LEAD PLASTIC TSSOP
TJMAX = 125°C, θJA = 28°C/W
EXPOSED PAD (PIN 39) IS SGND, MUST BE SOLDERED TO PCB
3865f
LTC3865/LTC3865-1
Pin Configuration
LTC3865-1
TG1
SW1
MODE/PLLIN
VID22
FREQ
RUN1
SENSE1+
SENSE1–
TOP VIEW
32 31 30 29 28 27 26 25
VOSENSE1 1
24 BOOST1
TK/SS1 2
23 BG1
ITH1 3
22 VIN
VID11 4
21 INTVCC
33
SGND
VID12 5
20 EXTVCC
19 BG2
ITH2 6
TK/SS2 7
18 PGND
VOSENSE2 8
17 BOOST2
TG2
SW2
PGOOD1
PGOOD2
RUN2
VID21
SENSE2+
SENSE2–
9 10 11 12 13 14 15 16
UH PACKAGE
32-LEAD (5mm s 5mm) PLASTIC QFN
TJMAX = 125°C, θJA = 34°C/W, θJC = 3°C/W
EXPOSED PAD (PIN 33) IS SGND, MUST BE SOLDERED TO PCB
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3865EUH#PBF
LTC3865EUH#TRPBF
3865
32-Lead (5mm × 5mm) Plastic QFN
–40°C to 85°C
LTC3865EUH-1#PBF
LTC3865EUH-1#TRPBF
38651
32-Lead (5mm × 5mm) Plastic QFN
–40°C to 85°C
LTC3865IUH#PBF
LTC3865IUH#TRPBF
3865
32-Lead (5mm × 5mm) Plastic QFN
–40°C to 125°C
LTC3865IUH-1#PBF
LTC3865IUH-1#TRPBF
38651
32-Lead (5mm × 5mm) Plastic QFN
–40°C to 125°C
LTC3865EFE#PBF
LTC3865EFE#TRPBF
LTC3865FE
38-Lead Plastic TSSOP
–40°C to 85°C
LTC3865IFE#PBF
LTC3865IFE#TRPBF
LTC3865FE
38-Lead Plastic TSSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3865f
LTC3865/LTC3865-1
Electrical Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TJ = 25°C. VIN = 15V, VRUN1,2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loops
VIN
Input Voltage
VOSENSE1,2
Output Voltage Sensing (E-Grade)
Output Voltage Sensing (I-Grade)
4.5
(Note 4) ITH1,2 Voltage = 1.2V
VID11 = VID21 = GND, VID12 = VID22 = GND
VID11 = VID21 = GND, VID12 = VID22 = Float
VID11 = VID21 = GND, VID12 = VID22 = INTVCC
VID11 = VID21 = Float, VID12 = VID22 = GND
VID11 = VID21 = Float, VID12 = VID22 = Float
VID11 = VID21 = Float, VID12 = VID22 = INTVCC
VID11 = VID21 = INTVCC, VID12 = VID22 = GND
VID11 = VID21 = INTVCC, VID12 = VID22 = Float
VID11 = VID21 = INTVCC, VID12 = VID22 = INTVCC
l
l
l
l
l
l
l
l
l
1.089
0.990
1.188
1.485
0.596
1.782
2.463
3.251
4.925
(Note 4) ITH1,2 Voltage = 1.2V
VID11 = VID21 = GND, VID12 = VID22 = GND
VID11 = VID21 = GND, VID12 = VID22 = Float
VID11 = VID21 = GND, VID12 = VID22 = INTVCC
VID11 = VID21 = Float, VID12 = VID22 = GND
VID11 = VID21 = Float, VID12 = VID22 = Float
VID11 = VID21 = Float, VID12 = VID22 = INTVCC
VID11 = VID21 = INTVCC, VID12 = VID22 = GND
VID11 = VID21 = INTVCC, VID12 = VID22 = Float
VID11 = VID21 = INTVCC, VID12 = VID22 = INTVCC
l
l
l
l
l
l
l
l
l
1.084
0.985
1.182
1.478
0.593
1.773
2.450
3.234
4.900
IOSENSE1,2
Feedback Current
(Note 4) VID11 = VID21 = VID12 = VID22 = Float
VREFLNREG
Reference Voltage Line Regulation
VIN = 4.5V to 38V (Note 4)
VLOADREG
Output Voltage Load Regulation
(Note 4)
Measured in Servo Loop; ∆ITH Voltage = 1.2V to 0.7V
Measured in Servo Loop; ∆ITH Voltage = 1.2V to 1.6V
l
l
38
V
1.100
1.000
1.200
1.500
0.602
1.800
2.500
3.300
5.000
1.111
1.010
1.212
1.515
0.608
1.818
2.538
3.350
5.075
V
V
V
V
V
V
V
V
V
1.100
1.000
1.200
1.500
0.602
1.800
2.500
3.300
5.000
1.117
1.015
1.218
1.523
0.611
1.827
2.550
3.366
5.100
V
V
V
V
V
V
V
V
V
–10
–50
nA
0.002
0.02
%/V
0.01
–0.01
0.1
–0.1
%
%
gm1,2
Transconductance Amplifier gm
ITH1,2 = 1.2V; Sink/Source 5µA; (Note 4)
2.2
IQ
Input DC Supply Current
Normal Mode
Shutdown
(Note 5)
VIN = 15V
VRUN1,2 = 0V
3
30
UVLO
Undervoltage Lockout on INTVCC
VINTVCC Ramping Down
UVLOHYS
UVLO Hysteresis
VOVL
Feedback Overvoltage Lockout
Measured at VOSENSE1,2 with VID Pins Floating
ISENSE
Sense Pins Total Current
(Each Channel); VSENSE1,2 = 3.3V
DFMAX
Maximum Duty Cycle
In Dropout
l
0.64
94
mmho
50
mA
µA
3.3
V
0.55
V
0.66
0.68
V
±1
±2
µA
95
%
ITK/SS1,2
Soft-Start Charge Current
VTK/SS1,2 = 0V
VRUN1,2
RUN Pin On Threshold
VRUN1, VRUN2 Rising
VRUN1,2HYS
RUN Pin On Hysteresis
80
mV
IRUN1,2HYS
RUN Pin Current Hysteresis
4.5
µA
l
0.9
1.3
1.7
µA
1.1
1.22
1.35
V
VSENSE(MAX) Maximum Current Sense Threshold
(E-Grade)
VITH1,2 = 3.3V, ILIM = 0V
VITH1,2 = 3.3V, ILIM = Float
VITH1,2 = 3.3V, ILIM = INTVCC
In Overvoltage Condition
l
l
l
l
24
44
68
–63
30
50
75
–53
36
56
82
–43
mV
mV
mV
mV
Maximum Current Sense Threshold
(I-Grade)
VITH1,2 = 3.3V, ILIM = 0V
VITH1,2 = 3.3V, ILIM = Float
VITH1,2 = 3.3V, ILIM = INTVCC
In Overvoltage Condition
l
l
l
l
22
42
66
–65
30
50
75
–53
38
58
84
–41
V
V
V
V
TG RUP
TG Driver Pull-Up On-Resistance
TG High
2.6
Ω
TG RDOWN
TG Driver Pull-Down On-Resistance
TG Low
1.5
Ω
BG RUP
BG Driver Pull-Up On-Resistance
BG High
3
Ω
3865f
LTC3865/LTC3865-1
Electrical Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TJ = 25°C. VIN = 15V, VRUN1,2 = 5V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
BG RDOWN
BG Driver Pull-Down On-Resistance
BG Low
MIN
TYP
1.4
MAX
UNITS
Ω
TG1,2 tr
TG1,2 tf
TG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
BG1,2 tr
BG1,2 tf
BG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
TG/BG t1D
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
BG/TG t2D
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
tON(MIN)
Minimum On-Time
(Note 7)
90
ns
INTVCC Linear Regulator
Internal VCC Voltage
6V < VIN < 38V
VLDO INT
INTVCC Load Regulation
ICC = 0mA to 20mA
VEXTVCC
EXTVCC Switchover Voltage
EXTVCC Ramping Positive
VLDO EXT
EXTVCC Voltage Drop
ICC = 20mA, VEXTVCC = 5V
VLDOHYS
EXTVCC Hysteresis
VINTVCC
4.8
l
4.5
5.0
5.2
V
0.5
2
%
4.7
50
V
100
200
mV
mV
Oscillator and Phase-Locked Loop
fNOM
Nominal Frequency
RFREQ = 162k
450
500
550
kHz
fLOW
Lowest Frequency
RFREQ = 0Ω
210
250
290
kHz
fHIGH
Highest Frequency
RFREQ ≥ 325k
650
770
880
kHz
RMODE/PLLIN MODE/PLLIN Input Resistance
IFREQ
250
Frequency Setting Current
6.5
kΩ
7.5
8.5
µA
0.1
0.3
V
±2
µA
–12.5
12.5
%
%
PGOOD OUTPUT
VPGL
PGOOD Voltage Low
IPGOOD = 2mA
IPGOOD
PGOOD Leakage Current
VPGOOD = 5V
VPG
PGOOD Trip Level
VOSENSE with Respect to Set Regulated Voltage
VID11 = VID12 = VID21 = VID22 = Float
VOSENSE Ramping Negative
VOSENSE Ramping Postitive
tPG
PGOOD Bad Blanking Time
Measured from VID Transitition Edge
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3865E/LTC3865E-1 are guaranteed to meet performance
specifications over the 0°C to 85°C operating junction temperature range.
Specifications over the –40°C to 85°C operating junction temperature
range are assured by design, characterization and correlation with
statistical process controls. The LTC3865I/LTC3865I-1 are guaranteed to
meet performance specifications over the full –40°C to 125°C operating
junction temperature range. Note that the maximum ambient temperature
is determined by specific operating conditions in conjunction with board
layout, the rated package thermal resistance and other environmental
factors.
–7
7
–10
10
100
µs
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC3865UH: TJ = TA + (PD • 34°C/W)
LTC3865FE: TJ = TA + (PD • 25°C/W)
Note 4: The LTC3865/LTC3865-1 are tested in a feedback loop that servos
VITH1,2 to a specified voltage and measures the resultant VOSENSE1,2.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 7: The minimum on-time condition is specified for an inductor peakto-peak ripple current 40% of IMAX (see Minimum On-Time Considerations
in the Applications Information section).
Note 8: VSENSE(MAX) defaults to 50mV for the LTC3865-1.
3865f
LTC3865/LTC3865-1
Typical Performance Characteristics
90
80
80
70
70
BURST
DCM
50
EFFICIENCY (%)
EFFICIENCY (%)
100
90
CCM
40
30
20
0
0.01
0.1
1
10
LOAD CURRENT (A)
BURST
50
DCM
40
CCM
30
70
3
60
POWER LOSS
50
0
0.01
0.1
1
10
LOAD CURRENT (A)
1
30
20
100
10
0
20
30
3865 G25
Load Step
(Forced Continuous Mode)
Load Step
(Pulse-Skipping Mode)
VOUT
100mV/DIV
ACCOUPLED
VOUT
100mV/DIV
ACCOUPLED
IL
5A/DIV
IL
5A/DIV
VOUT
100mV/DIV
ACCOUPLED
IL
5A/DIV
ILOAD
5A/DIV
500mA
TO 7A
ILOAD
5A/DIV
500mA
TO 7A
ILOAD
5A/DIV
500mA
TO 7A
3865 G01
40µs/DIV
VIN = 12V
VOUT = 1.5V
FIGURE 16 CIRCUIT
Inductor Current at Light Load
3865 G02
40µs/DIV
VIN = 12V
VOUT = 1.5V
FIGURE 16 CIRCUIT
RUN
2V/DIV
RUN1
2V/DIV
VOUT1 = 1.5V
1Ω LOAD
500mV/DIV
VOUT = 1.5V
500mV/DIV
Burst Mode
OPERATION
5A/DIV
VOUT2 = 1.2V
1Ω LOAD
500mV/DIV
VTRACK/SS
1V/DIV
PULSESKIPPING MODE
5A/DIV
3865 G04
3865 G03
Coincident Tracking
Prebiased Output at 1V
FORCED
CONTINUOUS
MODE
5A/DIV
0
INPUT VOLTAGE (V)
3865 G24
Load Step
(Burst Mode Operation)
VIN = 12V
2µs/DIV
VOUT = 1.5V
ILOAD = 100mA
FIGURE 16 CIRCUIT
2
40
VIN = 12V
VOUT = 1.2V
FIGURE 16 CIRCUIT
3865 G23
VIN = 12V
40µs/DIV
VOUT = 1.5V
FIGURE 16 CIRCUIT
4
EFFICIENCY
80
10
100
5
FIGURE 16 CIRCUIT
90
20
VIN = 12V
VOUT = 1.5V
FIGURE 16 CIRCUIT
10
60
100
POWER LOSS (W)
100
60
Efficiency and Power Loss
vs Input Voltage
Efficiency vs Load Current
EFFICIENCY (%)
Efficiency vs Load Current
40ms/DIV
3865 G05
40ms/DIV
3865 G06
3865f
LTC3865/LTC3865-1
Typical Performance Characteristics
Tracking Up and Down with
External Ramp
(Forced Continuous Mode)
5
VOUT2 = 1.2V
1Ω LOAD
500mV/DIV
5.00
4.75
3
2
4.50
4.25
4.00
3.75
3.50
1
3865 G07
20ms/DIV
VIN = 15V
INTVCC VOLTAGE (V)
VOUT1 = 1.5V
1Ω LOAD
500mV/DIV
INTVCC Line Regulation
5.25
4
QUIESCENT CURRENT (mA)
TK/SS1
TK/SS2
2V/DIV
Quiescent Current vs Input
Voltage without EXTVCC
3.25
0
5
10
25
15
30
20
INPUT VOLTAGE (V)
3.00
40
35
0
5
10
15 20 25 30
INPUT VOLTAGE (V)
3865 G08
ILIM = FLOAT
40
20
ILIM = GND
0
–20
0.5
1
1.5
VITH (V)
70
60
ILIM = FLOAT
50
40
ILIM = GND
30
20
2
10
0
1
4
3
VSENSE COMMON MODE VOLTAGE (V)
0
3865 G10
2
Maximum Current Sense
Voltage vs Feedback Voltage
(Current Foldback)
80
ILIM = INTVCC
70
60
ILIM = FLOAT
50
40
ILIM = GND
30
20
10
0
5
0
20
40
60
DUTY CYCLE (%)
80
100
3865 G12
TK/SS Pull-Up Current
vs Temperature
2.00
100
90
1.75
ILIM = INTVCC
80
70
60
ILIM = FLOAT
50
40
ILIM = GND
30
20
1.50
1.25
1.00
0.75
0.50
0.25
10
0
90
3865 G11
TK/SS CURRENT (µA)
0
MAXIMUM CURRENT SENSE VOLTAGE (mV)
VSENSE (mV)
100
ILIM = INTVCC
CURRENT SENSE THRESHOLD (mV)
60
–40
Maximum Current Sense
Threshold vs Duty Cycle
80
ILIM = INTVCC
CURRENT SENSE THRESHOLD (mV)
80
40
3865 G09
Maximum Current Sense
Threshold vs Common Mode
Voltage
Current Sense Threshold
vs ITH Voltage
35
0
0.1
0.3
0.4
0.5
0.2
FEEDBACK VOLTAGE (V)
0.6
3865 G13
0
–50 –25
75
50
25
TEMPERATURE (°C)
0
100
125
3865 G14
3865f
LTC3865/LTC3865-1
Typical Performance Characteristics
Shutdown (RUN) Threshold vs
Temperature
1.5
1.3
ON
1.2
OFF
1.1
1.0
–50
–25
50
0
75
25
TEMPERATURE (°C)
100
1000
900
0.606
OSCILLATOR FREQUENCY (kHz)
REGULATED FEEDBACK VOLTAGE (V)
0.608
1.4
RUN PIN VOLTAGE (V)
Oscillator Frequency vs
Temperature
Regulated Feedback Voltage
vs Temperature
0.604
0.602
0.600
0.598
50
25
75
0
TEMPERATURE (°C)
100
3865 G15
5
700
600
VFREQ = 1.2V
500
400
VFREQ = 0V
300
200
0.596
–50 –25
125
VFREQ = INTVCC
800
100
–50
125
–25
0
25
50
75
TEMPERATURE (°C)
3865 G16
Undervoltage Lockout Threshold
(INTVCC) vs Temperature
100
125
3865 G17
Oscillator Frequency
vs Input Voltage
Shutdown Current
vs Input Voltage
50
550
540
RISING
40
3
2
1
INPUT CURRENT (µA)
530
FALLING
FREQUENCY (kHz)
INTVCC VOLTAGE (V)
4
520
510
500
490
480
30
20
10
470
460
0
–50
–25
50
0
75
25
TEMPERATURE (°C)
100
125
450
10
4
22
28
16
INPUT VOLTAGE (V)
34
3865 G18
QUIESCENT CURRENT (mA)
SHUTDOWN CURRENT (µA)
5
40
30
20
10
–25
50
0
75
25
TEMPERATURE (°C)
0
5
10
15 20 25 30
INPUT VOLTAGE (V)
100
125
3865 G21
35
40
3865 G20
Quiescent Current vs Temperature
without EXTVCC
VIN = 15V
0
–50
0
3865 G19
Shutdown Current vs Temperature
50
40
VIN = 15V
4
3
2
1
0
–50
–25
50
0
75
25
TEMPERATURE (°C)
100
125
3865 G22
3865f
LTC3865/LTC3865-1
Pin Functions
(QFN/TSSOP)
VOSENSE1, VOSENSE2 (Pins 1, 8/Pins 5, 13): When the
internal programmable resistive divider is used, these
pins must be connected to their corresponding outputs.
When an external resistive divider is used, these pins are
used for error amplifier feedback inputs. They receive
the remotely sensed feedback voltages for each channel
directly from the outputs or from the external divider
across the outputs.
TK/SS1, TK/SS2 (Pins 2, 7/Pins 6, 12): Output Voltage
Tracking and Soft-Start Inputs. When one channel is
configured to be master of the two channels, a capacitor
to ground at this pin sets the ramp rate for the master
channel’s output voltage. When a channel is configured
to be the slave of the two channels, the output voltage
ramp of the master channel can be reproduced by a resistor divider and applied to this pin of the slave channel.
Internal soft-start currents of 1.3µA charge the soft-start
capacitors.
ITH1, ITH2 (Pins 3, 6/Pins 7, 11): Current Control Thresholds
and Error Amplifier Compensation Points. Each associated
channels’ current comparator tripping threshold increases
with its ITH control voltage.
VID11, VID12, VID21, VID22 (Pins 4, 5, 12, 28/ Pins 9,
10, 18, 37): VID Inputs for Output Voltage Programming.
Tie these pins to INTVCC, GND or leave them floating to
set the output voltages.
ILIM (Pin 13/Pin 19) (LTC3865 Only): Current Comparator Sense Voltage Range Inputs. This pin can be tied to
SGND, FLOAT or INTVCC to set the maximum current
sense threshold for each comparator. Current comparator sense voltage range of the LTC3865-1 is set to default
value of 50mV.
PGOOD (Pin 14 LTC3865/NA): Co-Bonded Power Good
Indicator Output for LTC3865 in QFN Package. Open-drain
logic output that is pulled to ground when either channel
output exceeds ±10% regulation window, after the internal
20µs power bad mask timer expires.
PGOOD1, PGOOD2 (Pins 14, 13 LTC3865-1/Pins 21, 20):
Separate Power Good Indicator Outputs for LTC3865-1 in
QFN package and LTC3865 in FE package. Open-drain logic
output that is pulled to ground when the corresponding
channel output exceeds ±10% regulation window, after
the internal 20µs power bad mask timer expires.
PGND (Pin 18/NA): Power Ground Pin. Connect this pin
closely to the sources of the bottom N-channel MOSFETs,
the (–) terminal of CVCC and the (–) terminal of CIN.
EXTVCC (Pin 20/Pin 28): External Power Input to an Internal Switch Connected to INTVCC. This switch closes and
supplies the IC power, bypassing the internal low dropout
regulator, whenever EXTVCC is higher than 4.7V. Do not
exceed 6V on this pin.
INTVCC (Pin 21/Pin 29): Internal 5V Regulator Output. The
control circuits are powered from this voltage. Decouple
this pin to PGND with a minimum of 4.7µF low ESR tantalum or ceramic capacitor.
VIN (Pin 22/Pin 30): Main Input Supply. Decouple this pin
to PGND with a capacitor (0.1µF to 1µF).
BG1, BG2 (Pins 23, 19/Pin 31, 27): Bottom Gate Driver
Outputs. These pins drive the gates of the bottom N-channel MOSFETs between PGND and INTVCC.
BOOST1, BOOST2 (Pins 24, 17/Pins 33, 25): Boosted
Floating Driver Supplies. The (+) terminal of the booststrap
capacitors connect to these pins. These pins swing from a
diode voltage drop below INTVCC up to VIN + INTVCC.
TG1, TG2 (Pins 25, 16/Pins 34, 23): Top Gate Driver
Outputs. These are the outputs of floating drivers with
a voltage swing equal to INTVCC superimposed on the
switch nodes voltages.
SW1, SW2 (Pins 26, 15/Pins 35, 22): Switch Node
Connections to Inductors. Voltage swing at these pins
is from a Schottky diode (external) voltage drop below
ground to VIN.
3865f
LTC3865/LTC3865-1
Pin Functions
(QFN/TSSOP)
MODE/PLLIN (Pin 27/Pin 36): Force Continuous Mode,
Burst Mode or Pulse-Skip Mode Selection Pin and External
Synchronization Input to Phase Detector Pin. Connect this
pin to SGND to force both channels in continuous mode of
operation. Connect to INTVCC to enable pulse-skip mode of
operation. Leaving the pin floating will enable Burst Mode
operation. A clock on the pin will force the controller into
continuous mode of operation and synchronize the internal
oscillator with the clock on this pin.
FREQ (Pin 29/Pin 38): This pin sets the frequency of the
internal oscillator. A constant current of 7.5µA is flowing
out of this pin and a resistor connected to this pin sets
its DC voltage, which in turn, sets the frequency of the
internal oscillator.
RUN1, RUN2 (Pins 30, 11/Pins 1, 17): Run Control
Inputs. A voltage above 1.2V on either pin turns on the IC.
However, forcing either of these pins below 1.2V causes
the IC to shut down the circuitry required for that particular
channel. There are 1µA pull-up currents for these pins.
Once the RUN pin rises above 1.2V, an additional 4.5µA
pull-up current is added to the pin.
SENSE1+, SENSE2+ (Pins 31, 10/Pins 2, 16): Current
Sense Comparator Inputs. The (+) inputs to the current
comparators are normally connected to DCR sensing
networks or current sensing resistors.
SENSE1–, SENSE2– (Pins 32, 9/Pin 3, 15): Current Sense
Comparator Inputs. The (–) inputs to the current comparators are connected to the outputs.
SGND (Pin 33/Pin 8): Signal Ground. All small-signal
components and compensation components should
connect to this ground, which in turn connects to PGND
at one point. Pin 33 is the Exposed Pad and available for
QFN package.
SGND (Exposed Pad Pin 33/ Exposed Pad Pin 39): Signal
Ground. Must be soldered to PCB, providing a local ground
for the control components of the IC, and be tied to the
PGND pin under the IC.
PGND1, PGND2 (NA/Pins 32, 26): Power Ground Pin.
Connect this pin closely to the sources of the bottom
N-channel MOSFETs, the (–) terminal of CVCC and the (–)
terminal of CIN.
3865f
10
LTC3865/LTC3865-1
FUNCTIONAL Diagram
FREQ/FREQ
MODE/PLLIN
VID1
VID2
EXTVCC
VIN
7.5µA
+
4.7V
INPUT VID LOGIC
AND
RESISTIVE DIVIDERS
MODE/SYNC
DETECT
PLL-SYNC
AND
LPF
+
–
F
0.6V
CIN
5V
REG
+
–
VIN
INTVCC
INTVCC
F
BOOST
OSC
BURSTEN
S
R
3k
+
ON
–
ICMP
+
–
TG
FCNT
Q
IREV
CB
M1
SW
SWITCH
LOGIC
AND
ANTISHOOT
THROUGH
SENSE+
L1
SENSE–
+
RUN
M2
CVCC
SLOPE COMPENSATION
PGND
PGOOD
INTVCC
UVLO
+
1
51k
ITHB
SLOPE RECOVERY
ACTIVE CLAMP
0.54V
VOSENSE
UV
–
SLEEP
VIN
SGND
+
VFB
–
0.66V
OV
–
–
+
SS
+
–
RUN
+
1.3µA
EA
– + +
0.6V
REF
COUT
BG
OV
ILIM
VOUT
DB
0.5V
1.2V
1µA
0.55V
ITH
RC
CC1
RUN
TK/SS
CSS
3865 FBD
3865f
11
LTC3865/LTC3865-1
Operation
Main Control Loop
The LTC3865/LTC3865-1 are constant-frequency, current
mode step-down controllers with two channels operating
180 degrees out-of-phase. During normal operation, each
top MOSFET is turned on when the clock for that channel
sets the RS latch, and turned off when the main current
comparator, ICMP, resets the RS latch. The peak inductor
current at which ICMP resets the RS latch is controlled by
the voltage on the ITH pin, which is the output of each error
amplifier, EA (refer to the Functional Diagram). VFB is the
voltage feedback signal, which is compared to the internal
reference voltage by the EA. When the load current increases, it causes a slight decrease in feedback voltage relative
to the 0.6V reference, which in turn causes the ITH voltage
to increase until the average inductor current matches the
new load current. After the top MOSFET has turned off, the
bottom MOSFET is turned on until either the inductor current starts to reverse, as indicated by the reverse current
comparator IREV, or the beginning of the next cycle.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin.
When the EXTVCC pin is left open or tied to a voltage less
than 4.7V, an internal 5V linear regulator supplies INTVCC
power from VIN. If EXTVCC is taken above 4.7V, the 5V
regulator is turned off and an internal switch is turned
on connecting EXTVCC. Using the EXTVCC pin allows the
INTVCC power to be derived from a high efficiency external
source such as one of the LTC3865/LTC3865-1 switching
regulator outputs.
Each top MOSFET driver is biased from the floating bootstrap capacitor, CB, which normally recharges during each
off cycle through an external diode when the top MOSFET
turns off. If the input voltage VIN decreases to a voltage
close to VOUT, the loop may enter dropout and attempt
to turn on the top MOSFET continuously. The dropout
detector detects this and forces the top MOSFET off for
about one-twelfth of the clock period every third cycle to
allow CB to recharge. However, it is recommended that a
load be present during the drop-out transition to ensure
CB is recharged.
Shutdown and Start-Up (RUN1, RUN2 and TK/SS1,
TK/SS2 Pins)
The two channels of the LTC3865/LTC3865-1 can be independently shut down using the RUN1 and RUN2 pins.
Pulling either of these pins below 1.2V shuts down the
main control loop for that controller. Pulling both pins
low disables both controllers and most internal circuits,
including the INTVCC regulator. Releasing either RUN pin
allows an internal 1µA current to pull up the pin and enable
that controller. Alternatively, the RUN pin may be externally
pulled up or driven directly by logic. Be careful not to exceed
the absolute maximum rating of 6V on this pin.
The start-up of each controller’s output voltage VOUT is
controlled by the voltage on the TK/SS1 and TK/SS2 pins.
When the voltage on the TK/SS pin is less than the 0.6V
internal reference, the LTC3865 regulates the VFB voltage
to the TK/SS pin voltage instead of the 0.6V reference.
This allows the TK/SS pin to be used to program a softstart by connecting an external capacitor from the TK/SS
pin to SGND. An internal 1.3µA pull-up current charges
this capacitor, creating a voltage ramp on the TK/SS pin.
As the TK/SS voltage rises linearly from 0V to 0.6V (and
beyond), the output voltage, VOUT , rises smoothly from
zero to its final value. Alternatively the TK/SS pin can
be used to cause the start-up of VOUT to “track” that of
another supply. Typically, this requires connecting to the
TK/SS pin an external resistor divider from the other supply to ground (see the Applications Information section).
When the corresponding RUN pin is pulled low to disable
a controller, or when INTVCC drops below its undervoltage
lockout threshold of 3.3V, the TK/SS pin is pulled low by
an internal MOSFET. When in undervoltage lockout, both
controllers are disabled and the external MOSFETs are
held off.
Light Load Current Operation (Burst Mode Operation,
Pulse-Skipping or Continuous Conduction)
The LTC3865/LTC3865-1 can be enabled to enter high
efficiency Burst Mode operation, constant-frequency pulseskipping mode, or forced continuous conduction mode. To
select forced continuous operation, tie the MODE/PLLIN
3865f
12
LTC3865/LTC3865-1
Operation
pin to a DC voltage below 0.6V (e.g., SGND). To select
pulse-skipping mode of operation, tie the MODE/PLLIN
pin to INTVCC. To select Burst Mode operation, float the
MODE/PLLIN pin. When a controller is enabled for Burst
Mode operation, the peak current in the inductor is set to
approximately one-third of the maximum sense voltage
even though the voltage on the ITH pin indicates a lower
value. If the average inductor current is higher than the
load current, the error amplifier EA will decrease the voltage
on the ITH pin. When the ITH voltage drops below 0.5V, the
internal sleep signal goes high (enabling “sleep” mode)
and both external MOSFETs are turned off.
In sleep mode, the load current is supplied by the output
capacitor. As the output voltage decreases, the EA’s output
begins to rise. When the output voltage drops enough, the
sleep signal goes low, and the controller resumes normal
operation by turning on the top external MOSFET on the
next cycle of the internal oscillator. When a controller is
enabled for Burst Mode operation, the inductor current is
not allowed to reverse. The reverse current comparator
(IREV) turns off the bottom external MOSFET just before the
inductor current reaches zero, preventing it from reversing and going negative. Thus, the controller operates in
discontinuous operation. In forced continuous operation,
the inductor current is allowed to reverse at light loads or
under large transient conditions. The peak inductor current is determined by the voltage on the ITH pin, just as
in normal operation. In this mode, the efficiency at light
loads is lower than in Burst Mode operation. However,
continuous mode has the advantages of lower output
ripple and less interference with audio circuitry.
When the MODE/PLLIN pin is connected to INTVCC, the
LTC3865 operates in PWM pulse-skipping mode at light
loads. At very light loads, the current comparator, ICMP ,
may remain tripped for several cycles and force the external
top MOSFET to stay off for the same number of cycles (i.e.,
skipping pulses). The inductor current is not allowed to
reverse (discontinuous operation). This mode, like forced
continuous operation, exhibits low output ripple as well as
low audio noise and reduced RF interference as compared
to Burst Mode operation. It provides higher low current
efficiency than forced continuous mode, but not nearly as
high as Burst Mode operation.
Frequency Selection and Phase-Locked Loop
(FREQ and MODE/PLLIN Pins)
The selection of switching frequency is a trade-off between
efficiency and component size. Low frequency operation increases efficiency by reducing MOSFET switching
losses, but requires larger inductance and/or capacitance
to maintain low output ripple voltage. The switching
frequency of the LTC3865’s controllers can be selected
using the FREQ pin. If the MODE/PLLIN pin is not being
driven by an external clock source, the FREQ pin can be
used to program the controller’s operating frequency from
250kHz to 770kHz.
There is a precision 7.5µA current flow out of FREQ pin
that user can program the controller’s switching frequency
with a single resistor to SGND. A curve is provided later in
the application section showing the relationship between
the voltage on the FREQ pin and switching frequency.
A phase-locked loop (PLL) is integrated on the LTC3865
to synchronize the internal oscillator to an external clock
source that is connected to the MODE/PLLIN pin. The
controller is operating in forced continuous mode when
it is synchronized.
The PLL loop filter network is integrated inside the
LTC3865/LTC3865-1. The phase-locked loop is capable
of locking any frequency within the range of 250kHz to
770kHz. The frequency setting resistor should always be
present to set the controller’s initial switching frequency
before locking to the external clock.
Power Good (PGOOD Pins)
On the LTC3865 (UH32 package), the PGOOD pin is
bonded to the open drains of two individual internal Nchannel MOSFETs. When either VOSENSE voltage is not
within ±10% of the programmed voltage, the PGOOD pin
is pulled low. The PGOOD pin is also pulled low when
either RUN pin is below 1.2V or when the LTC3865 is in
the soft-start or tracking phase. The PGOOD pin will flag
power good immediately when both VOSENSE are within
the ±10% of the programmed output voltage window.
However, there is an internal 20µs power bad mask when
either VOSENSE goes out the ±10% window. The internal
3865f
13
LTC3865/LTC3865-1
Operation
power bad mask is 100µs when there are any VID transitions. On the LTC3865‑1 (UH32 package) or the LTC3865
(FE38 package), each channel has its own PGOOD pin.
Therefore, the PGOOD pins now only respond to their own
channels. The PGOOD pins are allowed to be pulled up by
external resistors to sources of up to 6V.
Output Overvoltage Protection
An overvoltage comparator, OV, guards against transient
overshoots (>10%) as well as other more serious conditions that may overvoltage the output. In such cases,
the top MOSFET is turned off and the bottom MOSFET is
turned on until the reverse current limit for the overvoltage
condition is reached. The bottom MOSFET will be turned on
again at the next clock and be turned off when the reverse
current limit is reached again. This process repeats until
the overvoltage condition is cleared.
Output Voltage Programming
The output voltages of both channels of the LTC3865/
LTC3865-1 can be programmed to a preset value. There
are two VID pins for each channel and by connecting these
pins to INTVCC, GND, or by floating them, the output voltages can be set to the values in Table 1.
Table 1. Programming of Output Voltage
VID11/VID21
VID12/VID22
VOUT1/VOUT2 (V)
INTVCC
INTVCC
5.0
INTVCC
Float
3.3
INTVCC
GND
2.5
Float
INTVCC
1.8
Float
Float
0.6 or External Divider
Float
GND
1.5
GND
INTVCC
1.2
GND
Float
1.0
GND
GND
1.1
Applications Information
The Typical Application on the first page is a basic LTC3865
application circuit. The LTC3865 can be configured to use
either DCR (inductor resistance) sensing or low value resistor sensing. The choice between the two current sensing
schemes is largely a design trade-off between cost, power
consumption and accuracy. DCR sensing is becoming
popular because it saves expensive current sensing resistors and is more power efficient, especially in high current
applications. However, current sensing resistors provide
the most accurate current limits for the controller. Other
external component selection is driven by the load requirement, and begins with the selection of RSENSE (if RSENSE is
used) and inductor value. Next, the power MOSFETs are selected. Finally, input and output capacitors are selected.
Current Limit Programming
The ILIM pin is a tri-level logic input which sets the maximum current limit of the controller. When ILIM is either
grounded, floated or tied to INTVCC, the typical value for
the maximum current sense threshold will be 30mV, 50mV
or 75mV, respectively.
Which setting should be used? For the best current limit
accuracy, use the 75mV setting. The 30mV setting will
allow for the use of very low DCR inductors or sense
resistors, but at the expense of current limit accuracy.
The 50mV setting is a good balance between the two. For
single output dual phase applications, use the 50mV or
75mV setting for optimal current sharing.
SENSE+ and SENSE– Pins
The SENSE+ and SENSE– pins are the inputs to the current
comparators. The common mode input voltage range of
the current comparators is 0V to 5V. Both SENSE pins
are high impedance inputs with small base currents of
less than 1µA. When the SENSE pins ramp up from 0V to
1.4V, the small base currents flow out of the SENSE pins.
When the SENSE pins ramp down from 5V to 1.1V, the
small base currents flow into the SENSE pins. The high
impedance inputs to the current comparators allow accurate DCR sensing. However, care must be taken not to
float these pins during normal operation.
3865f
14
LTC3865/LTC3865-1
Applications Information
Filter components mutual to the sense lines should be
placed close to the LTC3865/LTC3865-1, and the sense
lines should run close together to a Kelvin connection
underneath the current sense element (shown in Figure 1).
Sensing current elsewhere can effectively add parasitic
inductance and capacitance to the current sense element,
degrading the information at the sense terminals and making the programmed current limit unpredictable. If DCR
sensing is used (Figure 2b), sense resistor R1 should be
placed close to the switching node, to prevent noise from
coupling into sensitive small-signal nodes. The capacitor
C1 should be placed close to the IC pins.
VIN
INTVCC
VIN
SENSE RESISTOR
PLUS PARASITIC
INDUCTANCE
BOOST
TG
RS
LTC3865 SW
BG
SENSE–
SGND
VOUT
CF • 2RF ≤ ESL/RS
POLE-ZERO
CANCELLATION
PGND
SENSE+
ESL
RF
CF
RF
FILTER COMPONENTS
PLACED NEAR SENSE PINS
3865 F02a
(2a) Using a Resistor to Sense Current
TO SENSE FILTER,
NEXT TO THE CONTROLLER
3865 F01
COUT
INDUCTOR OR RSENSE
Figure 1. Sense Lines Placement with Inductor or Sense Resistor
Low Value Resistors Current Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 2a. RSENSE is chosen based on the required
output current.
The current comparator has a maximum threshold
VSENSE(MAX) determined by the ILIM setting. The input
common mode range of the current comparator is 0V
to 5V. The current comparator threshold sets the peak of
the inductor current, yielding a maximum average output
current IMAX equal to the peak value less half the peak-topeak ripple current, ∆IL. To calculate the sense resistor
value, use the equation:
VSENSE(MAX )
∆I
I(MAX ) + L
2
Because of possible PCB noise in the current sensing loop,
the AC current sensing ripple of ∆VSENSE = ∆IL • RSENSE
also needs to be checked in the design to get a good
signal-to-noise ratio. In general, for a reasonably good
PCB layout, a 15mV ∆VSENSE voltage is recommended
as a conservative number to start with, either for RSENSE
or DCR sensing applications.
RSENSE =
VIN
INTVCC
VIN
BOOST
INDUCTOR
TG
LTC3865
L
SW
DCR
VOUT
BG
PGND
R1
SENSE+
C1*
R2
SENSE–
SGND
*PLACE C1 NEAR SENSE+,
SENSE– PINS
R1||R2 • C1 =
L
DCR
RSENSE(EQ) = DCR
R2
R1 + R2
3865 F02b
(2b) Using the Inductor DCR to Sense Current
Figure 2. Two Different Methods of Sensing Current
For previous generation current mode controllers, the
maximum sense voltage was high enough (e.g., 75mV for
the LTC1628 / LTC3728 family) that the voltage drop across
the parasitic inductance of the sense resistor represented
a relatively small error. For today’s highest current density
solutions, however, the value of the sense resistor can be
less than 1mΩ and the peak sense voltage can be as low
as 20mV. In addition, inductor ripple currents greater than
50% with operation up to 1MHz are becoming more common. Under these conditions the voltage drop across the
sense resistor’s parasitic inductance is no longer negligible.
A typical sensing circuit using a discrete resistor is shown
in Figure 2a. In previous generations of controllers, a small
RC filter placed near the IC was commonly used to reduce
the effects of capacitive and inductive noise coupled in
3865f
15
LTC3865/LTC3865-1
Applications Information
the sense traces on the PCB. A typical filter consists of
two series 10Ω resistors connected to a parallel 1000pF
capacitor, resulting in a time constant of 20ns.
This same RC filter, with minor modifications, can be used
to extract the resistive component of the current sense
signal in the presence of parasitic inductance. For example,
Figure 3 illustrates the voltage waveform across a 2mΩ
sense resistor with a 2010 footprint for the 1.2V/15A
converter operating at 100% load. The waveform is the
superposition of a purely resistive component and a
purely inductive component. It was measured using two
scope probes and waveform math to obtain a differential
measurement. Based on additional measurements of the
inductor ripple current and the on-time and off-time of
the top switch, the value of the parasitic inductance was
determined to be 0.5nH using the equation:
ESL =
VESL(STEP) tON • tOFF
∆IL
tON + tOFF
If the RC time constant is chosen to be close to the parasitic
inductance divided by the sense resistor (L/R), the resulting waveform looks resistive again, as shown in Figure 4.
For applications using low maximum sense voltages,
check the sense resistor manufacturer’s data sheet for
information about parasitic inductance. In the absence of
data, measure the voltage drop directly across the sense
resistor to extract the magnitude of the ESL step and use
the equation above to determine the ESL. However, do not
over filter. Keep the RC time constant less than or equal
to the inductor time constant to maintain a high enough
ripple voltage on VRSENSE.
The above generally applies to high density/high current
applications where I(MAX) >10A and low values of inductors are used. For applications where I(MAX) < 10A, set
RF to 10Ω and CF to 1000pF. This will provide a good
starting point.
The filter components need to be placed close to the IC.
The positive and negative sense traces need to be routed
as a differential pair and Kelvin connected to the sense
resistor.
Inductor DCR Sensing
VSENSE
20mV/DIV
VESL(STEP)
500ns/DIV
3865 F03
Figure 3. Voltage Waveform Measured
Directly Across the Sense Resistor
VSENSE
20mV/DIV
500ns/DIV
3865 F04
Figure 4. Voltage Waveform Measured After the
Sense Resistor Filter. CF = 1000pF, RF = 100Ω
For applications requiring the highest possible efficiency
at high load currents, the LTC3865/LTC3865-1 are capable
of sensing the voltage drop across the inductor DCR, as
shown in Figure 2b. The DCR of the inductor represents
the small amount of DC winding resistance of the copper,
which can be less than 1mΩ for today’s low value, high
current inductors. In a high current application requiring
such an inductor, conduction loss through a sense resistor would cost several points of efficiency compared to
DCR sensing.
If the external R1||R2 • C1 time constant is chosen to be
exactly equal to the L/DCR time constant, the voltage drop
across the external capacitor is equal to the drop across
the inductor DCR multiplied by R2/(R1 + R2). R2 scales the
voltage across the sense terminals for applications where
the DCR is greater than the target sense resistor value.
To properly dimension the external filter components, the
DCR of the inductor must be known. It can be measured
using a good RLC meter, but the DCR tolerance is not
3865f
16
LTC3865/LTC3865-1
Applications Information
always the same and varies with temperature; consult the
manufacturers’ data sheets for detailed information.
Using the inductor ripple current value from the Inductor Value Calculation section, the target sense resistor
value is:
RSENSE(EQUIV ) =
VSENSE(MAX )
∆I
I(MAX ) + L
2
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimum value for the Maximum Current Sense Threshold
(VSENSE(MAX)) in the Electrical Characteristics table (24mV,
44mV or 68mV, depending on the state of the ILIM pin).
Next, determine the DCR of the inductor. Where provided,
use the manufacturer’s maximum value, usually given at
20°C. Increase this value to account for the temperature
coefficient of resistance, which is approximately 0.4%/°C.
A conservative value for TL(MAX) is 100°C.
To scale the maximum inductor DCR to the desired sense
resistor value, use the divider ratio:
RD =
RSENSE(EQUIV )
DCR(MAX ) at TL(MAX )
C1 is usually selected to be in the range of 0.047µF to
0.47µF. This forces R1||R2 to around 2kΩ, reducing error
that might have been caused by the SENSE pins’ ±1µA
current.
The equivalent resistance R1||R2 is scaled to the room
temperature inductance and maximum DCR:
R1|| R2 =
L
(DCR at 20°C ) • C1
The sense resistor values are:
R1 =
R1|| R2
R1 • RD
; R2 =
RD
1 − RD
The maximum power loss in R1 is related to duty cycle,
and will occur in continuous mode at the maximum input
voltage:
PLOSS R1=
(V
IN(MAX ) − VOUT
R1
)• V
OUT
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
sense resistors. Light load power loss can be modestly
higher with a DCR network than with a sense resistor, due
to the extra switching losses incurred through R1. However,
DCR sensing eliminates a sense resistor, reduces conduction losses and provides higher efficiency at heavy loads.
Peak efficiency is about the same with either method.
To maintain a good signal to noise ratio for the current
sense signal, use a minimum ∆VSENSE of 10mV to 15mV.
For a DCR sensing application, the actual ripple voltage
will be determined by the equation:
∆VSENSE =
VIN − VOUT VOUT
R1• C1 VIN • fOSC
Slope Compensation and Inductor Peak Current
Slope compensation provides stability in constantfrequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally
by adding a compensating ramp to the inductor current
signal at duty cycles in excess of 40%. Normally, this
results in a reduction of maximum inductor peak current
for duty cycles >40%. However, the LTC3865/LTC3865-1
use a patented scheme that counteracts this compensating
ramp, which allows the maximum inductor peak current
to remain unaffected throughout all duty cycles.
Inductor Value Calculation
Given the desired input and output voltages, the inductor
value and operating frequency fOSC directly determine the
inductor’s peak-to-peak ripple current:
IRIPPLE =
VOUT  VIN – VOUT 
VIN  fOSC • L 
3865f
17
LTC3865/LTC3865-1
Applications Information
Lower ripple current reduces core losses in the inductor,
ESR losses in the output capacitors, and output voltage
ripple. Thus, highest efficiency operation is obtained at
low frequency with a small ripple current. Achieving this,
however, requires a large inductor.
A reasonable starting point is to choose a ripple current
that is about 40% of IOUT(MAX). Note that the largest ripple
current occurs at the highest input voltage. To guarantee
that ripple current does not exceed a specified maximum,
the inductor should be chosen according to:
L≥
VIN – VOUT VOUT
•
fOSC • IRIPPLE VIN
Inductor Core Selection
Once the inductance value is determined, the type of inductor must be selected. Core loss is independent of core
size for a fixed inductor value, but it is very dependent
on inductance selected. As inductance increases, core
losses go down. Unfortunately, increased inductance
requires more turns of wire and therefore copper losses
will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Power MOSFET and Schottky Diode
(Optional) Selection
Two external power MOSFETs must be selected for each
controller in the LTC3865/LTC3865-1: one N-channel
MOSFET for the top (main) switch, and one N-channel
MOSFET for the bottom (synchronous) switch.
The peak-to-peak drive levels are set by the INTVCC
voltage. This voltage is typically 5V during start-up
(see EXTVCC Pin Connection). Consequently, logic-level
threshold MOSFETs must be used in most applications.
The only exception is if low input voltage is expected
(VIN < 5V); then, sub-logic level threshold MOSFETs
(VGS(TH) < 3V) should be used. Pay close attention to the
BVDSS specification for the MOSFETs as well; most of the
logic-level MOSFETs are limited to 30V or less.
Selection criteria for the power MOSFETs include the onresistance, RDS(ON) , Miller capacitance, CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result is
then multiplied by the ratio of the application applied VDS
to the gate charge curve specified VDS. When the IC is
operating in continuous mode the duty cycles for the top
and bottom MOSFETs are given by:
Main Switch Duty Cycle =
VOUT
VIN
Synchronous Switch Duty Cycle =
VIN – VOUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
VOUT
2
IMAX ) (1+ d) RDS(ON) +
(
VIN

2 I
( VIN )  MAX
 (RDR )(CMILLER ) •
2 

1
1 

 • fOSC
+
 VINTVCC – VTH(MIN) VTH(MIN) 
PMAIN =
PSYNC =
VIN – VOUT
2
IMAX ) (1+ d) RDS(ON)
(
VIN
where d is the temperature dependency of RDS(ON) and
RDR (approximately 2Ω) is the effective driver resistance
at the MOSFET’s Miller threshold voltage. VTH(MIN) is the
typical MOSFET minimum threshold voltage.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
3865f
18
LTC3865/LTC3865-1
Applications Information
which are highest at high input voltages. For VIN < 20V
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short-circuit when the synchronous switch is on close
to 100% of the period.
The term (1 + d) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
d = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
The optional Schottky diodes conduct during the dead time
between the conduction of the two power MOSFETs. These
prevent the body diodes of the bottom MOSFETs from turning on, storing charge during the dead time and requiring
a reverse recovery period that could cost as much as 3%
in efficiency at high VIN. A 1A to 3A Schottky is generally
a good compromise for both regions of operation due to
the relatively small average current. Larger diodes result
in additional transition losses due to their larger junction
capacitance.
Soft-Start and Tracking
The LTC3865/LTC3865-1 have the ability to either soft-start
by themselves with a capacitor or track the output of another
channel or external supply. When one particular channel
is configured to soft-start by itself, a capacitor should be
connected to its TK/SS pin. This channel is in the shutdown
state if its RUN pin voltage is below 1.2V. Its TK/SS pin is
actively pulled to ground in this shutdown state.
Once the RUN pin voltage is above 1.2V, the channel powers up. A soft-start current of 1.3µA then starts to charge
its soft-start capacitor. Note that soft-start or tracking is
achieved not by limiting the maximum output current of
the controller but by controlling the output ramp voltage
according to the ramp rate on the TK/SS pin. Current
foldback is disabled during this phase to ensure smooth
soft-start or tracking. The soft-start or tracking range is
defined to be the voltage range from 0V to 0.6V on the
TK/SS pin. The total soft-start time can be calculated as:
t SOFTSTART = 0.6 •
CSS
1.3 µA
Regardless of the mode selected by the MODE/PLLIN pin,
the regulator will always start in pulse-skipping mode
up to TK/SS = 0.5V. Between TK/SS = 0.5V and 0.54V, it
will operate in forced continuous mode and revert to the
selected mode once TK/SS > 0.54V. The output ripple
is minimized during the 40mV forced continuous mode
window ensuring a clean PGOOD signal.
When the channel is configured to track another supply,
the feedback voltage of the other supply is duplicated by
a resistor divider and applied to the TK/SS pin. Therefore,
the voltage ramp rate on this pin is determined by the
ramp rate of the other supply’s voltage. Note that the small
soft-start capacitor charging current is always flowing,
producing a small offset error. To minimize this error, select
the tracking resistive divider value to be small enough to
make this error negligible.
In order to track down another channel or supply after
the soft-start phase expires, the LTC3865/LTC3865-1 are
forced into continuous mode of operation as soon as VFB
is below the undervoltage threshold of 0.54V regardless
of the setting on the MODE/PLLIN pin. However, the
LTC3865/LTC3865-1 should always be set in force continuous mode tracking down when there is no load. After
TK/SS drops below 0.1V, the corresponding channel will
operate in discontinuous mode.
Output Voltage Tracking
The LTC3865/LTC3865-1 allow the user to program how its
output ramps up and down by means of the TK/SS pins.
Through these pins, the output can be set up to either coincidentally or ratiometrically track another supply’s output,
as shown in Figure 5. In the following discussions, VOUT1
refers to the LTC3865/LTC3865-1’s output 1 as a master
channel and VOUT2 refers to the LTC3865/LTC3865‑1’s output 2 as a slave channel. In practice, though, either phase
can be used as the master. To implement the coincident
tracking in Figure 5a, connect an additional resistive divider
to VOUT1 and connect its midpoint to the TK/SS pin of the
slave channel. The ratio of this divider should be the same
as that of the slave channel’s internal feedback divider
3865f
19
LTC3865/LTC3865-1
Applications Information
VOUT1
OUTPUT VOLTAGE
OUTPUT VOLTAGE
VOUT1
VOUT2
VOUT2
TIME
TIME
(5a) Coincident Tracking
3865 F05
(5b) Ratiometric Tracking
Figure 5. Two Different Modes of Output Voltage Tracking
VOUT1
TO
TK/SS2
PIN
VOUT1
VOUT2
nR3
R1
R3
TO
TK/SS2
PIN
TO
TO
EA1 EA2
nR4
R2
R4
VOUT2
nR1
R1
R3
TO
TO
EA1 EA2
nR2
R2
R4
38551 F06
(6a) Coincident Tracking Setup
(6b) Ratiometric Tracking Setup
Figure 6. Setup for Coincident and Ratiometric Tracking
I
I
+
D1
D2
EA2
TK/SS2
0.6V
–
3865 F07
D3
VFB2
Figure 7. Equivalent Input Circuit of Error Amplifier
shown in Figure 6a. In this tracking mode, VOUT1 must
be set higher than VOUT2. To implement the ratiometric
tracking, the ratio of the VOUT2 divider should be exactly
the same as the master channel’s internal feedback divider.
By selecting different resistors, the LTC3865/LTC3865-1
can achieve different modes of tracking including the two
in Figure 5.
So which mode should be programmed? The coincident
mode offers better output regulation. This can be better
understood with the help of Figure 7. At the input stage
of the slave channel’s error amplifier, two common anode diodes are used to clamp the equivalent reference
20
voltage and an additional diode is used to match the shifted
common mode voltage. The top two current sources are
of the same magnitude. In coincident mode, the TK/SS
voltage is substantially higher than 0.6V at steady state and
effectively turns off D1. D2 and D3 will therefore conduct
the same current and offer tight matching between VFB2
and the internal precision 0.6V at steady state. In the
ratiometric mode, however, TK/SS equals 0.6V at steady
state. D1 will divert part of the bias current to make VFB2
slightly lower than 0.6V. Although this error is minimized
by the exponential I-V characteristics of the diode, it does
impose a finite amount of output voltage deviation.
3865f
LTC3865/LTC3865-1
Applications Information
When the master channel’s output experiences dynamic
excursion (under load transient, for example), the slave
channel output will be affected as well. For better output
regulation, use the coincident tracking mode instead of
ratiometric.
INTVCC Regulators and EXTVCC
The LTC3865 features a true PMOS LDO that supplies
power to INTVCC from the VIN supply. INTVCC powers the
gate drivers and much of the LTC3865/LTC3865-1’s internal
circuitry. The linear regulator regulates the voltage at the
INTVCC pin to 5V when VIN is greater than 5.5V. EXTVCC
connects to INTVCC through a P-channel MOSFET and can
supply the needed power when its voltage is higher than
4.7V. Each of these can supply a peak current of 80mA
and must be bypassed to ground with a minimum of 4.7µF
ceramic capacitor or low ESR electrolytic capacitor. No matter what type of bulk capacitor is used, an additional 0.1µF
ceramic capacitor placed directly adjacent to the INTVCC
and PGND pins is highly recommended. Good bypassing
is needed to supply the high transient currents required
by the MOSFET gate drivers and to prevent interaction
between channels.
High input voltage applications in which large MOSFETs are
being driven at high frequencies may cause the maximum
junction temperature rating for the LTC3865/LTC3865-1
to be exceeded. The INTVCC current, which is dominated
by the gate charge current, may be supplied by either the
5V linear regulator or EXTVCC. When the voltage on the
EXTVCC pin is less than 4.7V, the linear regulator is enabled.
Power dissipation for the IC in this case is highest and is
equal to VIN • IINTVCC. The gate charge current is dependent on operating frequency as discussed in the Efficiency
Considerations section. The junction temperature can be
estimated by using the equations given in Note 3 of the
Electrical Characteristics. For example, the LTC3865 INTVCC
current is limited to less than 42mA from a 38V supply in
the UH package and not using the EXTVCC supply:
TJ = 70°C + (42mA)(38V)(34°C/W) = 125°C
To prevent the maximum junction temperature from being
exceeded, the input supply current must be checked while
operating in continuous conduction mode (MODE/PLLIN
= SGND) at maximum VIN. When the voltage applied to
EXTVCC rises above 4.7V, the INTVCC linear regulator is
turned off and the EXTVCC is connected to the INTVCC.
The EXTVCC remains on as long as the voltage applied to
EXTVCC remains above 4.5V. Using the EXTVCC allows the
MOSFET driver and control power to be derived from one
of the LTC3865/LTC3865-1’s switching regulator outputs
during normal operation and from the INTVCC when the
output is out of regulation (e.g., start-up, short-circuit). If
more current is required through the EXTVCC than is specified, an external Schottky diode can be added between the
EXTVCC and INTVCC pins. Do not apply more than 6V to
the EXTVCC pin and make sure that EXTVCC < VIN.
Significant efficiency and thermal gains can be realized by
powering INTVCC from the output, since the VIN current
resulting from the driver and control currents will be scaled
by a factor of (Duty Cycle)/(Switcher Efficiency).
Tying the EXTVCC pin to a 5V supply reduces the junction
temperature in the previous example from 125°C to:
TJ = 70°C + (42mA)(5V)(34°C/W) = 77°C
However, for 3.3V and other low voltage outputs, additional circuitry is required to derive INTVCC power from
the output.
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 5V regulator resulting in an efficiency penalty of up to 10% at high input
voltages.
2. EXTVCC connected directly to VOUT. This is the normal
connection for a 5V regulator and provides the highest
efficiency.
3. EXTVCC connected to an external supply. If a 5V external
supply is available, it may be used to power EXTVCC
providing it is compatible with the MOSFET gate drive
requirements.
4. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage that has been boosted to greater
than 4.7V.
3865f
21
LTC3865/LTC3865-1
Applications Information
For applications where the main input power is below 5V,
tie the VIN and INTVCC pins together and tie the combined
pins to the 5V input with a 1Ω or 2.2Ω resistor as shown
in Figure 8 to minimize the voltage drop caused by the
gate charge current. This will override the INTVCC linear
regulator and will prevent INTVCC from dropping too low
due to the dropout voltage. Make sure the INTVCC voltage
is at or exceeds the RDS(ON) test voltage for the MOSFET
which is typically 4.5V for logic-level devices.
Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors, CB, connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs.
Capacitor CB in the Functional Diagram is charged through
external diode DB from INTVCC when the SW pin is low.
When one of the topside MOSFETs is to be turned on, the
driver places the CB voltage across the gate source of the
desired MOSFET. This enhances the MOSFET and turns on
the topside switch. The switch node voltage, SW, rises to
VIN and the BOOST pin follows. With the topside MOSFET
on, the boost voltage is above the input supply: VBOOST
= VIN + VINTVCC. The value of the boost capacitor, CB,
needs to be 100 times that of the total input capacitance
of the topside MOSFET(s). The reverse breakdown of the
external Schottky diode must be greater than VIN(MAX).
When adjusting the gate drive level, the final arbiter is the
total input current for the regulator. If a change is made
and the input current decreases, then the efficiency has
improved. If there is no change in input current, then there
is no change in efficiency.
VIN
LTC3865
INTVCC
RVIN
1Ω
CINTVCC
4.7µF
+
5V
CIN
3865 F08
Figure 8. Setup for a 5V Input
Undervoltage Lockout
The LTC3865/LTC3865-1 have two functions that help
protect the controller in case of undervoltage conditions.
A precision UVLO comparator constantly monitors the
INTVCC voltage to ensure that an adequate gate-drive
voltage is present. It locks out the switching action when
INTVCC is below 3.3V. To prevent oscillation when there is
a disturbance on the INTVCC, the UVLO comparator has
550mV of precision hysteresis.
Another way to detect an undervoltage condition is to
monitor the VIN supply. Because the RUN pins have a
precision turn-on reference of 1.2V, one can use a resistor
divider to VIN to turn on the IC when VIN is high enough.
An extra 4.5µA of current flows out of the RUN pin once
the RUN pin voltage passes 1.2V. One can program the
hysteresis of the run comparator by adjusting the values
of the resistive divider. For accurate VIN undervoltage
detection, VIN needs to be higher than 4.5V.
CIN and COUT Selection
The selection of CIN is simplified by the 2-phase architecture and its impact on the worst-case RMS current drawn
through the input network (battery/fuse/capacitor). It can be
shown that the worst-case capacitor RMS current occurs
when only one controller is operating. The controller with
the highest (VOUT)(IOUT) product needs to be used in the
formula below to determine the maximum RMS capacitor
current requirement. Increasing the output current drawn
from the other controller will actually decrease the input
RMS ripple current from its maximum value. The out-ofphase technique typically reduces the input capacitor’s RMS
ripple current by a factor of 30% to 70% when compared
to a single phase power supply solution.
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle (VOUT)/(VIN). To prevent
large voltage transients, a low ESR capacitor sized for the
3865f
22
LTC3865/LTC3865-1
Applications Information
maximum RMS current of one channel must be used. The
maximum RMS capacitor current is given by:
CIN Re quired IRMS ≈
1/ 2
IMAX
( VOUT )( VIN – VOUT ) 
VIN
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturers’
ripple current ratings are often based on only 2000 hours
of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature
than required. Several capacitors may be paralleled to meet
size or height requirements in the design. Due to the high
operating frequency of the LTC3865, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
The benefit of the LTC3865/LTC3865-1 2-phase operation
can be calculated by using the equation above for the higher
power controller and then calculating the loss that would
have resulted if both controller channels switched on at
the same time. The total RMS power lost is lower when
both controllers are operating due to the reduced overlap of
current pulses required through the input capacitor’s ESR.
This is why the input capacitor’s requirement calculated
above for the worst-case controller is adequate for the dual
controller design. Also, the input protection fuse resistance,
battery resistance, and PC board trace resistance losses
are also reduced due to the reduced peak currents in a 2phase system. The overall benefit of a multiphase design
will only be fully realized when the source impedance of the
power supply/battery is included in the efficiency testing.
The sources of the top MOSFETs should be placed within
1cm of each other and share a common CIN(s). Separating
the sources and CIN may produce undesirable voltage and
current resonances at VIN.
A small (0.1µF to 1µF) bypass capacitor between the chip VIN
pin and ground, placed close to the LTC3865/LTC3865-1,
is also suggested. A 2.2Ω to 10Ω resistor placed between
CIN (C1) and the VIN pin provides further isolation between
the two channels.
The selection of COUT is driven by the effective series
resistance (ESR). Typically, once the ESR requirement
is satisfied, the capacitance is adequate for filtering. The
output ripple (∆VOUT) is approximated by:

1 
∆VOUT ≈ IRIPPLE  ESR +
8 fCOUT 

where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage
since IRIPPLE increases with input voltage.
Setting Output Voltage
The LTC3865/LTC3865-1 output voltages are each set
by the voltages at VID pins. Each of the VID pins can be
floated, or INTVCC or grounded, depending on what preset
voltages are needed at the output (Table 1).
If the desired output voltage is not one of the preset
values, select 0.6V and use 1% resistors to divide VOUT ,
as shown in Figure 9. The regulated output voltage is
determined by:
 R 
VOUT = 0.6 V •  1+ B 
 RA 
To improve the frequency response, a feed-forward capacitor, CFF , may be used. Great care should be taken to
route the VOSENSE line away from noise sources, such as
the inductor or the SW line.
VOUT
1/2 LTC3865
RB
CFF
VOSENSE
RA
3865 F09
Figure 9. Setting Output Voltage
3865f
23
LTC3865/LTC3865-1
Applications Information
Fault Conditions: Current Limit and Current Foldback
The LTC3865/LTC3865-1 include current foldback to help
limit load current when the output is shorted to ground.
If the output falls below 50% of its nominal output level,
then the maximum sense voltage is progressively lowered
from its maximum programmed value to one-third of the
maximum value. Foldback current limiting is disabled
during the soft-start or tracking up. Under short-circuit
conditions with very low duty cycles, the LTC3865 will begin
cycle skipping in order to limit the short-circuit current.
In this situation the bottom MOSFET will be dissipating
most of the power but less than in normal operation. The
short-circuit ripple current is determined by the minimum
on-time tON(MIN) of the LTC3865/LTC3865-1 (≈ 90ns), the
input voltage and inductor value:
∆IL(SC) = tON(MIN) •
VIN
L
The resulting short-circuit current is:
ISC =
1/ 3 VSENSE(MAX )
RSENSE
1
– ∆IL(SC)
2
Phase-Locked Loop and Frequency Synchronization
The LTC3865/LTC3865-1 have a phase-locked loop (PLL)
comprised of an internal voltage-controlled oscillator (VCO)
and a phase detector. This allows the turn-on of the top
MOSFET of controller 1 to be locked to the rising edge of an
external clock signal applied to the MODE/PLLIN pin. The
turn-on of controller 2’s top MOSFET is thus 180 degrees
out-of-phase with the external clock. The phase detector
is an edge sensitive digital type that provides zero degrees
phase shift between the external and internal oscillators.
This type of phase detector does not exhibit false lock to
harmonics of the external clock.
The output of the phase detector is a pair of complementary
current sources that charge or discharge the internal filter
network. There is a precision 7.5µA of current flowing out
of FREQ pin. This allows the user to use a single resistor
to SGND to set the switching frequency when no external
clock is applied to the MODE/PLLIN pin. The internal switch
between FREQ pin and the integrated PLL filter network is
on, allowing the filter network to be at the same voltage
potential as of FREQ pin. The relationship between the voltage on the FREQ pin and the operating frequency is shown
in Figure 10 and specified in the Electrical Characteristic
table. If an external clock is detected on the MODE/PLLIN
pin, the internal switch mentioned above will turn off and
isolate the influence of FREQ pin. Note that the LTC3865 can
only be synchronized to an external clock whose frequency
is within range of the LTC3865/LTC3865-1’s internal VCO.
This is guaranteed to be between 250kHz and 770kHz. A
simplified block diagram is shown in Figure 11.
2.4V 5V
900
7.5µA
800
FREQ
700
FREQUENCY (kHz)
RSET
600
EXTERNAL
OSCILLATOR
500
400
MODE/
PLLIN
DIGITAL
SYNC
PHASE/
FREQUENCY
DETECTOR
VCO
300
200
100
0
3865 F11
0
0.5
1
1.5
FREQ PIN VOLTAGE (V)
2
2.5
3865 F10
Figure 11. Phase-Locked Loop Block Diagram
Figure 10. Relationship Between Oscillator
Frequency and Voltage at the FREQ Pin
3865f
24
LTC3865/LTC3865-1
Applications Information
If the external clock frequency is greater than the internal oscillator’s frequency, fOSC, then current is sourced
continuously from the phase detector output, pulling up
the filter network. When the external clock frequency is
less than fOSC, current is sunk continuously, pulling down
the filter network. If the external and internal frequencies
are the same but exhibit a phase difference, the current
sources turn on for an amount of time corresponding to
the phase difference. The voltage on the filter network is
adjusted until the phase and frequency of the internal and
external oscillators are identical. At the stable operating
point, the phase detector output is high impedance and
the filter capacitor holds the voltage.
Typically, the external clock (on MODE/PLLIN pin)
input high threshold is 1.6V, while the input low thres-hold
is 1V. The external clock should not be applied when the
IC is in shutdown.
Minimum On-Time Considerations
Minimum on-time, tON(MIN), is the smallest time duration
that the LTC3865/LTC3865-1 is capable of turning on the
top MOSFET. It is determined by internal timing delays
and the gate charge required to turn on the top MOSFET.
Low duty cycle applications may approach this minimum
on-time limit and care should be taken to ensure that
tON(MIN) <
VOUT
VIN( f)
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
The minimum on-time for the LTC3865/LTC3865-1 is
approximately 90ns, with reasonably good PCB layout,
minimum 30% inductor current ripple and at least 10mV
to 15mV ripple on the current sense signal. The minimum on-time can be affected by PCB switching noise in
the voltage and current loop. As the peak sense voltage
decreases the minimum on-time gradually increases to
130ns. This is of particular concern in forced continuous
applications with low ripple current at light loads. If the
duty cycle drops below the minimum on-time limit in this
situation, a significant amount of cycle skipping can occur
with correspondingly larger current and voltage ripple.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3865/LTC3865-1 circuits: 1) IC VIN current,
2) INTVCC regulator current, 3) I2R losses, 4) Topside
MOSFET transition losses.
1. The VIN current is the DC supply current given in
the Electrical Characteristics table, which excludes
MOSFET driver and control currents. VIN current typically results in a small (<0.1%) loss.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the
control circuit current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges of
the topside and bottom side MOSFETs.
3865f
25
LTC3865/LTC3865-1
Applications Information
Supplying INTVCC power through EXTVCC from an
output-derived source will scale the VIN current required for the driver and control circuits by a factor
of (Duty Cycle)/(Efficiency). For example, in a 20V
to 5V application, 10mA of INTVCC current results in
approximately 2.5mA of VIN current. This reduces the
mid-current loss from 10% or more (if the driver was
powered directly from VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor.
In continuous mode, the average output current flows
through L and RSENSE, but is “chopped” between the
topside MOSFET and the synchronous MOSFET. If the
two MOSFETs have approximately the same RDS(ON),
then the resistance of one MOSFET can simply be
summed with the resistances of L and RSENSE to obtain I2R losses. For example, if each RDS(ON) = 10mΩ,
RL = 10mΩ, RSENSE = 5mΩ, then the total resistance
is 25mΩ. This results in losses ranging from 2% to
8% as the output current increases from 3A to 15A for
a 5V output, or a 3% to 12% loss for a 3.3V output.
Efficiency varies as the inverse square of VOUT for the
same external components and output power level. The
combined effects of increasingly lower output voltages
and higher currents required by high performance digital
systems is not doubling but quadrupling the importance
of loss terms in the switching regulator system!
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high
input voltages (typically 15V or greater). Transition
losses can be estimated from:
Transition Loss = (1.7) VIN2 IO(MAX) CRSS f
Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is
very important to include these “system” level losses
during the design phase. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at the
switching frequency. A 25W supply will typically require a
minimum of 20µF to 40µF of capacitance having a maximum of 20mΩ to 50mΩ of ESR. The LTC3865 2-phase
architecture typically halves this input capacitance requirement over competing solutions. Other losses including
Schottky conduction losses during dead time and inductor core losses generally account for less than 2% total
additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to ∆ILOAD (ESR), where ESR is the effective
series resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. The
availability of the ITH pin not only allows optimization of
control loop behavior but also provides a DC coupled and
AC filtered closed loop response test point. The DC step,
rise time and settling at this test point truly reflects the
closed loop response. Assuming a predominantly second
order system, phase margin and/or damping factor can be
estimated using the percentage of overshoot seen at this
pin. The bandwidth can also be estimated by examining the
rise time at the pin. The ITH external components shown
in the Typical Application circuit will provide an adequate
starting point for most applications.
3865f
26
LTC3865/LTC3865-1
Applications Information
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1µs to 10µs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking the feedback loop. Placing a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This
is why it is better to look at the ITH pin signal which is in
the feedback loop and is the filtered and compensated
control loop response. The gain of the loop will be increased by increasing RC and the bandwidth of the loop
will be increased by decreasing CC. If RC is increased by
the same factor that CC is decreased, the zero frequency
will be kept the same, thereby keeping the phase shift the
same in the most critical frequency range of the feedback
loop. The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance.
A second, more severe transient is caused by switching
in loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT , causing a rapid drop in VOUT . No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • CLOAD . Thus a 10µF capacitor would
require a 250µs rise time, limiting the charging current
to about 200mA.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the IC. These items are also illustrated graphically in the
layout diagram of Figure 12. Figure 13 illustrates the
current waveforms present in the various branches of
the 2-phase synchronous regulators operating in the
continuous mode. Check the following in your layout:
1. Are the top N-channel MOSFETs M1 and M3 located
within 1 cm of each other with a common drain connection at CIN? Do not attempt to split the input decoupling for the two channels as it can cause a large
resonant loop.
2. Are the signal and power grounds kept separate? The
combined IC signal ground pin and the ground return
of CINTVCC must return to the combined COUT (–) terminals. The VOSENSE and ITH traces should be as short
as possible. The path formed by the top N-channel
MOSFET, Schottky diode and the CIN capacitor should
have short leads and PC trace lengths. The output
capacitor (–) terminals should be connected as close
as possible to the (–) terminals of the input capacitor
by placing the capacitors next to each other and away
from the Schottky loop described above.
3. Do the LTC3865 VOSENSE pins connect to the (+) terminals
of COUT? The connections between the VOSENSE pins
and COUT should not be along the high current input
feeds from the input capacitor(s).
4. Are the SENSE+ and SENSE– leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE+ and SENSE– should be as close as possible
to the IC. Ensure accurate current sensing with Kelvin
connections at the sense resistor or inductor, whichever
is used for current sensing.
5. Is the INTVCC decoupling capacitor connected close to
the IC, between the INTVCC and the power ground pins?
This capacitor carries the MOSFET drivers current peaks.
An additional 1µF ceramic capacitor placed immediately
next to the INTVCC and PGND pins can help improve
noise performance substantially.
3865f
27
LTC3865/LTC3865-1
Applications Information
VID11
ITH1
LTC3865
RPU2
PGOOD
PGOOD
VPULL-UP
VID12
VID21
L1
VID22
SENSE1+
–
CB1
M1
BOOST1
FREQ
ILIM
BG1
MODE/PLLIN
RIN
VIN
INTVCC
SENSE2+
BG2
VOSENSE2
BOOST2
ITH2
TK/SS2
CIN
CINTVCC
CERAMIC
+
SENSE2–
COUT1
M3
GND
M4
+
EXTVCC
D1
CERAMIC
CVIN
PGND
SGND
M2
+
VIN
RUN1
RUN2
VOUT1
SW1
SENSE1
fIN
RSENSE
TG1
+
VOSENSE1
TK/SS1
COUT2
D2
CB2
SW2
RSENSE
TG2
VOUT2
L2
3865 F12
Figure 12. Recommended Printed Circuit Layout Diagram
SW1
L1
D1
RSENSE1
VOUT1
COUT1
RL1
VIN
RIN
CIN
SW2
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
D2
L2
RSENSE2
VOUT2
COUT2
RL2
3865 F13
Figure 13. Branch Current Waveforms
28
3865f
LTC3865/LTC3865-1
Applications Information
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), and boost nodes (BOOST1, BOOST2) away
from sensitive small-signal nodes, especially from the
opposite channel’s voltage and current sensing feedback pins. All of these nodes have very large and fast
moving signals and therefore should be kept on the
“output side” of the LTC3865/LTC3865-1 and occupy
minimum PC trace area. If DCR sensing is used, place
the top resistor (Figure 2b, R1) close to the switching
node.
7. Use a modified “star ground” technique: a low impedance, large copper area central grounding point on
the same side of the PC board as the input and output
capacitors with tie-ins for the bottom of the INTVCC
decoupling capacitor, the bottom of the voltage feedback
resistive divider and the SGND pin of the IC.
PC Board Layout Debugging
Start with one controller at a time. It is helpful to use a
DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output
switching node (SW pin) to synchronize the oscilloscope
to the internal oscillator and probe the actual output
voltage as well. Check for proper performance over the
operating voltage and current range expected in the
application. The frequency of operation should be maintained over the input voltage range down to dropout and
until the output load drops below the low current operation threshold—typically 10% of the maximum designed
current level in Burst Mode operation.
The duty cycle percentage should be maintained from
cycle to cycle in a well-designed, low noise PCB implementation. Variation in the duty cycle at a subharmonic rate
can suggest noise pickup at the current or voltage sensing
inputs or inadequate loop compensation. Overcompensation of the loop can be used to tame a poor PC layout if
regulator bandwidth optimization is not required. Only after
each controller is checked for its individual performance
should both controllers be turned on at the same time.
A particularly difficult region of operation is when one
controller channel is nearing its current comparator trip
point when the other channel is turning on its top MOSFET.
This occurs around 50% duty cycle on either channel due
to the phasing of the internal clocks and may cause minor
duty cycle jitter.
Reduce VIN from its nominal level to verify operation
of the regulator in dropout. Check the operation of the
undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, Schottky and the top
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
SGND pin of the IC.
3865f
29
LTC3865/LTC3865-1
Applications Information
Design Example
As a design example for a 2-channel medium current regulator, assume VIN = 12V (nominal), VIN = 20V
(maximum), VOUT1 = 3.3V, VOUT2 = 1.8V, IMAX1,2 = 5A, and
f = 500kHz (see Figure 14).
The regulated output voltages are set by connecting VID11
and VID22 to INTVCC and floating VID12 and VID21.
The frequency is set by biasing the FREQ pin to 1.2V (see
Figure 9).
The inductance values are based on a 35% maximum
ripple current assumption (1.75A for each channel). The
highest value of ripple current occurs at the maximum
input voltage:

VOUT
VOUT 
L=
 1−

f • ∆IL (MAX ) 
VIN(MAX ) 
4.7µF
M1
0.1µF
D3
LTC3865
MODE/PLLIN
ILIM
0.1µF
RUN1
TK/SS1
100pF
VIN(MAX ) f
=
1.8 V
= 180ns
20 V(500kHz)
+
0.1µF
D4
VIN
7V TO 20V
22µF
50V
M2
0.1µF
L2
2.2µH
BOOST2
SW2
BG2
3.65k
1%
PGND
0.1µF
RUN2
SENSE2–
SENSE1
VID11
VID12
VOSENSE1
ITH1
1800pF
VOUT
SENSE2+
–
4.75k
1%
The minimum on-time occurs on Channel 1 at the maximum
VIN, and should not be less than 90ns:
FREQ
SENSE1+
VOUT1
3.3V
5A
COUT1
100µF
s2
Channel 1 will have 1.45A (29%) ripple, and Channel 2 will
have 1.4A (28%) ripple. The peak inductor current will be
the maximum DC value plus one-half the ripple current,
or 5.725A for Channel 1 and 5.7A for Channel 2.
VIN PGOOD EXTVCC INTVCC
TG1
TG2
BG1
5.49k
1%
VOUT 
VOUT 
 1−

f •L 
VIN(NOM) 
1µF
BOOST1
SW1
1.37k
1%
∆IL(NOM) =
tON(MIN) =
2.2Ω
L1
3.3µH
Channel 1 will require 3.2µH, and Channel 2 will require
1.9µH. The next highest standard values are 3.3µH
and 2.2µH. At the nominal input voltage (12V), the ripple
will be:
1.58k
1%
VID21
VID22
VOSENSE2
ITH2
SGND
TK/SS2
0.1µF
VOUT2
1.8V
5A
2200pF
162k
1%
5.49k
1%
L1, L2: COILTRONICS HCP0703
M1, M2: VISHAY SILICONIX Si4816BDY
COUT1, COUT2: TAIYO YUDEN JMK325BJ107MM
100pF
COUT2
100µF
s2
3865 F14
Figure 14. High Efficiency Dual 500kHz 3.3V/1.8V Step-Down Converter
3865f
30
LTC3865/LTC3865-1
Applications Information
With ILIM floating, the equivalent RSENSE resistor value
can be calculated by using the minimum value for the
maximum current sense threshold (44mV).
VSENSE(MIN)
RSENSE(EQUIV ) =
∆IL(NOM)
ILOAD(MAX ) +
2
44mV
=
@ 7.7mΩ
1.5A
5A +
2
The equivalent RSENSE is the same for Channel 2.
The Coiltronics (Cooper) HCP0703-2R2 (20mΩ DCRMAX
at 20°C) and HCP0703-3R3 (30mΩ DCRMAX at 20°C) are
chosen. At 100°C, the estimated maximum DCR values are
26.4mΩ and 39.6mΩ. The divider ratios are:
RD =
and
RSENSE(EQUIV )
DCRMAX at TL(MAX )
=
7.7mΩ
= 0.3;
26.4mΩ
7.7mΩ
@ 0.2
39.6 mΩ
For each channel, 0.1µF is selected for C1.
L
2.2µH
=
(DCRMAX at 20°C) • C1 20 mΩ • 0.1µF
3.3 µH
= 1.1k
= 1.1k ; and
30 mΩ • 0.1µF
R1|| R2 =
The power loss in R1 at the maximum input voltage is:
PLOSS R1=
R1
(20 V − 3.3V) • 3.3V
=
= 10mW
5.5k
The respective values for Channel 2 are R1 = 3.66k,
R2 = 1.57k; and PLOSS R1 = 8mW.
Burst Mode operation is chosen for high light load efficiency
(Figure 15) by floating the MODE/PLLIN pin. Power loss
due to the DCR sensing network is slightly higher at light
loads than would have been the case with a suitable sense
resistor (8mΩ). At heavier loads, DCR sensing provides
higher efficiency.
The power dissipation on the topside MOSFET can be easily
estimated. Choosing a Siliconix Si4816BDY dual MOSFET
results in: RDS(ON) = 0.023Ω/0.016Ω, CMILLER @ 100pF.
At maximum input voltage with T(estimated) = 50°C:
3.3V 2
(5) [1+ (0.005)(50°C – 25°C)] •
20 V
(0.023Ω) + (20V )2  52A  (2Ω)(100pF ) •
PMAIN =
1 
 1
 5 – 2.3 + 2.3  ( 500kHz ) = 186 mW


100
EFFICIENCY (%)
EFFICIENCY
1
80
70
60
POWER LOSS
50
40
0.01
0.1
DCR
8mΩ
0.1
1
LOAD CURRENT (mA)
10
POWER LOSS (mW)
R1|| R2 1.1k
=
@ 5.5 k ;
RD
0.2
R1 • RD 5.5 k • 0.2
=
R2 =
@ 1.37k
1 − RD
1 − 0.2
10
90
For channel 1, the DCRSENSE filter/divider values are:
R1 =
( VIN(MAX ) − VOUT ) • VOUT
0.01
3865 F16
Figure 15. Design Example Efficiency vs Load
3865f
31
LTC3865/LTC3865-1
Applications Information
A short-circuit to ground will result in a folded back current of:
ISC =
CIN is chosen for an RMS current rating of at least 2A at
temperature assuming only channel 1 or 2 is on. COUT is
chosen with an ESR of 0.02Ω for low output ripple. The
output ripple in continuous mode will be highest at the
maximum input voltage. The output voltage ripple due to
ESR is approximately:
(1/ 3) 50mV – 1  90ns(20V)  = 1.8 A
2 
0.008Ω
3.3 µH 
with a typical value of RDS(ON) and d = (0.005/°C)(20)
= 0.1. The resulting power dissipated in the bottom
MOSFET is:
VORIPPLE = RESR (∆IL) = 0.02Ω(1.5A) = 30mVP-P
20 V – 3.3V
2
1.8 A ) (1.125) ( 0.016Ω )
(
20 V
= 48mW
PSYNC =
which is less than under full-load conditions.
TYPICAL APPLICATIONS
10µF
35V
+
10µF
35V
2.2Ω
22µF
50V
VIN
4.5V TO 24V
1µF
4.7µF
L2
0.47µH
D1
Q1
RJK0305DPB
D3
0.1µF
Q2
RJK0330DPB
BG1
COUT1
100µF
100Ω
1000pF
RUN1
+
1000pF
10k
LTC3865
MODE/PLLIN
ILIM
100Ω
COUT2,3
TG2
BOOST1
SW1
2mΩ
VOUT1
1.5V
15A
VIN PGOOD EXTVCC INTVCC
TG1
100pF
SENSE1+
D4
BOOST2
SW2
Q4
RJK0330DPB
BG2
PGND
FREQ
SENSE2+
100Ω
RUN2
1000pF
SENSE1–
SENSE2–
VID11
VID21
VID12
VID22
VOSENSE2
VOSENSE1
ITH2
ITH1
TK/SS1
SGND TK/SS2
0.1µF
Q3
RJK0305DPB
0.1µF
0.1µF
L2
0.47µH
D2
2mΩ
100Ω
1000pF
162k
1%
15k
D1, D2: VISHAY B340A
L1, L2: VISHAY IHLP4040DZERR47M11
COUT1, COUT4: TDK C322JX5R0J107MT
COUT2, COUT3, COUT5, COUT6: SANYO 4TPE 220µF
COUT5,6
100pF
+
VOUT2
1.2V
15A
COUT4
100µF
3865 F16
Figure 16. 1.5V/15A, 1.2V/15A Converter Using Sense Resistors
3865f
32
LTC3865/LTC3865-1
TYPICAL APPLICATIONS
3.3µF
50V
CIN2
1µF
2.2Ω
4.7µF
D3
CMDSH-3
HAT2266H
0.004Ω
VIN PGOOD EXTVCC INTVCC
TG1
TG2
0.1µF
L1
2.2µH
BOOST1
SW1
HAT2266H
PLLIN
400kHz
100Ω
1nF
COUT1
220µF
s2
100pF
BOOST2
SW2
L2
1.2µH
BG2
PGND
100Ω
1nF
100Ω
–
SENSE2
VID21
VID22
VOSENSE2
ITH2
SGND
0.1µF
0.004Ω
HAT2266H
RUN2
TK/SS1
17.8k
1%
HAT2266H
0.1µF
FREQ
SENSE1
VID11
VID12
VOSENSE1
ITH1
VIN
7V TO 36V
CIN1
22µF
50V
D4
CMDSH-3
SENSE2+
–
1800pF
10k
1%
ILIM
RUN1
50k
1%
+
MODE/PLLIN
SENSE1+
100Ω
VOUT1
3.6V
10A
BG1
LTC3865
+
3.3µF
50V
23.2k
1%
2200pF
TK/SS2
0.1µF
121k
14.7k
1%
10k
1%
100pF
COUT1, COUT2: SANYO 4TPE 220µF
L1: TOKO FDA1055-2R2M
L2: TOKO FD1055-1R2M
VOUT2
2V
10A
+
COUT2
220µF
s2
3865 F17
Figure 17. High VIN, 3.6V/10A, 2V/10A Converter With External Sense Resistors Synchronized at 400kHz
10µF
35V
4.7µF
RJK0305DPB
D3
VIN PGOOD EXTVCC INTVCC
TG2
TG1
0.1µF
L1
0.47µH
1µF
2.2Ω
BOOST1
SW1
RJK0330DPB
BG1
LTC3865
MODE/PLLIN
ILIM
100Ω
2mΩ
+
100Ω
1.21k
1%
RUN1
L2
0.47µH
RJK0330DPB
PGND
100Ω
0.1µF
RUN2
SENSE2–
SENSE1
VID11
VID12
VOSENSE1
ITH1
TK/SS1
BG2
2mΩ
100Ω
VID21
VID22
1.2V
30A
VOSENSE2
ITH2
SGND
+
TK/SS2
162k
100pF
22µF
50V
RJK0305DPB
0.1µF
SENSE2+
–
6800pF
COUT1
220µF
BOOST2
SW2
VIN
7V TO 20V
D4
FREQ
SENSE1+
0.1µF
+
10µF
35V
COUT2
220µF
0.1µF
COUT1, COUT2: SANYO 4TPE 220µF
L1, L2: VISHAY IHLP4040DZERR47M11
RUN1
0Ω
3865 F18
RUN2
Figure 18. High Current, Single Output, 1.2V/30A Converter Using Sense Resistors
3865f
33
LTC3865/LTC3865-1
Package Description
UH Package
32-Lead Plastic QFN (5mm × 5mm)
(Reference LTC DWG # 05-08-1693 Rev D)
0.70 p0.05
5.50 p0.05
4.10 p0.05
3.50 REF
(4 SIDES)
3.45 p 0.05
3.45 p 0.05
PACKAGE OUTLINE
0.25 p 0.05
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
5.00 p 0.10
(4 SIDES)
BOTTOM VIEW—EXPOSED PAD
0.75 p 0.05
R = 0.05
TYP
0.00 – 0.05
R = 0.115
TYP
PIN 1 NOTCH R = 0.30 TYP
OR 0.35 s 45o CHAMFER
31 32
0.40 p 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
3.50 REF
(4-SIDES)
3.45 p 0.10
3.45 p 0.10
(UH32) QFN 0406 REV D
0.200 REF
NOTE:
1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE
M0-220 VARIATION WHHD-(X) (TO BE APPROVED)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
0.25 p 0.05
0.50 BSC
3865f
34
LTC3865/LTC3865-1
Package Description
FE Package
38-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1772 Rev A)
Exposed Pad Variation AA
4.75 REF
38
9.60 – 9.80*
(.378 – .386)
4.75 REF
(.187)
20
6.60 ±0.10
4.50 REF
2.74 REF
SEE NOTE 4
6.40
2.74
REF (.252)
(.108)
BSC
0.315 ±0.05
1.05 ±0.10
0.50 BSC
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.50 – 0.75
(.020 – .030)
0.09 – 0.20
(.0035 – .0079)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
2. DIMENSIONS ARE IN MILLIMETERS
(INCHES)
3. DRAWING NOT TO SCALE
1
0.25
REF
19
1.20
(.047)
MAX
0o – 8o
0.50
(.0196)
BSC
0.17 – 0.27
(.0067 – .0106)
TYP
0.05 – 0.15
(.002 – .006)
FE38 (AA) TSSOP 0608 REV A
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
3865f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
35
LTC3865/LTC3865-1
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
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LTC3611
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3865f
36 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
LT 0210 • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2010
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