NSC LP2975AIMMX-5.0 Mosfet ldo driver/controller Datasheet

LP2975
MOSFET LDO Driver/Controller
General Description
Features
A high-current LDO regulator is simple to design with the
LP2975 LDO Controller. Using an external P-FET, the
LP2975 will deliver an ultra low dropout regulator with extremely low quiescent current.
High open loop gain assures excellent regulation and ripple
rejection performance.
The trimmed internal bandgap reference provides precise
output voltage over the entire operating temperature range.
Dropout voltage is “user selectable” by sizing the external
FET: the minimum input-output voltage required for operation is the maximum load current multiplied by the RDS(ON)
of the FET.
Overcurrent protection of the external FET is easily implemented by placing a sense resistor in series with VIN. The
57 mV detection threshold of the current sense circuitry minimizes dropout voltage and power dissipation in the resistor.
The standard product versions available provide output voltages of 12V, 5V, or 3.3V with guaranteed 25˚C accuracy of
1.5% (“A” grade) and 2.5% (standard grade).
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Block Diagram
Connection Diagram
Simple to use, few external components
Ultra-small mini SO-8 package
1.5% (A grade) precision output voltage
Low-power shutdown input
< 1 µA in shutdown
Low operating current (180 µA typical @ VIN = 5V)
Wide supply voltage range (1.8V to 24V)
Built-in current limit amplifier
Overtemperature protection
12V, 5V, and 3.3V standard output voltages
Can be programmed using external divider
−40˚C to +125˚C junction temperature range
Applications
n High-current 5V to 3.3V regulator
n Post regulator for switching converter
n Current-limited switch
Surface Mount Mini SO-8 Package
DS100034-2
Top View
For Order Numbers
See Table 1 of this Document
See NS Package Number MUA08A
DS100034-1
*RSET values are: 208k for 12V part, 72.8k for 5V part, and 39.9k for 3.3V
part.
© 1999 National Semiconductor Corporation
DS100034
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LP2975 MOSFET LDO Driver/Controller
September 1997
Ordering Information
TABLE 1. Package Marking and Ordering Information
Output Voltage
Grade
Order Information
Package Marking
Supplied As:
12
A
LP2975AIMMX-12
L47A
3.5k Units on Tape and Reel
12
A
LP2975AIMM-12
L47A
250 Units on Tape and Reel
12
STD
LP2975IMMX-12
L47B
3.5k Units on Tape and Reel
12
STD
LP2975IMM-12
L47B
250 Units on Tape and Reel
5.0
A
LP2975AIMMX-5.0
L46A
3.5k Units on Tape and Reel
5.0
A
LP2975AIMM-5.0
L46A
250 Units on Tape and Reel
5.0
STD
LP2975IMMX-5.0
L46B
3.5k Units on Tape and Reel
5.0
STD
LP2975IMM-5.0
L46B
250 Units on Tape and Reel
3.3
A
LP2975AIMMX-3.3
L45A
3.5k Units on Tape and Reel
3.3
A
LP2975AIMM-3.3
L45A
250 Units on Tape and Reel
3.3
STD
LP2975IMMX-3.3
L45B
3.5k Units on Tape and Reel
3.3
STD
LP2975IMM-3.3
L45B
250 Units on Tape and Reel
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2
Absolute Maximum Ratings (Note 1)
Input Supply Voltage (Survival)
−0.3V to +26V
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Input Supply Voltage (Operating)
+1.8V to +24V
Storage Temperature Range
−65˚C to +150˚C
Operating Junction Temperature
Range
−40˚C to +125˚C
Lead Temp. (Soldering, 5 seconds)
Current Limit Pins (Survival)
−0.3V to +VIN
Comp Pin (Survival)
−0.3V to +2V
Gate Pin (Survival)
−0.3V to +VIN
ON/OFF Pin (Survival)
−0.3V to +20V
Feedback Pin (Survival)
−0.3V to +24V
260˚C
ESD Rating
2 kV
Power Dissipation
(Note 2)
Internally Limited
Electrical Characteristics
Limits in standard typeface are for TJ = 25˚C, and limits in boldface type apply over the full operating temperature range. Unless otherwise specified; VON/OFF = 1.5V, VIN = 15V.
Symbol
VREG
Parameter
Regulation Voltage
(12V Versions)
Regulation Voltage
(5V Versions)
Regulation Voltage
(3.3V Versions)
VCOMP
Comp Pin Voltage
Quiescent Current
IQ
Conditions
Typ
12.5 < VIN < 24V
12.0
(VIN - 0.5V) > VGATE
> (VIN - 5V)
5.5 < VIN < 24V
5.0
(VIN - 0.5V) > VGATE
> (VIN - 4.5V)
3.8 < VIN < 24V
3.3
(VIN - 0.5V) > VGATE
> (VIN - 3.3V)
VREG < VIN < 24V
VIN = 5V
VON/OFF
Current Limit
Sense Voltage
VIN = 15V
VFB = 0.9 X VREG
ON/OFF Threshold
Output = ON
0.94
Output = OFF
ION/OFF
IG
Min
Max
11.820
12.180
11.700
12.300
11.640
12.360
11.520
12.480
4.925
5.075
4.875
5.125
4.850
5.150
4.800
5.200
3.250
3.350
3.217
3.383
3.201
3.399
3.168
3.432
1.215
1.265
1.203
1.277
1.209
1.271
1.196
1.284
VON/OFF = 1.5V
Gate Drive Current (Sourcing)
VG = 7.5V
VFB= 1.1 X VREG
3.5
VG = 7.5V
VFB = 0.9 X VREG
1100
240
240
320
320
1
1
45
69
45
69
39
72
39
72
1.10
34
1.20
0.70
0.70
0.40
0.40
50
50
75
75
1.3
1.3
0.3
0.3
350
350
40
40
15
Units
V
V
µA
mV
1.10
1.20
0.87
ON/OFF
Input Bias Current
Gate Drive Current (Sinking)
Max
0.01
57
LM2975I-X.X
(Note 3)
Min
180
VON/OFF = 0V
VCL
1.240
LM2975AI-X.X
(Note 3)
19
15
V
µA
mA
µA
VG(MIN)
Gate Clamp Voltage
VIN = 24V
VFB = 0.9 X VREG
17
19
V
R(VIN-G)
Resistance from
Gate to VIN
VIN = 24V
VON/OFF = 0
500
kΩ
Open Loop
Voltage Gain
VIN = 15V
0.5V ≤ VGATE ≤ 13
5000
V/V
Note 1: Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the device outside of its rated operating conditions.
Note 2: The LP2975 has internal thermal shutdown which activates at a die temperature of about 150˚C. It should be noted that the power dissipated within the
LP2975 is low enough that this protection circuit should never activate due to self-heating, even at elevated ambient temperatures.
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Electrical Characteristics
(Continued)
Note 3: Limits are 100% production tested at 25˚C. Limits over the operating temperature range are guaranteed through correlation using Statistical Quality Control
(SQC) methods. The limits are used to calculate National’s Average Outgoing Quality Level (AOQL).
Typical Application Circuits
5V - 3.3V @ 5A LDO Regulator
DS100034-3
*See Application Hints.
**If current limiting is not required, short out this resistor.
Adjustable Voltage 5A LDO Regulator
DS100034-4
*See Application Hints.
***If current limiting is not required, short out this resistor.
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Typical Performance Characteristics
Unless otherwise specified: TA = 25˚C, CIN = 1 µF, ON/OFF pin
is tied to 1.5V.
VIN Referred Gate Clamp Voltage
Minimum Operating Voltage
DS100034-5
ON/OFF Threshold
DS100034-6
Current Limit Sense Voltage
DS100034-7
ON/OFF Pin Current
DS100034-8
Supply Current
DS100034-10
DS100034-9
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Typical Performance Characteristics
Unless otherwise specified: TA = 25˚C, CIN = 1 µF, ON/OFF pin
is tied to 1.5V. (Continued)
ON/OFF Input Resistance
Gate Current
DS100034-34
DS100034-11
Gate-Ground Saturation
Line Regulation
DS100034-14
DS100034-13
Load Regulation
Leakage Current
DS100034-35
DS100034-15
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Typical Performance Characteristics
Unless otherwise specified: TA = 25˚C, CIN = 1 µF, ON/OFF pin
is tied to 1.5V. (Continued)
Controller Gain and Phase Response
DS100034-36
maximum output voltage. The measured change in output
voltage is divided by the nominal output voltage and expressed as a percentage. VIN = 5.6V for this test.
Reference Designs
The LP2975 controller can be used with virtually any
P-channel MOSFET to build a wide variety of linear voltage
regulators.
Since it would be impossible to document all the different
voltage and current combinations that could be built, a number of reference designs will be presented along with performance data for each.
THE PERFORMANCE DATA SHOWN IS ACTUAL TEST
DATA, BUT IS NOT GUARANTEED.
5 mA ≤ IL ≤ 5A: LOAD REGULATION = 0.012%
0 ≤ IL ≤ 5A: LOAD REGULATION = 0.135%
Line Regulation
Line regulation is defined as the maximum change in output
voltage as the input voltage is varied. It is measured by
changing the input voltage and recording the minimum/
maximum output voltage. The measured change in output
voltage is divided by the nominal output voltage and expressed as a percentage. IL = 5A for this test.
5.4V ≤ VIN ≤ 10V: LINE REGULATION = 0.03%
DESIGN #1: VOUT = 5V @ 5A (Refer to Typical Application
Circuits)
COMPONENTS:
CIN = 82 µF Aluminum Electrolytic
COUT = 120 µF Aluminum Electrolytic
CF = 220 pF
RSC = 10 mΩ
P-FET = NDP6020P
Heatsink: (assuming VIN ≤ 7V and TA ≤ 60˚C) if protection
against a continuous short-circuit is required, a heatsink with
θS-A ≤ 1.5 ˚C/W must be used. However, if continuous shortcircuit survivability is not needed, a heatsink with θS-A ≤
6 ˚C/W is adequate.
Output Noise Voltage
Output noise voltage was measured by connecting a wideband AC voltmeter (HP 400E) directly across the output capacitor. VIN = 6V and IL = 5A for this test.
NOISE = 75 µV (rms)
Transient Response
Transient response is defined as the change in output voltage which occurs after the load current is suddenly changed.
VIN = 5.6V for this test.
The load resistor is connected to the regulator output using a
switch so that the load current increases from 0 to 5A
abruptly. The change in output voltage is shown in the scope
photo below (the vertical scale is 200 mV/division and the
horizontal scale is 10 µs/division). The regulator nominal output (5V) is located on the center line of the photo.
The output shows a maximum change of about −600 mV
compared to nominal. This is due to the relatively small output capacitor chosen for this design. Increasing COUT greatly
improves transient response (see Designs #2 and #3).
PERFORMANCE DATA:
Dropout Voltage
Dropout voltage is defined as the minumum input-to-output
differential voltage required by the regulator to keep the output in regulation. It is measured by reducing VIN until the output voltage drops below the nominal value (the nominal
value is the output voltage measured with VIN = 5.5V). IL =
5A for this test.
DROPOUT VOLTAGE = 323 mV
Load Regulation
Load regulation is defined as the maximum change in output
voltage as the load current is varied. It is measured by
changing the load resistance and recording the minimum/
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Reference Designs
maximum output voltage. The measured change in output
voltage is divided by the nominal output voltage and expressed as a percentage. VIN = 3.5V for this test.
(Continued)
0 ≤ IL ≤ 0.5A: LOAD REGULATION = 0.034%
Line Regulation
Line regulation is defined as the maximum change in output
voltage as the input voltage is varied. It is measured by
changing the input voltage and recording the minimum/
maximum output voltage. The measured change in output
voltage is divided by the nominal output voltage and expressed as a percentage. IL = 0.5A for this test.
3.5V ≤ VIN ≤ 6V: LINE REGULATION = 0.017%
Output Noise Voltage
Output noise voltage was measured by connecting a wideband AC voltmeter (HP 400E) directly across the output capacitor. VIN = 5V and IL = 0.5A for this test.
NOISE = 85 µV (rms)
DS100034-37
Transient Response
Transient response is defined as the change in output voltage which occurs after the load current is suddenly changed.
VIN = 3.5V for this test.
The load resistor is connected to the regulator output using a
switch so that the load current increases from 0 to 0.5A
abruptly. The change in output voltage is shown in the scope
photo (the vertical scale is 20 mV/division and the horizontal
scale is 50 µs/division). The regulator nominal output (3V) is
located on the center line of the photo. A maximum change
of about −50 mV is shown.
Transient Response for 0–5A Load Step
DESIGN #2: VOUT = 3V @ 0.5A (Refer to Typical
Application Circuits, Adjustable Voltage Regulator)
COMPONENTS:
CIN = 68 µF Tantalum
COUT = 2 X 68 µF Tantalum
CC = 470 pF
R1 = 237 kΩ, 1%
R2 = NOT USED
RSC = 0.1Ω
Tie feedback pin to VOUT
P-FET = NDT452P
Heatsink: Tab of N-FET is soldered down to 0.6 in2 copper
area on PC board.
Output Voltage Adjustment: For this application, a 3.3V part
is “trimmed” down to 3V by using a single external 237 kΩ resistor at R1, which parallels the internal 39.9 kΩ resistor (reducing the effective resistance to 34.2 kΩ).
Because the tempco of the external resistor will not match
the tempco of the internal resistor (which is typically 3000
ppm), this method of adjusting VOUT by using a single resistor is only recommended in cases where the output voltage
is adjusted ≤ 10% away from the nominal value.
PERFORMANCE DATA:
DS100034-38
Dropout Voltage
Transient Response for 0–0.5A Load Step
Dropout voltage is defined as the minimum input-to-output
differential voltage required by the regulator to keep the output in regulation. It is measured by reducing VIN until the output voltage drops below the nominal value (the nominal
value is the output voltage measured with VIN = 5V). IL =
0.5A for this test.
Minimizing COUT
It is often desirable to decrease the value of COUT to save
cost and reduce size. The design guidelines suggest selecting COUT to set the first pole ≤ 200 Hz (see later section Output Capacitor), but this is not an absolute requirement in all
cases.
The effect of reducing COUT is to decrease phase margin. As
phase margin is decreased, the output ringing will increase
when a load step is applied to the output. Eventually, if COUT
is made small enough, the regulator will oscillate.
To demonstrate these effects, the value of COUT in reference
design #2 is halved by removing one of the two 68 µF output
capacitors and the transient response test is repeated (see
DROPOUT VOLTAGE = 141 mV
Load Regulation
Load regulation is defined as the maximum change in output
voltage as the load current is varied. It is measured by
changing the load resistance and recording the minimum/
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8
Reference Designs
DESIGN #3: VOUT = 1.5V @ 6A. (Refer to Typical
Application Circuits, Adjustable Voltage Regulator)
(Continued)
photo below). The total overshoot increases from −50 mV to
about −75 mV, and the second “ring” on the transient is noticeably larger.
COMPONENTS:
CIN = 1000 µF Aluminum Electrolytic
COUT = 4 X 330 µF OSCON Aluminum Electrolytic
CC = NOT USED
R1 = 261Ω, 1%
R2 = 1.21 kΩ, 1%
RSC = 6 mΩ
P-FET = NDP6020P
Heatsink: (Assuming VIN ≤ 3.3V and TA ≤ 60˚C) if protection
against a continuous short-circuit is required, a heatsink with
θS-A < 2.5 ˚C/W must be used. However, if continuous shortcircuit survivability is not needed, a heatsink with θS-A <
7 ˚C/W is adequate.
PERFORMANCE DATA:
Dropout Voltage
Dropout voltage is defined as the minimum input-to-output
differential voltage required by the regulator to keep the output in regulation. It is measured by reducing VIN until the output voltage drops below the nominal value (the nominal
value is the output voltage measured with VIN = 3.3V). IL =
6A for this test.
DROPOUT VOLTAGE = 0.68V
DS100034-39
Transient Response with Output Capacitor Halved
The design is next tested with only a 4.7 µF output capacitor
(see scope photo below). Observe that the vertical scale has
been increased to 100 mV/division to accommodate the
−250 mV undershoot. More important is the severe ringing
as the transient decays. Most designers would recognize
this immediately as the warning sign of a marginally stable
design.
Load Regulation
Load regulation is defined as the maximum change in output
voltage as the load current is varied. It is measured by
changing the load resistance and recording the minimum/
maximum output voltage. The measured change in output
voltage is divided by the nominal output voltage and expressed as a percentage. VIN = 3.3V for this test.
0 ≤ IL ≤ 6A: LOAD REGULATION = 0.092%
Line Regulation
Line regulation is defined as the maximum change in output
voltage as the input voltage is varied. It is measured by
changing the input voltage and recording the minimum/
maximum output voltage. The measured change in output
voltage is divided by the nominal output voltage and expressed as a percentage. IL = 6A for this test.
3.3V ≤ VIN ≤ 5V: LINE REGULATION = 0.033%
Output Noise Voltage
Output noise voltage was measured by connecting a wideband AC voltmeter (HP 400E) directly across the output capacitor. VIN = 3.3V and IL = 6A for this test.
NOISE = 60 µV (rms)
DS100034-40
Transient Response with Only 4.7 µF Output Cap
The reason this design is marginally stable is that the 4.7 µF
output capacitor (along with the 6Ω output load) sets the pole
fp at 5 kHz. Analysis shows that the unity-gain frequency of
the loop is increased to about 100 kHz, allowing the FET’s
gate capacitance pole fpg to cause significant phase shift before the loop gain goes below unity. Also, because of the low
output voltage, the feedforward capacitor provides less than
10˚ of positive phase shift. For good stability, the output
capcitor needs to be larger than 4.7 µF.
Transient Response
Transient response is defined as the change in output voltage which occurs after the load current is suddenly changed.
VIN = 3.3V for this test.
The load resistor is connected to the regulator output using a
switch so that the load current increases from 0 to 6A
abruptly. The change in output voltage is shown in the scope
photo (the vertical scale is 50 mV/division and the horizontal
scale is 20 µs/division. The regulator nominal output (1.5V)
is located on the center line of the photo. A maximum change
of about −80 mV is shown.
For detailed information on stability and phase margin, see
the Application Hints section.
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Reference Designs
means faster changes in Gate voltage (which corresponds to
faster transient response) will occur with a smaller amount of
Gate capacitance.
(Continued)
2) The Gate capacitance forms a pole in the loop gain which
can reduce phase margin. When possible, this pole should
be kept at a higher frequency than the cross-over frequency
of the regulator loop (see later section CROSS-OVER FREQUENCY AND PHASE MARGIN).
A high value of Gate capacitance may require that a feedforward capacitor be used to cancel some of the excess phase
shift (see later section FEED-FORWARD CAPACITOR) to
prevent loop instability.
POWER DISSIPATION: The maximum power dissipated in
the FET in any application can be calculated from:
PMAX = (VIN − VOUT) x IMAX
Where the term IMAX is the maximum output current. It
should be noted that if the regulator is to be designed to withstand short-circuit, a current sense resistor must be used to
limit IMAX to a safe value (refer to section SHORT-CIRCUIT
CURRENT LIMITING).
The power dissipated in the FET determines the best choice
for package type. A TO-220 package device is best suited for
applications where power dissipation is less than 15W.
Power levels above 15W would almost certainly require a
TO-3 type device.
In low power applications, surface-mount package devices
are size-efficient and cost-effective, but care must be taken
to not exceed their power dissipation limits.
DS100034-41
Transient Response for 0–6A Load Step
Application Hints
SELECTING THE FET
The best choice of FET for a specific application will depend
on a number of factors:
VOLTAGE RATING: The FET must have a Drain-to-Source
breakdown voltage (sometimes called BVDSS) which is
greater than the input voltage.
DRAIN CURRENT: On-state Drain current must be specified
to be greater than the worst-case (short circuit) load current
for the application.
TURN-ON THRESHOLD: The Gate-to-Source voltage
where the FET turns on (called the Gate Threshold Voltage)
is very important. Many FET’s are intended for use with
G-to-S voltages in the 5V to 10V range. These should only
be used in applications where the input voltage is high
enough to provide > 5V of drive to the Gate.
Newer FET’s are becoming available with lower turn-on
thresholds (Logic-Level FET’s) which turn on fully with a gate
voltage of only 3V to 4V. Low threshold FET’s should be
used in applications where the input voltage is ≤ 5V.
ON RESISTANCE: FET on resistance (often called RDSON)
is a critical parameter since it directly determines the minimum input-to-output voltage required for operation at a given
load current (also called dropout voltage).
RDSON is highly dependent on the amount of Gate-toSource voltage applied. For example, the RDSON of a FET
with VG-S = 5V will typically decrease by about 25% as the
VG-S is increased to 10V. RDSON is also temperature dependent, increasing at higher temperatures.
POWER DISSIPATION AND HEATSINKING
Since the LP2975 controller is suitable for use with almost
any external P-FET, it follows that designs can be built which
have very high power dissipation in the pass FET. Since the
controller can not protect the FET from overtemperature
damage, thermal design must be carefully done to assure a
reliable design.
THERMAL DESIGN METHOD: The temperature of the FET
and the power dissipated is defined by the equation:
TJ = (θJ-A x PD) + TA
Where:
TJ is the junction temperature of the FET.
TA is the ambient temperature.
PD is the power dissipated by the FET.
θJ-A is the junction-to-ambient thermal resistance.
To ensure a reliable design, the following guidelines are recommended:
1) Design for a maximum (worst-case) FET junction temperature which does not exceed 150˚C.
2) Heatsinking should be designed for worst-case (maximum) values of TA and PD.
3) In designs which must survive a short circuit on the output,
the maximum power dissipation must be calculated assuming that the output is shorted to ground:
PD(MAX) = VIN x ISC
The dropout voltage of any LDO design is directly related to
RDSON, as given by:
VDROPOUT = ILOAD x (RDSON + RSC)
Where RSC is the short-circuit current limit set resistor (see
Application Circuit).
Where ISC is the short-circuit output current.
4) If the design is not intended to be short-circuit proof, the
maximum power dissipation for intended operation will be:
PD(MAX) = (VIN − VOUT) x IMAX
Where IMAX is the maximum output current.
LOW POWER ( < 2W) APPLICATIONS: In most cases,
some type of small surface-mount device will be used for the
GATE CAPACITANCE: Selecting a FET with the lowest possible Gate capacitance improves LDO performance in two
ways:
1) The Gate pin of the LP2975 (which drives the Gate of the
FET) has a limited amount of current to source or sink. This
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10
Application Hints
of 0.3–0.5˚C/W. The best source of information for this is
heatsink catalogs (Wakefield, AAVID, Thermalloy) since they
also sell mounting hardware.
θS-A is the heatsink-to-ambient thermal resistance, which
defines how well a heatsink transfers heat into the air. Once
this is determined, a heatsink must be selected which has a
value which is less than or equal to the computed value. The
value of θS-A is usually listed in the manufacturer’s data
sheet for a heatsink, but the information is sometimes given
in a graph of temperature rise vs. dissipated power.
(Continued)
FET in low power designs. Because of the increased cell
density (and tiny packages) used by modern FET’s, the current carrying capability may easily exceed the power dissipation limits of the package. It is possible to parallel two or
more FET’s, which divides the power dissipation among all
of the packages.
It should be noted that the “heatsink” for a surface mount
package is the copper of the PC board and the package itself
(direct radiation).
DESIGN EXAMPLE: A design is to be done which takes
3.3V in and provides 2.5V out at a load current of 7A. The
power dissipation will be calculated for both normal operation and short circuit conditions.
For normal operation:
PD = (VIN − VOUT) x ILOAD = 5.6W
Surface-mount devices have the value of θJ-A specified for a
typical PC board mounting on their data sheet. In most cases
it is best to start with the known data for the application (PD,
TA, TJ) and calculate the required value of θJ-A needed. This
value will define the type of FET and, possibly, the heatsink
required for cooling.
If the output is shorted to ground:
PD(SC) = VIN x ISC = 3.3 x 7.7 = 25.4W
(Assuming that a sense resistor is selected to set the value
of ISC 10% above the nominal 7A).
θJ-A will be calculated assuming a maximum TA of 70˚C and
a maximum TJ of 150˚C:
θJ-A = (TJ − TA)/PD(MAX)
For normal operation:
θJ-A = (150 − 70) / 5.6 = 14.3˚C/W
For designs which must operate with the output shorted to
ground:
θJ-A = (150 − 70) / 25.4 = 3.2˚C/W
θJ-A = (TJ − TA)/PD(MAX)
DESIGN EXAMPLE: A design is to be done with VIN = 5V
and VOUT = 3.3V with a maximum load current of 300 mA.
Based on these conditions, power dissipation in the FET during normal operation would be:
PD = (VIN − VOUT) x ILOAD
Solving, we find that PD = 0.51W. Assuming that the maximum allowable value of TJ is 150˚C and the maximum TA is
70˚C, the value of θJ-A is found to be 157˚C/W.
However, if this design must survive a continuous short on
the output, the power dissipated in the FET is higher:
PD(SC) = VIN x ISC = 5 x 0.33 = 1.65W
(This assumes the current sense resistor is selected for an
ISC value that is 10% higher than the required 0.3A).
The value of θJ-A required to survive continuous short circuit
is calculated to be 49˚C/W.
Having solved for the value(s) of θJ-A, a FET can be selected. It should be noted that a FET must be used with a
θJ-A value less than or equal to the calculated value.
HIGH POWER (≥2W) APPLICATIONS: As power dissipation increases above 2W, a FET in a larger package must be
used to obtain lower values of θJ-A. The same formulae derived in the previous section are used to calculate PD and
θJ-A.
Having found θJ-A, it becomes necessary to calculate the
value of θS-A (the heatsink-to-ambient thermal resistance) so
that a heatsink can be selected:
θS-A = θJ-A − (θJ-C + θC-S)
Where:
θJ-C is the junction-to-case thermal resistance. This parameter is the measure of thermal resistance between the
semiconductor die inside the FET and the surface of the
case of the FET where it mounts to the heatsink (the value of
θJ-C can be found on the data sheet for the FET). A typical
FET in a TO-220 package will have a θJ-C value of approximately 2–4˚C/W, while a device in a TO-3 package will be
about 0.5–2˚C/W.
θC-S is the case-to-heatsink thermal resistance, which
measures how much thermal resistance exists between the
surface of the FET and the heatsink. θC-S is dependent on
the package type and mounting method. A TO-220 package
with mica insulator and thermal grease secured to a heatsink
will have a θC-S value in the range of 1–1.5˚C/W. A TO-3
package mounted in the same manner will have a θC-S value
The value of 14.3˚C/W can be easily met using a TO-220 device. Calculating the value of θS-A required (assuming a
value of θJ-C = 3˚C/W and θC-S = 1˚C/W):
θS-A = θJ-A − (θJ-C + θC-S)
θS-A = 14.3 − (3 + 1) = 10.3˚C/W
Any heatsink may be used with a thermal resistance ≤
10.3˚C/W @ 5.6W power dissipation (refer to manufacturer’s
data sheet curves). Examples of suitable heatsinks are Thermalloy #6100B and IERC #LATO127B5CB.
However, if the design must survive a sustained short on the
output, the calculated θJ-A value of 3.2˚C/W eliminates the
possibility of using a TO-220 package device.
Assuming a TO-3 device is selected with a θJ-C value of
1.5˚C/W and θC-S = 0.4˚C/W, we can calculate the required
value of θS-A:
θS-A = θJ-A − (θJ-C + θC-S)
θS-A = 3.2 − (1.5 + 0.4) = 1.3˚C/W
A θS-A value ≤1.3˚C/W would require a relatively large heatsink, or possibly some kind of forced airflow for cooling.
SHORT-CIRCUIT CURRENT LIMITING
Short-circuit current limiting is easiliy implemented using a
single external resistor (RSC). The value of RSC can be calculated from:
RSC = VCL / ISC
Where:
ISC is the desired short circuit current.
VCL is the current limit sense voltage.
The value of VCL is 57 mV (typical), with guaranteed limits
listed in the Electrical Characteristics section. When doing a
worst-case calculation for power dissipation in the FET, it is
important to consider both the tolerance of VCL and the tolerance (and temperature drift) of RSC.
11
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Application Hints
For best results in most designs, the frequency of fz should
fall between 5 kHz and 50 kHz. It must be noted that the values of COUT and ESR usually vary with temperature (severely in the case of aluminum electrolytics), and this must
be taken into consideration.
(Continued)
For maximum accuracy, the INPUT and CURRENT LIMIT
pins must be Kelvin connected to RSC, to avoid errors
caused by voltage drops along the traces carrying the current from the input supply to the Source pin of the FET.
For the design example (VOUT = 5V @ 1A), select a capacitor
which meets the fz requirements. Solving the equation for
ESR yields:
EXTERNAL CAPACITORS
The best capacitors for use in a specific design will depend
on voltage and load current (examples of tested circuits for
several different output voltages and currents are provided in
a previous section.)
Information in the next sections is provided to aid the designer in the selection of the external capacitors.
INPUT CAPACITOR: Although not always required, an input
capacitor is recommended. Good bypassing on the input assures that the regulator is working from a source with a low
impedance, which improves stability. A good input capacitor
can also improve transient response by providing a reservoir
of stored energy that the regulator can utilize in cases where
the load current demand suddenly increases. The value
used for CIN may be increased without limit. Refer to the Reference Designs section for examples of input capacitors.
OUTPUT CAPACITOR: The output capacitor is required for
loop stability (compensation) as well as transient response.
During sudden changes in load current demand, the output
capacitor must source or sink current during the time it takes
the control loop of the LP2975 to adjust the gate drive to the
pass FET. As a general rule, a larger output capacitor will improve both transient response and phase margin (stability).
The value of COUT may be increased without limit.
OUTPUT CAPACITOR AND COMPENSATION: Loop compensation for the LP2975 is derived from COUT and, in some
cases, the feed-forward capacitor CF (see next section).
COUT forms a pole (referred to as fp) in conjuction with the
load resistance which causes the loop gain to roll off (decrease) at an additional −20 dB/decade. The frequency of
the pole is:
fp = 0.16 / [ (RL + ESR) x COUT]
ESR = 0.16 / (fz x COUT)
Assuming fz = 5 kHz and 50 kHz, the limiting values of ESR
for the 180 µF capacitor are found to be:
18 mΩ ≤ ESR ≤ 0.18Ω
A good-quality, low-ESR capacitor type such as the Panasonic HFQ is a good choice. However, the 10V/180 µF capacitor (#ECA-1AFQ181) has an ESR of 0.3Ω which is not in
the desired range.
To assure a stable design, some of the options are:
1) Use a different type capacitor which has a lower ESR
such as an organic-electrolyte OSCON.
2) Use a higher voltage capacitor. Since ESR is inversely
proportional to the physical size of the capacitor, a higher
voltage capacitor with the same C value will typically have a
lower ESR (because of the larger case size). In this example, a Panasonic ECA-1EFQ181 (which is a 180 µF/25V
part) has an ESR of 0.17Ω and would meet the desired ESR
range.
3) Use a feed-forward capacitor (see next section).
FEED-FORWARD CAPACITOR: Although not required in
every application, the use of a feed-forward capacitor (CF)
can yield improvements in both phase margin and transient
response in most designs.
The added phase margin provided by CF can prevent oscillations in cases where the required value of COUT and ESR
can not be easily obtained (see previous section).
CF can also reduce the phase shift due to the pole resulting
from the Gate capacitance, stabilizing applications where
this pole occurs at a low frequency (before cross-over) which
would cause oscillations if left uncompensated (see later
section GATE CAPACITANCE POLE FREQUENCY).
Even in a stable design, adding CF will typically provide more
optimal loop response (faster settling time). For these reasons, the use of a feed-forward capacitor is always recommended.
Where:
RL is the load resistance.
COUT is the value of the output capacitor.
ESR is the equivalent series resistance of COUT.
As a general guideline, the frequency of fp should be ≤ 200
Hz. It should be noted that higher load currents correspond
to lower values of RL, which requires that COUT be increased
to keep fp at a given frequency.
DESIGN EXAMPLE: Select the minimum required output
capacitance for a design whose output specifications are 5V
@ 1A:
fp = 0.16 / [ (RL + ESR) x COUT]
Re-written:
COUT = 0.16 / [fp x (RL + ESR) ]
Values used for the calculation:
fp = 200 Hz, RL = 5Ω, ESR = 0.1Ω (assumed).
Solving for COUT, we get 157 µF (nearest standard size
would be 180 µF).
The ESR of the output capacitor is very important for stability, as it creates a zero (fz) which cancels much of the phase
shift resulting from one of the poles present in the loop. The
frequency of the zero is calculated from:
fz = 0.16 / (ESR x COUT)
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CF is connected across the top resistor in the divider used to
set the output voltage (see Typical Application Circuit). This
forms a zero in the loop response (defined as fzf), whose frequency is:
fzf = 6.6 x 10−6 / [CF x (VOUT / 1.24 − 1) ]
When solved for CF, the fzf equation is:
CF = 6.6 x 10−6 / [fzf x (VOUT / 1.24 − 1) ]
For most applications, fzf should be set between 5 kHz and
50 kHz.
ADJUSTING THE OUTPUT VOLTAGE
If an output voltage is required which is not available as a
standard voltage, the LP2975 can be used as an adjustable
regulator (see Typical Application circuit). The external resistors R1 and R2 (along with the internal 24 kΩ resistor) set
the output voltage.
It is important to note that R2 is connected in parallel with the
internal 24 kΩ resistor. If we define REQ as the total resistance between the COMP pin and ground, then its value
will be the parallel combination of R2 and 24 kΩ:
12
Application Hints
Where:
RL is the load resistance.
ESR is the equivalent series resistance of the output capacitor.
(Continued)
REQ = (R2 x 24k) / (R2 + 24k)
It follows that the output voltage will be:
VOUT = 1.24 [ (R1 / REQ) + 1]
Some important considerations for an adjustable design:
The tolerance of the internal 24 kΩ resistor is about ± 20%.
Also, its temperature coefficient is almost certainly different
than the TC of the external resistor that is used for R2.
The term RL / / ESR is defined as:
(RL x ESR) / (RL + ESR)
It can be seen from these equations that CEFF varies with RL.
To get the worst-case (maximum) value for CEFF, use the
maximum value of load current, which also means the minimum value of load resistance RL. It should be noted that in
most cases, the ESR is the dominant term which determines
the value of RL / / ESR.
For these reasons, it is recommended that R2 be set at a
value that is not greater than 1.2k. In this way, the value of
R2 will dominate REQ, and the tolerance and TC of the internal 24k resistor will have a negligible effect on output voltage
accuracy.
Gate Pin Output Impedance
To determine the value for R1:
R1 = REQ [ (VOUT / 1.24) − 1]
External Capacitors (Adjustable Application)
All information in the previous section EXTERNAL CAPACITORS applies to the adjustable application with the exception of how to select the value of the feed-forward capacitor.
The feed-forward capacitor CC in the adjustable application
(see Typical Application Circuit) performs exactly the same
function as described in the previous section FEEDFORWARD CAPACITOR. However, because R1 is userselected, a different formula must be used to determine the
value of CC:
CC = 1 / (2 π x R1 x fzf)
As stated previously, the optimal frequency at which to place
the zero fzf is usually between 5 kHz and 50 kHz.
DS100034-20
Gate Capacitance Pole Frequency (fpg)
The pole frequency resulting from the Gate capacitance
CEFF is defined as fpg and can be approximated from:
fpg ≅ 0.16 / (RO x CEFF)
Where:
RO is the output impedance of the LP2975 Gate pin which
drives the Gate of the FET. It is important to note that RO is
a function of input supply voltage (see graph GATE PIN
OUTPUT IMPEDANCE). As shown, the minimum value of
RO is about 550Ω @ VIN = 24V, increasing to about 1.55 kΩ
@ VIN = 3V.
Using the equation for fpg, a family of curves are provided
showing how fpg varies with CEFF for several values of RO
(see graph fpg vs. CEFF):
OPTIMIZING DESIGN STABILITY
Because the LP2975 can be used with a variety of different
applications, there is no single set of components that are
best suited to every design. This section provides information which will enable the designer to select components that
optimize stability (phase margin) for a specific application.
Gate Capacitance
An important consideration of a design is to identify the frequency of the pole which results from the capacitance of the
Gate of the FET (this pole will be referred to as fpg). As fpg
gets closer to the loop crossover frequency, the phase margin is reduced. Information will now be provided to allow the
total Gate capacitance to be calculated so that fpg can be approximated.
fpg vs. CEFF
The first step in calculating fp is to determine how much effective Gate capacitance (CEFF) is present. The formula for
calculating CEFF is:
CEFF = CGS + CGD [1 + Gm (RL / / ESR) ]
Where:
CGS is the Gate-to-Source capacitance, which is found
from the values (refer to FET data sheet for values of CISS
and CRSS):
CGS = CISS − CRSS
GGD is the Gate-to-Drain capacitance, which is equal to:
CGD = CRSS
Gm is the transconductance of the FET. The FET data
sheet specifies forward transconductance (Gfs) at some
value of drain current (defined as ID). To find Gm at the desired value of load current (defined as IL), use the formula:
Gm = Gfs x (IL / ID)1/2
DS100034-21
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Application Hints
next section FEED-FORWARD COMPENSATION). This can
improve phase margin by cancelling some of the excess
phase shift.
(Continued)
As can be seen in the graph, values of CEFF in the
500 pF–2500 pF range produce values for fpg between
40 kHz and 700 kHz. To determine what effect fpg will have
on stability, the bandwidth of the regulator loop must be calculated (see next section CROSSOVER FREQUENCY AND
PHASE MARGIN).
Feed-Forward Compensation
Phase shift in the loop gain of the regulator results from fp
(the pole from the output capacitor and load resistance), fpg
(the pole from the FET gate capacitance), as well as the IC’s
internal controller pole (see typical curve). If the total phase
shift becomes excessive, instability can result.
Crossover Frequency and Phase Margin
The term fc will be used to define the crossover frequency of
the regulator loop (which is the frequency where the gain
curve crosses the 0 dB axis). The importance of this frequency is that it is the point where the loop gain goes below
unity, which marks the usable bandwidth of the regulator
loop.
It is the phase margin (or lack of it) at fc that determines
whether the regulator is stable. Phase margin is defined as
the total phase shift subtracted from 180˚. In general, a
stable loop requires at least 20˚-30˚ of phase margin at fc.
fc can be approximated by the following equation (all terms
have been previously defined):
The total phase shift can be reduced using feed-forward
compensation, which places a zero in the loop to reduce the
effects of the poles.
The feed-forward capacitor CF can accomplish this, provided
it is selected to set the zero at the correct frequency. It is important to point out that the feed-forward capacitor produces
both a zero and a pole. The frequency where the zero occurs
will be defined as fzf, and the frequency of the pole will be
defined as fpf. The equations to calculate the frequencies
are:
fzf = 6.6 x 10-6/ [CF x (VOUT/1.24 − 1) ]
fpf = 6.6 x 10-6/ [CF x (1 − 1.24/VOUT)]
In general, the feed-forward capacitor gives the greatest improvement in phase margin (provides the maximum reduction in phase shift) when the zero occurs at a frequency
where the loop gain is > 1 (before the crossover frequency).
The pole must occur at a higher frequency (the higher the
better) where most of the phase shift added by the new pole
occurs beyond the crossover frequency. For this reason, the
pole-zero pair created by CF become more effective at improving loop stability as they get farther apart in frequency.
In reviewing the equations for fzf and fpf, it can be seen that
they get closer together in frequency as VOUT decreases.
For this reason, the use of CF gives greatest benefit at higher
output voltages, declining as VOUT approaches 1.24V
(where CF has no effect at all).
In selecting a value of feed-forward capacitor, the crossover
frequency fc must first be calculated. In general, the frequency of the zero (fzf) set by this capacitor should be in the
range:
0.2 fc ≤ fzf ≤ 1.0 fc
DS100034-23
This equation assumes that no CF is used and fpg/fc > 1.
If the frequency of the Gate capacitance pole fpg has been
calculated (previous section), the amount of added phase
shift may now be determined. As shown in the graph below
(see graph PHASE SHIFT DUE TO fpg), the amount of
added phase shift increases as fpg approaches fc.
The amount of phase shift due to fpg that can occur before
oscillation takes place depends on how much added phase
shift is present as a result of the COUT pole (see previous
section OUTPUT CAPACITOR).
Phase Shift Due to fpg
The equation to determine the value of the feed-forward capacitor in fixed-voltage applications is:
CF = 6.6 x 10-6/ [fzf x (VOUT/1.24 − 1) ]
In adjustable applications (using an external resistive divider) the capacitor is found using:
CC = 1/(2 π x R1 x fzf)
SUMMARY OF STABILITY INFORMATION
This section will present an explanation of theory and terminology used to analyze loop stability, along with specific information related to stabilizing LP2975 applications.
BODE PLOTS AND PHASE SHIFT
Loop gain information is most often presented in the form of
a Bode Plot, which plots Gain (in dB) versus Frequency (in
Hertz).
A Bode Plot also conveys phase shift information, which can
be derived from the locations of the poles and zeroes.
DS100034-22
Because of this, there is no exact number for fpg/fc that can
be given as a fixed limit for stable operation. However, as a
general guideline, it is recommended that fpg ≥ 3 fc.
If this is not found to be true after inital calculations, the ratio
of fpg/fc can be increased by either reducing CEFF (selecting
a different FET) or using a larger value of COUT.
POLE: A pole causes the slope of the gain curve to decrease by an additional −20 dB/decade, and it also causes
phase lag (defined as negative phase shift) to occur.
A single pole will cause a maximum −90˚ of phase lag (see
graph EFFECTS OF A SINGLE POLE). It should be noted
Along with these two methods, another technique for improving loop stability is the use of a feed-forward capacitor (see
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Application Hints
STABILITY ANALYSIS OF TYPICAL APPLICATIONS
(Continued)
The first application to be analyzed is a fixed-output voltage
regulator with no feed-forward capacitor (see graph STABLE
PLOT WITHOUT FEED-FORWARD).
that when the total phase shift at 0 dB reaches (or gets close
to) −180˚, oscillations will result. Therefore, it can be seen
that at least two poles in the gain curve are required to
cause instability.
ZERO: A zero has an effect that is exactly opposite to a pole.
A zero will add a maximum +90˚ of phase lead (defined as
positive phase shift). Also, a zero causes the slope of the
gain curve to increase by an additional +20 dB/decade (see
graph EFFECTS OF A SINGLE ZERO).
Stable Plot without Feed-Forward
Effects of a Single Pole
DS100034-27
In this example, the value of COUT is selected so that the
pole formed by COUT and RL (previously defined as fp) is set
at 200 Hz. The ESR of COUT is selected so that zero formed
by the ESR and COUT (defined as fz) is set at 5 kHz (these
selections follow the general guidelines stated previously in
this document). Note that the gate capacitance is assumed
to be moderate, with the pole formed by the CGATE (defined
as fpg) occurring at 100 kHz.
To estimate the total phase margin, the individual phase shift
contributions of each pole and zero will be calculated assuming fp = 200 Hz, fz = 5 kHz, fc = 10 kHz and fpg = 100 kHz:
Controller pole shift = −90˚
fp shift = −arctan (10k/200) = −89˚
fz shift = arctan (10k/5k) = +63˚
fpg shift = −arctan (10k/100k) = −6˚
Summing the four numbers, the estimate for the total phase
shift is −122˚, which corresponds to a phase margin of 58˚.
This application is stable, but could be improved by using a
feed-forward capacitor (see next section).
EFFECT OF FEED-FORWARD: The example previously
used will be continued with the addition of a feed-forward capacitor CF (see graph IMPROVED PHASE MARGIN WITH
FEED-FORWARD). The zero formed by CF (previously defined as fzf) is set at 10 kHz and the pole formed by CF (previously defined as fpf) is set at 40 kHz (the 4X ratio of fpf/fzf
corresponds to VOUT = 5V).
DS100034-25
TOTAL PHASE SHIFT: The actual test of whether or not a
regulator is stable is the amount of phase shift that is present
when the gain curve crosses the 0 dB axis (the frequency
where this occurs was previously defined as fc).
The phase shift at fc can be estimated by looking at all of the
poles and zeroes on the Bode plot and adding up the contributions of phase lag and lead from each one. As shown in
the graphs, most of the phase lag (or lead) contributed by a
pole (or zero) occurs within one decade of the frequency of
the pole (or zero).
In general, a phase margin (defined as the difference between the total phase shift and −180˚) of at least 20˚ to 30˚
is required for a stable loop.
Effects of a Single Zero
DS100034-26
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Application Hints
(Continued)
High ESR Unstable without Feed-Forward
Improved Phase Margin with Feed-Forward
DS100034-29
As shown, moving the location of fz lower in frequency extends the bandwidth, pushing the crossover frequency fc out
to about 200 kHz. In viewing the plot, it can be seen that fp
and fz essentially cancel out, leaving only the controller pole
and fpg. However, since fpg now occurs well before fc, it will
cause enough phase shift to leave very little phase margin.
This application would either oscillate continuously or be
marginally stable (meaning it would exhibit severe ringing on
transient steps).
This can be improved by adding a feed-forward capacitor CF,
which adds a zero (fzf) and a pole (fpf) to the gain plot (see
graph HIGH ESR CORRECTED WITH FEED-FORWARD).
In this case, CF is selected to place fzf at about the same frequency as fpg (essentially cancelling out the phase shift due
to fpg). Assuming the added pole fpf is near or beyond the fc
frequency, it will add < 45˚ of phase lag, leaving a phase
margin of > 45˚ (adequate for good stability).
DS100034-28
To estimate the total phase margin, the individual phase shift
contributions of each pole and zero will be calculated assuming fp = 200 Hz, fz = 5 kHz, fzf = 10 kHz, fpf = 40 kHz,
fc = 50 kHz, and fpg = 100 kHz:
Controller pole shift = −90˚
fp shift = −arctan (50k/200) = −90˚
fz shift = arctan (50k/5k) = +84˚
fzf shift = arctan (50k/100k) = +79˚
fpf shift = −arctan (50k/40k) = −51˚
fpg shift = −arctan (50k/100k) = −27˚
Summing the six numbers, the estimate for the total phase
shift is −95˚, which corresponds to a phase margin of 85˚
(a 27˚ improvement over the same application without the
feed-forward capacitor).
For this reason, a feed-forward capacitor is recommended in
all applications. Although not always required, the added
phase margin typically gives faster settling times and provides some design guard band against COUT and ESR variations with temperature.
High ESR Corrected with Feed-Forward
CAUSES AND CURES OF OSCILLATIONS
The most common cause of oscillations in an LDO application is the output capacitor ESR. If the ESR is too high or too
low, the zero (fz) does not provide enough phase lead.
HIGH ESR: To illustrate the effect of an output capacitor with
high ESR, the previous example will be repeated except that
the ESR will be increased by a factor of 20X. This will cause
the frequency of the zero fz to decrease by 20X, which
moves it from 5 kHz down to 250 Hz (see graph HIGH ESR
UNSTABLE WITHOUT FEED-FORWARD).
DS100034-31
LOW ESR: To illustrate how an output capacitor with low
ESR can cause an LDO regulator to oscillate, the same example will be shown except that the ESR will be reduced sufficiently to increase the original fz from 5 kHz to 50 kHz.
The plot now shows (see graph LOW ESR UNSTABLE
WITHOUT FEED-FORWARD) that the crossover frequency
fc has moved down to about 8 kHz. Since fz is 6X fc, it means
that the zero fz can only provide about 9˚ of phase lead at fc,
which is not sufficient for stability.
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Application Hints
The use of a feed-forward capacitor CF will help reduce excess phase shift due to fpg, but its effectiveness depends on
output voltage (see next section).
(Continued)
Low ESR Unstable without Feed-Forward
LOW OUTPUT VOLTAGE AND CF
The feed-forward capacitor CF will provide a positive phase
shift (lead) which can be used to cancel some of the excess
phase lag from any of the various poles present in the loop.
However, it is important to note that the effectiveness of CF
decreases with output voltage.
This is due to the fact that the frequencies of the zero fzf and
pole fpf get closer together as the output voltage is reduced
(see equations in section FEED-FORWARD COMPENSATION).
CF is more effective when the pole-zero pair are farther
apart, because there is less self cancellation. The net benefit
in phase shift provided by CF is the difference between the
lead (positive phase shift) from fzf and the lag (negative
phase shift) from fpf which is present at the crossover frequency fc. As the pole and zero frequency approach each
other, that difference diminishes to nothing.
The amount of phase lead at fC provided by CF depends
both on the fzf/fpf ratio and the location of fz with respect to fc.
To illustrate this more clearly, a graph is provided which
shows how much phase lead can be obtained for VOUT =
12V, 5V, and 3.3V (see graph PHASE LEAD PROVIDED BY
CF).
DS100034-30
This application can also be improved by adding a feedforward capacitor. CF will add both a zero fzf and pole fpf to
the gain plot (see graph LOW ESR CORRECTED WITH
FEED-FORWARD).
The crossover frequency fc is now about 10 kHz. If CF is selected so that fzf is about 5 kHz, and fpf is about 20 kHz
(which means VOUT = 5V), the phase margin will be considerably improved. Calculating out all the poles and zeroes,
the phase margin is increased from 9˚ to 43˚ (adequate for
good stability).
Phase Lead Provided by CF
Low ESR Corrected with Feed-Forward
DS100034-33
The most important information on the graph is the frequency range of fzf which will provide the maximum benefit
(most positive phase shift):
For VOUT = 12V: 0.1 fc < fz < 1.0 fc
For VOUT = 5V: 0.2 fc < fz < 1.2 fc
For VOUT = 3.3V: 0.2 fc < fz < 1.3 fc
DS100034-32
EXCESSIVE GATE CAPACITANCE: Higher values of gate
capacitance shift the pole fpg to lower frequencies, which can
cause stability problems (see previous section GATE CAPACITANCE POLE FREQUENCY). As shown in the graph
fpg vs. CEFF, the pole fpg will likely fall somewhere between
40 kHz and 500 kHz. How much phase shift this adds depends on the crossover frequency fc.
The effect of gate capacitance becomes most important at
high values of ESR for the output capacitor (see graph HIGH
ESR UNSTABLE WITHOUT FEED-FORWARD). Higher values of ESR increase fc, which brings fpg more into the positive gain portion of the curve. As fpg moves to a lower frequency (corresponding to higher values of gate
capacitance), this effect becomes even worse.
This points out why FET’s should be selected with the lowest
possible gate capacitance: it makes the design more tolerant
of higher ESR values on the output capacitor.
It’s also important to note how the maximum available phase
shift that CF can provide drops off with VOUT. At 12V, more
than 50˚ can be obtained, but at 3.3V less than 30˚ is possible. The lesson from this is that higher voltage designs are
more tolerant of phase shifts from both fpg (the gate capacitance pole) and incorrect placement of fz (which means the
output capacitor ESR is not at its nominal value). At lower
values of VOUT, these parameters must be more precisely
selected since CF can not provide as much correction.
GENERAL DESIGN PROCEDURE
Assuming that VIN, VOUT, and RL are defined:
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Application Hints
maximum phase gain for the output voltage selected (see
previous section LOW OUTPUT VOLTAGE AND CF). The
formula for calculating CF is in the previous section FEEDFORWARD CAPACITOR.
Lower ESR electrolytics are available which use organic
electrolyte (OSCON types), but are more costly than typical
aluminum electrolytics.
If the calculated value of ESR is higher than what is found in
the selected capacitor, an external resistor can be placed in
series with COUT.
LOW VOLTAGE DESIGNS: Designs which have a low output voltage (where the positive effects of CF are very small)
may be marginally stable if the COUT and ESR values are not
carefully selected.
Also, if the FET gate capacitance is large (as in the case of
a high-current FET), the pole fpg could possibly get low
enough in frequency to cause a problem.
(Continued)
1) Calculate the required value of capacitance for COUT so
that the pole fp ≤ 200 Hz (see previous section OUTPUT CAPACITOR). For this calculation, an ESR of about 0.1Ω can
be assumed for the purpose of determining COUT.
IMPORTANT: If a smaller value of output capacitor is used
(so that the value of fp > 200 Hz), the phase margin of the
control loop will be reduced. This will result in increased ringing on the output voltage during a load transient. If the output
capacitor is made extremely small, oscillations will result.
To illustrate this effect, scope photos have been presented
showing the output voltage of reference design #2 as the
output capacitor is reduced to approximately 1/30 of the
nominal value (the value which sets fp = 200 Hz). As shown,
the effect of deviating from the nominal value is gradual and
the regulator is quite robust in resisting going into oscillations.
2) Approximate the crossover frequency fc using the equation in the previous section CROSSOVER FREQUENCY
AND PHASE MARGIN.
3) Calculate the required ESR of the output capacitor so that
the frequency of the zero fz is set to 0.5 fc (see previous section OUTPUT CAPACITOR).
4) Calculate the value of the feed-forward capacitor CF so
that the zero fzf occurs at the frequency which yields the
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The solution in both cases is to increase the amount of output capacitance which will shift fp to a lower frequency (and
reduce overall loop bandwidth). The ESR and CF calculations should be repeated, since this changes the crossover
frequency fc.
18
LP2975 MOSFET LDO Driver/Controller
Physical Dimensions
inches (millimeters) unless otherwise noted
Surface Mount Mini SO-8 Package
NS Package Number MUA08A
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