LINER LTC3854EDDBTRPBF Small footprint, wide vin range synchronous step-down dc/dc controller Datasheet

LTC3854
Small Footprint, Wide
VIN Range Synchronous
Step-Down DC/DC Controller
Description
Features
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Wide Operating VIN Range: 4.5V to 38V
RSENSE or DCR Current Sensing
±1% 0.8V Reference Accuracy Over Temperature
400kHz Switching Frequency
Dual N-channel MOSFET Synchronous Drive
Very Low Dropout Operation: 97% Duty Cycle
Starts Up Into Pre-Biased Output
Adjustable Output Voltage Soft-Start
Output Current Foldback Limiting
(Disabled During Soft-Start)
Output Overvoltage Protection
5V LDO for External Gate Drive
OPTI-LOOP® Compensation Minimizes COUT
Low Shutdown IQ: 15µA
Tiny Thermally Enhanced 12-Pin 2mm × 3mm DFN
and MSOP Packages
The LTC3854® is a high performance synchronous stepdown switching DC/DC controller that drives an all Nchannel synchronous power MOSFET stage.
The LTC3854 features a 400kHz constant frequency current
mode architecture. The LTC3854 operates from a 4.5V to
38V (40V absolute maximum) input voltage range and
regulates the output voltage from 0.8V to 5.5V.
The RUN/SS pin provides both soft-start and enable
features. OPTI-LOOP compensation allows the transient
response to be optimized over a wide range of output capacitance and ESR values. Current foldback limits MOSFET
dissipation during short circuit conditions. Current foldback
functions are disabled during soft-start.
The LTC3854 has a minimum on-time at 75ns, making it
well suited for high step-down ratios. The strong onboard
MOSFET drivers allow the use of high power external
MOSFETs to produce output currents up to 20A.
Applications
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Automotive Systems
Telecom Systems
Industrial Equipment
Distributed DC Power Systems
L, LT, LTC, LTM, Linear Technology, the Linear logo and OPTI-LOOP are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners. Protected by U.S. Patents, including 5705919, 6498466, 5408150, 6222231.
Typical Application
High Efficiency Synchronous Step-Down Converter
VIN
ITH
2200pF
10k
0.1µF
BOOST
100pF
INTVCC
5.49k
0.22µF
4.7µF
8.06k
FB
42.2k
4.7µH
AT 8.2mΩ DCR
SW
BG
SENSE–
SENSE+
EFFICIENCY
+
VOUT
5V
7A
1.6
95
1.2
94
0.8
POWER LOSS
150µF
93
0.4
4.99k
92
GND
POWER LOSS (W)
RUN/SS
2.0
96
EFFICIENCY (%)
0.1µF
97
VIN
6V TO 38V
47µF
50V
TG
LTC3854
Efficiency and Power Loss
vs Load Current
VIN = 12V
1
2
3
4
5
LOAD CURRENT (A)
6
7
0
3854 TA01b
3854 TA01
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LTC3854
Absolute Maximum Ratings
(Note 1)
Input Supply Voltage (VIN)......................... 40V to –0.3V
Top Side Driver Voltage (BOOST)............... 46V to –0.3V
Switch Voltage (SW)................................... 40V to –5.0V
INTVCC, BOOST-SW..................................... 6V to –0.3V
SENSE+, SENSE– . ........................................ 6V to –0.3V
RUN/SS......................................................... 6V to –0.3V
ITH, FB Voltages........................................ 2.7V to –0.3V
INTVCC Peak Output Current (Note 8).....................40mA
Operating Temperature Range
(Notes 2, 3)...........................................–40°C to 85°C
Maximum Junction Temperature....................... 125°C
Storage Temperature Range................... –65°C to 125°C
Pin Configuration
TOP VIEW
FB 1
12 SENSE+
ITH 2
11 SENSE–
RUN/SS 3
BOOST 4
13
TOP VIEW
FB
ITH
RUN/SS
BOOST
TG
SW
10 VIN
9 INTVCC
TG 5
8 BG
SW 6
7 GND
DDB PACKAGE
12-LEAD (3mm s 2mm) PLASTIC DFN
TJMAX = 125°C, θJA = 76°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 13) IS GND, MUST BE SOLDERED TO PCB
1
2
3
4
5
6
13
12
11
10
9
8
7
SENSE+
SENSE–
VIN
INTVCC
BG
GND
MSE PACKAGE
12-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 40°C/W, θJC = 16°C/W
EXPOSED PAD (PIN 13) IS GND, MUST BE SOLDERED TO PCB
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3854EDDB#PBF
LTC3854EDDB#TRPBF
LDPC
12-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
LTC3854IDDB#PBF
LTC3854IDDB#TRPBF
LDPC
12-Lead (3mm × 2mm) Plastic DFN
–40°C to 85°C
LTC3854EMSE#PBF
LTC3854EMSE#TRPBF
3854
12-Lead Plastic MSOP
–40°C to 85°C
LTC3854IMSE#PBF
LTC3854IMSE#TRPBF
3854
12-Lead Plastic MSOP
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3854fa
LTC3854
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, VRUN = 5V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
38
V
0.8
0.808
V
±5
±50
nA
Main Control Loop
VIN
Operating Input Voltage Range
VFB
Regulated Feedback Voltage
(Note 4); ITH Voltage = 1.2V
4.5
IFB
Feedback Current
(Note 4)
VREFLNREG
Reference Voltage Line Regulation
VIN = 6V to 38V (Note 4)
VLOADREG
Output Voltage Load Regulation
(Note 4) Measured in Servo Loop;
∆ITH Voltage = 0.7V to 1.2V
Measured in Servo Loop;
∆ITH Voltage = 1.2V to 2V
l
0.792
0.002
0.02
%/V
l
0.1
0.5
%
l
–0.1
–0.5
%
gm
Transconductance Amplifier gm
ITH = 1.2V; Sink/Source = 5µA (Note 4)
2.0
gmGBW
Transconductance Amplifier GBW
ITH = 1.2V; (Guaranteed by Design)
3
IQ
Input DC Supply Current
Normal Mode
Shutdown
(Note 5)
RUN = 0V
2
10
UVLO
Undervoltage Lockout
VIN Ramping Down; Measured at INTVCC
mmho
MHz
3
25
mA
µA
l
3.0
3.5
V
l
0.86
0.88
0.90
V
±0.5
±1
µA
2.0
µA
UVLOHYST
Undervoltage Lockout Hysteresis
VIN Ramping Down then Up; Measured at INTVCC
VOVL
Feedback Overvoltage Lockout
Measured at FB
350
ISENSE
Sense Pins Source Current
VSENSE– = VSENSE+ = 3.3V
DFMAX
Maximum Duty Factor
In Dropout
97
98
IRUN/SS
Soft-Start Charge Current
RUN/SS = 0V
0.6
1.25
VRUN/SS_SD
Shutdown Threshold
RUN/SS Pin Must be Taken Below this Value to
Reset Part (or Put into Shutdown Mode)
0.4
V
VRUN/SS_ON
Soft-Start Threshold
Soft-Start Mode
1.2
V
VSENSE(MAX)
Maximum Current Sense Threshold
FB = 0.7V, VSENSE– = 3.3V, VIN = 6V
TG RUP
TG Driver Pull-Up On Resistance
TG High
2.5
Ω
TG RDOWN
TG Driver Pull-Down On Resistance
TG Low
1.2
Ω
BG RUP
BG Driver Pull-Up On Resistance
BG High
2.5
Ω
BG RDOWN
BG Driver Pull-down On Resistance
BG Low
2.1
Ω
TG tr
TG tf
TG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
BG tr
BG tf
BG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
25
ns
ns
TG/BG t1D
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
BG/TG t2D
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
tON(MIN)
Minimum On-Time
(Note 7)
75
ns
40
50
mV
%
65
mV
INTVCC Linear Regulator
VINTVCC
Internal VCC Voltage
6V < VIN < 38V
VLDO INT
INTVCC Load Regulation
ICC = 0 to 20mA
4.8
5.0
5.2
V
0.2
1.0
%
400
440
kHz
Oscillator
fSW
Switching Frequency
360
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LTC3854
electrical Characteristics
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3854E is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LTC3854I is guaranteed to meet
performance specifications over the full –40°C to 85°C operating
temperature range.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC3854DDB: TJ = TA + (PD • 76°C/W)
LTC3854MSE: TJ = TA + (PD • 40°C/W)
Note 4: The LTC3854 is tested in a feedback loop that servos VITH to a
specified voltage and measures the resultant VFB.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels. Not 100% tested in production.
Note 7: The minimum on-time condition is specified for an inductor peakto-peak ripple current 40% of IMAX (see Minimum On-Time Considerations
in the Applications Information section).
Note 8: The LTC3854 maximum LDO current specification assumes there
is no external DC load current being pulled from INTVCC pin.
Typical Performance Characteristics
Efficiency vs Output Current
100
EFFICIENCY (%)
VIN = 24V
94
VIN = 32V
92
90
97
IOUT = 5A
96
95
IOUT = 10A
94
88
86
VOUT = 5V
98
VIN = 12V
96
EFFICIENCY (%)
VOUT = 5V
VIN = 6V
98
Efficiency vs Input Voltage
99
(SEE FIGURE 9)
1
2
3
4
6
5
7
8
LOAD CURRENT (A)
9
93
10
(SEE FIGURE 9)
5
10
3854 G01
25
20
VIN (V)
Top Gate and Bottom Gate in
Forced Continuous Mode
5A Load Step VOUT = 5V, VIN = 24V
5VOUT
200mV/DIV
15
30
35
40
3854 G02
Top Gate and Bottom Gate in
Dropout
BG
BG
ILOAD
10A/DIV
TG
TG
5A/DIV
LOAD STEP
50µs/DIV
3854 G03
4µs/DIV
3854 G04
4µs/DIV
3854 G05
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LTC3854
Typical Performance Characteristics
Switching Waveforms at High
Duty Cycle, No Load
Switching Waveforms at No Load
VOUT = 2.5V
VOUT = 5V
TG
TG
BG
BG
IL
IL
1µs/DIV
Quiescent Current vs VIN
Quiescent Current vs Temperature
1.30
1.14
1.13
1.12
1.11
1.10
1.15
1.10
1.05
1.00
6
11
16
26
21
VIN (V)
31
4.90
4.85
0.90
–45 –25
36
–5
15 35 55 75
TEMPERATURE (°C)
3854 G08
4.80
95
58
60
56
VSENSE(MAX) (mV)
52
50
48
46
44
55
50
45
40
42
0
1
3
5
2
4
6
COMMON MODE VOLTAGE (V)
7
3854 G11
35
0
20
60
40
80
DUTY CYCLE (%)
100
120
3854 G12
9
14
19
24
VIN (V)
29
34
39
3854 G10
Maximum Current Sense Voltage
vs Feedback Voltage
(Current Foldback)
Maximum Current Sense Threshold
vs Duty Cycle
65
54
4
3854 G09
60
VSENSE(MAX) (mV)
4.95
0.95
Maximum Current Sense Threshold
vs Sense Common Mode Voltage
40
INTVCC vs Input Voltage
5.00
1.20
INTVCC (V)
1.15
1.09
3854 G07
1.25
QUIESCENT CURRENT (mA)
QUIESCENT CURRENT (mA)
1.16
5.05
MAXIMUM CURRENT SENSE VOLTAGE (mV)
1.17
2µs/DIV
3854 G06
60
50
40
30
20
10
0
0
0.05 0.15 0.25 0.35 0.45 0.55 0.65 0.75
FB (V)
3854 G13
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LTC3854
Typical Performance Characteristics
Oscillator Frequency
vs Temperature
Undervoltage Lockout Falling
Threshold vs Temperature
440
440
3.9
430
3.8
400
FREQUENCY (kHz)
410
3.6
3.5
3.4
3.3
0
20 40 60 80
TEMPERATURE (°C)
3.0
–40 –20
100 120
0
3854 G15
360
100 120
19
20
10
15
10
0
5
–10
0
0
0.5
1
1.5
VITH (V)
2
2.5
3
SHUTDOWN CURRENT (µA)
30
20
4
8
12
16
3854 G18
INTVCC Load Regulation
5.20
5.15
5.10
5.10
5.05
5.05
INTVCC (V)
4.95
4.85
4.85
10
15 20 25 30
LDO CURRENT (mA)
35
40
3854 G21
32
36
15
25
20
VIN (V)
30
35
40
3854 G17
VIN = 15V
18
17
16
15
14
13
12
10
–45 –30 –15 0 15 30 45 60 75 90 105
TEMPERATURE (°C)
40
3854 G20
3854 G19
RUN/SS Shutdown Threshold
vs Temperature
1.40
1.35
1.30
4.95
4.90
5
28
5.00
4.90
0
20 24
VIN (V)
INTVCC vs Temperature
ILOAD = 10mA
5.15 VIN = 15V
VIN = 15V
5.00
10
11
RUN/SS (V)
SHUTDOWN CURRENT (µA)
25
40
5
Shutdown Current vs Temperature
20
30
50
CURRENT SENSE THRESHOLD (mV)
INTVCC (V)
20 40 60 80
TEMPERATURE (°C)
Shutdown Current
vs Input Voltage
60
4.80
390
3854 G16
Current Sense Threshold
vs ITH Voltage
5.20
400
370
3.1
380
–40 –20
410
380
3.2
390
Oscillator Frequency vs VIN
420
3.7
420
UVLO (V)
FREQUENCY (kHz)
430
4.0
1.25
1.20
1.15
4.80
–45 –30 –15 0 15 30 45 60 75 90 105
TEMPERATURE (°C)
3854 G22
1.10
–45 –30 –15 0 15 30 45 60 75 90 105
TEMPERATURE (°C)
3854 G23
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LTC3854
Pin Functions
FB (Pin 1): Error Amplifier Feedback Input. This pin receives
the remotely-sensed feedback voltage from an external
resistor divider across the output.
ITH (Pin 2): Error Amplifier Output and Switching Regulator
Compensation Point. The current comparator trip point
increases with this control voltage.
RUN/SS (Pin 3): Run Control, Soft-Start. If the voltage
on this pin is held below 0.4V, the part is in shutdown. If
the pin is released the capacitance to ground at this pin
sets the soft-start ramp rate. An internal 1.25µA soft-start
current is always charging this pin.
BOOST (Pin 4): Bootstrapped Supply to the Top Side Floating Driver. A low ESR capacitor is connected between the
BOOST and SW pins and an external Schottky diode is tied
between the BOOST and INTVCC pins. The voltage swing
on the BOOST pin is INTVCC to (VIN + INTVCC).
TG (Pin 5): High Current Gate Drive for Top N-channel
MOSFET. This is the output of floating driver with a voltage swing equal to INTVCC superimposed on the switch
node voltage.
SW (Pin 6): Switch Node Connection to Inductor. The voltage swing on this pin is from a Schottky diode (external)
forward voltage (when this diode is added across the Nchannel synchronous MOSFET) below ground to VIN.
GND (Pin 7): Small Signal and Power Ground. This is
the high current ground for the gate driver. The internal
signal ground is Kelvin connected to this pin for noise
suppression.
BG (Pin 8): High Current Gate Drive for Bottom (Synchronous) N-channel MOSFET. The voltage swing at this pin is
from ground to INTVCC.
INTVCC (Pin 9): Output of the Internal 5V Low Dropout
Regulator. The driver and control circuits are powered
from this voltage. Must be decoupled to power ground
with a minimum of 2.2µF low ESR ceramic capacitor (X5R
or better).
VIN (Pin 10): Main Supply Pin. A bypass capacitor should
be tied between this pin and the signal ground pin.
SENSE– (Pin 11): The (–) Input to the Differential Current
Comparator.
SENSE+ (Pin 12): The (+) Input to the Differential Current
Comparator. The ITH pin voltage and controlled offsets
between the SENSE– and SENSE+ pins in conjunction with
RSENSE (or RDCR) set the peak current trip threshold.
SGND (Exposed Pad Pin 13): The exposed pad must be
soldered to PCB ground for electrical contact and rated
thermal performance.
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LTC3854
Functional Diagram
VIN
1.2V
0.88V
0.80V
BANDGAP/
LDO
INTVCC
BOOST
MAIN
SWITCH
DRIVER
OSCILLATOR
INTVCC
UVLO
TG
DRIVE
CONTROL
AND
ANTI-SHOOT
THROUGH
SW
SYNCHRONOUS
SWITCH DRIVER
BG
ON
GND
CLK
S Q
+
R
SLOPE COMP
GENERATOR
SENSE–
SENSE+
V TO I
CONVERTER
VSENSE m ICS
OV
–
CURRENT
SENSE
COMPARATOR
VFB
0.88V
TG_OFF
+
+
RUN
ICMP
–
–
+
1.2V
SS
–
–
0.4V
1.25µA
SD
+
ITH
ERROR
AMP
VITH
+
+
–
LEVEL SHIFT
RUN/SS
0.8V
ITH
3854 FD
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LTC3854
Operation
Main Control Loop
Shutdown and Start-Up (RUN/SS)
The LTC3854 is a constant-frequency, peak current mode
step-down controller. During normal operation, the top
MOSFET is turned on when the clock sets the RS latch,
and is turned off when the main current comparator, ICMP,
resets the RS latch. The peak inductor current at which
ICMP resets the RS latch is controlled by the voltage on
the ITH pin, which is the output of the error amplifier EA.
The VFB pin receives the voltage feedback signal, which is
compared to the internal reference voltage by the EA. When
the load current increases, it causes a slight decrease in
VFB relative to the 0.8V reference, which in turn causes
the ITH voltage to increase until the average inductor current matches the new load current. After the top MOSFET
has turned off, the bottom MOSFET is turned on until the
beginning of the next cycle.
The LTC3854 is shut down using the RUN/SS pin. Pulling
this pin below 1.2V disables the controller and most of the
internal circuitry, including the INTVCC regulator.
INTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin. An
internal 5V low dropout linear regulator supplies INTVCC
power from VIN.
The top MOSFET driver is biased from a floating bootstrap
capacitor CB , which recharges during each off cycle through
an external Schottky diode when the top MOSFET turns
off. If the input voltage VIN decreases to a voltage close to
VOUT, the loop may enter dropout and attempt to turn on
the top MOSFET continuously. The dropout detector then
forces the top MOSFET off for 1/10 of the clock period
every fourth cycle to allow CB to recharge.
However, for RUN/SS>0.8V the internal bandgap is
functional and the input current will be greater than the
minimum shutdown current. To keep the part in a true
shutdown mode the RUN/SS pin should be held below 0.4V.
Releasing RUN/SS pin allows an internal 1.25µA current to
pull up the pin and enable the controller. Alternatively, the
RUN/SS pin may be externally pulled up or driven directly
by logic. Be careful not to exceed the Absolute Maximum
Rating of 6V on this pin.
The start-up of the controller’s output voltage VOUT
is governed by the voltage on the RUN/SS pin until
RUN/SS > 2V.
When the voltage on the RUN/SS pin is greater than 1.2V
and less than 2V the LTC3854 regulates the VFB voltage to
1.2V below the RUN/SS pin voltage. The RUN/SS pin programs the soft-start period through an external capacitor
from the RUN/SS pin to GND. An internal 1.25µA pull-up
current charges this capacitor creating a voltage ramp on
the RUN/SS pin. As the RUN/SS voltage rises linearly from
1.2V to 2V, VOUT rises smoothly from zero to the target
output voltage. When the LTC3854 is in undervoltage
lockout the external MOSFETs are held off.
Frequency of Operation
The LTC3854 operates at a fixed frequency of 400kHz.
Output Overvoltage Protection
An overvoltage comparator, OV, guards against transient
overshoots (>10%) as well as other more serious conditions that may overvoltage the output. In such cases, the
top MOSFET is turned off and the bottom MOSFET is turned
on until the overvoltage condition is cleared.
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LTC3854
Applications Information
The LTC3854 can be configured to use either DCR (inductor
winding resistance) sensing or low value resistor sensing.
The choice of the two current sensing schemes is largely
a design tradeoff between cost, power consumption, and
accuracy. DCR sensing is becoming popular because it
eliminates expensive current sensing resistors and is
more power efficient, especially in high current applications. However, current sensing resistors provide the most
accurate current limits for the controller. Other external
component selection is driven by the load requirement,
and begins with the selection of RSENSE (if RSENSE is used)
and inductor value. Next, the power MOSFETs and Schottky
diodes are selected. Finally, input and output capacitors are
selected. The Typical Application shown on the first page
can be configured for operation up to 38V on VIN.
SENSE+ and SENSE– Pins
The SENSE+ and SENSE– pins are the inputs to the current comparator. The common mode input voltage range
of the current comparator is 0V to 5.5V. Both SENSE pins
are high impedance inputs with small input bias currents
of less than 1μA. When the SENSE pins ramp up from 0V
to 1.4V, small bias currents flow out of the SENSE pins.
When the SENSE pins ramp down from 5.5V to 1.1V, the
small bias currents flow into the SENSE pins. The high
impedance inputs to the current comparator allow accurate
DCR sensing.
Using a Sense Resistor for Current Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 1. RSENSE is chosen based on the required output
current.
The current comparator has a maximum threshold of 50mV.
The input common mode range of the current comparator
is 0V to 5.5V. The current comparator threshold sets the
peak of the inductor current, yielding a maximum average output current IMAX equal to the peak value less half
the peak-to-peak ripple current, ΔIL. Allowing a margin
of 20% for variations in the IC and external component
values yields:
VSENSE(MAX)
∆I
IMAX + L
2
Inductor DCR Sensing
RSENSE = 0.8 •
For applications requiring the highest possible efficiency,
the LTC3854 is capable of sensing the voltage drop across
the inductor DCR, as shown in Figure 2. The DCR of the
inductor represents the small amount of DC copper winding resistance, which can be less than 1mΩ for today’s
low value, high current inductors. When the external
R1||R2•C1 time constant is chosen to be equal to the
L/DCR time constant, the voltage drop across the external
VIN
INTVCC
BOOST
LTC3854
TG
RSENSE
SW
VOUT
BG
GND
SENSE+
SENSE–
FILTER COMPONENTS
PLACED NEAR SENSE PINS
3854 F01
Figure 1. Using a Resistor to Sense Current with the LTC3854
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10
LTC3854
Applications Information
VIN
CIN
VIN
RUN/SS
M1
TG
CB
CSS
ITH
CC
CC2
BOOST
LTC3854
DBOOST
RC
RFB1
SW
FB
RFB2
INDUCTOR
INTVCC
L
DCR
VOUT
CVINT
SENSE–
BG
SENSE+
GND
M2
R1
C1
COUT
R2
3854 F02
Figure 2. Buck Regulator Using DCR Current Sense
capacitor is equal to the voltage drop across the inductor
DCR • R2/(R1+R2). R2 may be used to scale the voltage
across the same terminals when the DCR is greater than
the target sense resistance. Check the manufacturer’s
datasheet for specifications regarding the inductor DCR, in
order to properly dimension the external filter components.
The DCR of the inductor can also be measured using a
precision RLC meter.
Slope Compensation and Inductor Peak Current
Slope compensation provides stability in constant-frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally
by adding a compensating ramp to the inductor current
signal. Normally, this results in a reduction of maximum
inductor peak current for high duty cycles. However, the
LTC3854 uses a novel scheme that allows the maximum
inductor peak current to remain unaffected throughout
all duty cycles.
Inductor Value Calculation
The inductor value has a direct effect on ripple current.
The inductor ripple current ΔIL decreases with higher
inductance or frequency and increases with higher VIN.
LMIN =


V
1
• VOUT  1− OUT 
∆IL • fSW
 VIN(MAX) 
Accepting larger values of ΔIL allows the use of low value
inductors, but results in a higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is IL = 0.4 • (IMAX). The maximum
ΔIL occurs at the maximum input voltage.
Option 1: DCR within desired range
R1• C1=
L
(R2 not used)
DCR
Option 2: DCR > desired RSENSE
R1||R2 • C1 =
L
(at 20°C)
DCR
RSENSE (EQ) = DCR(MAX) •
R2
R1+ R2
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LTC3854
Applications Information
Inductor Core Selection
Once the value for L is determined, the type of inductor
must be selected. High efficiency converters generally
cannot afford the core loss found in low cost powdered
iron cores, forcing the use of more expensive ferrite or
molypermalloy cores. Actual core loss is independent of
core size for a fixed inductor value, but it is very dependent
on inductance selected. As inductance increases, core
losses decrease. Unfortunately, increased inductance
requires more turns of wire and therefore copper losses
will increase.
Ferrite designs have very low core loss and are preferred
at high switching frequencies; allowing design goals to
concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design current
is exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Power MOSFET and Schottky Diode (Optional)
Selection
then multiplied by the ratio of the applied VDS to the gate
charge curve specified VDS. When the IC is operating in
continuous mode the duty cycles for the top and bottom
MOSFETs are given by:
Main Switch Duty Cycle =
VOUT
=D
VIN
Synchronous Switch Duty Cycle =
VIN − VOUT
= 1− D
VIN
The MOSFET power dissipations at maximum output
current are given by:
PMAIN =
VOUT
2
IMAX ) (1+ δ ) RDS(ON) +
(
VIN

( VIN )2  IMAX
(R )(C
)•
2  DR MILLER

1 
1
+

 (f)
 VINTVCC − VTH(MIN) VTH(MIN) 
V −V
2
PSYNC = IN OUT (IMAX ) (1+ δ ) RDS(ON)
VIN
Two external power MOSFETs must be selected for the
LTC3854 controller: one N-channel MOSFET for the top
(main) switch, and one N-channel MOSFET for the bottom
(synchronous) switch.
where δ is the temperature dependency of RDS(ON) and
RDR (approximately 2Ω) is the effective driver resistance
at the MOSFET’s Miller threshold voltage. VTH(MIN) is the
typical MOSFET minimum threshold voltage.
The peak-to-peak drive levels are set by the INTVCC voltage.
This voltage is 5V during start-up. Consequently, logiclevel threshold MOSFETs can be used in most applications.
The only exception is if low input voltage is expected (VIN
< 5V); then, sub-logic level threshold MOSFETs (VGS(TH)
< 3V) should be used. Pay close attention to the BVDSS
specification for the MOSFETs as well; most of the logic
level MOSFETs are limited to 30V or less.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V,
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
short-circuit when the synchronous switch is on close to
100% of the period.
Selection criteria for the power MOSFETs include the
on-resistance RDS(ON), Miller capacitance CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result is
The term (1 + δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
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LTC3854
Applications Information
An optional Schottky diode connected from GND (anode)
to the SW node (cathode) conducts during the dead time
between the conduction of the two power MOSFETs. This
prevents the body diode of the bottom MOSFET from turning on, storing charge during the dead time and requiring
a reverse recovery period that could cost as much as 3%
in efficiency at high VIN. A 1A to 3A Schottky is generally
a good size due to the relatively small average current.
Larger diodes result in additional transition losses due to
their larger junction capacitance.
Soft-Start
When the LTC3854 is configured to soft-start by itself,
a capacitor must be connected to the RUN/SS pin. The
LTC3854 is in the shutdown state if the RUN/SS pin voltage is below 1.2V. The RUN/SS pin has an internal 1.25µA
pull-up current and should be externally pulled low (<0.4V)
to keep IC in shutdown mode.
Once the RUN/SS pin voltage reaches 1.2V, the LTC3854
is enabled. As the RUN/SS pin moves from 1.2V to 2V the
LTC3854 operates in a forced discontinuous mode with
the bottom gate turning on only one time for every four
clock cycles to allow the output to come up to its required
value. During this time the error amp compares the FB
pin to a level shifted version of the RUN/SS pin allowing
the output to come up in a controlled fashion. Current
foldback is disabled during this phase to ensure smooth
soft-start or tracking. Once the RUN/SS pin is greater
than 2V the LTC3854 operates in forced continuous mode.
The LTC3854 output voltage is soft-start controlled when
RUN/SS is between 1.2V and 2V. The total soft-start time
can be calculated as:
C
t SOFT-START = 0.8 • SS
1.25µA
If the RUN/SS pin is externally driven beyond 2V (5V
is recommended) the soft-start feature is disabled and
the LTC3854 will immediately go into forced continuous
mode. Care must be taken to insure the RUN/SS pin has
either a capacitor tied to it or is driven externally. Do not
let this pin float.
INTVCC Regulator
The LTC3854 features a PMOS low dropout linear regulator
(LDO) that supplies power to INTVCC from the VIN supply.
INTVCC powers the gate drivers and much of the LTC3854’s
internal circuitry. The LDO regulates the voltage at the
INTVCC pin to 5V when VIN is greater than 6V.
The LDO supplies a peak current of 40mA and must be
bypassed to ground with a minimum of 2.2μF low ESR
ceramic capacitor. Good bypassing is needed to supply
the high transient currents required by the MOSFET gatedrivers.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3854 to be
exceeded. The INTVCC current, which is dominated by the
gate-charge current, is supplied by the 5V LDO.
Power dissipation for the IC in this case is highest and is
equal to VIN •IINTVCC. The gate-charge current is dependent
on operating frequency (400kHz), and the QG of the power
MOSFETs, as discussed in the Efficiency Considerations
section. The junction temperature can be estimated by
using the equations given in Note 3 of the Electrical Characteristics section. For example, if the LTC3854 INTVCC
current is limited to less than 17mA from a 36V supply in
the DFN package; then the junction temperature is:
TJ = 70°C + [(17mA•36V)•(76°C/W)] = 116.5°C
To prevent the maximum junction temperature from being exceeded, the input supply current must be checked
during operation at maximum VIN.
Topside MOSFET Driver Supply (CB, DB)
An external bootstrap capacitor CB connected from the
BOOST pin to the SW pin and supplies the gate drive voltage for the topside MOSFET. Capacitor CB in the Functional
Diagram is charged though external diode DB from INTVCC
when the SW pin is low. When the topside MOSFET is to
be turned on, the driver places the CB voltage across the
gate source of the MOSFET. This enhances the MOSFET
and turns on the topside switch. The switch node voltage, SW, rises to VIN and the BOOST pin follows. With the
topside MOSFET on, the boost voltage is above the input
supply: VBOOST = VIN + VINTVCC. The value of the boost
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LTC3854
Applications Information
capacitor CB needs to be at least 100 times that of the
total input capacitance of the topside MOSFET. The reverse
breakdown of the external Schottky diode must be greater
than VIN(MAX). When adjusting the gate-drive level, the
final arbiter is the total input current for the regulator. If
a change is made and the input current decreases, then
the efficiency has improved. If there is no change in input
current, then there is no change in efficiency.
Undervoltage Lockout
The LTC3854 has two functions that help protect the
controller in case of undervoltage conditions. A precision
UVLO comparator constantly monitors the INTVCC voltage
to ensure that an adequate gate-drive voltage is present.
Switching action is disabled when INTVCC is below 3.5V.
To prevent oscillation caused by a disturbance on INTVCC,
the UVLO comparator has 350mV of hysteresis.
Another way to detect an undervoltage condition is to
monitor the VIN supply. The RUN/SS pin has a precision
turn-on reference of 1.2V, enabling a resistor divider to VIN
to turn on the IC when VIN is above the desired value.
It is recommended that the resistor divider be used if the
input voltage will be quickly cycled on and off.
CIN Selection
In forced continuous mode, the source current of the
top N-channel MOSFET is a square wave of duty cycle
VOUT/VIN. To prevent large voltage transients, a low ESR
input capacitor sized for the maximum RMS current must
be used.
IRMS =
IOUT
2
The maximum RMS capacitor current is:
IRMS =
1/ 2
IMAX
( VOUT ) • ( VIN − VOUT ))
VIN
This formula has a maximum at VIN = 2•VOUT, where
IRMS = IOUT/2.
This simple worst-case condition is commonly used for
design because even significant deviations do not offer
much relief. Note that capacitor manufacturers’ ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design. Always consult
the manufacturer if there is any question.
COUT Selection
The selection of COUT is primarily determined by the effective series resistance (ESR) to minimize voltage ripple.
The output ripple (∆VOUT) in continuous mode is:


1
∆VOUT = ∆IL  ESR +
8 • fSW • COUT 

Where fSW = 400kHz, COUT = output capacitance and ∆IL =
ripple current in the inductor. The output ripple is highest
at maximum input voltage since ∆IL increases with input
voltage. Typically, once the ESR requirement for COUT has
been met, the RMS current rating generally far exceeds
the IRIPPLE(P-P) requirement. With ∆IL= 0.3IOUT(MAX) and
allowing 2/3 of the ripple due to ESR, the output ripple
will be less than 50mV at max VIN assuming:
COUT Required ESR < 2.2 RSENSE
COUT >
1
8fSW RSENSE
The first condition relates to the ripple current into the
ESR of the output capacitance while the second term guarantees that the output capacitance does not significantly
discharge during the operating frequency period due to
ripple current. The choice of smaller output capacitance
increases the ripple voltage due to the discharging term
but can be compensated with capacitors of very low ESR
to maintain the ripple voltage at or below 50mV. The ITH pin
OPTI-LOOP compensation components can be optimized
to provide stable, high performance transient response
regardless of the output capacitors selected. The selection of output capacitors for applications with large load
current transients is primarily determined by the voltage
tolerance specifications of the load. The resistive component of the capacitor, ESR, multiplied by the load current
change plus any output voltage ripple must be within the
voltage tolerance of the load.
3854fa
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LTC3854
Applications Information
The required ESR due to a load current step is:
ESR ≤
∆V
∆I
where ∆I is the change in current from full load to zero load
(or minimum load) and ∆V is the allowed voltage deviation
(not including any droop due to finite capacitance).
The amount of capacitance needed is determined by the
maximum energy stored in the inductor. The capacitance
must be sufficient to absorb the change in inductor current
when a high current to low current transition occurs. The
opposite load current transition is generally determined by
the control loop OPTI-LOOP components, so make sure
not to over compensate and slow down the response. The
minimum capacitance to assure the inductors’ energy is
adequately absorbed is:
L ( ∆I)
COUT >
2 ( ∆V ) VOUT
2
Manufacturers such as Nichicon, United Chemi-Con
and Sanyo should be considered for high performance
through-hole capacitors. The OS-CON semiconductor
electrolyte capacitor available from Sanyo has the lowest
(ESR)(size) product of any aluminum electrolytic at a
somewhat higher price. An additional ceramic capacitor
in parallel with OS-CON capacitors is recommended to
reduce the inductive effects.
In surface mount applications, ESR, RMS current handling
and load step specifications may require multiple capacitors in parallel. Aluminum electrolytic, dry tantalum and
special polymer capacitors are available in surface mount
packages. Special polymer surface mount capacitors offer
very low ESR but have much lower capacitive density per
unit volume than other capacitor types. These capacitors
offer a very cost-effective output capacitor solution and are
an ideal choice when combined with a controller having
high loop bandwidth. Tantalum capacitors offer the highest
capacitance density and are often used as output capacitors for switching regulators having controlled soft-start.
Several excellent surge-tested choices are the AVX TPS,
AVX TPSV or the KEMET T51 0 series of surface mount
tantalums, available in case heights ranging from 1.5mm
to 4.1mm. Aluminum electrolytic capacitors can be used
in cost-driven applications, provided that consideration is
given to ripple current ratings, temperature and long-term
reliability. A typical application will require several aluminum electrolytic capacitors in parallel. A combination of the
above mentioned capacitors will often result in maximizing
performance and minimizing overall cost. Other capacitor
types include Nichicon PL series, NEC Neocap, Panasonic
SP and Sprague 595D series. Consult manufacturers for
other specific recommendations.
Like all components, capacitors are not ideal. Each
capacitor has its own benefits and limitations. Combinations of different capacitor types have proven to be a very
cost effective solution. Remember also to include high
frequency decoupling capacitors. They should be placed
as close as possible to the power pins of the load. Any
inductance present in the circuit board traces negates
their usefulness.
Setting Output Voltage
The LTC3854 output voltage is set by an external feedback resistive divider carefully placed across the output,
as shown in Figure 3. The regulated output voltage is
determined by:
 R 
VOUT = 0.8  1+ B 
 RA 
VOUT
LTC3854
RB
CFF
FB
RA
3854 F03
Figure 3. Feed-Forward Capacitor on FB Pin
To improve the frequency response, a feed-forward capacitor, CFF, may be used. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
Fault Conditions: Current Foldback
The LTC3854 includes current foldback to help limit load
current when the output is shorted to ground. If the output
falls below 40% of its nominal output level, the maximum
3854fa
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LTC3854
Applications Information
sense voltage is progressively lowered from its maximum
programmed value to 25% of the maximum value. Foldback
current limiting is disabled during soft-start.
large currents when VIN is in the normal operating range
yet still provide an adequate safety margin and good overall
performance in dropout with a slow ramping VIN.
Minimum and Maximum On-Time Considerations
Efficiency Considerations
Minimum on-time tON(MIN) is the smallest time duration
that the LTC3854 is capable of turning on the top MOSFET.
It is determined by internal timing delays and the gate
charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It
is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
VOUT
>t
VIN • fSW ON(MIN)
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
The minimum on-time for the LTC3854 is approximately
75ns. However, as the peak sense voltage decreases the
minimum on-time gradually increases. This is of particular concern in forced continuous applications with low
ripple current at light loads. If the duty cycle drops below
the minimum on-time limit in this situation, a significant
amount of cycle skipping can occur with correspondingly
larger current and voltage ripple.
Care should also be taken for applications where the duty
cycle can approach the maximum given in the data sheet
(98%). In all low dropout applications, such as VOUT = 5V
and VIN(MIN) = 4.5V, careful selection of the bottom synchronous MOSFET is required. For applications where the
input voltage can drop below the targeted output voltage,
and subsequently ramp up, a low threshold synchronous
MOSFET with a small total gate charge should be chosen.
This selection for the bottom synchronous MOSFET will
insure that the bottom gate minimum on-time is sufficient
in dropout to allow for the initial boost capacitor refresh
that is needed to adequately turn on the top side driver and
begin the switching cycle. Another method to guarantee
performance in this type of application is to increase the
minimum output load to 50mA. This minimum load will
allow the user to choose larger MOSFETs for delivery of
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3854 circuits: 1) IC VIN current, 2) INTVCC
regulator current, 3) I2R losses, 4) Topside MOSFET
transition losses.
1. The VIN current is the DC supply current given in the
Electrical Characteristics table, which excludes MOSFET
driver and control currents. VIN current typically results
in a small (<0.1%) loss.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the
control circuit current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges of
the topside and bottom side MOSFETs.
3. I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor.
In continuous mode, the average output current flows
through L and RSENSE, but is “chopped” between the
topside MOSFET and the synchronous MOSFET. If the
two MOSFETs have approximately the same RDS(ON),
then the resistance of one MOSFET can simply be
summed with the resistances of L and RSENSE to obtain
3854fa
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LTC3854
Applications Information
I2R losses. For example, if each RDS(ON) = 10mΩ, DCR
= 10mΩ, RSENSE = 5mΩ then the total resistance is
25mΩ. This results in losses ranging from 2% to 8%
as the output current increases from 3A to 15A for
a 5V output, or a 3% to 12% loss for a 3.3V output.
Efficiency varies as the inverse square of VOUT for the
same external components and output power level. The
combined effects of increasingly lower output voltages
and higher currents required by high performance digital
systems is not doubling but quadrupling the importance
of loss terms in the switching regulator system!
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high
input voltages (typically 15V or greater). Transition
losses can be estimated from:
Transition Loss = 1.7VIN2 • IO(MAX) • CRSS • fS
Other “hidden” losses such as copper trace and the battery internal resistance can account for an additional 5%
to 10% efficiency degradation in portable systems. It is
very important to include these “system” level losses
during the design phase. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at the
switching frequency. A 25W supply will typically require a
minimum of 20μF to 40μF of capacitance having a maximum of 20mΩ to 50mΩ of ESR. Other losses including
Schottky conduction losses during dead time and inductor core losses generally account for less than 2% total
additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to ∆ILOAD • ESR, where ESR is the effective
series resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. The
availability of the ITH pin not only allows optimization of
control loop behavior but also provides a DC coupled and
AC filtered closed loop response test point. The DC step,
rise time and settling at this test point truly reflects the
closed loop response. Assuming a predominantly second
order system, phase margin and/or damping factor can be
estimated using the percentage of overshoot seen at this
pin. The bandwidth can also be estimated by examining the
rise time at the pin. The ITH external components shown
in the Typical Application circuit will provide an adequate
starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1μs to 10μs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking the feedback loop. Placing a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This
is why it is better to look at the ITH pin signal which is
in the feedback loop and is the filtered and compensated
control loop response. The gain of the loop will be increased by increasing RC and the bandwidth of the loop
will be increased by decreasing CC. If RC is increased by
the same factor that CC is decreased, the zero frequency
will be kept the same, thereby keeping the phase shift the
same in the most critical frequency range of the feedback
loop. The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
alter its delivery of current quickly enough to prevent this
3854fa
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LTC3854
Applications Information
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • CLOAD. Thus a 10μF capacitor would
require a 250μs rise time, limiting the charging current
to about 200mA.
1) Are the signal and power grounds segregated? The
LTC3854 GND pin should tie to the ground plane close
to the output capacitor(s). The low current or signal
ground trace should make a single point connection
directly to the GND pin. The synchronous MOSFET
source pins should connect to the input capacitor(s)
ground.
PC Board Layout Checklist
2) Does the VFB pin connect directly to the feedback resistors? The resistive divider R1, R2 must be connected
between the (+) plate of COUT and signal ground. The
47pF to 100pF capacitor should be as close as possible
to the LTC3854. Be careful locating the feedback resistors too far away from the LTC3854. The VFB line should
not be routed close to any other nodes with high slew
rates.
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3854. These items are also illustrated graphically
in the layout diagram of Figure 4. Check the following in
your layout:
+
LTC3854
TG
CSS
RC
47pF
3
CC
CC2
2
1
11
1000pF
12
RUN/SS
BOOST
ITH
VIN
VFB
INTVCC
SENSE–
BG
SENSE+
GND
6
5
M1
CIN
+
SW
4
VIN
10
DB
9
8
+
CB
D1
M2
4.7µF
7
–
L1
–
R1
+
R2
COUT
VOUT
RSENSE
3854 F04
+
Figure 4. LTC3854 Layout Diagram
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LTC3854
Applications Information
3) Are the SENSE– and SENSE+ leads routed together
with minimum PC trace spacing? The filter capacitor
between SENSE+ and SENSE– should be as close
as possible to the LTC3854. Ensure accurate current
sensing with Kelvin connections as shown in Figure 5.
Series resistance can be added to the SENSE lines to
increase noise rejection.
4) Does the (+) terminal of CIN connect to the drain of
the topside MOSFET(s) as closely as possible? This
capacitor provides the AC current to the MOSFET(s).
5) Is the INTVCC decoupling capacitor connected closely
between INTVCC and GND? This capacitor carries the
MOSFET driver peak currents.
6) Keep the switching node (SW), top gate node (TG),
bottom gate node (BG) and boost node (BOOST) away
from sensitive small-signal nodes, especially from the
voltage and current sensing feedback pins. All of these
nodes have very large and fast moving signals and
therefore should be kept on the “output side” (Pins
4,5,6 and 8) of the LTC3854 GND and occupy minimum
PC trace area.
CURRENT SENSE
RESISTOR
(RSENSE)
SENSE+ SENSE–
Figure 5. Kelvin Sensing RSENSE
3854 F04
PC Board Layout Debugging
It is helpful to use a DC-50MHz current probe to monitor
the current in the inductor while testing the circuit. Monitor
the output switching node (SW pin) to synchronize the
oscilloscope to the internal oscillator and probe the actual
output voltage as well. Check for proper performance over
the operating voltage and current range expected in the application. The frequency of operation should be maintained
over the input voltage range down to dropout.
The duty cycle percentage should be maintained from cycle
to cycle in a well-designed, low noise PCB implementation.
Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs
or inadequate loop compensation. Overcompensation of
the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. A 1Ω to 10Ω
boost resistor may help to improve noise immunity. This
resistor is placed between the BOOST pin and the node
formed by the cathode of the boost Schottky and the
positive terminal of the boost capacitor.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, Schottky and the top
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
GND pin of the IC.
3854fa
19
LTC3854
Applications Information
Design Example
Consider the design of a 1.2V, 15A buck regulator with a
VIN range of 4.5V to 28V using a DCR sensing scheme.
Inductor Selection
Assuming an inductor ripple of 40% of IOUT, L can be
calculated for the worst case of VIN = VIN(MAX).
LMIN =


V
1
• VOUT  1− OUT 
∆IL • fSW
 VIN(MAX) 
1
 1.2V 
• 1.2V •  1−

0.40 • 15A • 400kHz
20V 
LMIN = 0.47µH
LMIN =
Next, determine the DCR of the inductor. When provided,
use the manufacturer’s maximum value, usually given at
25°C. Increase this value to account for the temperature
coefficient of resistance, which is approximately 0.4%/°C. A
conservative value for TLMAX is 100°C which corresponds
to a delta of 0.3. To allow the converter to source 15A with
an inductor temperature of 100°C without hitting maximum
current limit we need a DCR at 25°C of:
DCR(25°C) =
0.8 • VSENSE(MAX)
∆IL 

 IMAX + 2  • (1+ δ)
0.8 • 50mV
DCR(25°C) =
15A • 0.4 

 15A +
 • (1+ 0.3)
2
DCR(25°C) = 1.7mΩ
The 0.56µH inductor from the IHLP4040DZ-01 series has
a typical DCR of 1.7mΩ and a maximum of 1.8mΩ and
as ISAT of 49A. The saturation current is well above our
operating current maximum.
The maximum inductor will be the DC value plus one half
the ripple current. Using this inductor gives an inductor
ripple current of 6A (keeping the ripple current high will
also help insure the minimum on-time requirement of
75ns is not violated).
tON(MIN) =
VOUT
VIN(MAX) • fSW
1.2V
20V • 400kHz
tON(MIN) = 150ns
tON(MIN) =
To choose R1 for DCR sensing we use:
R1• C1=
L
at 25°C
DCR
Choosing C1 = 100nF and using the maximum DCR value
at 25°C, we get:
0.56µH
1.8mΩ • 100nF
R1= 3.11k
R1=
Choose 3.09k.
Output Capacitor Selection
The output voltage AC ripple due to capacitive impedance
and ESR in normal continuous mode operation can be
calculated from:


1
∆VOUT = ∆IL  ESR +
8 • fSW • COUT 

The second term is the AC capacitive impedance part of
the above equation and used alone will yield a minimum
COUT of:
COUT >
∆IL
8 • fSW • ∆VOUT
0.4 • 15A
8 • 400kHz • 0.01• 1.2V
COUT > 156µF
COUT >
3854fa
20
LTC3854
Applications Information
However, the amount of capacitance needed is determined
not only by the allowed ripple in steady state but by the
maximum energy stored in the inductor. The capacitance
must be sufficient in value to absorb the change in inductor current when a high current to low current transient
occurs. The minimum capacitance to assure the inductor’s
energy is adequately absorbed during a 5A load step for
a maximum overshoot of 2% is:
L • ∆IL 2
COUT ≥
2 • ∆VOUT • VOUT
0.56µH • (5A)2
0.02 • 1.2V
COUT ≥ 583µF
COUT ≥
A maximum overshoot or undershoot of 2% for a 5A load
step will require an ESR of:
ESR < 0.02 •
VOUT
1.2V
= 0.02 •
≤ 5mΩ
∆ILOAD
5A
Several quality capacitors are available with low enough
ESR.
Multilayer ceramic capacitors tend to have very low ESR
values. It is also a good practice to reduce the ESL by putting
several capacitors in parallel on the output (a parallel bank
of larger and smaller capacitors will improve performance
in both a DC and a transient condition).
To keep ripple very low and design for any possible large
excursions in current 2x 330µF (tantalum or polymer
surface) and 1x 47µF polymer low ESR type were connected in parallel.
Choosing FB Resistors (See Figure 3)
 R 
VOUT = 0.8  1+ B 
 RA 
RB = 0.5R A
Choosing CIN Capacitors
CIN is chosen for a RMS current rating of at least IOUT(MAX)/2
= 6A. Again, keeping ESR low will improve performance
and reduce power loss (several capacitors in parallel is
once again a good choice). We will use an 180µF 25V
electrolytic with 2x 10µF 25V low ESR ceramic capacitors
connected in parallel.
Choosing MOSFETs
The power dissipation in the main and synchronous FETs
can be easily estimated. Choosing a Renesas RJK0305DPB
for the main FET results in the following parameters:
BVDSS = 30V
RDS(ON) = 13mΩ maximum at 25°C, VGS = 4.5V
QGD = 1.5nC at VDS, test 10V results in CMILLER = 1.5nC/10V
= 150pF
QG = 8nC, typical, at VGS = 4.5V
VMILLER = 2.8V
At VIN = 20V, IOUT = 15A, estimated TJ = 100°C for the
top FET and given
VINTVCC = 5.0V
RDR,PULLUP = 2.6Ω
RDR,PULLDOWN = 1.5Ω
the total losses in the main FET will be:
PMAIN =
1.2V
• (15A)2 • 1+ 0.005 • (100°C – 25°C)
20V
2 15A
• 13mΩ + ( 20V ) •
• 150pF
2
1.2Ω 
 2.5Ω
•
+
•f
 5V − 2.8V 2.8V  SW
(
)
PMAIN = 0.55W
Using 1% 10.0k for RA gives 1% 4.99k for RB.
3854fa
21
LTC3854
Applications Information
Choosing an RJK0330DPB for the bottom FET will provide:
BVDSS = 30V
RDS(ON) = 3.9mΩ maximum at 25°C, VGS = 4.5V
Given a typical QG of 8nC for the RJK0305DPB and 27nC
for the RJK0330DPB and the 400kHz switching frequency,
the current supplied by INTVCC will be:
QG = 27nC, typical, at VGS = 4.5V
PSYNC =
Some airflow may be required for higher ambient temperatures. A maximum MOSFET junction temperature of
110°C at worst case ambient generally provides adequate
margin.
20V − 1.2V
2
• (15A )
20V
• (1+ 0.005 • (100°C – 25°C)) • 3.9mΩ
IGATECHG = (8nC + 27nC) • 400kHz = 14mA
The resulting controller temperature at 60°C and a 20V
input will be:
PSYNC = 1.1W
Assuming a thermal resistance of 40°C/W for the main and
synchronous FETs, the resulting junction temperatures at
an ambient of 60°C will be 82°C and 104°C, respectively.
TJ = 60°C + 20V • 14mA • 76°C/W = 81°C
which is well under the maximum junction temperature
of 125°C.
VIN
M1
LTC3854 TG
0.1µF
RUN/SS
ITH
2200pF
3.9k
0.1µF
BOOST
D1
3.09k
0.1µF
VIN
4.5V TO 28V
10µF
25V
s2
+
INTVCC
10k
4.7µF
FB
4.99k
L
0.56µH
1.8mΩ DCR
SW
100pF
180µF
25V
SENSE–
BG
SENSE+
GND
D1 = CMDSH-3
M1 = RENESAS RJK0305DPB
M2 = RENESAS RJK0330DPB
L = VISHAY IHLP4040DZ-01 0.56µH
COUT1 = SANYO 2R5TPE330M9
COUT2 = MURATA GRM31CR60J476K
VOUT
1.2V
15A
COUT1
330µF
s2
COUT2
47µF
M2
3854 F06
Figure 6. 1.2V/15A Converter from Design Example
3854fa
22
LTC3854
Typical Applications
VIN
M1
LTC3854 TG
0.1µF
RUN/SS
ITH
2200pF
3.9k
47µF
25V
s2
0.1µF
BOOST
L
0.56µH
SW
100pF
VIN
4.5V TO 14V
180µF
25V
RSENSE
2mΩ
D1
+
INTVCC
20k
4.7µF
FB
17.4k
SENSE–
BG
SENSE+
GND
M2
VOUT
1.5V
15A
COUT1
330µF
s2
COUT2
47µF
s2
1nF
47Ω
47Ω
D1 = CMDSH-3
M1 = RENESAS RJK0305DPB
M2 = RENESAS RJK0330DPB
L = TOKO FDA1055-R56M
COUT1 = SANYO 2R5TPE330M9
COUT2 = MURATA GRM31CR60J476K
3854 TA02
Figure 7. 1.5V/15A RSENSE Application
3854fa
23
LTC3854
typical Applications
VIN
LTC3854
0.1µF
RUN/SS
2200pF
15k
0.1µF
SW
100pF
47µF
25V
s2
L
0.22µH,
1.8mΩ DCR
BOOST
ITH
180µF
25V
M1
TG
VIN
4.5V TO 14V
D1
INTVCC
+
0.1µF
1.21k
4.7µF
32.4k
FB
COUT2
47µF
s2
M2
BG
SENSE–
8.06k
SENSE+
VOUT
1V
20A
COUT1
330µF
s2
GND
3854 TA03
D1 = CMDSH-3
M1, M2 = VISHAY Si7866ADP
L = SUMIDA CDEP104NP-0R2NC
COUT1 = SANYO 2RSTPE330M9
COUT2 = MURATA GRM31CR60J476K
Figure 8. 1.0V/20A DCR Sense Application
VIN
M1
LTC3854 TG
0.1µF
RUN/SS
ITH
2200pF
7.5k
47µF
50V
s2
0.1µF
BOOST
L
3.6µH
SW
100pF
VIN
6V TO 38V
150µF
50V
RSENSE
3mΩ
D1
+
INTVCC
8.06k
4.7µF
FB
42.2k
SENSE–
BG
SENSE+
GND
VOUT
5V
10A
COUT1
220µF
M2
4.7nF
16Ω
16Ω
D1 = ZETEX ZLLS1000
M1, M2 = INFINEON BSC093N04LS
L = COILTRONICS HC1-3R6-R
COUT1 = SANYO 6TPE220MI
3854 TA04
Figure 9. 5.0V/10A RSENSE Application
3854fa
24
LTC3854
package description
DDB Package
12-Lead Plastic DFN
(Reference LTC DWG # 05-08-1723 Rev Ø)
0.64 ±0.05
(2 SIDES)
0.70 ±0.05
2.55 ±0.05
1.15 ±0.05
PACKAGE
OUTLINE
0.25 ± 0.05
0.45 BSC
2.39 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
3.00 ±0.10
(2 SIDES)
R = 0.05
TYP
R = 0.115
TYP
7
0.40 ± 0.10
12
2.00 ±0.10
(2 SIDES)
PIN 1 BAR
TOP MARK
(SEE NOTE 6)
0.200 REF
0.75 ±0.05
0.64 ± 0.10
(2 SIDES)
6
0.23 ± 0.05
0 – 0.05
PIN 1
R = 0.20 OR
0.25 × 45°
CHAMFER
1
(DDB12) DFN 0106 REV Ø
0.45 BSC
2.39 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE
3854fa
25
LTC3854
Package Description
Package
MSE MSE
Package
12-Lead
Plastic
MSOP
, Exposed
12-Lead
Plastic
MSOP,
Exposed
Die Die
PadPad
(Reference
LTC DWG
# 05-08-1666
(Reference
LTC DWG
# 05-08-1666
Rev Rev
B) B)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 p 0.102
(.112 p .004)
5.23
(.206)
MIN
2.845 p 0.102
(.112 p .004)
0.889 p 0.127
(.035 p .005)
6
1
1.651 p 0.102 3.20 – 3.45
(.065 p .004) (.126 – .136)
0.12 REF
12
0.65
0.42 p 0.038
(.0256)
(.0165 p .0015)
BSC
TYP
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
4.039 p 0.102
(.159 p .004)
(NOTE 3)
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
7
NO MEASUREMENT PURPOSE
0.406 p 0.076
(.016 p .003)
REF
12 11 10 9 8 7
DETAIL “A”
0o – 6o TYP
3.00 p 0.102
(.118 p .004)
(NOTE 4)
4.90 p 0.152
(.193 p .006)
GAUGE PLANE
0.53 p 0.152
(.021 p .006)
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.22 – 0.38
(.009 – .015)
TYP
1 2 3 4 5 6
0.650
(.0256)
BSC
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.86
(.034)
REF
0.1016 p 0.0508
(.004 p .002)
MSOP (MSE12) 0608 REV B
3854fa
26
LTC3854
Revision History
REV
DATE
DESCRIPTION
PAGE NUMBER
A
10/09
Edits to Typical Application
Updated Efficiency Graph
Edit to Electrical Characteristics and Notes
Text Changes to Pin Functions
1
1
3, 4
7
Change to Functional Diagram
8
Updated Related Parts Table
28
3854fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC3854
Related Parts
PART NUMBER DESCRIPTION
COMMENTS
LTC3851A/
LTC3851A-1
No RSENSE™ Wide VIN Range Synchronous Step-Down
DC/DC Controller
Phase-Lockable Fixed Operating Frequency, 250kHz to 750kHz, 4V ≤ VIN ≤
38V, 0.8V ≤ VOUT ≤ 5.25V, MSOP-16E, 3mm × 3mm QFN-16, SSOP-16
LTC3878
No RSENSE Constant On-Time Synchronous Step-Down
DC/DC Controller
Very Fast Transient Response, tON(MIN) = 43ns, 4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤
0.9VIN, SSOP-16
LTC3879
No RSENSE Constant On-Time Synchronous Step-Down
DC/DC Controller
Very Fast Transient Response, tON(MIN) = 43ns, 4V ≤ VIN ≤ 38V, 0.6V ≤ VOUT ≤
0.9VIN, MSOP-16E, 3mm × 3mm QFN-16
LTC3850/
LTC3850-1/
LTC3850-2
Dual 2-Phase, High Efficiency Synchronous Step-Down
DC/DC Controllers, RSENSE or DCR Current Sensing and
Tracking
Phase-Lockable Fixed Operating Frequency, 250kHz to 780kHz, 4V ≤ VIN ≤
30V, 0.8V ≤ VOUT ≤ 5.25V
LTC3853
Triple Output, Multiphase Synchronous Step-Down DC/DC Phase-Lockable Fixed Operating Frequency, 250kHz to 750kHz, 4V ≤ VIN ≤
Controller, RSENSE or DCR Current Sensing and Tracking
24V, VOUT Up to 13.5V
LTM®4600HV
10A DC/DC µModule® Complete Power Supply
High Efficiency, Compact Size, Ultrafast Transient Response, 4.5V ≤ VIN ≤
28V, 0.8V ≤ VOUT ≤ 5V, 15mm × 15mm × 2.8mm
LTM4601AHV
12A DC/DC µModule Complete Power Supply
High Efficiency, Compact Size, Ultrafast Transient Response, 4.5V ≤ VIN ≤
28V, 0.8V ≤ VOUT ≤ 5V, 15mm × 15mm × 2.8mm
LTC3601
1.5A, 4MHz, Monolithic Synchronous Step-Down DC/DC
Converter
High Efficiency, Phase Lockable, IQ = 300µA, 4V ≤ VIN ≤ 15V, VOUT(MIN) 0.6V,
3mm × 3mm QFN-16, MSOP-16E
LTC3603
2.5A, 3MHz, Monolithic Synchronous Step-Down DC/DC
Converter
High Efficiency, Phase Lockable, IQ = 75µA, 4.5V ≤ VIN ≤ 15V, VOUT(MIN) 0.6V,
4mm × 4mm QFN-20
LTC3605
5A, 4MHz, Monolithic Synchronous Step-Down DC/DC
Converter
High Efficiency, Adjustable Frequency, 800kHz to 4MHz, 4V ≤ VIN ≤ 15V,
VOUT(MIN) 0.6V, 4mm × 4mm QFN-24
LTC3608
8A, 1MHz, Monolithic Synchronous Step-Down DC/DC
Converter
High Efficiency, Adjustable Constant On-Time, 4V ≤ VIN ≤ 18V, VOUT(MIN) 0.6V,
7mm × 8mm QFN-52
LTC3609
6A, 1MHz, Monolithic Synchronous Step-Down DC/DC
Converter
High Efficiency, Adjustable Constant On-Time, 4V ≤ VIN ≤ 32V, VOUT(MIN) 0.6V,
7mm × 8mm QFN-52
LTC3610
12A, 1MHz, Monolithic Synchronous Step-Down DC/DC
Converter
High Efficiency, Adjustable Constant On-Time, 4V ≤ VIN ≤ 24V, VOUT(MIN) 0.6V,
9mm × 9mm QFN-64
LTC3611
10A, 1MHz, Monolithic Synchronous Step-Down DC/DC
Converter
High Efficiency, Adjustable Constant On-Time, 4V ≤ VIN ≤ 32V, VOUT(MIN) 0.6V,
9mm × 9mm QFN-64
LTC3824
Low IQ, High Voltage DC/DC Controller, 100% Duty Cycle
Selectable Fixed Operating Frequency, 4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ VIN,
IQ = 40µA, MSOP-10E
LTC3834/
LTC3834-1
Low IQ, Synchronous Step-Down DC/DC Controller
Phase-Lockable Fixed Operating Frequency, 140kHz to 650kHz, 4V ≤ VIN ≤
36V, 0.8V ≤ VOUT ≤ 10V, IQ = 30µA,
LT®3845
Low IQ, High Voltage Synchronous Step-Down DC/DC
Controller
Adjustable Fixed Operating Frequency, 100kHz to 500kHz, 4V ≤ VIN ≤ 60V,
1.23V ≤ VOUT ≤ 36V, IQ = 30µA, TSSOP-16
µModule is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation.
3854fa
28 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
LT 1209 REV A • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2009
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