LINER LTC3780IG High efficiency, synchronous, 4-switch buck-boost controller Datasheet

LTC3780
High Efficiency, Synchronous,
4-Switch Buck-Boost Controller
DESCRIPTIO
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FEATURES
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Single Inductor Architecture Allows VIN Above,
Below or Equal to VOUT
Wide VIN Range: 4V to 36V Operation
Synchronous Rectification: Up to 98% Efficiency
Current Mode Control
±1% Output Voltage Accuracy: 0.8V < VOUT < 30V
Phase-Lockable Fixed Frequency: 200kHz to 400kHz
Power Good Output Voltage Monitor
Internal LDO for MOSFET Supply
Quad N-Channel MOSFET Synchronous Drive
VOUT Disconnected from VIN During Shutdown
Adjustable Soft-Start Current Ramping
Foldback Output Current Limiting
Selectable Low Current Modes
Output Overvoltage Protection
Available in 24-Lead SSOP and Exposed Pad
(5mm × 5mm) 32-Lead QFN Packages
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APPLICATIO S
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The operating mode of the controller is determined through
the FCB pin. For boost operation, the FCB mode pin can
select among Burst Mode® operation, Discontinuous mode
and Forced Continuous mode. During buck operation, the
FCB mode pin can select among Skip-Cycle mode, Discontinuous mode and Forced Continuous mode. Burst Mode
operation and Skip-Cycle mode provide high efficiency
operation at light loads while Forced Continuous mode
and Discontinuous mode operate at a constant frequency.
Fault protection is provided by an output overvoltage
comparator and internal foldback current limiting. A Power
Good output pin indicates when the output is within 7.5%
of its designed set point.
Automotive Systems
Telecom Systems
DC Power Distribution Systems
High Power Battery-Operated Devices
Industrial Control
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 5481178, 6304066, 5929620, 5408150, 6580258,
patent pending on current mode architecture and protection
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The LTC®3780 is a high performance buck-boost switching regulator controller that operates from input voltages
above, below or equal to the output voltage. The constant
frequency current mode architecture allows a phaselockable frequency of up to 400kHz. With a wide 4V to 30V
(36V maximum) input and output range and seamless
transfers between operating modes, the LTC3780 is ideal
for automotive, telecom and battery-powered systems.
TYPICAL APPLICATIO
High Efficiency Buck-Boost Converter
22µF
50V
CER
+
4.7µF
1µF
CER
VIN PGOOD INTVCC
TG2
0.1µF
100µF
16V
CER
VOUT
12V
5A
Efficiency and Power Loss
VOUT = 12V, ILOAD = 5A
TG1
BOOST2
BOOST1
SW2
10
100
0.1µF
9
95
SW1
8
2200pF
20k
BG1
ITH
PLLIN
SS
RUN
ON/OFF
VOSENSE
0.1µF
SGND
FCB
SENSE+ SENSE– PGND
105k
1%
7.5k
7
90
6
5
85
4
80
3
POWER LOSS (W)
LTC3780
BG2
EFFICIENCY (%)
VIN
4V TO 36V
2
75
1
1000pF
70
0.010Ω
0
0
2µH
5
10
20
15
VIN (V)
25
30
35
3780 TA01b
3780 TA01
3780f
1
LTC3780
W W
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ABSOLUTE MAXIMUM RATINGS (Note 1)
Input Supply Voltage (VIN)........................ –0.3V to 36V
Topside Driver Voltages
(BOOST1, BOOST2) .................................. –0.3V to 42V
Switch Voltage (SW1, SW2) ........................ –5V to 36V
INTVCC, EXTVCC, RUN, SS, (BOOST – SW1),
(BOOST2 – SW2), PGOOD .......................... –0.3V to 7V
PLLIN Voltage .......................................... –0.3V to 5.5V
PLLFLTR Voltage ...................................... –0.3V to 2.7V
FCB, STBYMD Voltages ....................... –0.3V to INTVCC
ITH, VOSENSE Voltages .............................. –0.3V to 2.4V
Peak Output Current <10ms (TG1, TG2, BG1, BG2) .. 3A
INTVCC Peak Output Current ................................ 40mA
Operating Temperature Range (Note 7)
LTC3780E ........................................... – 40°C to 85°C
LTC3780I ............................................ – 40°C to 85°C
Junction Temperature (Note 2) ............................ 125°C
Storage Temperature Range .................. –65°C to 125°C
Lead Temperature (Soldering, 10 sec)
SSOP Only ........................................................ 300°C
W
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PACKAGE/ORDER INFORMATION
ORDER PART
NUMBER
NC
TG1
BOOST1
NC
NC
SS
NC
TOP VIEW
PGOOD
TOP VIEW
ORDER PART
NUMBER
PGOOD
1
24 BOOST1
SS
2
23 TG1
SENSE+
3
22 SW1
–
4
21 VIN
ITH
5
20 EXTVCC
VOSENSE 4
VOSENSE
6
19 INTVCC
SGND 5
SGND
7
18 BG1
RUN 6
19 PGND
RUN
8
17 PGND
FCB 7
18 BG2
FCB
9
16 BG2
PLLFTR 8
17 SW2
PLLFLTR 10
15 SW2
PLLIN 11
14 TG2
24 SW1
SENSE– 2
23 VIN
ITH 3
22 EXTVCC
21 INTVCC
33
20 BG1
UH PART
MARKING
3780
3780I
NC
TG2
BOOST2
NC
NC
STBYMD
NC
9 10 11 12 13 14 15 16
13 BOOST2
STBYMD 12
LTC3780EUH
LTC3780IUH
32 31 30 29 28 27 26 25
SENSE+ 1
PLLIN
SENSE
LTC3780EG
LTC3780IG
UH PACKAGE
32-LEAD (5mm × 5mm) PLASTIC QFN
G PACKAGE
24-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 130°C/W
TJMAX = 125°C, θJA = 34°C/W
EXPOSED PAD (PIN 33) IS SGND
(MUST BE SOLDERED TO PCB)
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● indicates specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
0.792
0.800
0.808
V
–5
–50
nA
0.1
–0.1
0.5
–0.5
%
%
Main Control Loop
VOSENSE
Feedback Reference Voltage
ITH = 1.2V (Note 3)
IVOSENSE
Feedback Pin Input Current
(Note 3)
VLOADREG
Output Voltage Load Regulation
(Note 3)
∆ITH = 1.2V to 0.7V
∆ITH = 1.2V to 1.8V
●
●
●
3780f
2
LTC3780
ELECTRICAL CHARACTERISTICS
The ● indicates specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VREF(LINEREG)
Reference Voltage Line Regulation
gm(EA)
Error Amplifier Transconductance
gm(GBW)
Error Amplifier GBW
IQ
Input DC Supply Current
Normal
Standby
Shutdown Supply Current
MIN
TYP
MAX
UNITS
VIN = 4V to 30V, ITH = 1.2V (Note 3)
0.002
0.02
%/V
ITH = 1.2V, Sink/Source = 3µA (Note 3)
0.32
mS
0.6
MHz
(Note 4)
2400
1500
55
70
µA
µA
µA
0.76
0.800
0.84
V
–0.30
–0.18
–0.1
µA
5.3
5.5
V
VRUN = 0V, VSTBYMD > 2V
VRUN = 0V, VSTBYMD = Open
VFCB
Forced Continuous Threshold
IFCB
Forced Continuous Pin Current
VFCB = 0.85V
VBINHIBIT
Burst Inhibit (Constant Frequency)
Threshold
Measured at FCB Pin
UVLO
Undervoltage Reset
VIN Falling
VOVL
Feedback Overvoltage Lockout
Measured at VOSENSE Pin
ISENSE
Sense Pins Total Source Current
VSENSE– = VSENSE+ = 0V
VSTBYMD(START)
Start-Up Threshold
VSTBYMD Rising
VSTBYMD(KA)
Keep-Alive Power-On Threshold
VSTBYMD Rising, VRUN = 0V
DF MAX, BOOST Maximum Duty Factor
% Switch C On
DF MAX, BUCK
Maximum Duty Factor
% Switch A On (in Dropout)
VRUN(ON)
RUN Pin On Threshold
VRUN Rising
ISS
Soft-Start Charge Current
VRUN = 2V
VSENSE(MAX)
Maximum Current Sense Threshold
Boost: VOSENSE = VREF – 50mV
Buck: VOSENSE = VREF – 50mV
●
0.84
3.8
4
V
0.86
0.88
V
–380
0.4
µA
0.7
V
1.25
V
99
%
99
●
●
1
1.5
0.5
1.2
–95
160
–130
%
2
V
µA
185
–150
mV
mV
VSENSE(MIN,BUCK) Minimum Current Sense Threshold
Discontinuous Mode
–6
mV
TG1, TG2 tr
TG Rise Time
CLOAD = 3300pF (Note 5)
50
ns
TG1, TG2 tf
TG Fall Time
CLOAD = 3300pF (Note 5)
45
ns
BG1, BG2 tr
BG Rise Time
CLOAD = 3300pF (Note 5)
45
ns
BG1, BG2 tf
BG Fall Time
CLOAD = 3300pF (Note 5)
55
ns
TG1/BG1 t1D
TG1 Off to BG1 On Delay,
Switch C On Delay
CLOAD = 3300pF Each Driver
80
ns
BG1/TG1 t2D
BG1 Off to TG1 On Delay,
Synchronous Switch D On Delay
CLOAD = 3300pF Each Driver
80
ns
TG2/BG2 t3D
TG2 Off to BG2 On Delay,
Synchronous Switch B On Delay
CLOAD = 3300pF Each Driver
80
ns
BG2/TG2 t4D
BG2 Off to TG2 On Delay,
Switch A On Delay
CLOAD = 3300pF Each Driver
80
ns
Mode
Transition 1
BG1 Off to BG2 On Delay,
Switch A On Delay
CLOAD = 3300pF Each Driver
90
ns
Mode
Transition 2
BG2 Off to BG1 On Delay,
Synchronous Switch D On Delay
CLOAD = 3300pF Each Driver
90
ns
3780f
3
LTC3780
ELECTRICAL CHARACTERISTICS
The ● indicates specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 15V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
tON(MIN,BOOST)
Minimum On-Time for Main Switch in
Boost Operation
tON(MIN,BUCK)
Minimum On-Time for Synchronous
Switch in Buck Operation
MIN
TYP
MAX
UNITS
Switch C (Note 6)
200
240
ns
Switch B (Note 6)
180
220
ns
6
6.3
V
0.2
2
%
Internal VCC Regulator
VINTVCC
Internal VCC Voltage
7V < VIN < 30V, VEXTVCC = 5V
∆VLDO(LOADREG)
Internal VCC Load Regulation
ICC = 0mA to 20mA, VEXTVCC = 5V
VEXTVCC
EXTVCC Switchover Voltage
ICC = 20mA, VEXTVCC Rising
∆VEXTVCC(HYS)
EXTVCC Switchover Hysteresis
∆VEXTVCC
EXTVCC Switch Drop Voltage
●
●
5.7
5.4
5.7
V
200
ICC = 20mA, VEXTVCC = 6V
150
mV
300
mV
Oscillator and Phase-Locked Loop
fNOM
Nominal Frequency
VPLLFLTR = 1.2V
260
300
330
kHz
fLOW
Lowest Frequency
VPLLFLTR = 0V
170
200
220
kHz
fHIGH
Highest Frequency
VPLLFLTR = 2.4V
340
400
440
kHz
RPLLIN
PLLIN Input Resistance
IPLLLPF
Phase Detector Output Current
fPLLIN < fOSC
fPLLIN > fOSC
∆VFBH
PGOOD Upper Threshold
VOSENSE Rising
5.5
7.5
10
%
∆VFBL
PGOOD Lower Threshold
VOSENSE Falling
–5.5
–7.5
–10
%
∆VFB(HYST)
PGOOD Hysteresis
VOSENSE Returning
2.5
VPGL
PGOOD Low Voltage
IPGOOD = 2mA
0.1
IPGOOD
PGOOD Leakage Current
VPGOOD = 5V
50
kΩ
–15
15
µA
µA
PGOOD Output
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: TJ for the QFN package is calculated from the temperature TA and
power dissipation PD according to the following formula:
TJ = TA + (PD • 34°C/W)
Note 3: The IC is tested in a feedback loop that servos VITH to a specified
voltage and measures the resultant VOSENSE.
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
%
0.3
V
±1
µA
Note 5: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 6: The minimum on-time condition is specified for an inductor peakto-peak ripple current ≥ 40% of IMAX (see minimum on-time
considerations in the Applications Information section).
Note 7: The LTC3780E is guaranteed to meet performance specifications
from 0°C to 85°C. Performance over the –40°C to 85°C operating
temperature range is assured by design, characterization and correlation
with statistical process controls. The LTC3780I is guaranteed and tested
over the – 40°C to 85°C operating temperature range.
3780f
4
LTC3780
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Output Current
(Boost Operation)
TA = 25°C unless otherwise noted.
Efficiency vs Output Current
(Buck Operation)
Efficiency vs Output Current
100
100
100
BURST
BURST
90
90
90
80
80
70
CCM
60
DCM
70
EFFICIENCY (%)
SC
80
EFFICIENCY (%)
EFFICIENCY (%)
DCM
CCM
60
50
50
VIN = 6V
VOUT = 12V
40
0.01
0.1
1
0.1
1
VIN = 18V
VOUT = 12V
40
0.01
10
0.1
ILOAD (A)
ILOAD (A)
EXTVCC Voltage Drop
INTVCC VOLTAGE (V)
1000
500
6.5
120
6.0
100
EXTVCC VOLTAGE DROP (mV)
VFCB = 0V
2000
STANDBY
10
3780 G03
Internal 6V LDO Line Regulation
Supply Current vs Input Voltage
2500
1500
1
ILOAD (A)
3780 G02
3780 G01
SUPPLY CURRENT (µA)
DCM
60
50
VIN = 12V
VOUT = 12V
40
0.01
10
CCM
70
5.5
5.0
4.5
4.0
80
60
40
20
SHUTDOWN
0
3.5
0
0
5
20
15
10
25
INPUT VOLTAGE (V)
30
0
35
5
20
15
25
10
INPUT VOLTAGE (V)
INTVCC and EXTVCC Switch
Voltage vs Temperature
5.90
5.85
5.80
5.75
5.70
EXTVCC SWITCHOVER THRESHOLD
VIN = 18V
4
–0.1
3
2
1
–0.2
VIN = 12V
–0.3
VIN = 6V
–0.4
FCB = 0V
VOUT = 12V
5.60
5.55
–50 –25
50
25
0
75
TEMPERATURE (°C)
100
125
3780 G07
50
40
0
NORMALIZED VOUT (%)
5.95
EXTVCC SWITCH RESISTANCE (Ω)
INTVCC VOLTAGE
20
30
CURRENT (mA)
Load Regulation
5
6.00
10
3780 G06
EXTVCC Switch Resistance
vs Temperature
6.05
INTVCC AND EXTVCC SWITCH VOLTAGE (V)
1
35
3780 G05
3780 G04
5.65
30
0
–50
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3780 G08
–0.5
0
1
3
2
LOAD CURRENT (A)
4
5
3780 G09
3780f
5
LTC3780
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Discontinuous Current Mode
(DCM, VIN = 6V, VOUT = 12V)
Continuous Current Mode
(CCM, VIN = 12V, VOUT = 12V)
SW2
10V/DIV
SW2
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
VOUT
100mV/DIV
IL
2A/DIV
VIN = 6V
VOUT = 12V
5µs/DIV
IL
2A/DIV
3780 G10
VIN = 12V
VOUT = 12V
5µs/DIV
3780 G11
VIN = 18V
VOUT = 12V
Burst Mode Operation
(VIN = 12V, VOUT = 12V)
SW1
10V/DIV
SW1
10V/DIV
VOUT
500mV/DIV
VOUT
200mV/DIV
IL
2A/DIV
IL
2A/DIV
SW1
10V/DIV
SW1
10V/DIV
VOUT
100mV/DIV
IL
1A/DIV
10µs/DIV
3780 G14
VIN = 18V
VOUT = 12V
SW2
10V/DIV
SW1
10V/DIV
SW1
10V/DIV
VOUT
100mV/DIV
IL
1A/DIV
IL
2A/DIV
3780 G16
2.5µs/DIV
3780 G15
Discontinuous Current Mode
(DCM, VIN = 18V, VOUT = 12V)
SW2
10V/DIV
VOUT
100mV/DIV
3780 G12
Skip Cycle Mode
(VIN = 18V, VOUT = 12V)
Discontinuous Current Mode
(DCM, VIN = 12V, VOUT = 12V)
SW2
10V/DIV
5µs/DIV
SW2
10V/DIV
VIN = 12V
VOUT = 12V
3780 G13
Discontinuous Current Mode
(DCM, VIN = 6V, VOUT = 12V)
5µs/DIV
SW1
10V/DIV
IL
2A/DIV
SW2
10V/DIV
VIN = 6V
VOUT = 12V
SW2
10V/DIV
VOUT
100mV/DIV
SW2
10V/DIV
25µs/DIV
Continuous Current Mode
(CCM, VIN = 18V, VOUT = 12V)
VOUT
100mV/DIV
Burst Mode Operation
(VIN = 6V, VOUT = 12V)
VIN = 6V
VOUT = 12V
TA = 25°C unless otherwise noted.
VOUT
100mV/DIV
IL
1A/DIV
VIN = 12V
VOUT = 12V
5µs/DIV
3780 G17
VIN = 18V
VOUT = 12V
2.5µs/DIV
3780 G18
3780f
6
LTC3780
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Oscillator Frequency
vs Temperature
Undervoltage Reset
vs Temperature
Minimum Current Sense
Threshold vs Duty Factor (Buck)
4.0
450
400
–20
VPLLFLTR = 1.2V
300
250
VPLLFLTR = 0V
200
150
100
3.8
ISENSE+ (mV)
UNDERVOLTAGE RESET (V)
VPLLFLTR = 2.4V
350
FREQUENCY (kHz)
TA = 25°C unless otherwise noted.
3.6
3.4
–40
–60
3.2
50
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
3.0
–50 –25
125
100
50
25
0
75
TEMPERATURE (°C)
–80
100
125
100
60
40
DUTY FACTOR (%)
3780 G20
3780 G19
200
MAXIMUM ISNESE+ THRESHOLD (mV)
BOOST
ISNESE+ (mV)
160
140
130
120
120
150
100
50
0
–50
–100
–150
–50 –25
110
20
60
40
DUTY FACTOR (%)
80
100
0
20
40
60
DUTY FACTOR (%)
80
100
BUCK
50
25
75
0
TEMPERATURE (°C)
3780 G23
3780 G22
Peak Current Threshold
vs VITH (Boost)
100
125
3780 G24
Valley Current Threshold
vs VITH (Buck)
100
200
150
50
100
ISENSE+ (mV)
0
ISENSE+ (mV)
100
0
Minimum Current Sense
Threshold vs Temperature
140
180
20
3780 G21
Maximum Current Sense
Threshold vs Duty Factor (Buck)
Maximum Current Sense
Threshold vs Duty Factor (Boost)
ISENSE+ (mV)
80
50
0
–50
0
–100
–50
–100
0
0.4
0.8
1.2
1.6
VITH (V)
1.8
2.4
3780 G25
–150
0
0.4
0.8
1.2
1.6
VITH (V)
2.0
2.4
3780 G26
3780f
7
LTC3780
U W
TYPICAL PERFOR A CE CHARACTERISTICS
Load Step
TA = 25°C unless otherwise noted.
Load Step
VOUT
500mV/DIV
VOUT
500mV/DIV
IL
5A/DIV
IL
5A/DIV
3780 G27
VIN = 18V
200µs/DIV
VOUT = 12V
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
Load Step
VOUT
500mV/DIV
IL
5A/DIV
VIN = 12V
200µs/DIV
VOUT = 12V
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
Line Transient
3780 G28
VIN = 6V
200µs/DIV
VOUT = 12V
LOAD STEP: 0A TO 5A
CONTINUOUS MODE
3780 G29
Line Transient
VIN
10V/DIV
VIN
10V/DIV
VOUT
500mV/DIV
VOUT
500mV/DIV
IL
1A/DIV
IL
1A/DIV
VOUT = 12V
500µs/DIV
ILOAD = 1A
VIN STEP: 7V TO 20V
CONTINUOUS MODE
U
U
U
PI FU CTIO S
3780 G30
VOUT = 12V
500µs/DIV
ILOAD = 1A
VIN STEP: 20V TO 7V
CONTINUOUS MODE
3780 G31
(SSOP/QFN)
PGOOD (Pin 1/Pin 30): Open-Drain Logic Output. PGOOD
is pulled to ground when the output voltage is not within
±7.5% of the regulation point.
voltage and built-in offsets between SENSE– and SENSE+
pins, in conjunction with RSENSE, set the current trip
threshold.
SS (Pin 2/Pin 31): Soft-start reduces the input power
sources’ surge currents by gradually increasing the
controller’s current limit. A minimum value of 6.8nF is
recommended on this pin.
SENSE– (Pin 4/Pin 2): The (–) Input to the Current Sense
and Reverse Current Detect Comparators.
SENSE+ (Pin 3/Pin 1): The (+) Input to the Current Sense
and Reverse Current Detect Comparators. The ITH pin
ITH (Pin 5/Pin 3): Current Control Threshold and Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges
from 0V to 2.4V.
3780f
8
LTC3780
U
U
U
PI FU CTIO S
(SSOP/QFN)
VOSENSE (Pin 6/Pin 4): Error Amplifier Feedback Input. This
pin connects the error amplifier input to an external resistor divider from VOUT.
SGND (Pin 7/Pin 5): Signal Ground. All small-signal components and compensation components should connect to
this ground, which should be connected to PGND at a single
point.
RUN (Pin 8/Pin 6): Run Control Input. Forcing the RUN pin
below 1.5V causes the IC to shut down the switching regulator circuitry. There is a 100k resistor between the RUN pin
and SGND in the IC. Do not apply >6V to this pin.
FCB (Pin 9/Pin 7): Forced Continuous Control Input. The
voltage applied to this pin sets the operating mode of the
controller. When the applied voltage is less than 0.8V, the
forced continuous current mode is active. When this pin
is allowed to float, the burst mode is active in boost
operation and the skip cycle mode is active in buck
operation. When the pin is tied to INTVCC, the constant
frequency discontinuous current mode is active in buck or
boost operation.
PLLFLTR (Pin 10/Pin 8): The Phase-Locked Loop’s Lowpass Filter is Tied to This Pin. Alternatively, this pin can be
driven with an AC or DC voltage source to vary the frequency
of the internal oscillator.
PLLIN (Pin 11/Pin 10): External Synchronization Input to
Phase Detector. This pin is internally terminated to SGND
with 50kΩ. The phase-locked loop will force the rising
bottom gate signal of the controller to be synchronized with
the rising edge of the PLLIN signal.
STBYMD (Pin 12/Pin 11): LDO Control Pin. Determines
whether the internal LDO remains active when the controller is shut down. See Operation section for details. If the
STBYMD pin is pulled to ground, the SS pin is internally
pulled to ground, preventing start-up and thereby providing a single control pin for turning off the controller.
Decouple this pin with 0.1µF if not tied to a DC potential.
BOOST2, BOOST1 (Pins 13, 24/Pins 14, 27): Boosted
Floating Driver Supply. The (+) terminal of the bootstrap capacitor CA and CB (Figure 11) connects here. The BOOST2
pin swings from a diode voltage below INTVCC up to VIN +
INTVCC. The BOOST1 pin swings from a diode voltage below
INTVCC up to VOUT + INTVCC.
TG2, TG1 (Pins 14, 23/Pins 15, 26): Top Gate Drive. Drives
the top N-channel MOSFET with a voltage swing equal to
INTVCC superimposed on the switch node voltage SW.
SW2, SW1 (Pins 15, 22/Pins 17, 24): Switch Node. The
(–) terminal of the bootstrap capacitor CA and CB (Figure 11)
connects here. The SW2 pin swings from a Schottky diode
(external) voltage drop below ground up to VIN. The SW1
pin swings from a Schottky diode (external) voltage drop
below ground up to VOUT.
BG2, BG1 (Pins 16, 18/Pins 18, 20): Bottom Gate Drive.
Drives the gate of the bottom N-channel MOSFET between
ground and INTVCC.
PGND (Pin 17/Pin 19): Power Ground. Connect this pin
closely to the source of the bottom N-channel MOSFET, the
(–) terminal of CVCC and the (–) terminal of CIN (Figure 11).
INTVCC (Pin 19/Pin 21): Internal 6V Regulator Output. The
driver and control circuits are powered from this voltage.
Decouple this pin to ground with a minimum of 4.7µF low
ESR tantalum or ceramic capacitor.
EXTVCC (Pin 20/Pin 22): External VCC Input. When EXTVCC
exceeds 5.7V, an internal switch connects this pin to INTVCC
and shuts down the internal regulator so that the controller
and gate drive power is drawn from EXTVCC. Do not exceed
7V at this pin and ensure that EXTVCC < VIN.
VIN (Pin 21/Pin 23): Main Input Supply. Decouple this pin
to SGND with an RC filter (1Ω, 0.1µF).
Exposed Pad (Pin 33, QFN Only): This pin is SGND and
must be soldered to PCB ground.
3780f
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LTC3780
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BLOCK DIAGRA
INTVCC
VIN
BOOST2
STBYMD
FCB
+
TG2
FCB
ILIM
BUCK
LOGIC
SW2
INTVCC
–
BG2
RSENSE
+
PGND
IREV
BG1
–
FCB
BOOST
LOGIC
1.2V
4(VFB)
+
SW1
TG1
ICMP
BOOST1
1.2µA
OV
–
SS
INTVCC
–
0.86V
INTVCC
+
VOUT
RUN
SLOPE
EA
100k
VOSENSE
–
+
VFB
0.80V
ITH
SHDN
RST
4(VFB)
RUN/
SS
SENSE+
SENSE–
PLLIN
VREF
VIN
VIN
50k
5.7V
+
–
EXTVCC
PLLFLTR
CLK
6V
LDO
REG
0.86V
6V
+
INTVCC
SGND
FIN
PHASE DET
RLP
OSCILLATOR
CLP
–
+
PGOOD
INTERNAL
SUPPLY VOSENSE
–
0.74V
+
3780 BD
3780f
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SS voltage while CSS is slowly charged during start-up.
This “soft-start” clamping prevents abrupt current from
being drawn from the input power supply.
MAIN CONTROL LOOP
The LTC3780 is a current mode controller that provides an
output voltage above, equal to or below the input voltage.
The LTC proprietary topology and control architecture
employs a current-sensing resistor in Buck or Boost modes.
The sensed inductor current is controlled by the voltage on
the ITH pin, which is the output of the amplifier EA. The
VOSENSE pin receives the voltage feedback signal, which is
compared to the internal reference voltage by the EA.
POWER SWITCH CONTROL
Figure 1 shows a simplified diagram of how the four power
switches are connected to the inductor, VIN, VOUT and
GND. Figure 2 shows the regions of operation for the
LTC3780 as a function of duty cycle D. The power switches
are properly controlled so the transfer between modes is
continuous. When VIN approaches VOUT, the Buck-Boost
region is reached; the mode-to-mode transition time is
typically 200ns.
The top MOSFET drivers are biased from floating booststrap
capacitors CA and CB (Figure 11), which are normally
recharged through an external diode when the top MOSFET
is turned off. Schottky diodes across the synchronous
switch D and synchronous switch B are not required, but
provide a lower drop during the dead time. The addition of
the Schottky diodes will typically improve peak efficiency
by 1% to 2% at 400kHz.
Buck Region (VIN > VOUT)
Switch D is always on and Switch C is always off during
this mode. At the start of every cycle, Synchronous Switch
B is turned on first. Inductor current is sensed when
Synchronous Switch B is turned on. After the sensed
inductor current falls below the reference voltage, which is
proportional to VITH, Synchronous Switch B is turned off
The main control loop is shut down by pulling the RUN pin
low. When the RUN pin voltage is higher than 1.5V, an
internal 1.2µA current source charges soft-start capacitor
CSS at the SS pin. The ITH voltage is then clamped to the
VIN
TG2
VOUT
A
SW2
BG2
D
L
TG1
SW1
B
C
BG1
RSENSE
3780 F01
Figure 1. Simplified Diagram of the Output Switches
98%
DMAX
BOOST
DMIN
BOOST
DMAX
BUCK
3%
DMIN
BUCK
A ON, B OFF
PWM C, D SWITCHES
BOOST REGION
FOUR SWITCH PWM
BUCK/BOOST REGION
D ON, C OFF
PWM A, B SWITCHES
BUCK REGION
3780 F02
Figure 2. Operating Mode vs Duty Cycle
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and Switch A is turned on for the remainder of the cycle.
Switches A and B will alternate, behaving like a typical
synchronous buck regulator. The duty cycle of switch A
increases until the maximum duty cycle of the converter in
Buck mode reaches DMAX_BUCK, given by:
CLOCK
SWITCH A
SWITCH B
SWITCH C
DMAX_BUCK = (1 – DBUCK-BOOST) • 100%
SWITCH D
where DBUCK-BOOST = duty cycle of the Buck-Boost switch
range:
IL
3780 F04a
(4a) Buck-Boost Mode (VIN ≥ VOUT)
DBUCK-BOOST = (200ns • f) • 100%
and f is the operating frequency in Hz.
Figure 3 shows typical Buck mode waveforms. If VIN
approaches VOUT, the Buck-Boost region is reached.
CLOCK
SWITCH A
SWITCH B
CLOCK
SWITCH C
SWITCH A
SWITCH D
SWITCH B
0V
SWITCH C
2.4V
SWITCH D
I
IL
3780 F04b
(4b) Buck-Boost Mode (VIN ≤ VOUT)
Figure 4. Buck-Boost Mode
3780 F03
Figure 3. Buck Mode (VIN > VOUT)
Buck-Boost (VIN ≅ VOUT)
Boost Region (VIN < VOUT)
When VIN is close to VOUT, the controller is in Buck-Boost
mode. Figure 4 shows typical waveforms in this mode.
Every cycle, if the controller starts with Switches B and D
turned on, Switches A and C are then turned on. Finally,
Switches A and D are turned on for the remainder of the
time. If the controller starts with Switches A and C turned
on, Switches B and D are then turned on. Finally, Switches
A and D are turned on for the remainder of the time.
Switch A is always on and Synchronous Switch B is always
off in Boost mode. Every cycle, Switch C is turned on first.
Inductor current is sensed when Synchronous Switch C is
turned on. After the sensed inductor current exceeds the
reference voltage which is proportional to VITH, Switch C
is turned off and Synchronous Switch D is turned on for
the remainder of the cycle. Switches C and D will alternate,
behaving like a typical synchronous boost regulator.
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The duty cycle of Switch C decreases until the minimum
duty cycle of the converter in Buck mode reaches
DMIN_BOOST, given by:
DMIN_BOOST = (DBUCK-BOOST) • 100%
where DBUCK-BOOST is the duty cycle of the Buck-Boost
switch range:
DBUCK-BOOST = (200ns • f) • 100%
and f is the operating frequency in Hz.
Figure 5 shows typical boost mode waveforms. If VIN
approaches VOUT, the Buck-Boost region is reached.
CLOCK
2.4V
SWITCH A
0V
SWITCH B
SWITCH C
SWITCH D
I
3780 F05
Figure 5. Boost Mode (VIN < VOUT)
LOW CURRENT OPERATION
The FCB pin is a multifunction pin providing two functions:
1) to provide regulation for a secondary winding by
temporarily forcing continuous PWM operation in Buck
mode and 2) to select among three modes for both buck
and boost operations by accepting a logic input. Figure 6
shows the different modes.
FCB PIN
BUCK MODE
BOOST MODE
0V to 0.75V
Force Continuous Mode
Force Continuous Mode
0.85V to 5V
Skip-Cycle Mode
Burst Mode Operation
>5.3V
DCM with Constant Freq
DCM with Constant Freq
Figure 6. Different Operating Modes
When the FCB pin voltage is lower than 0.8V, the controller behaves as a continuous, PWM current mode synchronous switching regulator. In Boost mode, Switch A is
always on. Switch C and Synchronous Switch D are
alternately turned on to maintain the output voltage
independent of direction of inductor current. Every ten
cycles, Switch A is forced off for about 300ns to allow CA
to recharge. In Buck mode, Synchronous Switch D is
always on. Switch A and Synchronous Switch B are
alternately turned on to maintain the output voltage independent of direction of inductor current. Every ten cycles,
Synchronous Switch D is forced off for about 300ns to
allow CB to recharge. This is the least efficient operating
mode at light load, but may be desirable in certain applications. In this mode, the output can source or sink
current. The sunk current will be forced back into the main
power supply potentially boosting the input supply to
dangerous voltage levels—BEWARE!
When the FCB pin voltage is below VINTVCC – 1V, but
greater than 0.8V, the controller enters Burst Mode operation in Boost operation or enters Skip-Cycle mode in Buck
operation. During Boost operation, Burst Mode operation
sets a minimum output current level before inhibiting the
switch C and turns off Synchronous Switch D when the
inductor current goes negative. This combination of requirements will, at low currents, force the ITH pin below a
voltage threshold that will temporarily inhibit turn-on of
power switches C and D until the output voltage drops.
There is 100mV of hysteresis in the burst comparator tied
to the ITH pin. This hysteresis produces output signals to
the MOSFETs C and D that turn them on for several cycles,
followed by a variable “sleep” interval depending upon the
load current. The maximum output voltage ripple is limited
to 3% of the nominal DC output voltage as determined by
a resistive feedback divider. During buck operation, SkipCycle mode sets a minimum positive inductor current
level. When inductor current is lower than this level,
Synchronous Switch B is kept off. In every cycle, the body
3780f
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diode of Synchronous Switch B or the Schottky diode,
which is in parallel in with Synchronous Switch B, is used
to discharge inductor current. As a result, some cycles will
be skipped when the output load current drops below 1%
of the maximum designed load in order to maintain the
output voltage.
When the FCB pin voltage is tied to the INTVCC pin, the
controller enters constant frequency Discontinuous Current mode (DCM). For Boost operation, Synchronous
Switch D is held off whenever the ITH pin is below a
threshold voltage. In every cycle, Switch C is used to
charge inductor current. After the output voltage is high
enough, the controller will enter continuous current Buck
mode for one cycle to discharge inductor current. In the
following cycle, the controller will resume DCM Boost
operation. For Buck operation, constant frequency Discontinuous Current mode sets a minimum negative inductor current level. Synchronous Switch B is turned off
whenever inductor current is lower than this level. At very
light loads, this constant frequency operation is not as
efficient as Burst Mode operation or Skip-Cycle, but does
provide lower noise, constant frequency operation.
FREQUENCY SYNCHRONIZATION AND
FREQUENCY SETUP
The phase-locked loop allows the internal oscillator to be
synchronized to an external source via the PLLIN pin. The
phase detector output at the PLLFLTR pin is also the DC
frequency control input of the oscillator. The frequency
ranges from 200kHz to 400kHz, corresponding to a DC
voltage input from 0V to 2.4V at PLLFLTR. When locked,
the PLL aligns the turn on of the top MOSFET to the rising
edge of the synchronizing signal. When PLLIN is left open,
the PLLFLTR pin goes low, forcing the oscillator to its
minimum frequency.
INTVCC/EXTVCC POWER
Power for all power MOSFET drivers and most internal
circuitry is derived from the INTVCC pin. When the EXTVCC
pin is left open, an internal 6V low dropout linear regulator
supplies INTVCC power. If EXTVCC is taken above 5.7V, the
6V regulator is turned off and an internal switch is turned
on, connecting EXTVCC to INTVCC. This allows the INTVCC
power to be derived from a high efficiency external source.
POWER GOOD (PGOOD) PIN
The PGOOD pin is connected to an open drain of an internal
MOSFET. The MOSFET turns on and pulls the pin low when
the output is not within ±7.5% of the nominal output level
as determined by the resistive feedback divider. When the
output meets the ±7.5% requirement, the MOSFET is
turned off and the pin is allowed to be pulled up by an
external resistor to a source of up to 7V.
FOLDBACK CURRENT
Foldback current limiting is activated when the output
voltage falls below 70% of its nominal level, reducing power
waste. During start-up, foldback current limiting is disabled.
INPUT UNDERVOLTAGE RESET
The SS capacitor will be reset if the input voltage is allowed
to fall below approximately 4V. The SS capacitor will
attempt to charge through a normal soft-start ramp after
the input voltage rises above 4V.
3780f
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OUTPUT OVERVOLTAGE PROTECTION
An overvoltage comparator guards against transient overshoots (>7.5%) as well as other more serious conditions
that may overvoltage the output. In this case, Synchronous Switch B and Synchronous Switch D are turned on
until the overvoltage condition is cleared or the maximum
negative current limit is reached. When inductor current is
lower than the maximum negative current limit, Synchronous Switch B and Synchronous Switch D are turned off,
and Switch A and Switch C are turned on until the inductor
current reaches another negative current limit. If the
comparator still detects an overvoltage condition, Switch
A and Switch C are turned off, and Synchronous Switch B
and Synchronous Switch D are turned on again.
SHORT-CIRCUIT PROTECTION AND CURRENT LIMIT
Switch A on-time is limited by output voltage. When
output voltage is reduced and is lower than its nominal
level, Switch A on-time will be reduced.
In every Boost mode cycle, current is limited by a voltage
reference, which is proportional to the ITH pin voltage. The
maximum sensed current is limited to 160mV. In every
Buck mode cycle, the maximum sensed current is limited
to 130mV.
STANDBY MODE PIN
The STBYMD pin is a three-state input that controls
circuitry within the IC as follows: When the STBYMD pin
is held at ground, the SS pin is pulled to ground. When the
pin is left open, the internal SS current source charges the
SS capacitor, allowing turn-on of the controller and activating necessary internal biasing. When the STBYMD pin
is taken above 2V, the internal linear regulator is turned on
independent of the state on the RUN and SS pins, providing an output power source for “wake-up” circuitry.
Decouple the pin with a small capacitor (0.1µF) to ground
if the pin is not connected to a DC potential.
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Figure 11 is a basic LTC3780 application circuit. External
component selection is driven by the load requirement,
and begins with the selection of RSENSE and the inductor
value. Next, the power MOSFETs are selected. Finally, CIN
and COUT are selected. This circuit can be configured for
operation up to an input voltage of 36V.
RSENSE Selection and Maximum Output Current
RSENSE is chosen based on the required output current.
The current comparator threshold sets the peak of the
inductor current in Boost mode and the maximum inductor valley current in Buck mode. In Boost mode, the
maximum average load current is:
IOUT(MAX,BOOST) =
160mV • VIN ∆IL
–
2
RSENSE • VOUT
where ∆IL is peak-to-peak inductor ripple current. In Buck
mode, the maximum average load current is:
130mV ∆IL
IOUT(MAX,BUCK) =
+
RSENSE
2
Figure 7 shows how the load current (IMAXLOAD • RSENSE)
varies with input and output voltage
Allowing a margin for variations in LTC3780 and external
component values yields:
RSENSE =
2 • 160mV • VIN
2 • IOUT(MAX,BOOST) • VOUT + ∆IL(BOOST)
Selection of Operation Frequency
The LTC3780 uses a constant frequency architecture and
has an internal voltage controlled oscillator. The switching
frequency is determined by the internal oscillator capacitor. This internal capacitor is charged by a fixed current
plus an additional current that is proportional to the
voltage applied to the PLLFLTR pin. The frequency of this
oscillator can be varied over a 2-to-1 range. The PLLFLTR
pin can be grounded to lower the frequency to 200kHz or
tied to 2.4V to yield approximately 400kHz. When PLLIN is
left open, the PLLFLTR pin goes low, forcing the oscillator
to minimum frequency.
A graph for the voltage applied to the PLLFLTR pin vs
frequency is given in Figure 8. As the operating frequency
is increased the gate charge losses will be higher, reducing
efficiency. The maximum switching frequency is approximately 400kHz.
450
160
400
OPERATING FREQUENCY (kHz)
IMAX(LOAD) • RSENSE (mV)
150
140
130
120
110
350
300
250
200
150
100
50
100
0.1
1
10
VIN/VOUT (V)
3780 F07
Figure 7. Load Current vs VIN/VOUT
0
0
2
0.5
1.5
1
PLLFLTR PIN VOLTAGE (V)
2.5
3780 F08
Figure 8. Frequency vs PLLFLTR Pin Voltage
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Inductor Selection
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. The inductor
value has a direct effect on ripple current. The inductor
current ripple ∆IL is typically set to 20% to 40% of the
maximum inductor current. For a given ripple the inductance terms are as follows:
LBOOST >
LBUCK >
VIN(MIN)2 • VOUT – VIN(MIN) • 100
(
)
ƒ • IOUT(MAX) • % Ripple • VOUT2
(
)
VOUT • VIN(MAX) – VOUT • 100
ƒ • IOUT(MAX) • % Ripple • VIN(MAX)
H,
H
where:
In Boost mode, the discontinuous current shifts from the
input to the output, so COUT must be capable of reducing
the output voltage ripple. The effects of ESR (equivalent
series resistance) and the bulk capacitance must be considered when choosing the right capacitor for a given
output ripple voltage. The steady ripple due to charging
and discharging the bulk capacitance is given by:
Ripple (Boost,Cap) =
f is operating frequency, Hz
% Ripple is allowable inductor current ripple, %
VIN(MIN) is minimum input voltage, V
VIN(MAX) is maximum input voltage, V
VOUT is output voltage, V
IOUT(MAX) is maximum output load current
For high efficiency, choose an inductor with low core loss,
such as ferrite and molypermalloy (from Magnetics, Inc.).
Also, the inductor should have low DC resistance to reduce
the I2R losses, and must be able to handle the peak
inductor current without saturating. To minimize radiated
noise, use a toroid, pot core or shielded bobbin inductor.
CIN and COUT Selection
In Boost mode, input current is continuous. In Buck mode,
input current is discontinuous. In Buck mode, the selection of input capacitor CIN is driven by the need to filter the
input square wave current. Use a low ESR capacitor sized
to handle the maximum RMS current. For Buck operation,
the input RMS current is given by:
IRMS ≈ IOUT(MAX) •
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT(MAX)/2. This simple worst-case condition is
commonly used for design because even significant
deviations do not offer much relief. Note that ripple
current ratings from capacitor manufacturers are often
based on only 2000 hours of life which makes it advisable
to derate the capacitor.
VOUT
VIN
•
–1
VIN
VOUT
Ripple (Buck,Cap) =
(
IOUT(MAX) • VOUT – VIN(MIN)
COUT • VOUT • f
(
IOUT(MAX) • VIN(MAX ) – VOUT
COUT • VIN(MAX) • f
)V
)V
where COUT is the output filter capacitor.
The steady ripple due to the voltage drop across the ESR
is given by:
∆VBOOST,ESR = IL(MAX,BOOST) • ESR
∆VBUCK,ESR = IL(MAX,BUCK) • ESR
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic and
ceramic capacitors are all available in surface mount
packages. Ceramic capacitors have excellent low ESR
characteristics but can have a high voltage coefficient.
Capacitors are now available with low ESR and high ripple
current ratings such as OS-CON and POSCAP.
Power MOSFET Selection and
Efficiency Considerations
The LTC3780 requires four external N-channel power
MOSFETs, two for the top switches (Switch A and D,
shown in Figure 1) and two for the bottom switches
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(Switch B and C shown in Figure 1). Important parameters
for the power MOSFETs are the breakdown voltage VBR,DSS,
threshold voltage VGS,TH, on-resistance RDS(ON), reverse
transfer capacitance CRSS and maximum current IDS(MAX).
The drive voltage is set by the 6V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used in
LTC3780 applications. If the input voltage is expected to
drop below 5V, then the sub-logic threshold MOSFETs
should be considered.
In order to select the power MOSFETs, the power dissipated by the device must be known. For Switch A, the
maximum power dissipation happens in Boost mode,
when it remains on all the time. Its maximum power
dissipation at maximum output current is given by:
2
⎛V
⎞
PA,BOOST = ⎜ OUT • IOUT(MAX) ⎟ • ρT • RDS(ON)
⎝ VIN
⎠
where ρT is a normalization factor (unity at 25°C) accounting for the significant variation in on-resistance with
temperature, typically about 0.4%/°C as shown in Figure 9. For a maximum junction temperature of 125°C,
using a value ρT = 1.5 is reasonable.
Switch B operates in Buck mode as the synchronous
rectifier. Its power dissipation at maximum output current
is given by:
PB,BUCK
V –V
= IN OUT • IOUT(MAX)2 • ρT • RDS(ON)
VIN
Switch C operates in Boost mode as the control switch. Its
power dissipation at maximum current is given by:
PC,BOOST =
( VOUT – VIN )VOUT • I
VIN2
+ k • VOUT3 •
2
OUT(MAX)
• ρT • RDS(ON)
IOUT(MAX)
• CRSS • f
VIN
where CRSS is usually specified by the MOSFET manufacturers. The constant k, which accounts for the loss caused
by reverse recovery current, is inversely proportional to
the gate drive current and has an empirical value of 1.7.
For Switch D, the maximum power dissipation happens in
Boost mode, when its duty cycle is higher than 50%. Its
maximum power dissipation at maximum output current
is given by:
PD,BUCK
V
= IN
VOUT
2
⎛V
⎞
• ⎜ OUT • IOUT(MAX) ⎟ • ρT • RDS(ON)
⎝ VIN
⎠
For the same output voltage and current, Switch A has the
highest power dissipation and Switch B has the lowest
power dissipation unless a short occurs at the output.
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
TJ = TA + P • RTH(JA)
ρT NORMALIZED ON-RESISTANCE (Ω)
2.0
1.5
1.0
0.5
0
–50
50
100
0
JUNCTION TEMPERATURE (°C)
150
3780 F09
Figure 9. Normalized RDS(ON) vs Temperature
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The RTH(JA) to be used in the equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the case to the ambient temperature (RTH(JC)). This value
of TJ can then be compared to the original, assumed value
used in the iterative calculation process.
Schottky Diode (D1, D2) Selection
and Light Load Operation
The Schottky diodes D1 and D2 shown in Figure 1 conduct
during the dead time between the conduction of the power
MOSFET switches. They are intended to prevent the body
diode of Synchronous Switches B and D from turning on
and storing charge during the dead time. In particular, D2
significantly reduces reverse recovery current between
Switch D turn-off and Switch C turn-on, which improves
converter efficiency and reduces Switch C voltage stress.
In order for the diode to be effective, the inductance
between it and the synchronous switch must be as small
as possible, mandating that these components be placed
adjacently.
In Buck mode, when the FCB pin voltage is 0.85 < VFCB <
5V, the converter operates in Skip-Cycle mode. In this
mode, Synchronous Switch B remains off until the inductor peak current exceeds one-fifth of its maximum peak
current. As a result, D1 should be rated for about one-half
to one-third of the full load current.
In Boost mode, when the FCB pin voltage is higher than
5.3V, the converter operates in Discontinuous Current
mode. In this mode, Synchronous Switch D remains off
until the inductor peak current exceeds one-fifth of its
maximum peak current. As a result, D2 should be rated for
about one-third to one-fourth of the full load current.
In Buck mode, when the FCB pin voltage is higher than
5.3V, the converter operates in constant frequency Discontinuous Current mode. In this mode, Synchronous
Switch B remains on until the inductor valley current is
lower than the sense voltage representing the minimum
negative inductor current level (VSENSE = –5mV). Both
Switch A and B are off until next clock signal.
In Boost mode, when the FCB pin voltage is 0.85 < VFCB <
5.3V, the converter operates in Burst Mode operation. In
this mode, the controller clamps the peak inductor current
to approximately 20% of the maximum inductor current.
The output voltage ripple can increase during Burst Mode
operation.
INTVCC Regulator
An internal P-channel low dropout regulator produces 6V
at the INTVCC pin from the VIN supply pin. INTVCC powers
the drivers and internal circuitry within the LTC3780. The
INTVCC pin regulator can supply a peak current of 40mA
and must be bypassed to ground with a minimum of 4.7µF
tantalum, 10µF special polymer or low ESR type electrolytic capacitor. A 1µF ceramic capacitor placed directly
adjacent to the INTVCC and PGND IC pins is highly
recommended. Good bypassing is necessary to supply
the high transient current required by MOSFET gate
drivers.
Higher input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the
maximum junction temperature rating for the LTC3780 to
be exceeded. The system supply current is normally
dominated by the gate charge current. Additional external
loading of the INTVCC also needs to be taken into account
for the power dissipation calculations. The total INTV CC
current can be supplied by either the 6V internal linear
regulator or by the EXTVCC input pin. When the voltage
applied to the EXTVCC pin is less than 5.7V, all of the
INTVCC current is supplied by the internal 6V linear
regulator. Power dissipation for the IC in this case is
VIN • IINTVCC, and overall efficiency is lowered. The junction temperature can be estimated by using the equations
given in Note 2 of the Electrical Characteristics. For
example, LTC3780 VIN current is limited to less than
24mA from a 24V supply when not using the EXTVCC pin
as:
TJ = 70°C + 24mV • 24V • 95°C/W = 125°C
Use of the EXTVCC input pin reduces the junction temperature to:
TJ = 70°C + 24mV • 6V • 95°C/W = 84°C
To prevent maximum junction temperature from being
exceeded, the input supply current must be checked
operating in continuous mode at maximum VIN.
3780f
19
LTC3780
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EXTVCC Connection
Topside MOSFET Driver Supply (CA, DA, CB, DB)
The LTC3780 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
When the voltage applied to EXTVCC rises above 5.7V, the
internal regulator is turned off and a switch connects the
EXTVCC pin to the INTVCC pin thereby supplying internal
power. The switch remains closed as long as the voltage
applied to EXTVCC remains above 5.5V. This allows the
MOSFET driver and control power to be derived from the
output when (5.7V < VOUT < 7V) and from the internal
regulator when the output is out of regulation (start-up,
short-circuit). If more current is required through the
EXTVCC switch than is specified, an external Schottky
diode can be interposed between the EXTVCC and INTVCC
pins. Ensure that EXTVCC ≤ VIN.
Referring to Figure 11, the external bootstrap capacitors
CA and CB connected to the BOOST1 and BOOST2 pins
supply the gate drive voltage for the topside MOSFET
Switches A and D. When the top MOSFET Switch A turns
on, the switch node SW2 rises to VIN and the BOOST2 pin
rises to approximately VIN + INTVcc. When the bottom
MOSFET Switch B turns on, the switch node SW2 drops to
low and the boost capacitor CB is charged through DB from
INTVCC. When the top MOSFET Switch D turns on, the
switch node SW1 rises to VOUT and the BOOST1 pin rises
to approximately VOUT + INTVCC. When the bottom MOSFET Switch C turns on, the switch node SW1 drops to low
and the boost capacitor CA is charged through DA from
INTVCC. The boost capacitors CA and CB need to store
about 100 times the gate charge required by the top
MOSFET Switch A and D. In most applications a 0.1µF to
0.47µF, X5R or X7R dielectric capacitor is adequate.
The following list summarizes the three possible connections for EXTVCC:
1. EXTVCC left open (or grounded). This will cause INTVCC
to be powered from the internal 6V regulator at the cost
of a small efficiency penalty.
2. EXTVCC connected directly to VOUT (5.7V < VOUT < 7V).
This is the normal connection for a 6V regulator and
provides the highest efficiency.
3. EXTVCC connected to an external supply. If an external
supply is available in the 5.5V to 7V range, it may be
used to power EXTVCC provided it is compatible with
the MOSFET gate drive requirements.
Output Voltage
The LTC3780 output voltage is set by an external feedback
resistive divider carefully placed across the output capacitor. The resultant feedback signal is compared with the
internal precision 0.800V voltage reference by the error
amplifier. The output voltage is given by the equation:
⎛ R2 ⎞
VOUT = 0.8 V • ⎜ 1 + ⎟
⎝ R1⎠
Run Function
The RUN pin provides simple ON/OFF control for the
LTC3780. Driving the RUN pin above 1.5V permits the
controller to start operating. Pulling RUN below 1.5V puts
the LTC3780 into low current shutdown. Do not apply
more than 6V to the RUN pin.
Soft-Start Function
Soft-start reduces the input power sources’ surge currents by gradually increasing the controller’s current limit
(proportional to an internally buffered and clamped equivalent of VITH).
An internal 1.2µA current source charges up the CSS
capacitor. As the voltage on SS increases from 0V to 2.4V,
the internal current limit rises from 0V/RSENSE to
150mV/RSENSE. The output current limit ramps up slowly,
taking 1.5s/µF to reach full current. The output current
thus ramps up slowly, eliminating the starting surge
current required from the input power supply.
TIRMP =
2.4V
• CSS = (1.5s /µF ) • CSS
1.2µA
Do not apply more than 6V to the SS pin.
3780f
20
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The Standby Mode (STBYMD) Pin Function
The secondary output voltage VSEC is normally set as
shown in Figure 10 by turns ratio N of the transformer:
The Standby mode (STBYMD) pin provides several choices
for start-up and standby operational modes. If the pin is
pulled to ground, the SS pin is internally pulled to ground,
preventing start-up and thereby providing a single control
pin for turning off the controller. If the pin is left open or
decoupled with a capacitor to ground, the SS pin is
internally provided with a starting current, permitting
external control for turning on the controller. If the pin is
connected to a voltage greater than 1.25V, the internal
regulator (INTVCC) will be on even when the controller is
shut down (RUN pin voltage < 1.5V). In this mode, the
onboard 6V linear regulator can provide power to keepalive functions such as a keyboard controller.
VSEC ≈ (N + 1) • VOUT
However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current,
then VSEC will drop. An external resistive divider from VSEC
to the FCB pin sets a minimum voltage VSEC(MIN):
⎛ R6 ⎞
VSEC(MIN) ≈ 0.8 • ⎜ 1 + ⎟
⎝ R5 ⎠
If the VSEC drops below this level, the FCB voltage forces
temporary continuous switching operation until VSEC is
again above its minimum.
In order to prevent erratic operation if no external connections are made to FCB pin, the FCB pin has a 0.18µA
internal current source pulling the pin high. Include this
current when choosing resistor values R5 and R6.
FCB Pin Regulates Secondary Winding in Buck Mode
In Buck mode, the FCB pin can be used to regulate a
secondary winding or as a logic level input. Continuous
operation is forced when the FCB pin drops below 0.8V.
During continuous mode, current flows continuously in
the transformer primary. The secondary winding(s) draw
current only when Switch B and Switch D are on in Buck
mode. When primary load currents are low and/or the
VIN/VOUT ratio is low, the Synchronous Switch B may not
be on for a sufficient amount of time to transfer power
from the output capacitor to the secondary load. Forced
continuous operation will support secondary windings if
there is sufficient synchronous switch duty factor. Thus,
the FCB input pin removes the requirement that power
must be drawn from the auxiliary windings. With the loop
in continuous mode, the auxiliary outputs may nominally
be loaded without regard to the primary output load.
Fault Conditions: Current Limit and Current Foldback
The maximum inductor current is inherently limited in a
current mode controller by the maximum sense voltage. In
Boost mode, maximum sense voltage and the sense
resistance determines the maximum allowed inductor
peak current, which is:
IL(MAX,BOOST) =
160mV
RSENSE
VSEC
VOUT
VIN
R6
LTC3780
TG2
A
SW2
BG2
B
FCB
R5
•
T1
1:N
•
D
SW1
TG1
C
BG1
COUT
SGND
RSENSE
3780 F10
Figure 10. Secondary Output Loop
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21
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In Buck mode, maximum sense voltage and the sense
resistance determines the maximum allowed inductor
valley current, which is:
IL(MAX,BUCK) =
130mV
RSENSE
To further limit current in the event of a short circuit to
ground, the LTC3780 includes foldback current limiting. If
the output falls by more than 30%, then the maximum
sense voltage is progressively lowered to about one third
of its full value.
Fault Conditions: Overvoltage Protection
A comparator monitors the output for overvoltage conditions. The comparator (OV) detects overvoltage faults
greater than 7.5% above the nominal output voltage.
When the condition is sensed, Switches A and C are turned
off, and Switches B and D are turned on until the overvoltage condition is cleared. During an overvoltage condition,
a negative current limit (VSENSE = –60mV) is set to limit
negative inductor current. When the sensed current inductor current is lower than –60mV, Switch A and C are
turned on, and Switch B and D are turned off until the
sensed current is higher than –20mV. If the output is still
in overvoltage condition, Switch A and C are turned off,
and Switch B and D are turned on again.
Efficiency Considerations
2. Transition loss. This loss arises from the brief amount
of time Switch A or Switch C spends in the saturated
region during switch node transitions. It depends upon
the input voltage, load current, driver strength and
MOSFET capacitance, among other factors. The loss is
significant at input voltages above 20V and can be
estimated from:
Transition Loss ≈ 1.7A–1 • VIN2 • IOUT • CRSS • f
where CRSS is the reverse transfer capacitance.
3. INTVCC current. This is the sum of the MOSFET driver
and control currents. This loss can be reduced by
supplying INTVCC current through the EXTVCC pin from
a high efficiency source, such as an output derived
boost network or alternate supply if available.
4. CIN and COUT loss. The input capacitor has the difficult
job of filtering the large RMS input current to the regulator in Buck mode. The output capacitor has the more
difficult job of filtering the large RMS output current in
Boost mode. Both CIN and COUT are required to have low
ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries.
5. Other losses. Schottky diode D1 and D2 are responsible
for conduction losses during dead time and light load
conduction periods. Inductor core loss occurs predominately at light loads. Switch C causes reverse
recovery current loss in Boost mode.
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Although all dissipative
elements in circuit produce losses, four main sources
account for most of the losses in LTC3780 circuits:
When making adjustments to improve efficiency, the input
current is the best indicator of changes in efficiency. If you
make a change and the input current decreases, then the
efficiency has increased. If there is no change in input
current, then there is no change in efficiency.
1. DC I2R losses. These arise from the resistances of the
MOSFETs, sensing resistor, inductor and PC board
traces and cause the efficiency to drop at high output
currents.
As a design example, assume VIN = 5V to 18V (12V nominal), VOUT = 12V (5%), IOUT(MAX) = 5A and f = 400kHz.
Design Example
3780f
22
LTC3780
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Tie the PLLFLTR pin to INTVCC for 400kHz operation. The
inductance value is chosen first based on a 30% ripple
current assumption. In Buck mode, the ripple current is:
∆IL,BUCK =
VOUT
f •L
⎛ V ⎞
• ⎜ 1 – OUT ⎟
⎝
VIN ⎠
The highest value of ripple current occurs at the maximum
input voltage. In Boost mode, the ripple current is:
∆IL,BOOST =
VIN ⎛
V ⎞
• ⎜ 1 – IN ⎟
f • L ⎝ VOUT ⎠
A 6.8µH inductor will produce 13% ripple in Boost mode
(VIN = 6V) and 29% ripple in Buck mode (VIN = 18V).
The RSENSE resistor value can be calculated by using the
maximum current sense voltage specification with some
accommodation for tolerances.
(
TJ = 70°C + 1.94W • 40°C/W = 147.6°C
The maximum power dissipation of Switch B occurs in
Buck mode. Assuming a junction temperature of TJ = 80°C
with ρ80°C = 1.2, the power dissipation at VIN = 18V is:
PB,BUCK =
TJ = 70°C + 0.135W • 40°C/W = 75.4°C
The maximum power dissipation of Switch C occurs in
Boost mode. Assuming a junction temperature of TJ = 110°C
with ρ110°C = 1.4, the power dissipation at VIN = 5V is:
PC,BOOST =
2 • 160mV • VIN
2 • IOUT(MAX,BOOST) + ∆IL,BOOST • VOUT
Select an RSENSE of 10mΩ.
Output voltage is 12V. Select R1 as 20k. R2 is:
V
• R1
R2 = OUT
– R1
0.8
Select R2 as 280k. Both R1 and R2 should have a tolerance
of no more than 1%.
Next, choose the MOSFET switches. A suitable choice is
the Siliconix Si4840 (RDS(ON) = 0.009Ω (at VGS = 6V), CRSS
= 150pF, θJA = 40°C/W).
The maximum power dissipation of Switch A occurs
in Boost mode when Switch A stays on all the time.
Assuming a junction temperature of TJ = 150°C with
ρ150°C = 1.5, the power dissipation at VIN = 5V is:
⎛ 12 ⎞
PA,BOOST = ⎜ • 5⎟ • 1.5 • 0.009 = 1.94W
⎝ 5 ⎠
(12 – 5) • 12 • 52 • 1.4 • 0.009
52
+ 2 • 123 •
)
2
18 – 12 2
• 5 • 1.2 • 0.009 = 135mW
12
Double-check the TJ in the MOSFET at 70°C ambient
temperature:
The highest value of ripple current occurs at VIN = VOUT/2.
RSENSE =
Double-check the TJ in the MOSFET with 70°C ambient
temperature:
5
• 150p • 400k = 1.08W
5
Double-check the TJ in the MOSFET at 70°C ambient
temperature:
TJ = 70°C + 1.08W • 40°C/W = 113°C
The maximum power dissipation of Switch D occurs in
Boost mode when its duty cycle is higher than 50%.
Assuming a junction temperature of TJ = 100°C with
ρ100°C = 1.35, the power dissipation at VIN = 5V is:
2
PD,BUCK =
5 ⎛ 12 ⎞
• ⎜ • 5⎟ • 1.35 • 0.009 = 0.73W
12 ⎝ 5 ⎠
Double-check the TJ in the MOSFET at 70°C ambient
temperature:
TJ = 70°C + 0.73W • 40°C/W = 99°C
CIN is chosen to filter the square current in Buck mode. In
this mode, the maximum input current peak is:
IIN,PEAK(MAX,BUCK) = 5 • (1 + 29%) = 6.5A
3780f
23
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A low ESR (10mΩ) capacitor is selected. Input voltage
ripple is 65mV.
COUT is chosen to filter the square current in Boost mode.
In this mode, the maximum output current peak is:
IOUT,PEAK(MAX,BUCK) =
12
• 5 • (1 + 13%) = 13.6 A
5
A low ESR (5mΩ) capacitor is suggested. This capacitor
will limit output voltage ripple to 68mV.
PC Board Layout Checklist
The basic PC board layout requires a dedicated ground
plane layer. Also, for high current, a multilayer board
provides heat sinking for power components.
• The ground plane layer should not have any traces and
it should be as close as possible to the layer with power
MOSFETs.
• Place CIN, Switch A, Switch B and D2 in one compact
area. Place COUT, Switch C, Switch D and D1 in one
compact area.
• Use immediate vias to connect the components (including the LTC3780’s SGND and PGND pins) to the
ground plane. Use several large vias for each power
component.
• Use planes for VIN and VOUT to maintain good voltage
filtering and to keep power losses low.
• Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of
power components. Connect the copper areas to any
DC net (VIN or GND).
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3780. These items are also illustrated in Figure 11.
• Segregate the signal and power grounds. All small
signal components should return to the SGND pin at
one point which is then tied to the PGND pin close to the
sources of Switch B and Switch C.
• Place Switch B and Switch C as close to the controller
as possible, keeping the PGND, BG and SW traces
short.
• Keep the high dV/dT SW1, SW2, BOOST1, BOOST2, TG1
and TG2 nodes away from sensitive small-signal nodes.
• The path formed by Switch A, Switch B, D2 and the CIN
capacitor should have short leads and PC trace lengths.
The path formed by Switch C, Switch D, D1 and the COUT
capacitor also should have short leads and PC trace
lengths.
• The output capacitor (–) terminals should be connected
as close as possible the (–) terminals of the input
capacitor.
• Connect the INTVCC decoupling capacitor CVCC closely
to the INTVCC and PGND pins.
• Connect the top driver boost capacitor CA closely to the
BOOST1 and SW1 pins. Connect the top driver boost
capacitor CB closely to the BOOST2 and SW2 pins.
• Connect the input capacitors CIN and output capacitors
COUT close to the power MOSFETs. These capacitors
carry the MOSFET AC current in Boost and Buck mode.
• Connect VOSENSE pin resistive dividers to the (+) terminals of COUT and signal ground. A small VOSENSE
decoupling capacitor should be as close as possible to
the LTC3780 SGND pin. The R2 connection should not
be along the high current or noise paths, such as the
input capacitors.
• Route SENSE– and SENSE+ leads together with minimum PC trace spacing. The filter capacitor between
SENSE+ and SENSE– should be as close as possible to
the IC. Ensure accurate current sensing with Kelvin
connections at the SENSE resistor.
• Connect the ITH pin compensation network close to the
IC, between ITH and the signal ground pins. The capacitor helps to filter the effects of PCB noise and output
voltage ripple voltage from the compensation loop.
3780f
24
LTC3780
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APPLICATIO S I FOR ATIO
• Connect the INTVCC decoupling capacitor close to the
IC, between the INTVCC and the power ground pins.
This capacitor carries the MOSFET drivers’ current
peaks. An additional 1µF ceramic capacitor placed
immediately next to the INTVcc and PGND pins can help
improve noise performance substantially.
VOUT
RPU
VPULLUP
1
CSS
2
PGOOD BOOST1
TG1
SS
23
LTC3780
CC2
CC1
R1
RC
SENSE+
SW1
4
SENSE–
VIN
R2
6
7
8
9
10
ITH
EXTVCC
VOSENSE INTVCC
SGND
D
D2
DA
3
5
COUT
CA
24
BG1
RUN
PGND
FCB
BG2
PLLFLTR
SW2
PLLIN
TG2
22
CF
21
C
20
CVCC
19
L
18
RSENSE
17
16
B
D1
15
DB
fIN
11
12
STBYMD BOOST2
14
A
CB
13
CIN
RIN
3780 F11
VIN
Figure 11. LTC3780 Layout Diagram
3780f
25
LTC3780
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PACKAGE DESCRIPTIO
G Package
24-Lead Plastic SSOP (5.3mm)
(Reference LTC DWG # 05-08-1640)
7.90 – 8.50*
(.311 – .335)
24 23 22 21 20 19 18 17 16 15 14 13
1.25 ±0.12
7.8 – 8.2
5.3 – 5.7
7.40 – 8.20
(.291 – .323)
0.42 ±0.03
0.65 BSC
RECOMMENDED SOLDER PAD LAYOUT
1 2 3 4 5 6 7 8 9 10 11 12
2.0
(.079)
MAX
5.00 – 5.60**
(.197 – .221)
0° – 8°
0.09 – 0.25
(.0035 – .010)
0.55 – 0.95
(.022 – .037)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
0.65
(.0256)
BSC
0.22 – 0.38
(.009 – .015)
TYP
0.05
(.002)
MIN
G24 SSOP 0204
3. DRAWING NOT TO SCALE
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE
**DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE
3780f
26
LTC3780
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PACKAGE DESCRIPTIO
UH Package
32-Lead Plastic QFN (5mm × 5mm)
(Reference LTC DWG # 05-08-1693)
0.70 ±0.05
5.50 ±0.05
4.10 ±0.05
3.45 ±0.05
(4 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
BOTTOM VIEW—EXPOSED PAD
RECOMMENDED SOLDER PAD LAYOUT
5.00 ± 0.10
(4 SIDES)
PIN 1 NOTCH R = 0.30 TYP
OR 0.35 × 45° CHAMFER
R = 0.115
TYP
0.75 ± 0.05
0.00 – 0.05
31 32
0.40 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
3.45 ± 0.10
(4-SIDES)
(UH32) QFN 1004
0.200 REF
NOTE:
1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE
M0-220 VARIATION WHHD-(X) (TO BE APPROVED)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
0.25 ± 0.05
0.50 BSC
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3780f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC3780
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TYPICAL APPLICATIO
RPU
VPULLUP
CSS
6.8nF
1
2
CC2
47pF
RC
5.6k
R1
20k
TG1
SS
R2 280k
3
4
5
6
7
ON/OFF
8
9
INTVCC
10
11
12
+
23
SENSE
SW1
SENSE–
VIN
ITH
EXTVCC
VOSENSE INTVCC
SGND
BG1
RUN
PGND
FCB
BG2
PLLFLTR
PLLIN
SW2
TG2
STBYMD BOOST2
CSTBYMD
0.01µF
D
Si7884DP
DA
1N5819HW
22
CF 0.1µF
21
C
Si7884DP
20
L
6.8µF
CVCC 4.7µF
19
COUT
200µF
D2
B320A
24
LTC3780
68pF
CC1
3300pF
PGOOD BOOST1
CA
0.22µF
VOUT
12V
5A
D1
B320A
18
10mΩ
17
16
B
Si7884DP
15
DB
1N5819HW
14
A
Si7884DP
13
10Ω
CIN
47µF
CB 0.22µF
100Ω
3780 TA02
VIN
100Ω
Figure 12. LTC3780 12V/3A, Buck-Boost Regulator
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3780f
28
Linear Technology Corporation
LT/TP 0305 500 • PRINTED IN THE USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2005
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