Fairchild FSDL0365 Green mode fairchild power switch (fpstm) Datasheet

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FSDL0365RN, FSDM0365RN
Green Mode Fairchild Power Switch (FPSTM)
Features
• Internal Avalanche Rugged Sense FET
• Consumes only 0.65W at 240VAC & 0.3W load with
Advanced Burst-Mode Operation
• Frequency Modulation for low EMI
• Precision Fixed Operating Frequency
• Internal Start-up Circuit
• Pulse by Pulse Current Limiting
• Abnormal Over Current Protection
• Over Voltage Protection
• Over Load Protection
• Internal Thermal Shutdown Function
• Auto-Restart Mode
• Under Voltage Lockout
• Low Operating Current (3mA)
• Adjustable Peak Current Limit
• Built-in Soft Start
Applications
• SMPS for VCR, SVR, STB, DVD & DVCD
• SMPS for Printer, Facsimile & Scanner
• Adaptor for Camcorder
Description
The FSDx0365RN(x stands for L, M) are integrated Pulse
Width Modulators (PWM) and Sense FETs specifically
designed for high performance offline Switch Mode Power
Supplies (SMPS) with minimal external components. Both
devices are integrated high voltage power switching regulators which combine an avalanche rugged Sense FET with a
current mode PWM control block. The integrated PWM controller features include: a fixed oscillator with frequency
modulation for reduced EMI, Under Voltage Lock Out
(UVLO) protection, Leading Edge Blanking (LEB), optimized gate turn-on/turn-off driver, Thermal Shut Down
(TSD) protection, Abnormal Over Current Protection
(AOCP) and temperature compensated precision current
sources for loop compensation and fault protection circuitry.
When compared to a discrete MOSFET and controller or
RCC switching converter solution, the FSDx0365RN reduce
total component count, design size, weight and at the same
time increase efficiency, productivity, and system reliability.
Both devices are a basic platform well suited for cost effective designs of flyback converters.
OUTPUT POWER TABLE
230VAC ±15%(3)
85-265VAC
PRODUCT
Adapter(1)
Open
Frame(2)
Adapter(1)
Open
Frame(2)
FSDL321
11W
17W
8W
12W
FSDH321
11W
17W
8W
12W
FSDL0165RN
13W
23W
11W
17W
FSDM0265RN
16W
27W
13W
20W
FSDH0265RN
16W
27W
13W
20W
FSDL0365RN
19W
30W
16W
24W
FSDM0365RN
19W
30W
16W
24W
FSDL0165RL
13W
23W
11W
17W
FSDM0265RL
16W
27W
13W
20W
FSDH0265RL
16W
27W
13W
20W
FSDL0365RL
19W
30W
16W
24W
FSDM0365RL
19W
30W
16W
24W
Table 1. Notes: 1. Typical continuous power in a non-ventilated enclosed adapter measured at 50°C ambient. 2.
Maximum practical continuous power in an open frame
design at 50°C ambient. 3. 230 VAC or 100/115 VAC with
doubler.
Typical Circuit
AC
IN
DC
OUT
Vstr
Ipk
Drain
PWM
Vfb
Vcc
Source
Figure 1. Typical Flyback Application
Rev.1.0.4
©2004 Fairchild Semiconductor Corporation
FSDL0365RN, FSDM0365RN
Internal Block Diagram
Vcc
Vstr
5
2
Istart
+
V BURL /V BURH
-
Soft start
8V/12V
Vcc good
Vcc
V BURH
I B_PEAK
Vcc
Drain
6,7,8
Internal
Bias
Vref
Freq.
Modulation
Vcc
OSC
I delay
V FB
I FB
Normal
2.5R
Ipk
S
Q
R
Q
PWM
3
Burst
Gate
driver
R
4
LEB
V SD
Vcc
1 GND
S
Q
R
Q
Vovp
TSD
Vcc good
AOCP
Vocp
Figure 2. Functional Block Diagram of FSDx0365RN
2
FSDL0365RN, FSDM0365RN
Pin Definitions
Pin Number
Pin Name
1
GND
Sense FET source terminal on primary side and internal control ground.
Vcc
Positive supply voltage input. Although connected to an auxiliary transformer winding, current is supplied from pin 5 (Vstr) via an internal switch during
startup (see Internal Block Diagram section). It is not until Vcc reaches the
UVLO upper threshold (12V) that the internal start-up switch opens and device power is supplied via the auxiliary transformer winding.
Vfb
The feedback voltage pin is the non-inverting input to the PWM comparator.
It has a 0.9mA current source connected internally while a capacitor and optocoupler are typically connected externally. A feedback voltage of 6V triggers over load protection (OLP). There is a time delay while charging
between 3V and 6V using an internal 5uA current source, which prevents
false triggering under transient conditions but still allows the protection
mechanism to operate under true overload conditions.
Ipk
Pin to adjust the current limit of the Sense FET. The feedback 0.9mA current
source is diverted to the parallel combination of an internal 2.8kΩ resistor
and any external resistor to GND on this pin to determine the current limit.
If this pin is tied to Vcc or left floating, the typical current limit will be 2.15A.
Vstr
This pin connects directly to the rectified AC line voltage source. At start up
the internal switch supplies internal bias and charges an external storage
capacitor placed between the Vcc pin and ground. Once the Vcc reaches
12V, the internal switch is disabled.
Drain
The Drain pin is designed to connect directly to the primary lead of the transformer and is capable of switching a maximum of 650V. Minimizing the
length of the trace connecting this pin to the transformer will decrease leakage inductance.
2
3
4
5
6, 7, 8
Pin Function Description
Pin Configuration
8DIP
8LSOP
GND 1
8 Drain
Vcc 2
7 Drain
Vfb 3
6 Drain
Ipk 4
5 Vstr
Figure 3. Pin Configuration (Top View)
3
FSDL0365RN, FSDM0365RN
Absolute Maximum Ratings
(Ta=25°C, unless otherwise specified)
Characteristic
Drain Current Pulsed
(1)
Single Pulsed Avalanche
Energy(2)
Maximum Supply Voltage
Symbol
Value
Unit
IDM
12.0
ADC
EAS
127
mJ
VCC,MAX
20
V
Analog Input Voltage Range
VFB
-0.3 to VSD
V
Total Power Dissipation
PD
1.56
W
Operating Junction Temperature.
TJ
+150
°C
Operating Ambient Temperature.
TA
-25 to +85
°C
TSTG
-55 to +150
°C
Storage Temperature Range.
Note:
1. Repetitive rating: Pulse width limited by maximum junction temperature
2. L = 51mH, starting Tj = 25°C
3. L = 13µH, starting Tj = 25°C
4. Vsd is shutdown feedback voltage ( see Protection Section in Electrical Characteristics )
Thermal Impedance
Parameter
Symbol
Value
Unit
θJA(1)
θJC(2)
85.74
°C/W(3)
30.38
°C/W
8DIP
Junction-to-Ambient Thermal
Junction-to-Case Thermal
Note:
1. Free standing with no heatsink.
2. Measured on the GND pin close to plastic interface.
3. Soldered to 0.36 sq. inch(232mm2), 2 oz.(610g/m2) copper clad.
4
FSDL0365RN, FSDM0365RN
Electrical Characteristics
(Ta = 25°C unless otherwise specified)
Parameter
Symbol
Condition
Min.
Typ.
Max.
Unit
Sense FET SECTION
Startup Voltage (Vstr) Breakdown
BVSTR
VCC=0V, ID=1mA
650
-
-
V
Drain-Source Breakdown Voltage
BVDSS
VGS=0V, ID=50µA
650
-
-
V
VDS=660V, VGS=0V
-
-
50
µA
VDS=0.8Max.Rating,
VGS=0V, TC=125°C
-
-
200
µA
VGS=10V, ID=0.5A
-
3.6
4.5
Ω
-
315
-
pF
-
47
-
pF
-
9
-
pF
-
11.2
-
ns
-
34
-
ns
-
28.2
-
ns
-
32
-
ns
61
67
73
KHz
±1.5
±2.0
±2.5
KHz
45
50
55
KHz
±1.0
±1.5
±2.0
KHz
-
±5
±10
%
Off-State Current
(Max.Rating =660V)
IDSS
On-State Resistance(1)
RDS(ON)
Input Capacitance
CISS
Output Capacitance
COSS
Reverse Transfer Capacitance
CRSS
Turn On Delay Time
Rise Time
Turn Off Delay Time
Fall Time
VGS=0V, VDS=25V,
F=1MHz
TD(ON)
TR
TD(OFF)
TF
VDS=325V, ID=1.0A
(Sense FET switching
time is essentially
independent of
operating temperature)
CONTROL SECTION
Output Frequency
FOSC
Output Frequency Modulation
FMOD
Output Frequency
FOSC
Output Frequency Modulation
Frequency Change With Temperature(2)
FMOD
-
FSDM0365R
FSDL0365R
-25°C ≤ Ta ≤ 85°C
Maximum Duty Cycle
DMAX
71
77
83
%
Minimum Duty Cycle
DMIN
0
0
0
%
Start threshold voltage
VSTART
VFB=GND
11
12
13
V
Stop threshold voltage
VSTOP
VFB=GND
7
8
9
V
IFB
VFB=GND
0.7
0.9
1.1
mA
VFB=4V
10
15
20
ms
VBURH
-
0.4
0.5
0.6
V
VBURL
-
0.25
0.35
0.45
V
IOVER
Max. inductor current
1.89
2.15
2.41
A
Feedback Source Current
Internal Soft Start Time
TS/S
BURST MODE SECTION
Burst Mode Voltages
PROTECTION SECTION
Drain to Source Peak Current Limit
5
FSDL0365RN, FSDM0365RN
Current Limit Delay(3)
TCLD
Thermal Shutdown
TSD
Shutdown Feedback Voltage
Over Voltage Protection
Shutdown Feedback Delay Current
Leading Edge Blanking Time
-
500
-
ns
125
140
-
°C
VSD
5.5
6.0
6.5
V
VOVP
18
19
-
V
3.5
5.0
6.5
µA
200
-
-
ns
IDELAY
-
VFB=4V
TLEB
TOTAL DEVICE SECTION
Operating Current
Start Up Current
Vstr Supply Voltage
IOP
VCC=14V
1
3
5
mA
ISTART
VCC=0V
0.7
0.85
1.0
mA
VSTR
VCC=0V
35
-
-
V
Note:
1. Pulse test: Pulse width ≤ 300uS, duty ≤ 2%
2. These parameters, although guaranteed, are tested in EDS (wafer test) process
3. These parameters, although guaranteed, are not 100% tested in production
6
FSDL0365RN, FSDM0365RN
Comparison Between KA5x0365RN and FSDx0365RN
Function
KA5x0365RN
FSDx0365RN
FSDx0365RN Advantages
Soft-Start
not applicable
15mS
• Gradually increasing current limit
during soft-start further reduces peak
current and voltage component
stresses
• Eliminates external components used
for soft-start in most applications
• Reduces or eliminates output
overshoot
External Current Limit
not applicable
Programmable of
default current limit
• Smaller transformer
• Allows power limiting (constant overload power)
• Allows use of larger device for lower
losses and higher efficiency.
Frequency Modulation
not applicable
±2.0KHz @67KHz
±1.5KHz @50KHz
• Reduced conducted EMI
Burst Mode Operation
not applicable
Yes-built into
controller
• Improve light load efficiency
• Reduces no-load consumption
• Transformer audible noise reduction
Drain Creepage at
Package
1,02mm
7.62mm
• Greater immunity to arcing as a result
of build-up of dust, debris and other
contaminants
7
FSDL0365RN, FSDM0365RN
Typical Performance Characteristics (Sense FET part)
1
10
VGS
15.0 V
10.0 V
8.0 V
7.0 V
6.5 V
6.0 V
Bottom : 5.5 V
ID, Drain Current [A]
Top :
0
10
-1
10
※ Note :
1. 250µs Pulse Test
2. TC = 25℃
0
1
10
10
VDS, Drain-Source Voltage [V]
Output Characteristics
8.0
IDR , Reverse Drain Current [A]
RDS(ON) [Ω ],
Drain-Source On-Resistance
7.5
7.0
6.5
VGS = 10V
6.0
5.5
VGS = 20V
5.0
4.5
4.0
3.5
0
10
150℃
25℃
-1
0
1
2
3
4
5
6
10
7
0.2
0.4
On-Resistance vs. Drain Current
0.8
1.0
1.2
1.4
Source-Drain Diode Forward Voltage
700
500
Ciss
400
Coss
300
Crss
200
※ Note ;
1. VGS = 0 V
2. f = 1 MHz
100
-1
10
0
10
1
10
VDS, Drain-Source Voltage [V]
Capacitance vs. Drain-Source Voltage
VGS, Gate-Source Voltage [V]
12
Ciss = Cgs + Cgd (Cds = shorted)
Coss = Cds + Cgd
Crss = Cgd
600
Capacitances [pF]
0.6
VSD , Source-Drain Voltage [V]
ID, Drain Current [A]
8
※ Note :
1. VGS = 0V
2. 250µ s Pulse Test
※ Note : TJ = 25℃
3.0
2.5
1
10
10
VDS = 130V
V DS = 325V
8
VDS = 520V
6
4
2
0
※ Note : ID = 3.0 A
0
2
4
6
8
10
QG, Total Gate Charge [nC]
Gate Charge vs. Gate-Source Voltage
12
FSDL0365RN, FSDM0365RN
Typical Performance Characteristics (Continued)
BVDSS, (Normalized)
Drain-Source Breakdown Voltage
1.15
1.05
1.00
0.95
※ Note :
1. VGS = 0 V
2. ID = 250 µ A
0.90
-50
0
50
100
RDS(ON), (Normalized)
Drain-Source On-Resistance
2.5
1.10
2.0
1.5
1.0
※ Note :
1. VGS = 10 V
2. ID = 1.5 A
0.5
150
-50
0
o
50
100
150
o
TJ, Junction Temperature [ C]
TJ, Junction Temperature [ C]
On-Resistance vs. Temperature
Breakdown Voltage vs. Temperature
2.0
1
Operation in This Area
is Limited by R DS(on)
10
1.5
DC 10 s
-1
10
1s
10 ms
100 ms
ID, Drain Current [A]
10
1 ms
-2
10
-3
10
0
10
1
1.0
0.5
0.0
25
2
10
10
50
VDS, Drain-Source Voltage [V]
0.2
75
100
125
150
TC, Case Temperature [℃]
Max. Drain Current vs. Case Temperature
Max. Safe Operating Area
Zθ JC(t), Thermal Response
ID, Drain Current [A]
10 µs
100 µs
0
D=0.5
0.2
10
0.1
0.05
0.02
1
0.01
※ Notes :
1. Zθ JC(t) = 80 ℃/W Max.
2. Duty Factor, D=t1/t2
3. TJM - TC = PDM * Zθ JC(t)
single pulse
0.1
1E-5
1E-4
1E-3
0.01
0.1
1
10
100
1000
t1, Square Wave Pulse Duration [sec]
Thermal Response
9
FSDL0365RN, FSDM0365RN
Typical Performance Characteristics (Control Part)
1.20
1.20
1.00
1.00
Normalized
Normalized
(These characteristic graphs are normalized at Ta = 25°C)
0.80
0.60
0.40
0.80
0.60
0.40
0.20
0.20
0.00
0.00
-50
0
50
100
-50
150
0
1.20
1.20
1.00
1.00
Normalized
Normalized
150
Frequency Modulation (FMOD)
Operating Frequency (Fosc)
0.80
0.60
0.40
0.20
0.80
0.60
0.40
0.20
0.00
0.00
-50
0
50
100
150
-50
0
T emp[ ℃]
50
100
150
T emp[ ℃]
Operating supply current (Iop)
Maximum duty cycle (Dmax)
1.20
1.20
1.00
1.00
Normalized
Nomalized
100
T emp[ ℃]
T emp[ ℃]
0.80
0.60
0.40
0.20
0.80
0.60
0.40
0.20
0.00
0.00
-50
0
50
100
T emp[ ℃]
Start Threshold Voltage (Vstart)
10
50
150
-50
0
50
100
T emp[ ℃]
Stop Threshold Voltage (Vstop)
150
FSDL0365RN, FSDM0365RN
1.20
1.20
1.00
1.00
Normalized
Normalized
Typical Performance Characteristics (Continued)
0.80
0.60
0.40
0.80
0.60
0.40
0.20
0.20
0.00
0.00
-50
0
50
100
-50
150
0
Feedback Source Current (Ifb)
150
Peak current limit (Iover)
1.20
1.20
1.00
1.00
Normalized
Normalized
100
T emp[ ℃]
T emp[ ℃]
0.80
0.60
0.40
0.20
0.80
0.60
0.40
0.20
0.00
0.00
-50
0
50
100
150
-50
0
T emp[ ℃]
50
100
150
T emp[ ℃]
J-FET Start up current (Istr)
Start up Current (Istart)
1.20
1.20
1.00
1.00
Normalized
Normalized
50
0.80
0.60
0.40
0.80
0.60
0.40
0.20
0.20
0.00
0.00
-50
0
50
100
T emp[ ℃]
Burst peak current (Iburst)
150
-50
0
50
100
150
Temp[℃]
Over Voltage Protection (Vovp)
11
FSDL0365RN, FSDM0365RN
Functional Description
1. Startup : In previous generations of Fairchild Power
Switches (FPS) the Vstr pin had an external resistor to the
DC input voltage line. In this generation the startup resistor
is replaced by an internal high voltage current source and a
switch that shuts off when 15mS goes by after the supply
voltage, Vcc, gets above 12V. The source turns back on if
Vcc drops below 8V.
Vcc
Vfb
Vo
0.9mA
FB
3
OSC
D1
Cfb
D2
28R
Vfb*
Gate
driver
R
431
VSD
Vin,dc
Vref
2uA
OLP
Istr
Figure 5. Pulse width modulation (PWM) circuit
Vstr
Vcc
UVLO <8V
on
J-FET
15m S After UVLO
start(>12V)
off
Figure 4. High voltage current source
2. Feedback Control : The FSDx0365RN employs current
mode control, shown in figure 5. An opto-coupler (such as
the H11A817A) and shunt regulator (such as the KA431) are
typically used to implement the feedback network. Comparing the feedback voltage with the voltage across the Rsense
resistor plus an offset voltage makes it possible to control the
switching duty cycle. When the reference pin voltage of the
KA431 exceeds the internal reference voltage of 2.5V, the
H11A817A LED current increases, thus pulling down the
feedback voltage and reducing the duty cycle. This event
typically happens when the input voltage is increased or the
output load is decreased.
3. Leading edge blanking (LEB) : At the instant the internal
Sense FET is turned on, there usually exists a high current
spike through the Sense FET, caused by the primary side
capacitance and secondary side rectifier diode reverse recovery. Excessive voltage across the Rsense resistor would lead
to incorrect feedback operation in the current mode PWM
control. To counter this effect, the FPS employs a leading
edge blanking (LEB) circuit. This circuit inhibits the PWM
comparator for a short time (TLEB) after the Sense FET is
turned on.
12
4. Protection Circuit : The FPS has several protective functions such as over load protection (OLP), over voltage protection (OVP), abnormal over current protection (AOCP),
under voltage lock out (UVLO) and thermal shutdown
(TSD). Because these protection circuits are fully integrated
inside the IC without external components, the reliability is
improved without increasing cost. Once the fault condition
occurs, switching is terminated and the Sense FET remains
off. This causes Vcc to fall. When Vcc reaches the UVLO
stop voltage, 8V, the protection is reset and the internal high
voltage current source charges the Vcc capacitor via the Vstr
pin. When Vcc reaches the UVLO start voltage,12V, the FPS
resumes its normal operation. In this manner, the auto-restart
can alternately enable and disable the switching of the power
Sense FET until the fault condition is eliminated.
4.1 Over Load Protection (OLP) : Overload is defined as the
load current exceeding a pre-set level due to an unexpected
event. In this situation, the protection circuit should be activated in order to protect the SMPS. However, even when the
SMPS is in the normal operation, the over load protection
circuit can be activated during the load transition. In order to
avoid this undesired operation, the over load protection circuit is designed to be activated after a specified time to determine whether it is a transient situation or an overload
situation. In conjunction with the Ipk current limit pin (if
used) the current mode feedback path would limit the current
in the Sense FET when the maximum PWM duty cycle is
attained. If the output consumes more than this maximum
power, the output voltage (Vo) decreases below the set voltage. This reduces the current through the opto-coupler LED,
which also reduces the opto-coupler transistor current, thus
increasing the feedback voltage (Vfb). If Vfb exceeds 3V, the feedback input diode is blocked and the 5uA Idelay current source starts
to charge Cfb slowly up to Vcc. In this condition, Vfb continues
increasing until it reaches 6V, when the switching operation is terminated as shown in figure 6. The delay time for shutdown is the
time required to charge Cfb from 3V to 6V with 5uA.
FSDL0365RN, FSDM0365RN
monitors the current through the sensing resistor. The voltage across the resistor is then compared with a preset AOCP
level. If the sensing resistor voltage is greater than the AOCP
level, pulse by pulse AOCP is triggered regardless of uncontrollable LEB time. Here, pulse by pulse AOCP stops Sense
FET within 350nS after it is activated.
Vcc
8V
OLP
6V
FPS switching
Following Vcc
3V
Delay current (5uA) charges the Cfb
t1
t2
t1 = −
1
RC
t 2 = C fb
In (1 −
fb
t3
t4
V ( t 1)
); V ( t1) = 3V , R = 2 . 8 K Ω , C fb = C
R
t
fb _ fig . 2
(V (t1 + t 2) − V (t1))
; I delay = 5uA,V (t1 + t 2) − V (t1) = 3V
I delay
Figure 6. Over load protection
4.2 Thermal Shutdown (TSD) : The Sense FET and the control IC are integrated, making it easier for the control IC to
detect the temperature of the Sense FET. When the temperature exceeds approximately 140°C, thermal shutdown is acti-
4.4 Over Voltage Protection (OVP) : In case of malfunction in the secondary side feedback circuit, or feedback loop
open caused by a defect of solder, the current through the
opto-coupler transistor becomes almost zero. Then, Vfb
climbs up in a similar manner to the over load situation, forcing the preset maximum current to be supplied to the SMPS
until the over load protection is activated. Because excess
energy is provided to the output, the output voltage may
exceed the rated voltage before the over load protection is
activated, resulting in the breakdown of the devices in the
secondary side. In order to prevent this situation, an over
voltage protection (OVP) circuit is employed. In general,
Vcc is proportional to the output voltage and the FPS uses
Vcc instead of directly monitoring the output voltage. If
VCC exceeds 19V, OVP circuit is activated resulting in termination of the switching operation. In order to avoid undesired activation of OVP during normal operation, Vcc should
be properly designed to be below 19V.
vated.
4.3 Abnormal Over Current Protection (AOCP) :
PWM
COMPARATOR
Vfb
CLK
LEB
Drain
Out Driver
Vsense
AOCP
COMPARATOR
S
Q
R
5. Soft Start : The FPS has an internal soft start circuit that
increases the feedback voltage together with the Sense FET
current slowly after it starts up. The typical soft start time is
15msec, as shown in figure 8, where progressive increments
of Sense FET current are allowed during the start-up phase.
The pulse width to the power switching device is progressively increased to establish the correct working conditions
for transformers, inductors, and capacitors. The voltage on
the output capacitors is progressively increased with the
intention of smoothly establishing the required output voltage. It also helps to prevent transformer saturation and
reduce the stress on the secondary diode.
Rsense
VAOCP
Drain current
[A]
Figure 7. AOCP Function & Block
2.15A
1mS
Even though the FPS has OLP (Over Load Protection) and
current mode PWM feedback, these are not enough to protect the FPS when a secondary side diode short or a transformer pin short occurs. In addition to start-up, soft-start is
also activated at each restart attempt during auto-restart and
when restarting after latch mode is activated. The FPS has an
internal AOCP (Abnormal Over Current Protection) circuit
as shown in figure 7. When the gate turn-on signal is applied
to the power Sense FET, the AOCP block is enabled and
15steps
Current limit
0.98A
t
13
FSDL0365RN, FSDM0365RN
D R A IN
5V
Burst Operation
Burst Operation
Feedback
Normal Operation
S W IT C H
OFF
GND
I_ o v e r
0.5V
Rsense
0.3V
Current
waveform
Switching OFF
Switching OFF
Figure 8. Soft Start Function
Figure 10. Circuit for Burst Operation
6. Burst operation :In order to minimize power dissipation in
standby mode, the FPS enters burst mode operation.
+
0.3/0.5V
-
0.5V
Vcc
IB_PEAK
Vcc
Idelay
FB
Vcc
IFB
Normal
PWM
3
2.5R
Burst
R
7. Frequency Modulation : EMI reduction can be accomplished by modulating the switching frequency of a switched
power supply. Frequency modulation can reduce EMI by
spreading the energy over a wider frequency range than the
band width measured by the EMI test equipment. The
amount of EMI reduction is directly related to the depth of
the reference frequency. As can be seen in Figure 11, the frequency changes from 65KHz to 69KHz in 4mS for the
FSDM0265RN. Frequency modulation allows the use of a
cost effective inductor instead of an AC input mode choke to
satisfy the requirements of world wide EMI limits.
MOSFET
Current
Internal
O scillator
Figure 9. Circuit for Burst operation
69kH z
As the load decreases, the feedback voltage decreases. As shown in
figure 10, the device automatically enters burst mode when the
feedback voltage drops below VBURH(500mV). Switching still continues but the current limit is set to a fixed limit internally to minimize flux density in the transformer. The fixed current limit is
larger than that defined by Vfb = VBURH and therefore, Vfb is
driven down further. Switching continues until the feedback
voltage drops below VBURL(300mV). At this point switching
stops and the output voltages start to drop at a rate dependent
on the standby current load. This causes the feedback voltage to rise. Once it passes VBURH(500mV) switching resumes.
The feedback voltage then falls and the process repeats. Burst
mode operation alternately enables and disables switching of
the power Sense FET thereby reducing switching loss in
Standby mode.
14
D rain to
S ourc e
voltage
D rain to
S ource
current
V ds
W aveform
4k H z
65kH z
67kH z
69kH z
T urn-on
T urn-off
point
Figure 11. Frequency Modulation Waveform
FSDL0365RN, FSDM0365RN
5uA
Amplitude (dBµV)
900uA
Feed
Back
CISPR2QB
3
CISPR2AB
2 KΩ
Current
Limit
4
AK Ω
PWM
comparator
0.8 KΩ
Rsense
SenseFET
Sense
Figure 14. Peak current adjustment
Frequency (MHz)
Figure 12. KA5-series FPS Full Range EMI scan(67KHz,
no Frequency Modulation) with DVD Player SET
For example, FSDx0265RN has a typical Sense FET current
limit (IOVER) of 2.15A. The Sense FET current can be limited to
1A by inserting a 2.8kΩ between the current limit pin and
ground which is derived from the following equations:
2.15: 1 = 2.8KΩ : XKΩ ,
Amplitude (dBµV)
CISPR2QB
CISPR2AB
X = 1.3KΩ,
Since X represents the resistance of the parallel network, Y
can be calculated using the following equation:
Y = X / (1 - (X/2.8KΩ))
Frequency (MHz)
Figure 13. FSDX-series FPS Full Range EMI Scan (67KHz,
with Frequency Modulation) with DVD Player SET
8. Adjusting Current limit function: As shown in fig 14, a
combined 2.8KΩ internal resistance is connected into the
non-inverting lead on the PWM comparator. A external
resistance of Y on the current limit pin forms a parallel resistance with the 2.8KΩ when the internal diodes are biased by
the main current source of 900uA.
15
FSDL0365RN, FSDM0365RN
Typical application circuit
1. Set Top Box Example Circuit (20W Output Power)
12
2A/250V
C7
400V
/47u
FUSE
85VAC
~275VAC
100pF
/400V
C1
R3
56K/1/
4W
LF1
40mH
R1
47K
1
C8
6.8n/
1kV
D5
UF4007
KBP06M
100pF
/400V
C2
GreenFPS
PERFORMANCE SUMMARY
Output Power:
20W
Regulation
3.3V:
±5%
5.0V:
±5%
17.0V:
±7%
23.0:
±7%
Efficiency:
≥75%
No load Consumption:
0.12W at 230Vac
D
D
+23.0V
D12
R15
EGP20D 20R
D14
EGP20D
FSDM0365RN
6kR
C6
50V
47uF
4
D6
UF4004
5
C9
33n
50V
100uF
/50V
L3
100uF
/50V
C15
470uF
/35V
C16
220uF
/35V
+17.0V
D13
EGP20D
10
R4
30R
0.005~0.45A
11
3
5
D start
S VccVfb I_pk
1
R5
C17
6
SB360
D15
R21
330R
PC817
L2
C14
470uF
/10V
C13
1000uF
/16V
Q1 FOD2741A
PC817
R20
+3.3V
0.4~1.4A
C12
470uF
/10V
R14
R22
1KR 800R
R19
R13
2.7K
0.1uF/
monolithic
C209
R15 6.9K
TL431AZ
+5.0V
0.2~0.85A
L1
C11
1000uF
/16V
8
0.01~0.5A
R12
1.5K
Figure15. 20W multiple power supply using FSDM0365RN
Multiple Output, 20W, 85-265VAC Input Power supply:
Figure 15 shows a multiple output supply typical for high
end set-top boxes containing high capacity hard disks for
recording or LIPS(LCD Inverter Power Supply) for 15"
LCD monitor. The supply delivers an output power of 20W
cont./24 W peak (thermally limited) from an input voltage of
85 to 265 VAC. Efficiency at 20W, 85VAC is ≥75%.
The 3.3 V and 5 V outputs are regulated to ±5% without the
need for secondary linear regulators. DC stacking (the secondary winding reference for the other output voltages is
connected to the anode of D15. For more accuracy, connection to the cathode of D15 will be better.) is used to minimize
the voltage error for the higher voltage outputs. Due to the
high ambient operating temperature requirement typical of a
set-top box (60 °C) the FSDL0165RN is used to reduce conduction losses without a heatsink. Resistor R5 sets the device
current limit to limit overload power.
Leakage inductance clamping is provided by R1 and C8,
keeping the DRAIN voltage below 650 V under all conditions. Resistor R1 and capacitor C8 are selected such that R1
dissipates power to prevent rising of DRAIN Voltage caused
by leakage inductance. The frequency modulation feature of
FSDL0165RN allows the circuit shown to meet CISPR2AB
with simple EMI filtering (C1, LF1 and C2) and the output
grounded. The secondaries are rectified and smoothed by
D12, D13, D14,and D15. Diode D15 for the 3.4V output is a
Schottky diode to maximize efficiency. Diode D14 for the 5
V output is a PN type to center the 5 V output at 5 V. The 3.3
V and 5 V output voltage require two capacitors in parallel to
meet the ripple current requirement. Switching noise filter-
16
ing is provided by L3, L2 and L1. Resistor R15 prevents
peak charging of the lightly loaded 23V output. The outputs
are regulated by the reference (TL431) voltage in secondary.
Both the 3.3 V and 5 V outputs are sensed via R13 and R14.
Resistor R22 provides bias for TL431and R21 sets the overall DC gain. Resistor R21, C209, R14 and R13 provide loop
compensation.
FSDL0365RN, FSDM0365RN
2. Transformer Specification
1.
-
TR AN SFO RM ER SPECIFICATIO N
SCHEM ATIC DIAG R AM (TR ANSFO RM ER)
3mm
6mm
12
1
11
2
8
7
5
top
bottom
6
2.
W INDING SPEC IFIC ATIO N
PIN(S → F)
W IRE
TURNS
W INDING METHOD
N P/2
3 → 2
0.25 Φ × 1
22
SOLENOID W INDING
N 3.3V
6
→ 8
0.3 Φ × 8
2
STACK W INDING
N 5V
10 → 6
0.3 Φ × 2
1
STACK W INDING
N 16V
11 → 6
0.3 Φ × 4
7
SOLENOID W INDING
N 23V
12 → 11
0.3 Φ × 2
3
SOLENOID W INDING
N P/2
2 → 1
0.25 Φ × 1
22
SOLENOID W INDING
NB
4 → 5
0.25 Φ × 1
10
CENTER W INDING
NO.
3.
NB
N P/2
N 23V
N 17V
N 5V
N 3.3V
N P/2
10
3
4
ELECTRIC CH AR ACTERISTIC
CLOSURE
PIN
SPEC.
REM ARKS
INDUCTANCE
1-3
800uH ± 10%
1KHz, 1V
LEAKAGE L
1-3
15uH MAX.
2nd ALL SHORT
4. BOBBIN & CO R E.
CORE:
BO BBIN:
EER 2828
EER 2828
17
FSDL0365RN, FSDM0365RN
Layout Considerations
SURFACE MOUNTED
COPPER AREA FOR HEAT
SINKING
DC_link Capacitor
#1 : GND
#2 : VCC
#3 : Vfb
#4 : Ipk
#5 : Vstr
#6 : Drain
#7 : Drain
#8 : Drain
Y1CAPACITOR
- +
DC
OUT
Figure 15. Layout Considerations for FSDx0365RN using 8DIP
18
FSDL0365RN, FSDM0365RN
Package Dimensions
8DIP
19
FSDL0365RN, FSDM0365RN
Package Dimensions (Continued)
8LSOP
20
FSDL0365RN, FSDM0365RN
Ordering Information
Product Number
Package
Marking Code
BVDSS
FOSC
RDS(on)
FSDM0365RN
8DIP
DM0365R
650V
67KHz
3.6Ω
FSDL0365RN
8DIP
DL0365R
650V
50KHz
3.6Ω
FSDM0365RL
8LSOP
DM0365R
650V
67KHz
3.6Ω
FSDL0365RL
8LSOP
DL0365R
650V
50KHz
3.6Ω
21
FSDL0365RN, FSDM0365RN
DISCLAIMER
FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY
PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY
LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN; NEITHER
DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS.
LIFE SUPPORT POLICY
FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES
OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body,
or (b) support or sustain life, and (c) whose failure to
perform when properly used in accordance with
instructions for use provided in the labeling, can be
reasonably expected to result in a significant injury of the
user.
2. A critical component in any component of a life support
device or system whose failure to perform can be
reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.
www.fairchildsemi.com
6/17/04 0.0m 001
 2004 Fairchild Semiconductor Corporation
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