STMICROELECTRONICS TSM108ID

TSM108
AUTOMOTIVE SWITCH MODE
VOLTAGE AND CURRENT CONTROLLER
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APPLICATION DIAGRAM
MOSFET P
or PNP
TSM108
D
SO14
(Plastic Micropackage)
DESCRIPTION
TSM108 is a P-channel MOSFET controller which
ensures constant voltage and constant current in
Switching Mode Power Supply (step down) like in
automotive battery charging applications.
TSM108 can easily be configured for very wide
voltage and current needs.
TSM108 is built in rugged BCD technology and
includes a PWM generator, Voltage and Current
control loops, a precision Voltage Reference, and
a P-Mosfet Gate Drive output. TSM108 can
sustain 60V on Vcc, and therefore meet the
standard Load Dump requirements in the
Automotive field.
TSM108 includes security functions which lock the
PMosfet in OFF state: OVLO (Over Voltage
Lockout) and UVLO (Under Voltage Lockout). The
P-Mosfet Gate is also protected from over voltage
drive thanks to a 12V clamping protection circuit.
TSM108 includes a standby feature which allows
very low quiescent current when activated, as well
as safe P-Mosfet Off state.
TSM108 is suitable for car environment
accessories, as well as numerous other DC/DC
step down regulation.
November 2001
BATTERY
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POSITIVE LINE
CONSTANT VOLTAGE MODE CONTROL
CONSTANT CURRENT MODE CONTROL
PRECISION VOLTAGE AND CURRENT
CONTROL LOOPS
ADJUSTABLE SWITCHING FREQUENCY
ADJUSTABLE OVER VOLTAGE LOCKOUT
ADJUSTABLE UNDER VOLTAGE LOCKOUT
STANDBY MODE (LOW QUIESCENT
CURRENT)
SUSTAINS 60V
MINIMAL EXTERNAL COMPONENTS
COUNT
DRIVING ABILITY FOR EITHER P-MOSFET
OR PNP TRANSISTORS
DC INPUT
■ CURRENT MEASUREMENT ON OUTPUT
ORDER CODE
Package
Part Number
Temperature Range
D
TSM108I
•
-40°, +125°C
D = Small Outline Package (SO) - also available in Tape & Reel (DT)
PIN CONNECTIONS (top view)
VCC
1
14
GD
!STBY
2
13
VS
GND
3
12
ICTRL
UV
4
11
VCTRL
OV
5
10
VREF
G
6
9
ICOMP
OSC
7
8
VCOMP
1/13
TSM108
PIN DESCRIPTION
Name
Pin
Type
Description
VCC
1
Power Supply
Power Supply Line of the Device - Source of the P-MOSFET
GND
GD
3
14
Power Ground
Gate Drive
VREF
10
Output
0V Reference for all Voltages
Gate Drive Pin of the P-MOSFET - Middle Point of the MOSFET
Push Pull Output Stage
Voltage Reference Output
VS
13
HZ Input
Voltage Sense Resistor Input
ICTRL
12
HZ Input
Current Regulation Input
VCTRL
11
HZ Input
Voltage Regulation Input
VCOMP
8
Output
Compensation pin - Output of Voltage Control Op-Amp
ICOMP
9
Output
Compensation pin - Output of Current Control Op-Amp
OSC
!STBY
UV
7
2
4
Input
Input
I/O
OV
5
I/O
G
6
Test Pin
Oscillator Frequency Set Capacitor
Standby Command (Command = 0V ===> Device Standby)
Programmable Under Voltage Lockout. The middle point of the
integrated resistor bridge is accessible. Preset value is 8V min.
Programmable Over Voltage Lockout. The middle point of the
integrated resistor bridge is accessible. Preset value is 33V max.
Internally Connected to Ground
ABSOLUTE MAXIMUM RATINGS
Symbol
VCC
Tj
Parameter
Value
Unit
Supply Voltage
60
V
Maximum Junction Temperature
150
°C
Rthja
Thermal Resistance Junction to Ambient (SO package)
Tamb
Ambient Temperature
Vmax
Out Terminal Voltage (ICTRL, VS)
130
°C/W
-55 to +125
°C
10
V
Value
Unit
UVLO to OVLO
V
OPERATING CONDITIONS
Symbol
Parameter
VCC
Supply Voltage
Vter1
Out Terminal Voltage (ICTRL, VS)
0 to 9
V
Vter2
Out Terminal Voltage (UV, OV, OSC)
0 to 6
V
2/13
TSM108
ELECTRICAL CHARACTERISTICS
Tamb = 25°C, VCC = 12V (unless otherwise specified)
Symbol
Parameter
Test Condition
Min.
Typ.
Max.
Unit
4
7
mA
CURRENT CONSUMPTION
ICC
Current Consumption
STANDBY
Istby
Current Consumption in Standby Mode
Vsh
Input Standby Voltage High Impedance
Vsl
Input Standby Voltage Low
µA
150
Internal Pull up resistor.
Stby pin should be left
open
2
V
0.8
V
130
kHz
OSCILLATOR
FOSC
Frequency of the Oscillator
VOLTAGE CONTROL
Vref
COSC = 220pF
70
100
1) 2)
Voltage Control Reference
Tamb = 25°C
-25°C < Tamb < 85°C
2.450
Tamb = 25°C
-25°C < Tamb < 85°C
196
191
Tamb = 25°C
-25°C < Tamb < 85°C
15
Tamb = 25°C
-25°C < Tamb < 85°C
30
2.520
V
2.590
CURRENT CONTROL 3) 4) 5)
Vsense
Current Control Reference Voltage
206
216
221
mV
GATE DRIVE - P CHANNEL MOSFET DRIVE
Isink
Isource
Cload
Sink Current - Switch ON
Source Current - Switch OFF
40
mA
80
mA
Input Capacitance of the PMOSFET 6)
1
1.5
nF
Maximum Duty Cycle of the PWM function
95
100
%
9
V
PWM
∆max.
UVLO
UV
UVhyst
Ruvu
Ruvl
Under Voltage Lock Out 7)
UVLO Voltage Hysteresis - low to high
-25°C < Tamb < 85°C
8
200
mV
Upper Resistor of UVLO bridge
Lower Resistor of UVLO bridge (see note 8)
Tamb = 25°C
184
kΩ
Tamb = 25°C
76.5
kΩ
Over Voltage Lock Out (see note 7)
-25°C < Tamb < 85°C
8)
OVLO
OV
OVhyst
OVLO Voltage Hysteresis - low to high
32
35
V
400
mV
Rovu
Upper Resistor of OVLO bridge (see note 8)
Tamb = 25°C
275
kΩ
Rovl
Lower Resistor of OVLO bridge (see note 8)
Tamb = 25°C
23.2
kΩ
1. Vref parameter indicates global precision of the voltage control loop.
2. Control Gain : A v = 95dB ; Input Resistance : Rin = infinite ; Output Resistance : Rout = 700MΩ ; Output Source/Sink Current :
Iso, Isi = 150µA ; Recommended values for the compensation network are : 22nF & 22kΩ in series between output and ground.
3. Vsense parameter indicated global precision of the current control loop.
4. Control Gain : A v = 105dB ; Input Resistance : Rin =380kΩ ; Output Resistance : Rout = 105MΩ ; Output Source/Sink Current :
Iso, Isi = 150µA ; Recommended values for the compensation network are : 22nF & 22kΩ in series between output and ground.
5. A current foldback function is implemented thanks to a systematic -6mV negative offset on the current amplifier inputs which
protects the battery from over charging current under low battery voltage conditions, or output short circuit conditions.
6. The Gate Drive output stage has been optimized for PMosfets with input capacitance equal to Cload. A bigger Mosfet (with input
capacitance higher than Cload) can be used with TSM108, but the gate drive performances will be reduced (in particular when
reaching the Dmax. PWM mode).
7. The given limits comprise the hysteresis (UV hyst).
8. It is possible to modify the UVLO and OVLO limits by adding a resistor (to ground or to V CC) on the pins UV and OV.
The internal values of the resistor should be taken into account
3/13
TSM108
DETAILED INTERNAL SCHEMATIC
14
GD
15V
VCC
1
Protection block
maximum duty cycle = 95%
Vsense
200mV
!STBY
VS
2
13
10kΩ
20µA
VCC
UV
ICTRL
Ruvu
184kΩ
4
ICOMP
Ruvl
76,5kΩ
VCOMP
VREF
VCTRL
VCC
OV
VREF
Rovu
275kΩ
5
VREF
2,52V
Rovl
23,2kΩ
Oscillator block
4,5V
TSM108
4/13
GND
OSC
7
3
G
6
12
9
8
11
10
TSM108
OSCILLATOR FREQUENCY VERSUS TIMING CAPACITOR
Cosc Timing Capacitor (pF)
350
300
250
200
150
100
50
0
10
100
Oscilla tor fre que ncy (kHz )
1000
TSM108 AS A STAND ALONE DC/DC CONVERTER FOR CIGARETTE LIGHTER ACCESSORIES
DC INPUT
1
Q1
MOSFET P
L1
Rsense
1
D
Csupply
1
load
L2
Rf
DC OUTPUT
Cf
1
Vcc
+
Gnd
-
UV
OV
Cosc
Osc
Vs
Ictrl
Icomp
-
R1
Vcomp
Oscillator
!Stby
Uv/Ov/Stby
+
Vs
+
G
Vref
Vctrl
Comp
Comp
R2
5/13
TSM108
PRINCIPLE OF OPERATION AND APPLICATION HINTS
Description of a DC/DC step down battery
charging application
1. Voltage and Current Controller
TSM108 is designed to drive a P-Channel
MOSFET transistor in Switch Mode Step Down
Converter applications. Its two integrated
operational amplifiers ensure accurate Voltage
and Current Regulation.
The Voltage Control dedicated operational
amplifier acts as an error amplifier and compares
a part of the output voltage (external resistor
bridge) to an integrated highly precise voltage
reference (Vref).
The Current Control dedicated operational
amplifier acts as an error amplifier and compares
the drop voltage through the sense resistor to an
integrated low value voltage reference (Vs).
These two amplified errors are ORed through
diodes, and the resulting signal (“max of”) is a
reference for the PWM generator to set the
switching duty cycle of the P-Channel MOSFET
transistor.
The PWM generator comprises an oscillator (saw
tooth) and a comparator which gives a variable
duty cycle from 0 to 95%. This PWM signal is the
direct command of the output Push Pull stage to
drive the Gate of the P-Channel MOSFET.
Thanks to this architecture, the TSM108 is ideal to
be used from a DC power supply to control the
charging Voltage and Current of a battery in
applications such as Automotive accessories for
Portable Phone charging and power supplies.
2. Voltage Control
The Voltage Control loop is to be set thanks to an
external resistor bridge connected between the
output positive line and the Ground reference. The
middle point is to be connected to the Vctrl pin of
TSM108, and, if R1 is the upper resistor, and R2,
the lower resistor of the bridge, the values of R1
and R2 should follow:
❑ eq1: Vref = Vout x R2 / (R1 + R2)
When under Constant Voltage Control mode, the
output voltage is fixed thanks to the R1/R2 resistor
bridge.
The total value of R1 + R2 resistor bridge will
determine the necessary bleeding current to keep
the Voltage Control loop effective, even under “no
load” conditions.
The voltage compensation loop is directly
accessible from the pins Vcomp and Vref
(negative input of the Voltage Control dedicated
operational amplifier). The compensation network
is highly dependant of the conditions of use of the
TSM108
(switching
frequency,
external
components (R, L, C), MOSFET, output
capacitor...).
3. Current Control
The Current control loop is to be set thanks to the
Sense resistor which is to be placed in series on
the output positive line. The output side of the
Sense resistor should be connected to the Ictrl pin
of TSM108, and the common point between
Rsense and the filtering self L should be
connected to the Vs pin of TSM108. If Ilim is the
value of the charging current limit The value of
Rsense should verify:
❑ eq2: V s = Rsense x Ilim
When under Constant Current Control mode, the
output current is fixed thanks to the Rsense
resistor (under output short circuit conditions,
please refer to this corresponding section).
The wattage calibration (W) of the sense resistor
should be chosen according to:
❑ eq2a: W > R sense x Ilim2
The current compensation loop is directly
accessible from the pins Icomp and Ictrl (negative
input of the Current Control dedicated operational
amplifier.
The compensation network is highly dependant of
the conditions of use of the TSM108 (switching
frequency, external components (R, L, C),
MOSFET, output capacitor...).
4. PWM frequency
The internal oscillator of TSM108 is a saw tooth
waveform that can be frequency adjusted.
In automotive accessory battery charging
applications, it is recommended to set the
switching frequency at a typical 100kHz in order to
6/13
TSM108
obtain the best compromise between electrical
noise, and size of the filtering self.
An external capacitor is to be connected between
ground and the Osc pin of TSM108 to set the
switching frequency.
The maximum duty cycle of the PWM function is
limited to 95% in order to ensure safe driving of
the MOSFET.
OVLO is internally programmed to ensure 32V
min. and 33V max., but the middle point of the
integrated resistor bridge is accessible and the
value of the OVLO is therefore adjustable by
adding an external resistor to modify the bridge
ratio.
The resistor typical values of the bridge are given
(Rovh, Rovl).
5. Gate Drive
Examples:
The Gate Drive stage is directly commanded from
the PWM output signal. The Gate Drive stage is a
Push Pull Mosfet stage which bears different On
resistances in order to ensure a slower turn ON
than turn OFF of the P-Channel MOSFET. The
values of the output Gate Drive currents are given
by Isink (switch ON) and Isource (switch OFF).
The Gate Drive stage bears an integrated voltage
clamp which will prevent the P-Channel MOSFET
gate to be driven with voltages higher than 15V
(acting like a zener diode between Vcc and GD
(Gate Drive) pin.
6. Under Voltage Lock-Out, Over Voltage
Lock-Out
The UVLO and OVLO security functions aim at the
global application security.
When the Power supply decreases, there is the
inherent risk to drive the P-Channel MOSFET with
insufficient Gate voltage, and therefore to lead the
MOSFET to linear operation, and to its
destruction.
The UVLO is an input power supply voltage
detection which imposes a complete switch OFF
of the P-Channel MOSFET as soon as the Power
Supply decreases below UV. To avoid unwanted
oscillation of the MOSFET, a fixed hysteresis
margin is integrated (UVhyst).
UVLO is internally programmed to ensure 8V min
and 9V max, but the middle point of the integrated
resistor bridge is accessible and the value of the
UVLO is therefore adjustable by adding an
external resistor to modify the bridge ratio. The
resistor typical values of the bridge are given
(Ruvh, Ruvl).
When the Power supply increases, there is the
inherent risk to dissipate too much conduction
energy through the P-Channel MOSFET, and
therefore to lead to its destruction.
The OVLO is an input power supply voltage
detection which imposes a complete switch OFF
of the P-Channel MOSFET as soon as the Power
Supply increases above OV. To avoid unwanted
oscillation of the MOSFET, a fixed hysteresis
margin is integrated (OVhyst).
Let’s suppose that the internally set value of the
UVLO and / or OVLO level should be modified in a
specific
application,
or
under
specific
requirements.
6.1. UVLO decrease:
If the UVLO level needs to be lowered (UV1), an
additional resistor (Ruvh1) must be connected
between UV and Vcc following the equation:
❑ UV = Vref (Ruvh/Ruvl +1)
❑ UV1 = Vref ((Ruvh//Ruvh1)/Ruvl +1)
(i)
where Ruvh//Ruvh1 means that Ruvh1 is in
parallel to Ruvh
Solving i. we obtain:
❑ Ruvh1 = Ruvl x Ruvh (UV1 - Vref) / (Vref x
Ruvh - Ruvl (UV1 - Vref))
As an example, if UV1 needs to be set to 6V,
Ruvh1 = 256kΩ
6.2. UVLO increase:
If the UVLO level needs to be increased (UV2), an
additional resistor (Ruvl2) must be connected
between UV and Gnd following the equation.
❑ UV = Vref (Ruvh/Ruvl +1)
❑ UV1 = Vref (Ruvh/(Ruvl//Ruvl2) +1)
(ii)
where Ruvl//Ruvl2 means that Ruvl2 is in parallel
to Ruvl
Solving ii. we obtain:
❑ Ruvl2 = Vref x Ruvh Ruvl / (UV2 x Ruvl Vref x (Ruvh + Ruvl))
As an example, if UV2 needs to be set to 12V,
Ruvl2 = 132kΩ
6.3. OVLO decrease:
If the OVLO level needs to be lowered (OV1), an
additional resistor (Rovh1) must be connected
between OV and Vcc following the equation:
❑ OV = Vref (Rovh/Rovl +1)
❑ OV1 = Vref ((Rovh//Rovh1)/Rovl +1)
(iii)
where Rovh//Rovh1 means that Rovh1 is in
parallel to Rovh
Solving iii. we obtain:
❑ Rovh1 = Rovl x Rovh (OV1 - Vref) / (Vref x
Rovh - Rovl (OV1 - Vref))
As an example, if OV1 needs to be set to 25V,
Rovh1 = 867kΩ
7/13
TSM108
6.4. OVLO increase:
If the OVLO level needs to be increased (OV2), an
additional resistor (Rovl2) must be connected
between OV and Gnd following the equation.
❑ OV = Vref (Rovh/Rovl +1)
❑ OV2 = Vref (Rovh/(Rovl//Rovl2) +1)
(iv)
where Rovl//Rovl2 means that Rovl2 is in parallel
to Rovl
Solving iv. we obtain:
❑ Rovl2 = Vref x Rovh Rovl / (OV2 x Rovl Vref x (Rovh + Rovl))
As an example, if OV2 needs to be set to 40V,
Rovl2 = 87kΩ
Q1
D1
L1
GD
TSM108
Q1
D1
L1
7. Standby Mode
In order to reduce to a minimum the current
consumption of the TSM108 when in inactive
phase, the Standby mode (!STBY pin of TSM108)
imposes a complete OFF state of the P-Channel
MOSFET, as well as a complete shut off of the
main functions of the TSM108 (operational
amplifier, PWM generator and oscillator, UVLO
and OVLO) and therefore reduces the
consumption of the TSM108 to the Istby value.
This !STBY command is TTL compatible, which
means that it can be directly commanded from
whatever logic signal.
8.Power Transistor: P-MOSFET or PNP
Transistor?
The TSM108 can drive, with minor external
components change, either a P-channel
MOSFET, or a PNP transistor. The choice of the
transistor is completely to the user’s responsibility,
nevertheless, here follows a few elements which
will help to decide which is the most adapted
transistor to drive depending on the application
characteristics in terms of power and
performances.
The following figures shows two different
schematics where both driving abilities of TSM108
are shown. The third schematic shows how to
improve the switch off commutation when using a
bipolar PNP transistor.
P- MOSFET? PNP Transistor?
MOSFET P
Q1
GD
TSM108
8/13
D1
L1
GD
TSM108
The most immediate way to choose from a
P-channel MOSFET or a PNP transistor is to
consider the ratio between the output power of the
application and the expected components price:
the lower the power, the more suitable the PNP
transistor is; the higher the power, the more
suitable the P-channel MOSFET is. As an
example, for a DC/DC adaptor built for 12V/6V,
the recommended limit to choose from one to the
other is situated around 200mA.
Below 200mA, the price/performance ratio of the
PNP transistor is very attractive, whereas above
200mA, the P-channel Mosfet takes the
advantage.
9. Calculation of the Passive Elements
Let’s consider the following characteristics for a
Cigarette Lighter Cellular Phone Battery Charger:
Vin = 12V - input voltage of the converter
Vout = 6V - output voltage of the converter
F = 100kHz - switching frequency of the converter
adjustable with an external capacitor
Iout = 625mA - output current limitation
9.1. Inductor
The minimum inductor value to choose should
apply to
Lmin = (1 - D) R / 2F
where R = Vout / Iout = 9.6Ω
and where D = Vout / Vin = 0.5
Therefore, Lmin = 24µH.
TSM108
This component value is valid if the above
described characteristics are fixed... but in the
automotive field, the input voltage of the converter
is dependant of the car battery conditions. Also,
the frequency may vary depending on the
temperature, due to the fact that the frequency is
fixed by an external capacitor. Therefore, we must
calculate the inductor value considering the worst
case condition in order to avoid the saturation of
the inductor, which is when the battery voltage is
at it’s highest, and the switching frequency at it’s
lowest. Thanks to the OVLO function integrated in
TSM108, the operation of the DC/DC converter
will be stopped as soon as the voltage exceeds
the OVLO level. Let’s suppose the OVLO pin has
been left open, therefore, the maximum input
voltage of the DC/DC converter will be Vin max. =
32V. Frequency min stands in the range of 75kHz
In this case, D = 6 / 32 = 0.1875, therefore Lmin =
52µH.
If we allow a 25% security margin
Lmin = 68µH
Losses in the switch are:
Pswitch = Prise + Pfall + Pon
where Prise + Pfall represent the switching losses
and where Pon represents the conduction losses.
Prise + Pfall = Iout x Vin x (Trise + Tfall) x F / 2
Pon = Ron x Iout² x d
where Trise is the switching on time, and Tfall is
the switching off time, and where d is the duty
cycle of the switching profile, which can be
approximated to 1 under full load conditions.
With the two last equations, we can see easily that
what we may gain by choosing a performing low
Rdson P-channel MOSFET (for example) may be
jeopardized by the long on and off switching times
required when using a large input gate
capacitance.
10. Electromagnetic Compatibility
The small schematic hereafter shows how to
reduce the EMC noise when used in an EMC
sensitive environment:
EMC Improvement
9.2. Capacitor
The capacitor choice will depend mainly on the
accepted voltage ripple on the output
Ripple = DVout / Vout = (1-D) / 8LCF²
Therefore, C = (1-D) / 8LRippleF². If C = 22µF,
then Ripple = 0.4% which should be far
acceptable.
Here again, the worst conditions for the ripple are
set when the input voltage is at the highest (32V)
and the frequency at it's lowest (75kHz).
with C = 22µF, Ripple = 1.2%
9.3. Ratings for the Inductor, Capacitor,
Transistor and Diode
The inductor wire must be rated at the rms current,
and the core should not saturate for peak inductor
current. The capacitor must be selected to limit the
output ripple to the design specifications, to
withstand peak output voltage, and to carry the
required rms current.
The transistor and the diode should be rated for
the maximum input voltage (up to 60V in
automotive applications). The diode recovery time
must be in accordance with the time period and
the maximum authorized switching time of the
power transistor.
A compromise between the switching and
conducting performances of the transistor must be
found, because choosing a very low ohmic Mosfet
aiming at the benefit of low conduction losses may
bring much higher switching losses than the
expected benefit.
MOSFET P
Q1
D1
L1
GD
TSM108
The RC components should realize a time
constant corresponding to one tenth of the
switching time constant of the TSM108 (i.e. in our
example, the oscillator frequency is set to 10µs
corresponding to 100kHz, therefore, the RC
couple should realise a time constant close to
1µs).
Choosing the components must privilege a rather
small resistivity (between 10 to 100W). A guess
couple of values for RC in our example would be:
R= 22W, C= 47nF
11. Efficiency Calculations (rough estimation)
The following gives a rough estimation of the
efficiency of a car phone charger, knowing that the
exact calculations depend on a lot of parameters,
as well as on a wide choice of external
components.
Let’s consider the following characteristics of a
classical car phone charger application:
9/13
TSM108
❑ Vin = Vcc = 12V, Iout = 625mA, Vout = 6V
❑ Mosfet: Pchannel Mosfet: Rdson = 100mΩ,
Ciss = 1nF.
Driver: TSM108
PWM frequency: 100kHz
Free wheel diode: Vf = 0.7V
Shunt: Rsense = 330mΩ
The efficiency (η) of a regulator is defined as the
ratio of the charging power (Pout) to the total
power from the supply (Pin).
❑ Eq3: η = Pout/Pin
The output power is:
Pout=Iout x Vout where Iout is the charging
current (Vsense/Rsense = 625mA at full load) and
Vout is the regulated voltage (Vref(1+R1/R2) =
6V).
Pout = 3.75W
The input power can be found by adding the
output power (Pout) to the total power loss in the
circuit (Plosses) i.e.
❑ Pin = Pout + Plosses
The power is lost partly on the chip and partly on
the external components which are mainly the
diode, the switch and the shunt. Plosses = Pchip +
Pswitch + Pdiode + Pshunt.
In Plosses, we neglect the losses in the inductor
(because the current through the inductor is
smoothened making the serial resistor of the
inductor very low), and the losses in the Gate
(charge and discharge).
a. The power lost in the chip is Pchip = Vcc x Icc.
(Vcc = 12V, Icc = 6mA) Pchip = 72mΩ
b. The power lost in the switch depends on the ON
resistance of the switch and the current passing
through it. Also there is power loss in the switch
during switching time (commutation losses) and
that depends on the switching frequency and the
rise and fall time of the switching signal.
Rise time (Pchannel goes off) depends on the
output source current of the TSM108 and the input
gate capacitance of the Mosfet.
❑
❑
❑
❑
Trise = Ciss x Vgate / Isource
Fall time (Pchannel goes on) depends on the
output sink current of the TSM108 and the input
gate capacitance of the Mosfet .
Tfall = Ciss x Vgate / Isink
Trise = 150ns and Tfall = 300ns (Vgate is approx
12V).
❑ Pswitch = Prise + Pfall + Pon
where:
Prise = Iout x (Vcc+Vf) x Trise x PWMfreq / 2
Prise = 625mA x 12.7 x 150ns x 100kHz / 2.
Prise = 59.5mW
where:
Pfall = Iout x (Vcc+Vf) x Tfall x PWMfreq / 2
Pfall = 625mA x 12.7 x 300ns x 100kHz / 2.
Pfall = 119.1mW
where:
Pon = Rdson x Iout² x D (where D is the duty cycle
- at full charge, D can be approximated to 1)
Pon = 100mΩ x 625mA². Pon = 39.1mW
❑ Pswitch = 217.7mW
c. The power lost in the fly back diode is Pdiode =
Vf x Iout(1-D) where D = Vout/Vcc = 6/12. D = 0.5
❑ Pdiode=219mW
d. the power lost in the sense resistor (shunt
resistor) is Pshunt = Rsense x Iout²
❑ Pshunt = 129mW
Therefore,
Plosses = Pchip+Pswitch+Pdiode+Pshunt
= 72mW + 217.7mW + 219mW + 129mW
❑ Plosses = 638mW
The yield (efficiency) is
❑ Pout / Pin = 3.75 / (3.75 + 0.638) = 85.5%
η = 85.5%
The following table gives a tentative efficiency
improvement view following the choice of some
external components. Be aware that some of the
following choices have non negligible cost effects
on the total application.
Improved efficiency - by changing the external components value one by one
Rsense
Iout
Vout (R1/R2)
Rdson
Ciss
PWM Freq
Free Wheel
Yield
Cost influence
10/13
330mΩ
625mA
6V
100mΩ
0nF
100kHz
0.7V
85.5%
-
220mΩ
936mA
85.6%
=
7.5V
88.9%
=
140mΩ
0.85nF
85.7%
<
50kHz
87.3%
>
0.3V
88.1%
>>
TSM108
12. Measured Performances
The few following curves show the measured
performances of TSM108 used in DC/DC step
down converter, either with a Pchannel MOSFET
or with a PNP bipolar transistor.
12.1. Voltage and Current Control, Efficiency Performances using a Pchannel MOSFET:
CV & CC Regulation - Switching duty cycle vs Iout
7
70%
6
60%
5
50%
4
40%
3
30%
2
20%
1
10%
0
0
0.1
0.2
0.3
0.4
0.5
0.6
duty cycle on (%)
Vout (V)
Vin = 12V, Vout = 6V, Iout = 600mA
0%
0.7
Iout (A)
Converter efficiency & Switching duty cycle vs Iout
100%
90%
90%
80%
80%
70%
70%
60%
60%
50%
50%
40%
40%
30%
30%
20%
20%
10%
10%
0%
duty cycle on (%)
Efficiency (%)
Vin = 12V, Vout = 6V, Iout = 600mA
100%
0%
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
Iout (A)
Vout & Efficiency versus Vin
6.001
86%
6
85%
5.999
84%
5.998
83%
5.997
82%
5.996
81%
5.995
80%
5.994
79%
5.993
78%
5.992
Efficiency (%)
Vout (V)
Vout = 6V, Iout = 600mA
77%
5
10
15
20
25
30
35
Vin (V)
11/13
TSM108
12.2. Voltage and Current Control, Efficiency Performances using a PNP bipolar transistor
Vout & duty cycle versus Iout
Vin=12V, Vout=6V, Iout=200mA
70%
6
60%
5
50%
4
40%
3
30%
2
20%
1
10%
0
0
0.05
0.1
0.15
duty cycle on (%)
Vout (V)
PNP transistor Rbase = 220 L=150µH
7
0%
0.25
0.2
Iout (A)
Efficiency & duty cycle versus Iout
Vin=12V, Vout=6V, Iout=200mA
80%
70%
70%
60%
60%
50%
50%
40%
40%
30%
30%
20%
20%
10%
10%
0%
0
0.05
0.1
0.15
0%
0.25
0.2
Iout (A)
Vout & Efficiency versus Vin
Vout = 6V, Iout = 200mA
6.05
85%
6.045
80%
6.04
75%
6.035
70%
6.03
65%
6.025
60%
5
10
15
20
Vin (V)
12/13
25
30
35
Efficicency (%)
Vout (V)
PNP transistor Rbase =220 L=150µH
duty cycle on (%)
Efficiency (%)
PNP transistor Rbase = 220 L=150µH
80%
TSM108
PACKAGE MECHANICAL DATA
14 PINS - PLASTIC MICROPACKAGE (SO)
Dim.
A
a1
a2
b
b1
C
c1
D (1)
E
e
e3
F (1)
G
L
M
S
Millimeters
Min.
Typ.
Inches
Max.
Min.
1.75
0.2
1.6
0.46
0.25
0.1
0.35
0.19
Typ.
0.004
0.014
0.007
0.5
Max.
0.069
0.008
0.063
0.018
0.010
0.020
45° (typ.)
8.55
5.8
8.75
6.2
0.336
0.228
1.27
7.62
3.8
4.6
0.5
0.344
0.244
0.050
0.300
4.0
5.3
1.27
0.68
0.150
0.181
0.020
0.157
0.208
0.050
0.027
8° (max.)
Note : (1) D and F do not include mold flash or protrusions - Mold flash or protrusions shall not exceed 0.15mm (.066 inc) ONLY FOR DATA BOOK.
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences
of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted
by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject
to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not
authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics.
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