OPA2634 ® OPA 263 4 For most current data sheet and other product information, visit www.burr-brown.com Dual, Wideband, Single-Supply OPERATIONAL AMPLIFIER TM FEATURES DESCRIPTION ● ● ● ● ● ● ● The OPA2634 is a dual, low power, voltage-feedback, high-speed operational amplifier designed to operate on +3V or +5V single-supply voltage. Operation on ±5V or +10V supplies is also supported. The input range extends below ground and to within 1.2V of the positive supply. Using complementary common-emitter outputs provides an output swing to within 30mV of ground and 140mV of positive supply. Low distortion operation is ensured by the high gain bandwidth (140MHz) and slew rate (250V/µs). This makes the OPA2634 an ideal differential input buffer stage to 3V and 5V CMOS converters. Unlike other low power, single-supply operational amplifiers, distortion performance improves as the signal swing is decreased. A low 5.6nV/√Hz input voltage noise supports wide dynamic range operation. The OPA2634 is available in an industry-standard dual pinout SO-8 package. Where a single channel, singlesupply operational amplifier is required, consider the OPA634 and OPA635. Where lower supply current and speed are required, consider the OPA2631. HIGH BANDWIDTH: 150MHz (G = +2) +3V AND +5V OPERATION INPUT RANGE INCLUDES GROUND 4.8V OUTPUT SWING ON +5V SUPPLY HIGH OUTPUT CURRENT: 80mA HIGH SLEW RATE: 250V/µs LOW INPUT VOLTAGE NOISE: 5.6nV/√HZ APPLICATIONS ● ● ● ● ● DIFFERENTIAL RECEIVERS/DRIVERS ACTIVE FILTERS MATCHED I AND Q CHANNEL AMPLIFIERS CCD IMAGING CHANNELS LOW POWER ULTRASOUND +3V 2.26kΩ +3V 374Ω Q 1/2 OPA2634 100Ω ADS900 10-Bit 20Msps RELATED PRODUCTS 22pF 562Ω 750Ω SINGLES DUALS Medium Speed, No Disable With Disable OPA631 OPA632 OPA2631 — High Speed, No Disable With Disable OPA634 OPA635 OPA2634 — +3V 2.26kΩ +3V 374Ω I 1/2 OPA2634 100Ω ADS900 10-Bit 20Msps 22pF 562Ω 750Ω International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 Twx: 910-952-1111 • Internet: http://www.burr-brown.com/ • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132 © 1999 Burr-Brown Corporation PDS-1466A Printed in U.S.A. September, 1999 SPECIFICATIONS: VS = +5V At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS /2, unless otherwise noted (see Figure 1). OPA2634U TYP PARAMETER AC PERFORMANCE (Figure 1) Small-Signal Bandwidth Gain Bandwidth Product Peaking at a Gain of +1 Slew Rate Rise Time Fall Time Settling Time to 0.1% Spurious Free Dynamic Range Input Voltage Noise Input Current Noise NTSC Differential Gain NTSC Differential Phase Channel-to-Channel Crosstalk DC PERFORMANCE Open-Loop Voltage Gain Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Offset Current Input Offset Current Drift INPUT Least Positive Input Voltage Most Positive Input Voltage Common-Mode Rejection Ratio (CMRR) Input Impedance Differential-Mode Common-Mode OUTPUT Least Positive Output Voltage Most Positive Output Voltage Current Output, Sourcing Current Output, Sinking Short-Circuit Current (output shorted to either supply) Closed-Loop Output Impedance POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power Supply Rejection Ratio (PSRR) GUARANTEED CONDITIONS +25°C +25°C 0°C to 70°C –40°C to +85°C UNITS MIN/ MAX TEST LEVEL(1) G = +2, VO ≤ 0.5Vp-p G = +5, VO ≤ 0.5Vp-p G = +10, VO ≤ 0.5Vp-p G ≥ +10 VO ≤ 0.5Vp-p G = +2, 2V Step 0.5V Step 0.5V Step G = +2, 1V Step VO = 2Vp-p, f = 5MHz f > 1MHz f > 1MHz 150 36 16 140 5 250 2.4 2.4 15 63 5.6 2.8 0.10 0.16 < –90 100 24 11 100 — 170 3.4 3.5 19 56 6.2 3.8 — — — 84 20 10 82 — 125 4.7 4.5 22 51 7.3 4.2 — — — 78 18 8 75 — 115 5.2 4.8 23 50 7.7 5 — — — MHz MHz MHz MHz dB V/µs ns ns ns dB nV/√Hz pA/√Hz % degrees dB min min min min typ min max max max min max max typ typ typ B B B B C B B B B B B B C C C 63 — 45 ±2 — 60 ±8 — 55 ±2.3 — 53 ±10 4.6 80 ±4 15 dB mV µV/°C µA µA nA/°C min max max max max max A A B A A B –0.24 3.8 78 –0.1 3.5 75 –0.05 3.45 73 –0.01 3.4 65 V V dB max min min B A A 10 || 2.1 400 || 1.2 — — — — — — kΩ || pF kΩ || pF typ typ C C G = +2, f ≤ 100kHz 0.03 0.1 4.86 4.65 80 100 100 0.2 0.08 0.16 4.8 4.55 50 80 — — 0.09 0.17 4.75 4.5 45 65 — — 0.10 0.24 4.7 4.4 20 20 — — V V V V mA mA mA Ω max max min min min min typ typ A A A A A A C C VS = +5V, Each Channel VS = +5V, Each Channel Input Referred — — 12 12 55 2.7 10.5 12.5 11.3 52 2.7 10.5 13 9.75 50 2.7 10.5 13.25 8.5 49 V V mA mA dB min max max min min A A A A A –40 to +85 — — — °C typ C 125 — — — °C/W typ C Input Referred, 5MHz RL = 150Ω 66 ±3 — 25 ±0.6 — VCM = 2.0V VCM = 2.0V Input Referred RL = 1kΩ to 2.5V RL = 150Ω to 2.5V RL = 1kΩ to 2.5V RL = 150Ω to 2.5V THERMAL CHARACTERISTICS Specification: U Thermal Resistance U SO-8 ±7 NOTE: (1) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. ® OPA2634 2 SPECIFICATIONS: VS = +3V At TA = 25°C, G = +2 and RL = 150Ω to VS/2, unless otherwise noted (see Figure 2). OPA2634U TYP PARAMETER AC PERFORMANCE (Figure 2) Small-Signal Bandwidth Gain Bandwidth Product Peaking at a Gain of +1 Slew Rate Rise Time Fall Time Settling Time to 0.1% Spurious Free Dynamic Range Input Voltage Noise Input Current Noise Channel-to-Channel Crosstalk DC PERFORMANCE Open-Loop Voltage Gain Input Offset Voltage Average Offset Voltage Drift Input Bias Current Input Offset Current Input Offset Current Drift INPUT Least Positive Input Voltage Most Positive Input Voltage Common-Mode Rejection Ratio (CMRR) Input Impedance Differential-Mode Common-Mode OUTPUT Least Positive Output Voltage Most Positive Output Voltage Current Output, Sourcing Current Output, Sinking Short-Circuit Current (output shorted to either supply) Closed-Loop Output Impedance POWER SUPPLY Minimum Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power Supply Rejection Ratio (PSRR) GUARANTEED CONDITIONS +25°C +25°C 0°C to 70°C –40°C to +85°C UNITS G = +2, VO ≤ 0.5Vp-p G = +5, VO ≤ 0.5Vp-p G = +10, VO ≤ 0.5Vp-p G ≥ +10 VO ≤ 0.5Vp-p 1V Step 0.5V Step 0.5V Step 1V Step VO = 1Vp-p, f = 5MHz f > 1MHz f > 1MHz Input Referred, 5MHz 110 39 16 150 5 215 2.8 3.0 14 65 5.6 2.8 < –90 77 24 12 100 — 160 4.3 4.4 30 56 6.2 3.7 — 65 20 10 85 — 123 4.5 4.6 32 52 7.3 4.2 — 58 19 8 80 — 82 6.3 6.0 38 47 7.7 4.4 — MHz MHz MHz MHz dB V/µs ns ns ns dB nV/√Hz pA/√Hz dB min min min min typ min max max max min max max typ B B B B C B B B B B B B C RL = 150Ω 65 ±1.5 — 25 ±0.6 — 61 — 42 ±2 — 59 ±5 — 55 ±2.3 — 55 ±6 46 60 ±4 40 dB mV µV/°C µA µA nA/°C min max max max max max A A B A A B –0.25 1.8 75 –0.1 1.6 67 –0.05 1.55 64 –0.01 1.5 61 V V dB max min min B A A 10 || 2.1 400 || 1.2 — — — — — — kΩ || p kΩ || p typ typ C C Figure 2, f < 100kHz 0.03 0.08 2.9 2.8 45 65 100 0.2 0.06 0.16 2.86 2.7 35 30 — — 0.07 0.19 2.85 2.67 30 27 — — 0.08 0.43 2.45 2.2 12 10 — — V V V V mA mA mA Ω max max min min min min typ typ A A A A A A C C VS = +3V, Each Channel VS = +3V, Each Channel Input Referred — — 10.8 10.8 50 2.7 10.5 11.1 10.1 48 2.7 10.5 11.4 8.6 45 2.7 10.5 11.6 8.0 44 V V mA mA dB min max max min min A A A A A –40 to +85 — — — °C typ C 125 — — — °C/W typ C VCM = 1.0V VCM = 1.0V Input Referred RL = 1kΩ to 1.5V RL = 150Ω to 1.5V RL = 1kΩ to 1.5V RL = 150Ω to 1.5V THERMAL CHARACTERISTICS Specification: U Thermal Resistance U SO-8 ±4 MIN/ TEST MAX LEVEL(1) NOTE: (1) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems. ® 3 OPA2634 ABSOLUTE MAXIMUM RATINGS ELECTROSTATIC DISCHARGE SENSITIVITY Power Supply ................................................................................ +11VDC Internal Power Dissipation .................................... See Thermal Analysis Differential Input Voltage .................................................................. ±1.2V Input Voltage Range ..................................................... –0.5 to +VS +0.3V Storage Temperature Range ......................................... –40°C to +125°C Lead Temperature (soldering, 10s) .............................................. +300°C Junction Temperature (TJ ) ........................................................... +175°C Electrostatic discharge can cause damage ranging from performance degradation to complete device failure. Burr-Brown Corporation recommends that all integrated circuits be handled and stored using appropriate ESD protection methods. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet published specifications. PIN CONFIGURATIONS Top View SO-8 OPA2634 Out A 1 8 +VS –In A 2 7 Out B +In A 3 6 –In B GND 4 5 +In B PACKAGE/ORDERING INFORMATION PRODUCT PACKAGE PACKAGE DRAWING NUMBER(1) OPA2634U SO-8 Surface-Mount 182 –40°C to +85°C OPA2634U " " " " " SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER(2) TRANSPORT MEDIA OPA2634U OPA2634U/2K5 Rails Tape and Reel NOTES: (1) For detailed drawing and dimension table, please see end of data sheet, or Appendix C of Burr-Brown IC Data Book. (2) Models with a slash (/) are available only in Tape and Reel in the quantities indicated (e.g., /2K5 indicates 2500 devices per reel). Ordering 2500 pieces of “OPA2634U/2K5” will get a single 2500-piece Tape and Reel. For detailed Tape and Reel mechanical information, refer to Appendix B of Burr-Brown IC Data Book. ® OPA2634 4 TYPICAL PERFORMANCE CURVES: VS = +5V At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS /2, unless otherwise noted (see Figure 1). LARGE-SIGNAL FREQUENCY RESPONSE SMALL-SIGNAL FREQUENCY RESPONSE 12 6 VO = 0.2Vp-p 9 6 0 G = +5 –3 Gain (dB) Normalized Gain (dB) VO = 0.2Vp-p G = +2 3 –6 –9 G = +10 –12 3 0 VO = 1Vp-p –3 VO = 2Vp-p –6 VO = 4Vp-p –9 –15 –12 –18 1 10 100 300 1 VO VIN VO VIN Time (10ns/div) Time (10ns/div) OUTPUT SWING vs LOAD RESISTANCE 0.8 4.7 0.7 4.6 0.6 4.5 0.5 4.4 0.4 4.3 0.3 Minimum VO 4.1 0.2 Input-Referred Crosstalk (dB) 0.9 4.8 4.2 –40 Minimum Output Voltage (V) Maximum Output Voltage (V) CHANNEL-TO-CHANNEL CROSSTALK 1.0 Maximum VO 300 VO = 2Vp-p Input and Output Voltage (500mV/div) Input and Output Voltage (50mV/div) VO = 200mVp-p 4.9 100 LARGE-SIGNAL PULSE RESPONSE SMALL-SIGNAL PULSE RESPONSE 5.0 10 Frequency (MHz) Frequency (MHz) –50 –60 –70 –80 –90 0.1 4.0 50 0.0 1000 100 –100 1 10 100 Frequency (MHz) RL (Ω) ® 5 OPA2634 TYPICAL PERFORMANCE CURVES: VS = +5V (Cont.) At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 1). HARMONIC DISTORTION vs OUTPUT VOLTAGE HARMONIC DISTORTION vs NON-INVERTING GAIN –50 –50 VO = 2Vp-p f = 5MHz Harmonic Distortion (dBc) Harmonic Distortion (dBc) f = 5MHz –60 3rd Harmonic –70 2nd Harmonic –80 –90 3rd Harmonic –60 2nd Harmonic –70 –80 –90 0.1 1 1 4 10 Gain Magnitude (V/V) Output Voltage (Vp-p) HARMONIC DISTORTION vs INVERTING GAIN HARMONIC DISTORTION vs FREQUENCY –50 –50 VO = 2Vp-p 3rd Harmonic Harmonic Distortion (dBc) Harmonic Distortion (dBc) VO = 2Vp-p f = 5MHz –60 2nd Harmonic –70 –80 –60 3rd Harmonic –70 –80 2nd Harmonic –90 –90 1 10 0.1 1 Gain Magnitude (V/V) HARMONIC DISTORTION vs LOAD RESISTANCE HARMONIC DISTORTION vs SUPPLY VOLTAGE –50 –50 VO = 2Vp-p fO = 5MHz VO = 2Vp-p fO = 5MHz Harmonic Distortion (dBc) Harmonic Distortion (dBc) 10 Frequency (MHz) –60 –70 2nd Harmonic –80 –60 3rd Harmonic –70 2nd Harmonic –80 3rd Harmonic –90 –90 100 1000 3 RL (Ω) 5 6 7 Supply Voltage (V) ® OPA2634 4 6 8 9 10 TYPICAL PERFORMANCE CURVES: VS = +5V (Cont.) At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 1). TWO-TONE, 3rd-ORDER INTERMODULATION SPURIOUS CMRR AND PSRR vs FREQUENCY 80 Rejection Ratio, Input Referred (dB) 3rd-Order Spurious Level (dBc) –40 fO = 20MHz –45 –50 fO = 10MHz –55 –60 –65 –70 –75 fO = 5MHz –80 Load Power at Matched 50Ω Load –85 CMRR 75 70 65 PSRR 60 55 50 45 40 35 30 –90 –16 –14 –12 –10 –8 –6 –4 –2 100 0 1k 10k Single-Tone Load Power (dBm) 1M 10M RECOMMENDED RS vs CAPACITIVE LOAD INPUT NOISE DENSITY vs FREQUENCY 1000 100 100 RS (Ω) Voltage Noise (nV/√Hz) Current Noise (pA/√Hz) 100k Frequency (Hz) 10 Voltage Noise, eni = 5.6nV/√Hz 10 Current Noise, ini = 2.8pA/√Hz 1 1 1k 10k 100k 1M 1 10M 10 Frequency (Hz) VO = 0.2Vp-p CL = 1000pF CL = 10pF Open-Loop Gain (dB) Normalized Gain (dB) 0 –1 CL = 100pF –2 –3 –4 1/2 OPA2634 –5 RS VO CL –6 –7 1kΩ +VS/2 –8 1 10 1000 OPEN-LOOP GAIN AND PHASE FREQUENCY RESPONSE vs CAPACITIVE LOAD 2 1 100 Capacitive Load (pF) 100 100 90 80 70 60 50 40 30 20 10 0 –10 –20 Open-Loop Phase Open-Loop Gain 1k 300 0 –30 –60 –90 –120 –150 –180 –210 –240 –270 –300 –330 –360 10k 100k 1M 10M 100M Open-Loop Phase (°) 100 1G Frequency (Hz) Frequency (MHz) ® 7 OPA2634 TYPICAL PERFORMANCE CURVES: VS = +5V (Cont.) At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 1). CLOSED-LOOP OUTPUT IMPEDANCE vs FREQUENCY INPUT DC ERRORS vs TEMPERATURE 5.0 50 Input Offset Voltage (mV) 10 1 45 Input Offset Voltage 4.0 40 3.5 35 3.0 30 2.5 25 Input Bias Current 2.0 20 1.5 15 10X Input Offset Current 1.0 10 0.5 5 0.0 0.1 1k 10k 100k 1M 10M 0 –40 100M –20 0 40 60 80 100 Temperature (°C) Frequency (Hz) SLEW RATE AND GAIN BANDWIDTH PRODUCT vs SUPPLY VOLTAGE SUPPLY AND OUTPUT CURRENT vs TEMPERATURE 16 300 160 150 Gain Bandwidth Product Sinking Output Current 14 140 250 120 10 100 Sourcing Output Current 8 80 6 60 4 40 2 20 0 Slew Rate (V/µs) 12 Output Current (mA) Power Supply Current –20 0 20 40 60 80 100 150 75 100 50 50 25 0 0 3 100 4 5 6 7 Supply Voltage (V) Temperature (°C) SUPPLY AND OUTPUT CURRENTS vs SUPPLY VOLTAGE 18 180 160 Quiescent Supply Current 14 140 12 120 10 100 8 80 Output Current, Sourcing 6 60 Output Current, Sinking 4 40 2 20 0 3 4 5 6 7 Supply Voltage (V) ® 8 8 9 10 Output Current (mA) 16 0 OPA2634 125 Slew Rate 200 0 –40 Quiescent Supply Current (mA/chan) Power Supply Current (mA/chan) 20 8 9 10 Gain Bandwidth Product (MHz) Output Impedance (Ω) 4.5 Input Bias Current (µA) 10x Input Offset Current (µA) 100 TYPICAL PERFORMANCE CURVES: VS = +3V At TA = 25°C, G = +2, RF = 750Ω, and RL = 150Ω to VS/2, unless otherwise noted (see Figure 2). SMALL-SIGNAL FREQUENCY RESPONSE LARGE-SIGNAL FREQUENCY RESPONSE 6 12 VO = 0.2Vp-p VO = 0.2Vp-p G = +2 9 0 6 G = +5 –3 Gain (dB) –6 –9 G = +10 –12 3 VO = 1Vp-p 0 VO = 2Vp-p –3 –6 –15 –9 –18 –12 1 10 100 300 1 10 Frequency (MHz) TWO-TONE, 3rd-ORDER INTERMODULATION SPURIOUS fO = 20MHz –45 –50 –55 100 –60 RS (Ω) 3rd-Order Spurious Level (dBc) 300 RECOMMENDED RS vs CAPACITIVE LOAD 1000 –40 fO = 10MHz –65 –70 10 –75 –80 Load Power at Matched 50Ω Load fO = 5MHz –85 1 –90 –16 –14 –12 –10 –8 –6 1 –4 10 100 FREQUENCY RESPONSE vs CAPACITIVE LOAD OUTPUT SWING vs LOAD RESISTANCE 2 3.0 CL = 1000pF CL = 10pF 0 –1 CL = 100pF –2 –3 –4 1/2 OPA2634 –5 RS VO CL –6 –7 1.0 Maximum VO 2.9 Maximum Output Voltage (V) VO = 0.2Vp-p 1 1kΩ 0.9 2.8 0.8 2.7 0.7 2.6 0.6 2.5 0.5 2.4 0.4 2.3 0.3 2.2 0.2 Minimum VO 2.1 +VS/2 0.1 2.0 –8 1 10 1000 Capacitive Load (pF) Single-Tone Load Power (dBm) Normalized Gain (dB) 100 Frequency (MHz) 100 300 50 Frequency (MHz) Minimum Output Voltage (V) Normalized Gain (dB) 3 0.0 1000 100 RL (Ω) ® 9 OPA2634 APPLICATIONS INFORMATION +VS = 3V WIDEBAND VOLTAGE FEEDBACK OPERATION 6.8µF + The OPA2634 is a unity-gain stable, very high-speed, voltage-feedback op amp designed for single-supply operation (+3V to +5V). The input stage supports input voltages below ground, and within 1.2V of the positive supply. The complementary common-emitter output stage provides an output swing to within 30mV of ground and 140mV of the positive supply. It is compensated to provide stable operation with a wide range of resistive loads. 374Ω VIN 562Ω +VS FIGURE 2. DC-Coupled Signal—Resistive Load to Supply Midpoint. supply capability of both the ADC and its driver. The OPA2634 provides excellent performance in this demanding application. Its large input and output voltage ranges, and low distortion, support converters such as the ADS900 shown in this figure. The input level-shifting circuitry was designed so that VIN can be between 0V and 0.5V, while producing a 1V to 2V at the output pin for the ADS900. 0.1µF + ANTI-ALIASING FILTER Figure 3 shows an anti-aliasing filter, with a 5th-order Inverse Chebyshev response, based on a single OPA2634. This filter cascades two 2nd-order Sallen-Key sections with transmission zeros, and a real pole section. It has a –3dB 0.1µF VIN 1/2 OPA2634 750Ω 2 6.8µF + 1.50kΩ VOUT RL 150Ω +VS = 5V 53.6Ω 1/2 OPA2634 57.6Ω Figure 1 shows the AC-coupled, gain of +2 configuration used for the +5V Specifications and Typical Performance Curves. For test purposes, the input impedance is set to 50Ω with a resistor to ground. Voltage swings reported in the Specifications are taken directly at the input and output pins. For the circuit of Figure 1, the total effective load on the output at high frequencies is 150Ω || 1500Ω. The 1.50kΩ resistors at the non-inverting input provide the commonmode bias voltage. Their parallel combination equals the DC resistance at the inverting input, minimizing the DC offset. 1.50kΩ 0.1µF + 2.26kΩ VOUT RL 150Ω 0.1µF 750Ω 750Ω +VS 2 FIGURE 1. AC-Coupled Signal—Resistive Load to Supply Midpoint. 100Ω 215Ω 680pF 10pF 20Ω 56pF VIN Figure 2 shows the DC-coupled, gain of +2 configuration used for the +3V Specifications and Typical Performance Curves. For test purposes, the input impedance is set to 50Ω with a resistor to ground. Though not strictly a “rail-to-rail” design, this part comes very close, while maintaining excellent performance. It will deliver up to 2.8Vp-p on a single +3V supply with > 100MHz small-signal bandwidth. The 374Ω and 2.26kΩ resistors at the input level-shift VIN so that VOUT is within the allowed output voltage range when VIN = 0. See the Typical Performance Curves for information on driving capacitive loads. 1/2 OPA2634 210Ω 68.1Ω 330pF 191Ω VOUT 1/2 OPA2634 115Ω 93.1Ω 18pF 330pF 220pF 20Ω 150pF 205Ω 48.7Ω SINGLE-SUPPLY ADC INTERFACE The front page shows a DC-coupled, single-supply, dual ADC driver circuit. Many systems are now requiring +3V FIGURE 3. Inverse Chebyshev Anti-Aliasing Filter. ® OPA2634 1% Resistors 5% Capacitors 680pF 10 frequency of 5MHz, and a –60dB stopband starting at 12MHz. This filter works well on +5V or ±5V supplies, and with an A/D converter at 20MSPS (e.g., ADS900). VIN needs to be a very low impedance source, such as an op amp. +VS R2 The filter transfer function was designed using Burr-Brown’s FilterPro 42 design program (available at www.burrbrown.com in the Applications section) with a nominal stopband attenuation of 60dB. Table I gives the results (H0 = DC gain, fP = pole frequency, QP = pole quality, and fZ = zero frequency). Note that the parameters were generated at f–3dB = 5Hz, and then scaled to f–3dB = 5MHz. R1 VIN 1/2 OPA2634 R3 SECTION NO. H0 fP QP fZ 1 1V/V 5.04MHz 1.77 12.6MHz 2 1V/V 5.31MHz 0.64 20.4MHz 3 1V/V 5.50MHz — — TABLE I. Nominal Filter Parameters. Make sure that VIN and VOUT stay within the specified input and output voltage ranges. The front page circuit is a good example of this type of application. It was designed to take VIN between 0V and 0.5V, and produce VOUT between 1V and 2V, when using a +3V supply. This means G = 2.00, and ∆VOUT = 1.50V – G • 0.25V = 1.00V. Plugging into the above equations gives: NG = 2.33, R1 = 375Ω, R2 = 2.25kΩ, and R3 = 563Ω. The resistors were changed to the nearest standard values. 0 –10 Gain (dB) –20 –30 –40 NON-INVERTING AMPLIFIER WITH REDUCED PEAKING Figure 6 shows a non-inverting amplifier that reduces peaking at low gains. The resistor RC compensates the OPA2634 to have higher Noise Gain (NG), which reduces the AC response peaking (typically 5dB at G = +1 without RC) without changing the DC gain. VIN needs to be a low impedance source, such as an op amp. The resistor values are low to reduce noise. Using both RT and RF helps minimize the impact of parasitic impedances. –50 –60 –70 –80 10 R4 FIGURE 5. DC Level-Shifting Circuit. The components were chosen to give this transfer function. The 20Ω resistors isolate the amplifier outputs from capacitive loading, but affect the response at very high frequencies only. Figure 4 shows the nominal response simulated by SPICE; it is very close to the ideal response. 1 VOUT 100 Frequency (MHz) FIGURE 4. Nominal Filter Response. DC LEVEL-SHIFTING Figure 5 shows a DC-coupled, non-inverting amplifier that level-shifts the input up to accommodate the desired output voltage range. Given the desired signal gain (G), and the amount VOUT needs to be shifted up (∆VOUT) when VIN is at the center of its range, the following equations give the resistor values that produce the best DC offset: RT VIN RC 1/2 OPA2634 VOUT NG = G + ∆VOUT/VS R1 = R4/G RG R2 = R4/(NG – G) RF R3 = R4/(NG –1) where: NG = 1 + R4/R3 (Noise Gain) VOUT = (G)VIN + (NG – G)VS FIGURE 6. Compensated Non-Inverting Amplifier. ® 11 OPA2634 A good rule of thumb is to target the parallel combination of RF and RG (Figure 6) to be less than approximately 400Ω. The combined impedance RF || RG interacts with the inverting input capacitance, placing an additional pole in the feedback network and thus, a zero in the forward response. Assuming a 3pF total parasitic on the inverting node, holding RF || RG <400Ω will keep this pole above 130MHz. By itself, this constraint implies that the feedback resistor (RF) can increase to several kΩ at high gains. This is acceptable as long as the pole formed by RF and any parasitic capacitance appearing in parallel is kept out of the frequency range of interest. The noise gain can be calculated as follows: G1 = 1 + RF RG G2 = 1 + R T + R F /G1 RC NG = G1G 2 A unity-gain buffer can be designed by selecting RT = RF = 20.0Ω and RC = 40.2Ω (do not use RG ). This gives a noise gain of 2, therefore, its response will be similar to the characteristics plots with G = +2. Decreasing RC to 20.0Ω will increase the noise gain to 3, which typically gives a flat frequency response, but with less bandwidth. BANDWIDTH VS GAIN: NON-INVERTING OPERATION Voltage-feedback op amps exhibit decreasing closed-loop bandwidth as the signal gain is increased. In theory, this relationship is described by the Gain Bandwidth Product (GBP) shown in the specifications. Ideally, dividing GBP by the non-inverting signal gain (also called the Noise Gain, or NG) will predict the closed-loop bandwidth. In practice, this only holds true when the phase margin approaches 90°, as it does in high gain configurations. At low gains (increased feedback factors), most amplifiers will exhibit a more complex response with lower phase margin. The OPA2634 is compensated to give a slightly peaked response in a noninverting gain of 2 (Figure 1). This results in a typical gain of +2 bandwidth of 150MHz, far exceeding that predicted by dividing the 140MHz GBP by 2. Increasing the gain will cause the phase margin to approach 90° and the bandwidth to more closely approach the predicted value of (GBP/NG). At a gain of +10, the 16MHz bandwidth shown in the Typical Specifications is close to that predicted using the simple formula and the typical GBP. The circuit in Figure 2 can be redesigned to have less peaking by increasing the noise gain to 3. This is accomplished by adding RC = 2.55kΩ between the op amp’s inputs. DESIGN-IN TOOLS DEMONSTRATION BOARDS A PC board is available to assist in the initial evaluation of circuit performance using the OPA2634. It is available free as an unpopulated PC board delivered with descriptive documentation. The summary information for this board is shown below: PRODUCT PACKAGE BOARD PART NUMBER OPA2634U SO-8 DEM-OPA268xU LITERATURE REQUEST NUMBER MKT-352 The OPA2634 exhibits minimal bandwidth reduction going to +3V single-supply operation as compared with +5V supply. This is because the internal bias control circuitry retains nearly constant quiescent current as the total supply voltage between the supply pins is changed. Contact the Burr-Brown Applications support line to request this board. OPERATING SUGGESTIONS OPTIMIZING RESISTOR VALUES INVERTING AMPLIFIER OPERATION Since the OPA2634 is a voltage-feedback op amp, a wide range of resistor values may be used for the feedback and gain setting resistors. The primary limits on these values are set by dynamic range (noise and distortion) and parasitic capacitance considerations. For a non-inverting unity-gain follower application, the feedback connection should be made with a 25Ω resistor, not a direct short. This will isolate the inverting input capacitance from the output pin and improve the frequency response flatness. Usually, for G > 1 application, the feedback resistor value should be between 200Ω and 1.5kΩ. Below 200Ω, the feedback network will present additional output loading which can degrade the harmonic distortion performance. Above 1.5kΩ, the typical parasitic capacitance (approximately 0.2pF) across the feedback resistor may cause unintentional bandlimiting in the amplifier response. Since the OPA2634 is a general purpose, wideband voltage feedback op amp, all of the familiar op amp application circuits are available to the designer. Figure 7 shows a typical inverting configuration where the I/O impedances and signal gain from Figure 1 are retained in an inverting circuit configuration. Inverting operation is one of the more common requirements and offers several performance benefits. The inverting configuration shows improved slew rate and distortion. It also biases the input at VS/2 for the best headroom. The output voltage can be independently moved with bias-adjustment resistors connected to the input. In the inverting configuration, three key design consideration must be noted. The first is that the gain resistor (RG) becomes part of the signal channel input impedance. If input impedance matching is desired (which is beneficial whenever the signal is coupled through a cable, twisted pair, long ® OPA2634 12 non-inverting input (parallel combination of RT = 750Ω). If this resistor is set equal to the total DC resistance looking out of the inverting node, the output DC error, due to the input bias currents, will be reduced to (Input Offset Current) • RF. Because of the 0.1µF capacitor, the inverting input’s bias current flows through RF. Thus, RT = 750Ω = 1.50kΩ || 1.50kΩ is needed for the minimum output offset voltage. To reduce the additional high frequency noise introduced by RT, and power supply feedthrough, it is bypassed with a 0.1µF capacitor. At a minimum, the OPA2634 should see a source resistance of at least 50Ω to damp out parasitic-induced peaking—a direct short to ground on the non-inverting input runs the risk of a very high frequency instability in the input stage. +5V + 0.1µF 2RT 1.50kΩ 0.1µF 50Ω Source 0.1µF 2RT 1.50kΩ RG 374Ω 1/2 OPA2634 6.8µF RO 50Ω 50Ω Load RF 750Ω OUTPUT CURRENT AND VOLTAGE The OPA2634 provides outstanding output voltage capability. Under no-load conditions at +25°C, the output voltage typically swings closer than 140mV to either supply rail; the guaranteed swing limit is within 450mV of either rail (VS = +5V). RM 57.6Ω FIGURE 7. Gain of –2 Example Circuit. The minimum specified output voltage and current specifications over temperature are set by worst-case simulations at the cold temperature extreme. Only at cold start-up will the output current and voltage decrease to the numbers shown in the guaranteed tables. As the output transistors deliver power, their junction temperatures will increase, decreasing their VBE’s (increasing the available output voltage swing) and increasing their current gains (increasing the available output current). In steady-state operation, the available output voltage and current will always be greater than that shown in the over-temperature specifications since the output stage junction temperatures will be higher than the minimum specified operating ambient. PC board trace or other transmission line conductor), RG may be set equal to the required termination value and RF adjusted to give the desired gain. This is the simplest approach and results in optimum bandwidth and noise performance. However, at low inverting gains, the resultant feedback resistor value can present a significant load to the amplifier output. For an inverting gain of 2, setting RG to 50Ω for input matching eliminates the need for RM but requires a 100Ω feedback resistor. This has the interesting advantage that the noise gain becomes equal to 2 for a 50Ω source impedance—the same as the non-inverting circuits considered above. However, the amplifier output will now see the 100Ω feedback resistor in parallel with the external load. In general, the feedback resistor should be limited to the 200Ω to 1.5kΩ range. In this case, it is preferable to increase both the RF and RG values as shown in Figure 7, and then achieve the input matching impedance with a third resistor (RM) to ground. The total input impedance becomes the parallel combination of RG and RM. To maintain maximum output stage linearity, no output short-circuit protection is provided. This will not normally be a problem since most applications include a series matching resistor at the output that will limit the internal power dissipation if the output side of this resistor is shorted to ground. DRIVING CAPACITIVE LOADS The second major consideration, touched on in the previous paragraph, is that the signal source impedance becomes part of the noise gain equation and hence, influences the bandwidth. For the example in Figure 7, the RM value combines in parallel with the external 50Ω source impedance, yielding an effective driving impedance of 50Ω || 57.6Ω = 26.8Ω. This impedance is added in series with RG for calculating the noise gain. The resultant is 2.87 for Figure 7, as opposed to only 2 if RM could be eliminated as discussed above. The bandwidth will, therefore, be lower for the gain of –2 circuit of Figure 7 (NG = +2.9) than for the gain of +2 circuit of Figure 1. One of the most demanding and yet very common load conditions for an op amp is capacitive loading. Often, the capacitive load is the input of an A/D converter—including additional external capacitance which may be recommended to improve A/D linearity. A high-speed, high open-loop gain amplifier like the OPA2634 can be very susceptible to decreased stability and closed-loop response peaking when a capacitive load is placed directly on the output pin. When the primary considerations are frequency response flatness, pulse response fidelity and/or distortion, the simplest and most effective solution is to isolate the capacitive load from the feedback loop by inserting a series isolation resistor between the amplifier output and the capacitive load. The third important consideration in inverting amplifier design is setting the bias current cancellation resistors on the ® 13 OPA2634 The Typical Performance Curves show the recommended RS versus capacitive load and the resulting frequency response at the load. Parasitic capacitive loads greater than 2pF can begin to degrade the performance of the OPA2634. Long PC board traces, unmatched cables, and connections to multiple devices can easily exceed this value. Always consider this effect carefully, and add the recommended series resistor as close as possible to the output pin (see Board Layout Guidelines). ENI 4kT RG DC ACCURACY AND OFFSET CONTROL The balanced input stage of a wideband voltage-feedback op amp allows good output DC accuracy in a wide variety of applications. The power supply current trim for the OPA2634 gives even tighter control than comparable products. Although the high-speed input stage does require relatively high input bias current (typically 25µA out of each input terminal), the close matching between them may be used to reduce the output DC error caused by this current. This is done by matching the DC source resistances appearing at the two inputs. Evaluating the configuration of Figure 1 (which has matched DC input resistances), using worst-case +25°C input offset voltage and current specifications, gives a worstcase output offset voltage equal to (NG = non-inverting signal gain at DC): Equation 1: ±(NG • VOS(MAX)) ± (RF • IOS(MAX)) = ±(1 • 7.0mV) ± (750Ω • 2.0µA) = ±8.5mV = Output Offset Range for Figure 1 + ( I BN R S ) + 4kTR S NG 2 + ( I BI R F ) + 4kTR F NG 2 Dividing this expression by the noise gain (NG = (1+RF /RG)) will give the equivalent input-referred spot noise voltage at the non-inverting input, as shown in Equation 2. ® OPA2634 4kT = 1.6E –20J at 290°K Evaluating these two equations for the circuit and component values shown in Figure 1 will give a total output spot noise voltage of 12.5nV/√Hz and a total equivalent input spot noise voltage of 6.3nV/√Hz. This is including the noise added by the resistors. This total input-referred spot noise voltage is not much higher than the 5.6nV/√Hz specification for the op amp voltage noise alone. This will be the case as long as the impedances appearing at each op amp input are limited to the previously recommend maximum value of 400Ω, and the input attenuation is low. The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 1 shows the general form for the output noise voltage using the terms shown in Figure 8. ) √ 4kTRF I R 2 4kTR F 2 E N = E NI 2 + ( I BN R S ) + 4kTR S + BI F + NG NG High slew rate, unity gain stable, voltage-feedback op amps usually achieve their slew rate at the expense of a higher input noise voltage. The 5.6nV/√Hz input voltage noise for the OPA2634 is, however, much lower than comparable amplifiers. The input-referred voltage noise, and the two input-referred current noise terms (2.8pA/√Hz), combine to give low output noise under a wide variety of operating conditions. Figure 8 shows the op amp noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz. 2 IBI Equation 2: NOISE PERFORMANCE 2 RG FIGURE 8. Noise Analysis Model. The OPA2634 provides good distortion performance into a 150Ω load. Relative to alternative solutions, it provides exceptional performance into lighter loads and/or operating on a single +3V supply. Generally, until the fundamental signal reaches very high frequency or power levels, the 2nd harmonic will dominate the distortion with a negligible 3rd harmonic component. Focusing then on the 2nd harmonic, increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network; in the non-inverting configuration (Figure 1) this is sum of RF + RG, while in the inverting configuration, it is just RF. NI RF √ 4kTRS DISTORTION PERFORMANCE (E EO IBN ERS The criterion for setting this RS resistor is a maximum bandwidth, flat frequency response at the load. For a gain of +2, the frequency response at the output pin is already slightly peaked without the capacitive load, requiring relatively high values of RS to flatten the response at the load. Increasing the noise gain will also reduce the peaking, reducing the required RS value (see Figure 6). EO = 1/2 OPA2634 RS 14 BOARD LAYOUT GUIDELINES A fine scale output offset null, or DC operating point adjustment, is often required. Numerous techniques are available for introducing DC offset control into an op amp circuit. Most of these techniques are based on adding a DC current through the feedback resistor. In selecting an offset trim method, one key consideration is the impact on the desired signal path frequency response. If the signal path is intended to be non-inverting, the offset control is best applied as an inverting summing signal to avoid interaction with the signal source. If the signal path is intended to be inverting, applying the offset control to the non-inverting input may be considered. Bring the DC offsetting current into the inverting input node through resistor values that are much larger than the signal path resistors. This will insure that the adjustment circuit has minimal effect on the loop gain and hence the frequency response. Achieving optimum performance with a high frequency amplifier like the OPA2634 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability: on the noninverting input, it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. b) Minimize the distance (< 0.25") from the power supply pins to high frequency 0.1µF decoupling capacitors. At the device pins, the ground and power plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power supply connections should always be decoupled with these capacitors. An optional supply decoupling capacitor (0.1µF) across the two power supplies (for bipolar operation) will improve 2nd harmonic distortion performance. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. THERMAL ANALYSIS Maximum desired junction temperature will set the maximum allowed internal power dissipation as described below. In no case should the maximum junction temperature be allowed to exceed 175°C. Operating junction temperature (TJ) is given by TA + PD•θJA. The total internal power dissipation (PD) is the sum of quiescent power (PDQ) and additional power dissipated in the output stage (PDL) to deliver load power. Quiescent power is simply the specified no-load supply current times the total supply voltage across the part. PDL will depend on the required output signal and load but would, for resistive load connected to mid-supply (VS/2), be at a maximum when the output is fixed at a voltage equal to VS/4 or 3VS/4. Under this condition, PDL = VS2/(16 • RL), where RL includes feedback network loading. c) Careful selection and placement of external components will preserve the high frequency performance. Resistors should be a very low reactance type. Surfacemount resistors work best and allow a tighter overall layout. Metal film or carbon composition axially-leaded resistors can also provide good high frequency performance. Again, keep their leads and PC board traces as short as possible. Never use wirewound type resistors in a high frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. Other network components, such as non-inverting input termination resistors, should also be placed close to the package. Where double-side component mounting is allowed, place the feedback resistor directly under the package on the other side of the board between the output and inverting input pins. Even with a low parasitic capacitance shunting the external resistors, excessively high resistor values can create significant time constants that can degrade performance. Good axial metal film or surfacemount resistors have approximately 0.2pF in shunt with the resistor. For resistor values >1.5kΩ, this parasitic capacitance can add a pole and/or zero below 500MHz that can effect circuit operation. Keep resistor values as low as possible consistent with load driving considerations. The 750Ω feedback used in the typical performance specifications is a good starting point for design. Note that it is the power in the output stage and not into the load that determines internal power dissipation. As a worst-case example, compute the maximum TJ using the circuit of Figure 1 operating at the maximum specified ambient temperature of +85°C and driving a 150Ω load at mid-supply, for both channels: PD = 2 (10V • 13.25mA + 52/(16 • (150Ω || 1500Ω))) = 288mW Maximum TJ = +85°C + (0.29W • 125°C/W) = 121°C Although this is still well below the specified maximum junction temperature, system reliability considerations may require lower guaranteed junction temperatures. The highest possible internal dissipation will occur if the load requires current to be forced into the output at high output voltages or sourced from the output at low output voltages. This puts a high current through a large internal voltage drop in the output transistors. ® 15 OPA2634 d) Connections to other wideband devices on the board may be made with short direct traces or through on-board transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50mils to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the plot of Recommended RS vs Capacitive Load. Low parasitic capacitive loads (< 5pF) may not need an RS since the OPA2634 is nominally compensated to operate with a 2pF parasitic load. Higher parasitic capacitive loads without an RS are allowed as the signal gain increases (increasing the unloaded phase margin). If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary on board, and in fact, a higher impedance environment will improve distortion as shown in the distortion versus load plots. With a characteristic board trace impedance defined (based on board material and trace dimensions), a matching series resistor into the trace from the output of the OPA2634 is used as well as a terminating shunt resistor at the input of the destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and the input impedance of the destination device; this total effective impedance should be set to match the trace impedance. If the 6dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in the plot of Recommended RS vs Capacitive Load. This will not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. e) Socketing a high speed part is not recommended. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA2634 onto the board. INPUT AND ESD PROTECTION The OPA2634 is built using a very high-speed complementary bipolar process. The internal junction breakdown voltages are relatively low for these very small geometry devices. These breakdowns are reflected in the Absolute Maximum Ratings table. All device pins are protected with internal ESD protection diodes to the power supplies as shown in Figure 9. These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (e.g., in systems with ±15V supply parts driving into the OPA2634), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible since high values degrade both noise performance and frequency response. +V CC External Pin –V CC FIGURE 9. Internal ESD Protection. ® OPA2634 Internal Circuitry 16