LTC3605A 20V, 5A Synchronous Step-Down Regulator FEATURES DESCRIPTION n n n The LTC®3605A is a high efficiency, monolithic synchronous buck regulator using a phase lockable controlled on-time constant frequency, current mode architecture. PolyPhase operation allows multiple LTC3605A regulators to run out of phase while using minimal input and output capacitance. The operating supply voltage range is from 20V down to 4V, making it suitable for dual, triple or quadruple lithium-ion battery inputs as well as point of load power supply applications from a 12V or 5V rail. High Efficiency: Up to 96% 5A Output Current 4V to 20V VIN Range Integrated Power N-Channel MOSFETs (70mΩ Top and 35mΩ Bottom) n Adjustable Frequency 800kHz to 4MHz nPolyPhase® Operation (Up to 12 Phases) n Output Tracking n 0.6V ±1% Reference Accuracy n Current Mode Operation for Excellent Line and Load Transient Response n Shutdown Mode Draws Less Than 15µA Supply Current n LTC3605: 15V Absolute Maximum V IN n LTC3605A: 22V Absolute Maximum V IN n The LTC3605A Is Pin Compatible with the LTC3605 n Available in 24-Pin (4mm × 4mm) QFN Package n APPLICATIONS n n n n Point of Load Power Supply Portable Instruments Distributed Power Systems Battery-Powered Equipment The operating frequency is programmable from 800kHz to 4MHz with an external resistor. The high frequency capability allows the use of small surface mount inductors. For switching noise sensitive applications, it can be externally synchronized from 800kHz to 4MHz. The PHMODE pin allows user control of the phase of the outgoing clock signal. The unique constant frequency/controlled on-time architecture is ideal for high step-down ratio applications that are operating at high frequency while demanding fast transient response. Two internal phase-lock loops synchronize the internal oscillator to the external clock and also servos the regulator on-time to lock on to either the internal clock or the external clock if it’s present. L, LT, LTC, LTM, PolyPhase, OPTI-LOOP, Linear Technology and the Linear logo are registered trademarks and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 5847554, 6580258, 6304066, 6476589, 6774611. TYPICAL APPLICATION Efficiency and Power Loss High Efficiency 1MHz, 5A Step-Down Regulator 90 80 PVIN CLKOUT SVIN INTVCC 2.2µF CLKIN BOOST PGOOD LTC3605A SW VON VIN 10 VOUT = 3.3V RUN 0.1µF 1µH 11.5k FB RT ITH 47µF ×2 2.55k 16k 162k VOUT 3.3V 220pF 3605A TA01a PGND 70 1 60 50 40 0.1 30 20 VIN = 8V VIN = 12V VIN = 20V 10 0 0.01 0.1 1 LOAD CURRENT (A) 10 POWER LOSS (W) 22µF ×2 EFFICIENCY (%) VIN 4V TO 20V 100 0.01 3605A TA01b 3605af 1 LTC3605A ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) PVIN, SVIN, SW Voltage............................... –0.3V to 22V SW Transient Voltage.................................. –2V to 24.5V BOOST Voltage........................... –0.3V to PVIN + INTVCC RUN Voltage.............................................. –0.3V to SVIN VON Voltage................................................ –0.3V to SVIN INTVCC Voltage........................................... –0.3V to 3.6V ITH, RT, CLKOUT, PGOOD Voltage.......... –0.3V to INTVCC CLKIN, PHMODE, MODE Voltage........... –0.3V to INTVCC TRACK/SS, FB Voltage........................... –0.3V to INTVCC Operating Junction Temperature Range (Note 2)................................................... –40°C to 125°C Storage Temperature Range.................... –65°C to 125°C SVIN BOOST INTVCC SGND CLKOUT CLKIN TOP VIEW 24 23 22 21 20 19 RT 1 18 PVIN PHMODE 2 17 PVIN MODE 3 16 SW 25 PGND FB 4 15 SW 13 SW SW SW 9 10 11 12 PGND 8 VON 7 RUN 14 SW ITH 6 PGOOD TRACK/SS 5 UF PACKAGE 24-LEAD (4mm × 4mm) PLASTIC QFN TJMAX = 125°C, θJA = 37°C/W EXPOSED PAD (PIN 25) IS PGND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3605AEUF#PBF LTC3605AEUF#TRPBF 3605A 24-Lead (4mm × 4mm) Plastic QFN –40°C to 125°C LTC3605AIUF#PBF LTC3605AIUF#TRPBF 3605A 24-Lead (4mm × 4mm) Plastic QFN –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TJ ≈ TA = 25°C. (Note 2) SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS SVIN SVIN Supply Range PVIN VIN Power Supply Range IQ Input DC Supply Current Active Shutdown (Note 3) Mode = 0, RT = 162k VIN =12V, RUN = 0 VFB Feedback Reference Voltage ITH =1.2V (Note 4) l DVFB(LINE) Feedback Voltage Line Regulation VIN = 4V to 20V, ITH = 1.2V DVFB(LOAD) Feedback Voltage Load Regulation IFB Feedback Pin Input Current gm (EA) Error Amplifier Transconductance tON(MIN) Minimum On-Time 40 ns tOFF(MIN) Minimum Off-Time 70 ns ITH = 1.2V 4 20 V 1.2 20 V 1.5 11 5 40 mA µA 0.600 0.606 V l 0.001 0.03 %/V l 0.1 0.594 1.15 1.35 0.3 % ±30 nA 1.6 mS 3605af 2 LTC3605A ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TJ ≈ TA = 25°C. (Note 2) SYMBOL PARAMETER CONDITIONS MIN TYP MAX ILIM Positive Inductor Valley Current Limit VFB = 0.57V 5 6 7.5 A RTOP Top Power NMOS On-Resistance INTVCC = 3.3V 70 150 mW RBOTTOM Bottom Power NMOS On-Resistance INTVCC = 3.3V 35 60 mW VUVLO INTVCC Undervoltage Lockout Threshold INTVCC Falling INTVCC Hysteresis (Rising) 2.4 2.6 0.25 2.8 V V VRUN Run Threshold 2 (IQ = 2mA) Run Threshold 1 (IQ = 400µA) RUN Rising RUN Rising 1.2 0.45 1.25 0.6 1.3 0.75 V V VINTVCC Internal VCC Voltage 4V < VIN < 20V 3.2 3.3 3.4 V 0.5 UNITS DVINTVCC INTVCC Load Regulation ILOAD = 0mA to 20mA OV Output Overvoltage PGOOD Upper Threshold VFB Rising 7 10 13 % UV Output Undervoltage PGOOD Lower Threshold VFB Falling –13 –10 –7 % DVFB(HYS) PGOOD Hysteresis VFB Returning 1.5 RPGOOD PGOOD Pull-Down Resistance 1mA Load 12 0.54V < VFB < 0.66V IPGOOD PGOOD Leakage ITRACK/SS TRACK Pull-Up Current fOSC Oscillator Frequency RT = 162k CLKIN CLKIN Threshold CLKIN VIL CLKIN VIH 1 VVIN_OV VIN Overvoltage Lockout Threshold VIN Rising VIN Falling 23 21 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3605A is tested under pulsed load conditions such that TJ ≈ TA. The LTC3605AE (E-grade) is guaranteed to meet specifications from 0°C to 85°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3605AI (I-grade) is guaranteed over the full –40°C to 125°C operating temperature range. The junction temperature (TJ) is calculated from the ambient temperature (TA) and power dissipation (PD) according to the formula: TJ = TA + (PD • θJA°C/W) where θJA is the package thermal impedance. Note that the maximum ambient temperature is determined by specific operating conditions in conjunction with board layout, the rated thermal package thermal resistance and other environmental factors. l 0.85 % % 25 W 2 µA 2.5 4 µA 1 1.2 MHz 0.3 V V 24 22 V V 23.5 21.5 Note 3: Dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency. Note 4: The LTC3605A is tested in a feedback loop that adjusts VFB to achieve a specified error amplifier output voltage (ITH). Note 5: TJ is calculated from the ambient temperature TA and power dissipation as follows: TJ = TA + PD • (37°C/W). See Thermal Considerations section. Note 6: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active Continuous operation above the specified maximum operating junction temperature may impair device reliability. 3605af 3 LTC3605A TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C unless otherwise specified. 2.0 70 60 50 40 30 20 VOUT = 3.3V VOUT = 2.5V VOUT = 1.8V 10 0 0.01 0.1 1 LOAD CURRENT (A) 1.5 1.0 0.5 0 10 0 4 12 8 INPUT VOLTAGE (V) 3605A G18 10 5 4 0 MODE = 3.3V NO LOAD INTERNAL ITH COMPENSATION (ITH = 3.3V) 1.0 TOP FET 10 5 80 60 BOTTOM FET 40 20 0 0.5 1 1.5 2 2.5 3 3.5 FREQUENCY (MHz) 4 4.5 16 20 Load Regulation 1.5 100 15 12 8 INPUT VOLTAGE (V) 3605A G02 RDS(ON) vs Temperature RDS(ON) (mΩ) IINTVCC (mA) 15 0 20 120 20 0 16 Shutdown Current vs VIN 3605A G01 IINTVCC Current vs Frequency 25 20 NORMALIZED (%) EFFICIENCY (%) 80 QUIESCENT CURRENT (mA) VIN = 12V 90 fSW = 1MHz Quiescent Current vs VIN SHUTDOWN CURRENT (µA) Efficiency vs Load Current 100 0.5 VIN = 12V VOUT = 1.2V f = 1MHz MODE = INTVCC 0 EXTERNAL ITH COMPENSATION –0.5 –1.0 0 –45 –20 5 –1.5 30 55 80 105 130 TEMPERATURE (°C) 0 1 2 4 3 IOUT (A) 3605A G04 3605A G03 Load Step (External ITH Compensation) VOUT 100mV/DIV AC-COUPLED IL 5A/DIV IL 5A/DIV ILOAD 5A/DIV ILOAD 5A/DIV 6 7 3605A G05 Load Step (Internal ITH Compensation) VOUT 100mV/DIV AC-COUPLED 5 Output Tracking VTRACK VFB VIN = 12V VOUT = 1.2V ILOAD = 0.4A 20µs/DIV 3605A G06 VIN = 12V VOUT = 1.2V ILOAD = 0.4A ITH = 3.3V 20µs/DIV 3605A G07 VIN = 12V VOUT = 1.2V 500µs/DIV 3605A G08 3605af 4 LTC3605A TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C unless otherwise specified. Switching Frequency vs RT Switch Leakage vs VIN 4.5 350 96 4.0 300 94 3.0 2.5 2.0 1.5 1.0 0 200 150 100 0 50 100 150 200 250 300 350 400 450 500 RT (kΩ) 0 0 2 4 6 8 10 12 VIN (V) 1.0 0.5 2 4 6 8 10 VON (V) 12 14 16 18 ILOAD = 5A 0 5 10 15 VIN (V) 20 25 3505A G11 101 120 INTVCC Load Regulation TA = 25°C 100 100 99 NORMALIZED INTVCC (%) FREQUENCY (MHz) 1.5 NORMALIZED MAXIMUM OUTPUT CURRENT (%) VIN = 20V RT = 162k 0 80 14 16 18 20 Current Limit Foldback 2.0 0 86 3605A G10 Frequency vs VON Voltage 2.5 88 82 3605A G09 3.0 ILOAD = 1A 90 84 50 0.5 VOUT = 3.3V 92 250 EFFICIENCY (%) SWITCH LEAKAGE (nA) FREQUENCY (MHz) 3.5 Efficiency vs VIN 80 60 40 98 97 96 95 94 93 92 20 91 0 0 0.1 0.2 0.3 0.4 VFB (V) 0.5 0.6 0.7 90 0 80 100 120 40 60 20 INTVCC OUTPUT CURRENT (mA) 3605A G13 3605A G12 3605A G14 DCM Operation RUN Pin Threshold vs Temperature 140 CCM Operation 1.30 RUN THRESHOLD (V) 1.25 1.20 CLKOUT 2V/DIV CLKOUT 2V/DIV VSW 5V/DIV VSW 5V/DIV 1.15 IL 2A/DIV 1.10 1.05 1.00 –40 –15 60 35 85 10 TEMPERATURE (°C) 110 IL 2A/DIV VIN = 20V VOUT = 2.5V MODE = 0V IOUT = 0A L1 = 0.5µH 400ns/DIV 3605A G16 VIN = 20V VOUT = 2.5V MODE = 3.3V IOUT = 0A L1 = 0.5µH 400ns/DIV 3605A G17 3605A G15 3605af 5 LTC3605A PIN FUNCTIONS RT (Pin 1): Oscillator Frequency Programming Pin. Connect an external resistor (between 200k to 40k) from RT to SGND to program the frequency from 800kHz to 4MHz. Since the synchronization range is ±30% of set frequency, be sure that the set frequency is within this percentage range of the external clock to ensure frequency lock. VON (Pin 9): On-Time Voltage Input. Voltage trip point for the on-time comparator. Tying this pin to the output voltage makes the on-time proportional to VOUT and keeps the switching frequency constant at different VOUT. However, when VON is <0.6V or >6V, then switching frequency will no longer remain constant. PHMODE (Pin 2): Control Input to Phase Selector. Determines the phase relationship between internal oscillator and CLKOUT. Tie it to INTVCC for 2-phase operation, tie it to SGND for 3-phase operation, and tie it to INTVCC/2 for 4-phase operation. PGND (Pin 10, Exposed Pad Pin 25): Power Ground. Return path of internal power MOSFETs. Connect this pin to the negative terminals of the input capacitor and output capacitor. The exposed pad must be soldered to the PCB ground for electrical contact and rated thermal performance. MODE (Pin 3): Operation Mode Select. Tie this pin to INTVCC to force continuous synchronous operation at all output loads. Tying it to SGND enables discontinuous mode operation at light loads. Tying this pin to INTVCC /2 shuts off the internal clock during discontinuous intervals. FB (Pin 4): Output Feedback Voltage. Input to the error amplifier that compares the feedback voltage to the internal 0.6V reference voltage. This pin is normally connected to a resistive divider from the output voltage. TRACK/SS (Pin 5): Output Tracking and Soft-Start Pin. Allows the user to control the rise time of the output voltage. Putting a voltage below 0.6V on this pin bypasses the internal reference input to the error amplifier, instead it servos the FB pin to the TRACK voltage. Above 0.6V, the tracking function stops and the internal reference resumes control of the error amplifier. There’s an internal 2µA pull-up current from INTVCC on this pin, so putting a capacitor here provides soft-start function. ITH (Pin 6): Error Amplifier Output and Switching Regulator Compensation Point. The current comparator’s trip threshold is linearly proportional to this voltage, whose normal range is from 0.3V to 1.8V. Tying this pin to INTVCC activates internal compensation and output voltage positioning, raising VOUT to 1.5% higher than the nominal value at IOUT = 0 and 1.5% lower at IOUT = 5A. RUN (Pin 7): Run Control Input. Enables chip operation by tying RUN above 1.2V. Tying it below 1.1V shuts down the part. PGOOD (Pin 8): Output Power Good with Open-Drain Logic. PGOOD is pulled to ground when the voltage on the FB pin is not within ±10% of the internal 0.6V reference. 6 SW (Pins 11 to 16): Switch Node Connection to External Inductor. Voltage swing of SW is from a diode voltage drop below ground to PVIN. PVIN (Pins 17, 18): Power VIN. Input voltage to the onchip power MOSFETs. SVIN (Pin 19): Signal VIN. Filtered input voltage to the on-chip 3.3V regulator. Connect a (1Ω to 10Ω) resistor between SVIN and PVIN and bypass to GND with a 0.1µF capacitor. BOOST (Pin 20): Boosted Floating Driver Supply for Internal Top Power MOSFET. The (+) terminal of the bootstrap capacitor connects here. This pin swings from a diode voltage drop below INTVCC up to PVIN + INTVCC. INTVCC (Pin 21): Internal 3.3V Regulator Output. The internal power drivers and control circuits are powered from this voltage. Decouple this pin to power ground with a minimum of 1µF low ESR ceramic capacitor. SGND (Pin 22): Signal Ground Connection. CLKOUT (Pin 23): Output Clock Signal for PolyPhase Operation. The phase of CLKOUT with respect to CLKIN is determined by the state of the PHMODE pin. CLKOUT’s peak-to-peak amplitude is INTVCC to GND. CLKIN (Pin 24): External Synchronization Input to Phase Detector. This pin is internally terminated to SGND with 20k. The phase-locked loop will force the top power NMOS’s turn on signal to be synchronized with the rising edge of the CLKIN signal. 3605af LTC3605A BLOCK DIAGRAM VOUT VON 100K 0.6V 3pF 9 3 MODE 6V 35pF SVIN 3.3V REG ION PLL-SYNC (±30%) CIN2 CIN 17-18 ION INTVCC OST 21 V tON = VON (0.64pF) IION VIN INTVCC R S BOOST Q 20 CB TG 12 x OSC RT 1 + PHMODE 2 RT CLKIN 24 + IREV ICMP – OSC PLL-SYNC (±30%) – RUN CLKOUT 11–16 SENSE– BG CVCC 10, 25 FOLDBACK DISABLED AT START-UP 3.3µA + 0µA TO 10µA M2 PGND 3pF 35pF COUT DB SENSE+ –3.3µA TO 6.7µA 100k L1 VOUT SWITCH LOGIC AND ANTISHOOT THROUGH OV 23 M1 SW ON 20k 8 PGOOD 0.3V FOLDBACK x 4 + 0.6 – 1 180k – Q2 Q4 ITHB R2 0.66V OV FB + Q6 Q1 4 R1 – SGND UV 22 + – EA – + + x= PVIN 1Ω 19 SS + 0.6V 0.6V REF ITH 6 RC – RUN + 0.54V INTVCC 2µA 1.2V CC1 TRACK/SS RUN 7 5 CSS 3605A BD 3605af 7 LTC3605A OPERATION Main Control Loop The LTC3605A is a current mode monolithic step-down regulator. In normal operation, the internal top power MOSFET is turned on for a fixed interval determined by a one-shot timer, OST. When the top power MOSFET turns off, the bottom power MOSFET turns on until the current comparator, ICMP , trips, restarting the one-shot timer and initiating the next cycle. Inductor current is determined by sensing the voltage drop across the bottom power MOSFET ’s VDS. The voltage on the ITH pin sets the comparator threshold corresponding to the inductor valley current. The error amplifier, EA, adjusts this ITH voltage by comparing the feedback signal, VFB, from the output voltage with that of an internal 0.6V reference. If the load current increases, it causes a drop in the feedback voltage relative to the internal reference. The ITH voltage then rises until the average inductor current matches that of the load current. At low load current, the inductor current can drop to zero and become negative. This is detected by current reversal comparator, IREV , which then shuts off the bottom power MOSFET, resulting in discontinuous operation. Both power MOSFETs will remain off with the output capacitor supplying the load current until the ITH voltage rises above the zero current level (0.6V) to initiate another cycle. Discontinuous mode operation is disabled by tying the MODE pin to INTVCC, which forces continuous synchronous operation regardless of output load. The operating frequency is determined by the value of the RT resistor, which programs the current for the internal oscillator. An internal phase-lock loop servos the oscillator frequency to an external clock signal if one is present on the CLKIN pin. Another internal phase-lock loop servos the switching regulator on-time to track the internal oscillator to force constant switching frequency. Overvoltage and undervoltage comparators OV and UV pull the PGOOD output low if the output feedback voltage, VFB, exits a ±10% window around the regulation point. Continuous operation is forced during OV and UV condition except during start-up when the TRACK pin is ramping up to 0.6V. Foldback current limiting is provided if the output is shorted to ground. As VFB drops to zero, the maximum sense voltage allowed across the bottom power MOSFET is lowered to approximately 40% of the original value to reduce the inductor valley current. Pulling the RUN pin to ground forces the LTC3605A into its shutdown state, turning off both power MOSFETs and most of its internal control circuitry. Bringing the RUN pin above 0.6V turns on the internal reference only, while still keeping the power MOSFETs off. Further increasing the RUN voltage above 1.25V turns on the entire chip. INTVCC Regulator An internal low dropout (LDO) regulator produces the 3.3V supply that powers the drivers and the internal bias circuitry. The INTVCC can supply up to 100mA RMS and must be bypassed to ground with a minimum of 1µF ceramic capacitor. Good bypassing is necessary to supply the high transient currents required by the power MOSFET gate drivers. Applications with high input voltage and high switching frequency will increase die temperature because of the higher power dissipation across the LDO. Connecting a load to the INTVCC pin is not recommended since it will further push the LDO into its RMS current rating while increasing power dissipation and die temperature. VIN Overvoltage Protection In order to protect the internal power MOSFET devices against transient voltage spikes, the LTC3605A constantly monitors the VIN pin for an overvoltage condition. When VIN rises above 23.5V, the regulator suspends operation by shutting off both power MOSFETs. Once VIN drops below 21.5V, the regulator immediately resumes normal operation. The regulator does not execute its soft-start function when exiting an overvoltage condition. 3605af 8 LTC3605A OPERATION Output Voltage Programming Output Power Good The output voltage is set by an external resistive divider according to the following equation: When the LTC3605A’s output voltage is within the ±10% window of the regulation point, which is reflected back as a VFB voltage in the range of 0.54V to 0.66V, the output voltage is good and the PGOOD pin is pulled high with an external resistor. Otherwise, an internal open-drain pulldown device (12Ω) will pull the PGOOD pin low. To prevent unwanted PGOOD glitches during transients or dynamic VOUT changes, the LTC3605A’s PGOOD falling edge includes a blanking delay of approximately 52 switching cycles. VOUT = 0.6V • (1 + R2/R1) The resistive divider allows the VFB pin to sense a fraction of the output voltage as shown in Figure 1. VOUT R2 CFF FB LTC3605A R1 SGND 3605A F01 Figure 1. Setting the Output Voltage Programming Switching Frequency Connecting a resistor from the RT pin to SGND programs the switching frequency from 800kHz to 4MHz according to the following formula: Frequency (Hz) = 1.6e11 R T (W) The internal PLL has a synchronization range of ±30% around its programmed frequency. Therefore, during external clock synchronization be sure that the external clock frequency is within this ±30% range of the RT programmed frequency. Output Voltage Tracking and Soft-Start The LTC3605A allows the user to program its output voltage ramp rate by means of the TRACK/SS pin. An internal 2µA pulls up the TRACK/SS pin to INTVCC. Putting an external capacitor on TRACK/SS enables soft starting the output to prevent current surge on the input supply. For output tracking applications, TRACK/SS can be externally driven by another voltage source. From 0V to 0.6V, the TRACK/SS voltage will override the internal 0.6V reference input to the error amplifier, thus regulating the feedback voltage to that of TRACK/SS pins. During this start-up time, the LTC3605A will operate in discontinuous mode. When TRACK/SS is above 0.6V, tracking is disabled and the feedback voltage will regulate to the internal reference voltage. Multiphase Operation For output loads that demand more than 5A of current, multiple LTC3605As can be cascaded to run out of phase to provide more output current. The CLKIN pin allows the LTC3605A to synchronize to an external clock (±30% of frequency programmed by RT) and the internal phaselocked-loop allows the LTC3605A to lock onto CLKIN’s phase as well. The CLKOUT signal can be connected to the CLKIN pin of the following LTC3605A stage to line up both the frequency and the phase of the entire system. Tying the PHMODE pin to INTVCC, SGND or INTVCC/2 generates a phase difference (between CLKIN and CLKOUT) of 180 degrees, 120 degrees, or 90 degrees respectively, which corresponds to 2-phase, 3-phase or 4-phase operation. A total of 12 phases can be cascaded to run simultaneously out of phase with respect to each other by programming the PHMODE pin of each LTC3605A to different levels. Internal/External ITH Compensation During single phase operation, the user can simplify the loop compensation by tying the ITH pin to INTVCC to enable internal compensation. This connects an internal 30k resistor in series with a 40pF capacitor to the output of the error amplifier (internal ITH compensation point) while also activating output voltage positioning such that the output voltage will be 1.5% above regulation at no load and 1.5% below regulation at full load. This is a trade-off for simplicity instead of OPTI-LOOP® optimization, where ITH components are external and are selected to optimize the loop transient response with minimum output capacitance. 3605af 9 LTC3605A OPERATION Minimum Off-Time and Minimum On-Time Considerations The minimum off-time, tOFF(MIN), is the smallest amount of time that the LTC3605A is capable of turning on the bottom power MOSFET, tripping the current comparator and turning the power MOSFET back off. This time is generally about 70ns. The minimum off-time limit imposes a maximum duty cycle of tON/(tON + tOFF(MIN)). If the maximum duty cycle is reached, due to a dropping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is: VIN(MIN) = VOUT • tON + tOFF(MIN) tON Conversely, the minimum on-time is the smallest duration of time in which the top power MOSFET can be in its “on” state. This time is typically 40ns. In continuous mode operation, the minimum on-time limit imposes a minimum duty cycle of: DCMIN = f • tON(MIN) where tON(MIN) is the minimum on-time. As the equation shows, reducing the operating frequency will alleviate the minimum duty cycle constraint. In the rare cases where the minimum duty cycle is surpassed, the output voltage will still remain in regulation, but the switching frequency will decrease from its programmed value. This is an acceptable result in many applications, so this constraint may not be of critical importance in most cases. High switching frequencies may be used in the design without any fear of severe consequences. As the sections on inductor and capacitor selection show, high switching frequencies allow the use of smaller board components, thus reducing the size of the application circuit. CIN and COUT Selection The input capacitance, CIN, is needed to filter the trapezoidal wave current at the drain of the top power MOSFET. To prevent large voltage transients from occurring, a low ESR input capacitor sized for the maximum RMS current should be used. The maximum RMS current is given by: IRMS ≅IOUT(MAX) VOUT VIN VIN –1 VOUT This formula has a maximum at VIN = 2VOUT, where IRMS ≅ IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. For low input voltage applications, sufficient bulk input capacitance is needed to minimize transient effects during output load changes. The selection of COUT is determined by the effective series resistance (ESR) that is required to minimize voltage ripple and load step transients as well as the amount of bulk capacitance that is necessary to ensure that the control loop is stable. Loop stability can be checked by viewing the load transient response. The output ripple, DVOUT, is determined by: 1 DVOUT < DIL + ESR 8 • f • COUT The output ripple is highest at maximum input voltage since DIL increases with input voltage. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic, and ceramic capacitors are all available in surface mount packages. Special polymer capacitors are very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-sensitive applications provided that consideration is given to ripple current ratings and long-term reliability. Ceramic capacitors have excellent low ESR characteristics and small footprints. Their relatively low value of bulk capacitance may require multiples in parallel. 3605af 10 LTC3605A OPERATION Using Ceramic Input and Output Capacitors Inductor Selection Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. However, care must be taken when these capacitors are used at the input and output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the VIN input. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN large enough to damage the part. Given the desired input and output voltages, the inductor value and operating frequency determine the ripple current: When choosing the input and output ceramic capacitors, choose the X5R and X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. Since the ESR of a ceramic capacitor is so low, the input and output capacitor must instead fulfill a charge storage requirement. During a load step, the output capacitor must instantaneously supply the current to support the load until the feedback loop raises the switch current enough to support the load. The time required for the feedback loop to respond is dependent on the compensation and the output capacitor size. Typically, 3 to 4 cycles are required to respond to a load step, but only in the first cycle does the output drop linearly. The output droop, VDROOP , is usually about 2 to 3 times the linear drop of the first cycle. Thus, a good place to start with the output capacitor value is approximately: COUT ≈ 2.5 DIOUT fO • VDROOP More capacitance may be required depending on the duty cycle and load step requirements. In most applications, the input capacitor is merely required to supply high frequency bypassing, since the impedance to the supply is very low. A 22µF ceramic capacitor is usually enough for these conditions. Place this input capacitor as close to the PVIN pins as possible. DIL = VOUT VOUT 1– f • L VIN(MAX) Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and output voltage ripple. Highest efficiency operation is obtained at low frequency with small ripple current. However, achieving this requires a large inductor. There is a trade-off between component size, efficiency and operating frequency. A reasonable starting point is to choose a ripple current that is about 50% of IOUT(MAX). This is especially important at low VOUT operation where VOUT is 1.8V or below. Care must be given to choose an inductance value that will generate a big enough current ripple (40% to 50%) so that the chip’s valley current comparator has enough signal-to-noise ratio to force constant switching frequency. Meanwhile, also note that the largest ripple current occurs at the highest VIN. To guarantee that ripple current does not exceed a specified maximum, the inductance should be chosen according to: L= VOUT V • 1– OUT f • DIL(MAX) VIN(MAX) Once the value for L is known, the type of inductor must be selected. Actual core loss is independent of core size for a fixed inductor value, but is very dependent on the inductance selected. As the inductance or frequency increases, core losses decrease. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard”, which means that 3605af 11 LTC3605A OPERATION inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent output voltage ripple. Do not allow the core to saturate! Table 1. Inductor Selection Table INDUCTANCE DCR MAX CURRENT DIMENSIONS HEIGHT 6.7mm × 7mm 3mm Vishay IHLP-2525CZ-01 Series 0.33µH 4.1mW 18A 0.47µH 6.5mW 13.5A 0.68µH 9.4mW 11A 0.82µH 11.8mW 10A 1.0µH 14.2mW 9A Checking Transient Response Vishay IHLP-1616BZ-11 Series 0.22µH 4.1mW 12A 0.47µH 15mW 7A 4.3mm × 4.7mm 2.0mm 7mm × 7.7mm 2.0mm 6.9mm × 7.7mm 3.0mm Toko FDV0620 Series 0.20µH 4.5mW 12.4A 0.47µH 8.3mW 9A 1µH 18.3mW 5.7A NEC/Tokin MLC0730L Series 0.47µH 4.5mW 16.6A 0.75µH 7.5mW 12.2A 1µH 9mW 10.6A Cooper HCP0703 Series 0.22µH 2.8mW 23A 0.47µH 4.2mW 17A 0.68µH 5.5mW 15A 0.82µH 8mW 13A 1µH 10mW 11A 1.5µH 14mW 9A 7mm × 7.3mm 3.0mm 6.9mm × 7.3mm 3.2mm 7mm × 7.7mm 3.8mm TDK RLF7030 Series 1µH 8.8mW 6.4A 1.5µH 9.6mW 6.1A 2.2µH 12mW 5.4A Wurth Electronik WE-HC 744312 Series 0.25µH 2.5mW 18A 0.47µH 3.4mW 16A 0.72µH 7.5mW 12A 1µH 9.5mW 11A 1.5µH 10.5mW 9A Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar characteristics. The choice of which style inductor to use mainly depends on the price versus size requirements and any radiated field/EMI requirements. New designs for surface mount inductors are available from Toko, Vishay, NEC/Tokin, Cooper, TDK and Wurth Electronik. Refer to Table 1 for more details. The OPTI-LOOP compensation allows the transient response to be optimized for a wide range of loads and output capacitors. The availability of the ITH pin not only allows optimization of the control loop behavior but also provides a DC-coupled and AC-filtered closed-loop response test point. The DC step, rise time and settling at this test point truly reflects the closed-loop response. Assuming a predominantly second order system, phase margin and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The ITH external components shown in the circuit on the first page of this data sheet provides an adequate starting point for most applications. The series R-C filter sets the dominant pole zero loop compensation. The values can be modified slightly (from 0.5 to 2 times their suggested values) to optimize transient response once the final PC layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because their various types and values determine the loop feedback factor gain and phase. An output current pulse of 20% to 100% of full load current having a rise time of 1µs to 10µs will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to DILOAD • ESR, where ESR is the effective series resistance of COUT. DILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its 3605af 12 LTC3605A OPERATION steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second order overshoot/DC ratio cannot be used to determine phase margin. The gain of the loop increases with the R and the bandwidth of the loop increases with decreasing C. If R is increased by the same factor that C is decreased, the zero frequency will be kept the same, thereby keeping the phase the same in the most critical frequency range of the feedback loop. In addition, a feedforward capacitor, CFF , can be added to improve the high frequency response, as shown in Figure 1. Capacitor CFF provides phase lead by creating a high frequency zero with R2 which improves the phase margin. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Linear Technology Application Note 76. In some applications, a more severe transient can be caused by switching in loads with large (>10µF) input capacitors. The discharged input capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem, if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot Swap™ controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection and soft-starting. where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, three main sources usually account for most of the losses in LTC3605A circuits: 1) I2R losses, 2) switching and biasing losses, 3) other losses. 1. I2R losses are calculated from the DC resistances of the internal switches, RSW, and external inductor, RL. In continuous mode, the average output current flows through inductor L but is “chopped” between the internal top and bottom power MOSFETs. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1-DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus to obtain I2R losses: I2R losses = IOUT2(RSW + RL) 2. The INTVCC current is the sum of the power MOSFET driver and control currents. The power MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a power MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from INTVCC to ground. The resulting dQ/dt is a current out of INTVCC that is typically much larger than the DC control bias current. In continuous mode, IGATECHG = f(QT + QB), where QT and QB are the gate charges of the internal top and bottom power MOSFETs and f is the switching frequency. Since INTVCC is a low dropout regulator output powered by VIN, its power loss equals: Efficiency Considerations PLDO = VIN • IINTVCC The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: Refer to the IINTVCC vs Frequency curve in the Typical Performance Characterics for typical INTVCC current at various frequencies. 3. Other “hidden” losses such as transition loss and copper trace and internal load resistances can account for additional efficiency degradations in the overall power % Efficiency = 100%–(L1 + L2 + L3 +…) 3605af 13 LTC3605A OPERATION system. It is very important to include these “system” level losses in the design of a system. Transition loss arises from the brief amount of time the top power MOSFET spends in the saturated region during switch node transitions. The LTC3605A internal power devices switch quickly enough that these losses are not significant compared to other sources. Other losses including diode conduction losses during dead-time and inductor core losses which generally account for less than 2% total additional loss. and internal biasing current loss, transition loss, inductor core loss and other losses in the application. Therefore, the total power dissipated by the part is: PD = IOUT2 • RSW + VIN • IVIN (No Load) = 25A2 • 40.25mΩ + 12V • 11mA = 1.14W The QFN 4mm × 4mm package junction-to-ambient thermal resistance, θJA, is around 37°C/W. Therefore, the junction temperature of the regulator operating in a 25°C ambient temperature is approximately: Thermal Considerations TJ = 1.14W • 37°C/W + 25°C = 67°C In a majority of applications, the LTC3605A does not dissipate much heat due to its high efficiency and low thermal resistance of its exposed-back QFN package. However, in applications where the LTC3605A is running at high ambient temperature, high VIN, high switching frequency and maximum output current load, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 160°C, both power switches will be turned off until the temperature drops about 15°C cooler. Remembering that the above junction temperature is obtained from an RDS(ON) at 25°C, we might recalculate the junction temperature based on a higher RDS(ON) since it increases with temperature. Redoing the calculation assuming that RSW increased 15% at 67°C yields a new junction temperature of 72°C. If the application calls for a higher ambient temperature and/or higher switching frequency, care should be taken to reduce the temperature rise of the part by using a heat sink or air flow. Figure 2 is a temperature derating curve based on the DC1215 demo board. TRISE = PD • θJA As an example, consider the case when the LTC3605A is used in applications where VIN = 12V, IOUT = 5A, f = 1MHz, VOUT = 1.8V. The equivalent power MOSFET resistance RSW is: RSW = RDS(ON)Top • V VOUT + RDS(ON)Bot 1– OUT VIN VIN 1.8 10.2 + 35mW • 12 12 = 40.25mW = 70mW • The VIN current during 1MHz force continuous operation with no load is about 11mA, which includes switching 6 5 LOAD CURRENT (A) To avoid the LTC3605A from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: 4 3 2 VIN = 12V VOUT = 3.3V fSW = 1MHz DC1215 DEMO BOARD 1 0 20 60 80 100 120 40 AMBIENT TEMPERATURE (°C) 140 3605A F02 Figure 2. Load Current vs Ambient Temperature Junction Temperature Measurement The junction-to-ambient thermal resistance will vary depending on the size and amount of heat sinking copper on the PCB board where the part is mounted, as well as the amount of air flow on the device. One of the ways to 3605af 14 LTC3605A OPERATION measure the junction temperature directly is to use the internal junction diode on one of the pins (PGOOD) to measure its diode voltage change based on ambient temperature change. First remove any external passive component on the PGOOD pin, then pull out 100µA from the PGOOD pin to turn on its internal junction diode and bias the PGOOD pin to a negative voltage. With no output current load, measure the PGOOD voltage at an ambient temperature of 25°C, 75°C and 125°C to establish a slope relationship between the delta voltage on PGOOD and delta ambient temperature. Once this slope is established, then the junction temperature rise can be measured as a function of power loss in the package with corresponding output load current. Keep in mind that doing so will violate absolute maximum voltage ratings on the PGOOD pin, however, with the limited current, no damage will result. Board Layout Considerations When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3605A (refer to Figure 3). Check the following in your layout: 1.Do the capacitors CIN connect to the power PVIN and power PGND as close as possible? These capacitors provide the AC current to the internal power MOSFETs and their drivers. CIN 2. Are COUT and L1 closely connected? The (–) plate of COUT returns current to PGND and the (–) plate of CIN. 3.The resistive divider, R1 and R2, must be connected between the (+) plate of COUT and a ground line terminated near SGND. The feedback signal VFB should be routed away from noisy components and traces, such as the SW line, and its trace should be minimized. Keep R1 and R2 close to the IC. 4.Solder the Exposed Pad (Pin 25) on the bottom of the package to the PGND plane. Connect this PGND plane to other layers with thermal vias to help dissipate heat from the LTC3605A. 5. Keep sensitive components away from the SW pin. The RT resistor, the compensation capacitor CC and CITH and all the resistors R1, R3 and RC, and the INTVCC bypass capacitor, should be placed away from the SW trace and the inductor L1. Also, the SW pin pad should be kept as small as possible. 6.A ground plane is preferred, but if not available, keep the signal and power grounds segregated with smallsignal components returning to the SGND pin which is then connected to the PGND pin at the negative terminal of the output capacitor, COUT. Flood all unused areas on all layers with copper, which reduces the temperature rise of power components. These copper areas should be connected to PGND. L1 VIN GND VOUT VIN VOUT COUT GND 3605A F03a Figure 3a. Sample PCB Layout—Topside 3605A F03b Figure 3b. Sample PCB Layout—Bottom Side 3605af 15 LTC3605A OPERATION Design Example COUT will be selected based on the ESR that is required to satisfy the output voltage ripple requirement and the bulk capacitance needed for loop stability. For this design, two 47µF ceramic capacitors will be used. As a design example, consider using the LTC3605A in an application with the following specifications: VIN = 10.8V to 13.2V, VOUT = 1.8V, IOUT(MAX) = 5A, IOUT(MIN) = 500mA, f = 2MHz CIN should be sized for a maximum current rating of: 1.8V 13.2V IRMS = 5A – 1 13.2V 1.8V Because efficiency is important at both high and low load current, discontinuous mode operation will be utilized. First select from the characteristic curves the correct RT resistor value for 2MHz switching frequency. Based on that RT should be 80.6k. Then calculate the inductor value for about 50% ripple current at maximum VIN: 1/2 = 1.7A Decoupling the PVIN pins with two 22µF ceramic capacitors is adequate for most applications. 1.8V 1.8V L= 1– = 0.31µH 2MHz • 2.5A 13.2V The nearest standard value inductor would be 0.33µH. TYPICAL APPLICATIONS 12V to 1.2V 1MHz Buck Regulator C1 2.2µF 0.1µF D1 10Ω 24 1 162k 2 3 4 5 12k 330pF 6 10pF 0.1µF 22 23 21 20 CLKIN CLKOUT SGND INTVCC BOOST SVIN RT PVIN PHMODE PVIN MODE SW LTC3605A FB SW TRACK/SS SW ITH SW RUN PGOOD 7 SVIN VON PGND 9 10 8 SW 11 CIN 22µF ×2 19 18 VIN 4V TO 20V 0.1µF 17 16 15 L1 0.68µH 47µF 14 COUT 47µF ×2 13 SW 4.99k VOUT 1.2V 5A 4.99k 12 100k PGND SGND 3605A TA02 C1: AVX 0805ZD225MAT2A CIN: TDK C4532X5RIC226M COUT: TDK C3216X5ROJ476M D1: CENTRAL SEMI CMDSH-3 L1: VISHAY IHLP-2525CZERR68-M01 3605af 16 LTC3605A TYPICAL APPLICATIONS 12V, 10A 2-Phase Single Output Regulator 0.1µF C1 2.2µF D1 10Ω 24 1 162k 2 3 4 5 6 5.4k 470pF 0.1µF 23 22 21 20 CLKIN CLKOUT SGND INTVCC BOOST SVIN RT PVIN PHMODE PVIN MODE 18 SW TRACK/SS SW 15 10pF PGOOD 7 VON PGND 9 10 8 SVIN SW 11 L1 1.5µH 14 10k COUT1 47µF ×2 13 SW RUN 0.1µF 16 SW FB VIN 4V TO 20V 17 LTC3605A ITH CIN1 22µF ×2 19 SW 100pF VOUT 3.3V 10A 2.21k 12 100k PGND SGND 0.1µF C2 2.2µF D2 10Ω 24 1 162k 2 3 4 5 6 5.4k 470pF 12V, 10A, 2-Phase Efficiency 100 90 EFFICIENCY (%) 80 70 20 PVIN PHMODE PVIN SW MODE LTC3605A FB SW TRACK/SS SW ITH SW PGOOD 8 VON PGND 9 10 SW 11 CIN2 22µF ×2 19 0.1µF 18 17 16 15 L2 1.5µH 14 COUT2 47µF 13 SW 12 SVIN PGND SGND 3605A TA03 12V, 10A, 2-Phase Load Step VOUT 100mV/DIV AC-COUPLED DCM CCM 60 IL1 2A/DIV 50 40 IL2 2A/DIV 30 20 20µs/DIV 10 0 0.1 21 RT 7 C1, C2: AVX 0805ZD225MAT2A CIN1, CIN2: TDK C4532X5RIC226M COUT1, COUT2: TDK C3216X5ROJ476M D1, D2: CENTRAL SEMI CMDSH-3 L1, L2: VISHAY IHLP-2525CZER1R5-M01 22 CLKIN CLKOUT SGND INTVCC BOOST SVIN RUN 10pF 23 1 LOAD CURRENT (A) 3605A TA03c 10 3605A TA03b 3605af 17 LTC3605A TYPICAL APPLICATIONS Dual Output Tracking Application C1 2.2µF 0.1µF D1 10Ω 23 24 1 162k 2 3 4 5 16.2k 100pF 6 10pF 0.1µF 22 21 20 CLKIN CLKOUT SGND INTVCC BOOST SVIN RT PVIN PHMODE PVIN MODE 18 SW TRACK/SS SW ITH SW PGOOD 7 VON PGND 9 10 8 SVIN1 SW 15 L1 0.33µH 14 7.5k COUT1 47µF 13 11 4.99k PGND SGND 0.1µF D2 1 162k 2 3 4 5 16.2k 100pF 6 10pF 23 22 21 20 RT PVIN PHMODE PVIN MODE SW LTC3605A FB SW TRACK/SS SW ITH SW 7 SVIN2 PGOOD VON PGND 9 10 8 SW CIN2 22µF ×2 19 CLKIN CLKOUT SGND INTVCC BOOST SVIN RUN 2.49k 12 10Ω 24 VOUT1 1.8V 5A SW 100k C2 2.2µF 0.1µF 16 SW FB VIN1 4V TO 20V 17 LTC3605A RUN CIN1 22µF ×2 19 18 VIN2 4V TO 20V 0.1µF 17 16 15 L2 0.33µH VOUT2 1.2V 5A 4.99k 14 13 COUT2 47µF 4.99k SW 11 12 100k PGND SGND 3605A TA04 C1, C2: AVX 0805ZD225MAT2A CIN1, CIN2: TDK C4532X5RIC226M COUT1, COUT2: TDK C3216X5ROJ476M D1, D2: CENTRAL SEMI CMDSH-3 L1, L2: VISHAY IHLP-2525CZERR33-M01 Dual Output Tracking Waveform VOUT1 500mV/DIV VOUT2 500mV/DIV 500µs/DIV VIN = 12V VOUT1 = 1.8V, VOUT2 = 1.2V IOUT1 = 80mA, IOUT2 = 80mA 3605A TA04b 3605af 18 LTC3605A PACKAGE DESCRIPTION Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. UF Package 24-Lead Plastic QFN (4mm × 4mm) (Reference LTC DWG # 05-08-1697 Rev B) 0.70 ±0.05 4.50 ±0.05 2.45 ±0.05 3.10 ±0.05 (4 SIDES) PACKAGE OUTLINE 0.25 ±0.05 0.50 BSC RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS 4.00 ±0.10 (4 SIDES) BOTTOM VIEW—EXPOSED PAD R = 0.115 TYP 0.75 ±0.05 PIN 1 NOTCH R = 0.20 TYP OR 0.35 × 45° CHAMFER 23 24 PIN 1 TOP MARK (NOTE 6) 0.40 ±0.10 1 2 2.45 ±0.10 (4-SIDES) (UF24) QFN 0105 REV B 0.200 REF 0.00 – 0.05 0.25 ±0.05 0.50 BSC NOTE: 1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WGGD-X)—TO BE APPROVED 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE, IF PRESENT 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3605af Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LTC3605A TYPICAL APPLICATION –3.6V Negative Converter C1 2.2µF CIN 22µF ×2 0.1µF D1 10Ω 24 1 2 162k 3 4 5 16.2k 470pF 6 47pF 0.1µF 23 22 21 20 CLKIN CLKOUT SGND INTVCC BOOST SVIN RT PVIN PHMODE PVIN MODE SW LTC3605A FB SW TRACK/SS SW ITH SW RUN 7 SVIN PGOOD VON PGND 9 10 8 SW 11 VIN 3V TO 16V 19 0.1µF 18 17 16 15 L1 1µH 14 24.9k 13 COUT 47µF ×2 SW 4.99k 12 100k 3605A TA05 –3.6V Negative Converter Efficiency VOUT –3.6V 2A –3.6V Negative Converter 100 90 EFFICIENCY (%) 80 DCM 70 SW 60 CCM 50 IL 2A/DIV 40 30 20 10 0 0.01 0.1 1 LOAD CURRENT (A) 10 VIN = 12V VOUT = –3.6V ILOAD = 2A 400ns/DIV 3605A TA05c 3605A TA05b RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC3605 15V, 5A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 15V, VOUT(MIN) = 0.6V, IQ = 2mA, ISD < 1µA, 4mm × 4mm QFN24 LTC3603 15V, 2.5A (IOUT), 3MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4.5V to 15V, VOUT(MIN) = 0.6V, IQ = 75µA, ISD < 1µA, 3mm × 3mm QFN16, MSE16 LTC3414/LTC3416 4A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converters 95% Efficiency, VIN: 2.25V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64µA, ISD < 1µA, TSSOP20E LTC3415 7A (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 450µA, ISD < 1µA, 5mm × 7mm QFN38 LTC3608 18V, 8A (IOUT) 1MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 18V, VOUT(MIN) = 0.6V, IQ = 900µA, ISD < 15µA, 7mm × 8mm QFN52 LTC3610 24V, 12A (IOUT), 1MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 24V, VOUT(MIN) = 0.6V, IQ = 900µA, ISD < 15µA, 9mm × 9mm QFN64 LTC3611 32V, 10A (IOUT), 1MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN: 4V to 32V, VOUT(MIN) = 0.6V, IQ = 900µA, ISD < 15µA, 9mm × 9mm QFN64 3605af 20 Linear Technology Corporation LT 0512 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2012