MIC5190 Micrel MIC5190 Ultra High-Speed, High-Current Active Filter/LDO Controller General Description Features The MIC5190 is an ultra high-speed linear regulator. It uses an external N-Channel FET as its power device. The MIC5190 offers ultra high-speed to cope with the fast load demands of microprocessor cores, ASICs, and other high-speed devices. Signal bandwidths of greater than 500kHz can be achieved with a minimum amount of capacitance while at the same time keeping the output voltage clean, regardless of load demand. A powerful output driver delivers large MOSFETs into their linear regions, achieving ultra-low dropout voltage. 1.25VIN±10% can be turned into 0.9V ±1% without the use of a large amount of capacitance. MIC5190 (0.5V reference) is optimized for output voltages of below 1.0V. The MIC5190 is offered in 10-lead 3mm × 3mm MLF™ and 10-lead MSOP-10 packages and has an operating junction temperature range of –40°C to +125°C. All support documentation can be found on Micrel’s web site at www.micrel.com. • Input voltage range: VIN = 0.9V to 5.5V • +1.0% initial output tolerance • Dropout down to 25mV@10A • Filters out switching frequency noise on input • Very high large signal bandwidth >500kHz • PSRR >40dB at 500kHz • Adjustable output voltage down to 0.5V • Stable with any output capacitor • Excellent line and load regulation specifications • Logic controlled shutdown • Current limit protection • 3mm × 3mm 10-lead MLF™ and MSOP-10 packages • Available –40°C to +125°C junction temperature Applications • Distributed power supplies • ASIC power supplies • DSP, µP, and µC power supplies Typical Application VCC = 12V C1 0.01µF VIN =1.2V VOUT = 0.9V@7A IR3716S MIC5190 IS OUT VIN VCC1 FB VCC2 PGND EN C3 0.01µF R1 100Ω R2 125Ω C2 10µF SGND COMP R3 12.5kΩ GND GND MicroLeadFrame and MLF are trademarks of Amkor Technology, Inc. PowerPAK is a trademark of Siliconix, Inc. Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com December 2005 1 M9999-120105 MIC5190 Micrel Ordering Information FB Output Output Part Number Voltage Current Voltage Standard Pb-Free MIC5190BML MIC5190YML 0.5V ADJ ADJ MIC5190BMM MIC5190YMM 0.5V ADJ ADJ Junction Temp. Range Package –40°C to +125°C –40°C to +125°C 10-pin MLF™ MSOP-10 Pin Configuration VIN 1 FB 2 SGND 3 VCC1 4 COMP 5 10 IS VIN 1 9 PGND FB 2 8 OUT SGND 3 7 VCC2 VCC1 4 6 EN COMP 5 MLF™-10 (ML) 10 IS 9 PGND 8 OUT 7 VCC2 6 EN MSOP-10 (MM) Pin Description Pin Number Pin Name 1 VIN Input voltage (Current Sense +). 2 FB Feedback input to error amplifier. 3 SGND Signal ground. 4 VCC1 Supply to the internal voltage regulator. 5 COMP Error amplifier output for external compensation. 6 EN 7 VCC2 Power to output driver. 8 OUT Output drive to gate of power MOSFET. 9 PGND Power ground. 10 IS Current sense. December 2005 Pin Function Enable (Input): CMOS-compatible. Logic high = Enable, Logic low = Shutdown. Do not float pin. 2 M9999-120105 MIC5190 Micrel Absolute Maximum Ratings(1) Operating Ratings(3) Supply Voltage (VIN) .................................................. +6.0V Enable Input Voltage (VEN) ......................................... +14V VCC1, VCC2 ............................................................... +14V Junction Temperature (TJ) ................ –40°C ≤ TJ ≤ +125°C ESD ......................................................................... Note 2 Supply Voltage (VIN) ................................... +0.9V to +5.5V Enable Input Voltage (VEN) ................................. 0V to VCC VCC1,VCC2 ............................................... +4.5V to +13.2V Junction Temperature (TJ) ................ –40°C ≤ TJ ≤ +125°C Package Thermal Resistance MLF™ (θJA)(4) ..................................................... 60°C/W MSOP (θJA) (5) .............................................................. 200°C/W Electrical Characteristics(6) TA = 25°C with VIN = 1.2V, VCC = 12V, VOUT = 0.5V; bold values indicate –40°C < TJ < +125°C; unless otherwise specified. Parameter Condition Output Voltage Accuracy At 25°C Over temperature range Output Voltage Line Regulation Min VIN = 1.2V to 5.5V Feedback Voltage Typ Max Units –1 +1 % –2 +2 % –0.1 0.005 +0.1 %/V 0.495 0.5 0.505 V 0.02 0.5 % Output Voltage Load Regulation IL = 10mA to 1A VCC Pin Current (VCC1 + VCC2) Enable = 0V 40 VCC Pin Current (VCC1 + VCC2) Enable = 5V 15 20 mA VIN Pin Current Current from VIN 10 15 µA 13 30 µA 50 70 mV 25 100 µs FB Bias Current Current Limit Threshold 35 Start-up Time VEN = VIN Enable Input Threshold Regulator enable 0.8 Regulator shutdown Enable Pin Input Current 0.6 0.5 Enable Hysteresis µA V 0.2 V 100 mV VIL < 0.2V (Regulator shutdown) 100 nA VIH > 0.8V (Regulator enabled) 100 nA Notes: 1. Exceeding the absolute maximum ratings may damage the device. 2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF. 3. The device is not guaranteed to function outside its operating ratings. 4. Per JESD 51-5 (1S2P Direct Attach Method). 5. Per JESD 51-3 (1S0P). 6. Specification for packaged product only. December 2005 3 M9999-120105 MIC5190 Micrel Typical Characteristics 0.501 0.5 0.499 0.498 0.499 0.498 12.5 11.5 0.495 9.5 20 40 60 80 100 120 Temp (C) 10.5 0.497 0.496 8.5 0.497 0.496 0.495 -40 -20 0 0.501 0.5 7.5 0.496 0.495 0 1 2 3 4 5 6 7 8 9 10 Output Current (A) 0.504 0.503 0.502 6.5 0.498 0.497 VOUT vs. VCC Voltage 0.505 5.5 0.5 0.499 VOUT vs. Temperature Vout (V) 0.503 0.502 0.501 Vout (V) Output Voltage (V) 0.505 0.504 0.503 0.502 4.5 Load Regulation 0.505 0.504 VCC Current vs. VCCVoltage 0.8 0.5 10 8 6 Feedback Current vs. Temperature 13.5 40 VCC (V) 13.5 12.5 11.5 10.5 9.5 8.5 7.5 6.5 Enable Time vs. VCC Voltage 5.5 4.5 50 45 40 35 30 25 20 15 10 5 0 0 -40 -20 0 20 40 60 80 100 120 Temperature ( °C) 45 12.5 VCC Voltage(V) 5 50 11.5 13.5 12.5 11.5 10.5 9.5 8.5 7.5 6.5 4.5 5.5 10 10 55 10.5 11 15 60 9.5 12 20 8.5 13 65 CURRENT LIMIT (mA) Feedback Current (µA) 14 Current Limit Threshold vs. Vcc Voltage 7.5 25 9 4 2 0 -40 -20 0 20 40 60 80 100 120 Temperature (°C) 6.5 13.5 12.5 11.5 9.5 10.5 8.5 7.5 6.5 VCC Voltage (V) CC 15 5.5 4.5 13.5 12.5 11.5 9.5 10.5 8.5 7.5 6.5 5.5 0.3 5.5 0.4 Feedback Current Voltage vs. V Feedback Current (µA) Input Current ( µA) ENTH (V) 0.6 0.2 Input Current vs. Temperature 20 18 16 14 12 0.7 VCC Voltage (V) Enable Time (µsec) Enable Threshold vs. VCC Voltage 4.5 20 18 16 14 12 10 8 6 4 2 0 4.5 Input Current (mA) Vcc (V) VCC (V) Voltage December 2005 4 M9999-120105 MIC5190 Micrel Functional Characteristics Disable Transient ENABLE (1V/div) ENABLE (1V/div) OUTPUT (500mV/div) OUTPUT (500mV/div) Enable Transient TIME (100µs/div) TIME (10µs/div) 10A Load Transient December 2005 LOAD CURRENT OUTPUT (5A/div) (10mV/div) LOAD CURRENT (5A/div) OUTPUT (10mV/div) INPUT (100mV/div) INPUT (100mV/div) Transient Response TIME (100µs/div) 5 TIME (100µs/div) M9999-120105 MIC5190 Micrel Functional Diagram VCC1 INTERNAL VOLTAGE REGULATOR 50mV VIN IS CURRENT LIMIT AMPLIFIER VCC2 EN ENABLE OUTPUT CONTROL AND LEVEL SHIFT OUT PGND FB 0.5V ERROR AMPLIFIER SGND COMP Figure 1. MIC5190 Block Diagram Enable The MIC5190 comes with an active-high enable pin that allows the regulator to be disabled. Forcing the enable pin low disables the regulator and sends it into a low off-modecurrent state. Forcing the enable pin high enables the output voltage. The enable pin cannot be left floating; a floating enable pin may cause an indeterminate state on the output. FB The feedback pin is used to sense the output voltage for regulation. The feedback pin is compared to an internal 0.5V reference and the output adjusts the gate voltage accordingly to maintain regulation. Since the feedback biasing current is typically 13µA, smaller feedback resistors should be used to minimize output voltage error. COMP COMP is the external compensation pin. This allows complete control over the loop to allow stability for any type of output capacitor, load currents and output voltage. A detailed explanation of how to compensate the MIC5190 is in the “Designing with the MIC5190” section. SGND, PGND SGND is the internal signal ground which provides an isolated ground path from the high current output driver. The signal ground provides the grounding for noise sensitive circuits such as the current limit comparator, error amplifier and the internal reference voltage. PGND is the power ground and is the grounding path for the output driver. Functional Description VIN The VIN pin is connected to the N-Channel drain. VIN is the input power being supplied to the output. This pin is also used to power the internal current limit comparator and compare the ISENSE voltage for current limit. The voltage range is from 0.9V min to 5.5V max. ISENSE The ISENSE pin is the other input to the current limit comparator. The output current is limited when the ISENSE pin's voltage is 50mV less than the VIN pin. In cases where there is a current limited source and there isn’t a need for current limit, this pin can be tied directly to VIN. Its operating voltage range, like the VIN pin, is 0.9V min to 5.5V max. VCC1, VCC2 VCC1 supplies the error amplifier and internal reference, while VCC2 supplies the output gate drive. For this reason, ensure these pins have good input capacitor bypassing for better performance. The operating range is from 4.5V to 13.2V and both VCC pins should be tied together. Ensure that the voltage supplied is greater than a gate-source threshold above the output voltage for the N-Channel MOSFET selected. Output The output drives the external N-Channel MOSFET and is powered from VCC. The output can sink and source over 150mA of current to drive either an N-Channel MOSFET or an external NPN transistor. The output drive also has short circuit current protection. December 2005 6 M9999-120105 MIC5190 Micrel Applications Information ∆V = Designing with the MIC5190 Anatomy of a transient response Output voltage variation will depend on two factors: loop bandwidth and output capacitance. The output capacitance will determine how far the voltage will fall over a given time. With more capacitance, the drop in voltage will fall at a decreased rate. This is the reason that more capacitance provides a better transient response for the same given bandwidth. Load Current The measure of a regulator is how accurately and effectively it can maintain a set output voltage, regardless of the load's power demands. One measure of regulator response is the load step. The load step gauges how the regulator responds to a change in load current. Figure 2 is a look at the transient response to a load step. Output Voltage AC-Coupled 1 ∫ idt C ∆V ↓= ∆V = L di V= The time it takes for the regulator to respond is directly proportional to its bandwidth gain. Higher bandwidth control loops respond quicker causing a reduced drop on the supply for the same amount of capacitance. 1 ∫ idt C dt 1 BW Output voltage vs. time during recovery is directly proportional to gain vs. frequency. 1 ∫ idt ↓ C Final recovery back to the regulated voltage is the final phase of transient response and the most important factors are gain and time. Higher gain at higher frequency will get the output voltage closer to its regulation point quicker. The final settling point will be determined by the load regulation, which is proportional to DC (0Hz) gain and the associated loss terms. ∆V ↓= Time Figure 2. Typical Transient Response At the start of a circuit's power demand, the output voltage is regulated to its set point, while the load current runs at a constant rate. For many different reasons, a load may ask for more current without warning. When this happens, the regulator needs some time to determine the output voltage drop. This is determined by the speed of the control loop. So, until enough time has elapsed, the control loop is oblivious to the voltage change. The output capacitor must bear the burden of maintaining the output voltage. There are other factors that contribute to large signal transient response, such as source impedance, phase margin, and PSRR. For example, if the input voltage drops due to source impedance during a load transient, this will contribute to the output voltage deviation by filtering through to the output reduced by the loops PSRR at the frequency of the voltage transient. It is straightforward: good input capacitance reduces the source impedance at high frequencies. Having between 35° and 45° of phase margin will help speed up the recovery time. This is caused by the initial overshoot in response to the loop sensing a low voltage. Compensation The MIC5190 has the ability to externally control gain and bandwidth. This allows the MIC5190 design to be individually tailored for different applications. In designing the MIC5190, it is important to maintain adequate phase margin. This is generally achieved by having the gain cross the 0dB point with a single pole 20dB/decade roll-off. The compensation pin is configured as Figure 3 demonstrates. di dt Since this is a sudden change in voltage, the capacitor will try to maintain voltage by discharging current to the output. The first voltage drop is due to the output capacitor's ESL (equivalent series inductance). The ESL will resist a sudden change in current from the capacitor and drop the voltage quickly. The amount of voltage drop during this time will be proportional to the output capacitor's ESL and the speed at which the load steps. Slower load current transients will reduce this effect. ∆V = L di dt ↑ Placing multiple small capacitors with low ESL in parallel can help reduce the total ESL and reduce voltage droop during high speed transients. For high speed transients, the greatest voltage deviation will generally be caused by output capacitor ESL and parasitic inductance. ∆V ↓= L Internal Error Amplifier 3.42MΩ Driver 20pF di dt After the current has overcome the effects of the ESL, the output voltage will begin to drop proportionally to time and inversely proportional to output capacitance. ∆V ↓= L ↓ December 2005 1 ∫ idt ↑C External Comp Figure 3. Internal Compensation 7 M9999-120105 MIC5190 Micrel This places a pole at 2.3 kHz at 80dB and calculates as follows. 1 2π × 3.42MΩ × 20pF FP = 2.32kHz 100 FP = 225 The Dominant Pole 1 Fp = 2 × 3 .42 M × Ccomp 80 180 135 180 60 135 40 90 20 45 0 0 -20 0.01 1 10 100 1000 Frequency (KHz) 90 R LOAD × COUT Pole 20 45 0 0 -45 0.01 0.1 1 10 100 1000 Frequency (KHz) 10000 100000 Figure 6. External Compensation Frequency Response It is recommended that the gain bandwidth should be designed to be less than 1 MHz. This is because most capacitors lose capacitance at high frequency and becoming resistive or inductive. This can be difficult to compensate for and can create high frequency ringing or worse, oscillations. By increasing the amount of output capacitance, transient response can be improved in multiple ways. First, the rate of voltage drop vs. time is decreased. Also, by increasing the output capacitor, the pole formed by the load and the output capacitor decreases in frequency. This allows for the increasing of the compensation resistor, creating a higher mid-band gain. 10000 100000 Figure 4. Internal Compensation Frequency Response There is single pole roll off. For most applications, an output capacitor is required. The output capacitor and load resistance create another pole. This causes a two-pole system and can potentially cause design instability with inadequate phase margin. External compensation is required. By providing a dominant pole and zero–allowing the output capacitor and load to provide the final pole–a net single pole roll off is created, with the zero canceling the dominant pole. Figure 5 demonstrates placing an external capacitor (CCOMP) and resistor (RCOMP) for the external pole-zero combination. Where the dominant pole can be calculated as follows: 100 225 80 180 Gain (dB) 60 Internal Error Amplifier 3.42MΩ 2 × Rcomp × Ccomp -20 -45 0.1 Fz = 40 1 Driver Increasing COUT reduces the load resistance and output capacitor pole allowing for an increase in mid-band gain. 40 135 90 20 45 0 0 Phase (Deg) 80 Phase (Deg) 225 Gain (dB) 100 Gain (dB) External Zero Phase (Deg) 60 20pF -20 0.01 External Comp Figure 5. External Compensation 1 2π × 3.42MΩ × CCOMP And the zero can be calculated as follows: 1 2π × RCOMP × CCOMP This allows for high DC gain, and high bandwidth with the output capacitor and the load providing the final pole. December 2005 10 100 1000 Frequency (KHz) 10000 100000 This will have the effect of both decreasing the voltage drop as well as returning closer and faster to the regulated voltage during the recovery time. MOSFET Selection The typical pass element for the MIC5190 is an N-Channel MOSFET. There are multiple considerations when choosing a MOSFET. These include: • VIN to VOUT differential • Output current • Case size/thermal characteristics • Gate capacitance (CISS<10nF) • Gate to source threshold CCOMP FZ = 1 Figure 7. Increasing Output Capacitance RCOMP FP = -45 0.1 8 M9999-120105 MIC5190 Micrel The VIN(min) to VOUT ratio and current will determine the maximum RDSON required. For example, for a 1.8V (±5%) to 1.5V conversion at 5A of load current, dropout voltage can be calculated as follows (using VIN(min)): RDSON (V = RDSON (1.71V − 1.5V) = RDSON IN − VOUT ) θ JA TSSOP-8 <950mW TSSOP-8 <1W <1.1W <1.125W TO-220/TO-263 (D2Pack) <1.4W >1.4W Table 1. Power Dissipation and Package Recommendation In our example, our power dissipation is greater than 1.4W, so we’ll choose a TO-263 (D2Pack) N-Channel MOSFET. θJA is calculated as follows. θJA = θJC + θCS + θSA Where θJC is the junction-to-case resistance, θCS is the case-to-sink resistance and the θSA is the sink-to-ambient air resistance. In the D2 package we’ve selected, the θJC is 2°C/W. The θCS, assuming we are using the PCB as the heat sink, can be approximated to 0.2°C/W. This allows us to calculate the minimum θSA: θSA= θJA– θCS – θJC θSA= 31°C/W – 0.2°C/W – 2°C/W θSA= 28.8°C/W Referring to Application Hint 17, Designing PCB Heat Sinks, the minimum amount of copper area for a D2Pack at 28.8°C/W is 2750mm2 (or 0.426in2 ). The solid line denotes convection heating only (2 oz. copper) and the dotted line shows thermal resistance with 250LFM airflow. The copper area can be significantly reduced by increasing airflow or by adding external heat sinks. Now that we know the amount of power we will be dissipating, we will need to know the maximum ambient air temperature. For our case we’re going to assume a maximum of 65°C ambient temperature. Different MOSFETs have different maximum operating junction temperatures. Most MOSFETs are rated to 150°C, while others are rated as high as 175°C. In this case, we’re going to limit our maximum junction temperature to 125°C. The MIC5190 has no internal thermal protection for the MOSFET so it is important that the design provides margin for the maximum junction temperature. Our design will maintain better than 125°C junction temperature with 1.95W of power dissipation at an ambient temperature of 65°C. Our thermal resistance calculates as follows: θ JA <850mW PowerPAK™ SO-8 D-Pack PD = (VIN – VOUT) × IOUT PD = (1.89V – 1.5V) × 5A PD = 1.95W ( ) ( TSOP-6 SO-8 Running the N-Channel in dropout will seriously affect transient response and PSRR (power supply ripple rejection). For this reason, we want to select a MOSFET that has lower than 42mΩ for our example application. Size is another important consideration. Most importantly, the design must be able to handle the amount of power being dissipated. The amount of power dissipated can be calculated as follows (using VIN(max)): TJ max − TJ ambient Power Dissipation PowerPAK™1212-8 IOUT 5A = 42mΩ θ JA = Package PC Board Heat Sink Thermal Resistance vs. Area ) PD 125°C − 65°C = 1.95W = 31°C / W So our package must have a thermal resistance less than 31°C /W. Table 1. shows a good approximation of power dissipation and package recommendation. Figure 8. PC Board Heat Sink Another important characteristic is the amount of gate capacitance. Large gate capacitance can reduce transient performance by reducing the ability of the MIC5190 to slew the gate. It is recommended that the MOSFET used has an input capacitance <10nF (CISS). December 2005 9 M9999-120105 MIC5190 Micrel Deviations on the input voltage will be reduced by the MIC5190’s PSRR, but nonetheless appear on the output. There really is no minimum input capacitance, but it is recommended that the input capacitance be equal to or greater than the output capacitance for best performance. Output Capacitor The MIC5190 is stable with any type or value of output capacitor (even without any output capacitor!). This allows the output capacitor to select which parameters of the regulator are important. In cases where transient response is the most important, low ESR and low ESL ceramic capacitors are recommended. Also, the more capacitance on the output, the better the transient response. The gate-source threshold specified in most MOSFET data sheets refers to the minimum voltage needed to fully enhance the MOSFET. Although for the most part, the MOSFET will be operating in the linear region and the VGS (gate-source voltage) will be less than the fully enhanced VGS, it is recommended the VCC voltage has 2V over the minimum VGS and output voltage. This is due to the saturation voltage of the MIC5190 output driver. VCC1,2 ≥ 2V + VGS + VOUT For our example, with a 1.5V output voltage, our MOSFET is fully enhanced at 4.5VGS, and so our VCC voltage should be greater or equal to 8V. Input Capacitor Good input bypassing is important for improved performance. Low ESR and low ESL input capacitors reduce both the drain of the N-Channel MOSFET, as well as the source impedance to the MIC5190. When a load transient on the output occurs, the load step will also appear on the input. VIN J1 +VIN 330µF 16V 10µF 10µF 10µF 10Ω 100k 22µF U1 MIC2198-BML 1µF 25V J2 EN CSH 6 2 HSD EN/UVLO VSW IRF7821 11 L1 CSH VOUT 1.8µH CDEP134-1R8MC-H 10k VOUT 1VOUT @10A 0.1µF 10Ω VOUT VIN 12 4 CSH BST 10µF 10 100pF 10Ω MIC5190 5 8 VOUT VOUT 3 10k 1 330µF Tantalum D2 1N5819HW OUT VCC1 VIN VCC2 ISENSE 7 GND 560pF 11.5k D1 SD103BWS VDD GND 9 100Ω IRF7821 LSD FB COMP 10µF 2.2µF 10V 1µF FB COMP 10nF 8.06k 100Ω 12.4k Figure 9. Post Regulator December 2005 10 M9999-120105 MIC5190 Micrel Active Filter Another application for the MIC5190 is as an active filter on the output of a switching regulator. This improves the power supply in several ways. First, using the MIC5190 as a filter on the output can significantly reduce high frequency noise. Switching power supplies tends to create noise at the switching frequency in the form of a triangular voltage ripple. High frequency noise is also created by the high-speed switching transitions. A lot of time, effort , and money are thrown into the design of switching regulators to minimize these effects as much as possible. Figure 9 shows the MIC5190 as a post regulator. Feedback Resistors VOUT IR3716S MIC5190 R1 FB R2 COUT GND INPUT RIPPLE (100mV/div) Figure 10. Adjustable Output The feedback resistors adjust the output to the desired voltage and can be calculated as follows: OUTPUT (10mV/div) R1 VOUT = VREF 1 + R2 VREF is equal to 0.5V for the MIC5190. The minimum output voltage (R1=0) is 0.5V. For output voltages greater than 1V, use the MIC5191. The resistor tolerance adds error to the output voltage. These errors are accumulative for both R1 and R2. For example, our resistors selected have a ±1% tolerance. This will contribute to a ±2% additional error on the output voltage. The feedback resistors must also be small enough to allow enough current to the feedback node. Large feedback resistors will contribute to output voltage error. VOUT = 1V ILOAD = 10A TIME (1µs/div) Figure 11. Ripple Reduction Figure 11 shows the amount of ripple reduction for a 500 kHz switching regulator. The fundamental switching frequency is reduced from greater than 100mV to less than 10mV. INPUT (100mV/div) VERROR = R1× IFB VERROR = 1kΩ × 12µA VERROR = 12mV OUTPUT (10mV/div) For our example application, this will cause an increase in output voltage of 12mV. For the percentage increase, VERROR × 100 VOUT 12mV × 100 VERROR % = 1.5V VERROR % = 0.8% LOAD CURRENT (5A/div) VERROR % = By reducing R1 to 100Ω, the error contribution by the feedback resistors and feedback current is reduced to less than 0.1%. This is the reason R1 should not be greater than 100Ω. Figure 12. 10A Load Transient The transient response also contributes to the overall AC output voltage deviation. Figure 12 shows a 1A to 10A load transient. The top trace is the output of the switching regulator (same circuit as Figure10). The output voltage undershoots by 100mV. Just by their topology, linear regulators have the ability to respond at much higher speeds than a switching regulator. Linear regulators do not have the limitation or restrictions of switching regulators which must reduce their bandwidth to less than their switching frequency. Applying the MIC5190 Linear Regulator The primary purpose of the MIC5190 is as a linear regulator, which enables an input supply voltage to drop down through the resistance of the pass element to a regulated output voltage. December 2005 TIME (100µs/div) 11 M9999-120105 MIC5190 Micrel lower voltages these parasitic values can easily bump the output voltage out of a usable tolerance. Using the MIC5190 as a filter for a switching regulator reduces output noise due to ripple and high frequency switching noise. It also reduces undershoot (Figure 12) and overshoot (Figure 13) due to load transients with decreased capacitance. Circuit Board Load INPUT (100mV/div) Load LOAD CURRENT OUTPUT (5A/div) (10mV/div) Load Switching Power Supply Long Traces Load Figure 14. Board Layout TIME (100µs/div) Figure 13. Transient Response But by placing multiple small MIC5190 circuits close to each load, the parasitic trace elements caused by distance to the power supply are almost completely negated. By adjusting the switching supply voltage to 1.2V, for our example, the MIC5190 will provide accurate 1V output, efficiently and with very little noise. Due to the high DC gain (80dB) of the MIC5190, it also adds increased output accuracy and extremely high load regulation. Distributed Power Supply As technology advances and processes move to smaller and smaller geometries, voltage requirements go down and current requirements go up. This creates unique challenges when trying to supply power to multiple devices on a board. When there is one load to power, the difficulties are not quite as complex; trying to distribute power to multiple loads from one supply is much more problematic. If a large circuit board has multiple small-geometry ASICs, it will require the powering of multiple loads with its one power source. Assuming that the ASICs are dispersed throughout the board and that the core voltage requires a regulated 1V, Figure 14 shows the long traces from the power supply to the ASIC loads. Not only do we have to contend with the tolerance of the supply (line regulation, load regulation, output accuracy, and temperature tolerances), but the trace lengths create additional issues with resistance and inductance. With Circuit Board Load MIC5190 MIC5190 Load Load MIC5190 Switching Power Supply MIC5190 Load Figure 15. Improved Distributed Supplies December 2005 12 M9999-120105 MIC5190 Micrel 3.15 (0.122) 2.85 (0.114) DIMENSIONS: MM (INCH) 4.90 BSC (0.193) 3.10 (0.122) 2.90 (0.114) 1.10 (0.043) 0.94 (0.037) 0.30 (0.012) 0.15 (0.006) 0.15 (0.006) 0.05 (0.002) 0.50 BSC (0.020) 0.26 (0.010) 0.10 (0.004) 6° MAX 0° MIN 0.70 (0.028) 0.40 (0.016) 10-Pin MS0P (MM) 10-Lead MLF™ (ML) MICREL, INC. 1849 FORTUNE DRIVE SAN JOSE, CA 95131 TEL + 1 (408) 944-0800 FAX USA + 1 (408) 474-1000 WEB http://www.micrel.com The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2004 Micrel, Incorporated. December 2005 13 M9999-120105