STMICROELECTRONICS L6726ATR

L6726A
Single phase PWM controller
Feature
■
Flexible power supply from 5 V to 12 V
■
Power conversion input as low as 1.5 V
■
1% output voltage accuracy
■
High-current integrated drivers
■
Adjustable output voltage
■
0.8 V internal reference
■
Sensorless and programmable OCP across
Low-side RdsON
■
Oscillator internally fixed at 270 kHz
■
Programmable soft-start
■
Ls-less start up
■
Disable function
■
FB disconnection protection
■
SO-8 package
SO-8
Description
L6726A is a single-phase step-down controller
with integrated high-current drivers that provides
complete control logic, protections and reference
voltage to realize in an easy and simple way
general DC-DC converters by using a compact
SO-8 package.
Device flexibility allows managing conversions
with power input VIN as low as 1.5 V and device
supply voltage ranging from 5 V to 12 V.
L6726A provides simple control loop with transconductance error amplifier. The integrated 0.8 V
reference allows regulating output voltage with
±1% accuracy over line and temperature
variations. Oscillator is internally fixed to 270 kHz.
Applications
■
Subsystem power supply (MCH, IOCH, PCI...)
■
Memory and termination supply
■
CPU and DSP power supply
■
Distributed power supply
■
General DC / DC converters
L6726A provides programmable over current
protection. Current information is monitored
across the low-side MOSFET RdsON saving the
use of expensive and space-consuming sense
resistors.
FB disconnection protection prevents excessive
and dangerous output voltages in case of floating
FB pin.
Table 1.
March 2010
Device summary
Order codes
Package
Packaging
L6726A
SO-8
Tube
L6726ATR
SO-8
Tape and reel
Doc ID 12754 Rev 4
1/35
www.st.com
35
Contents
L6726A
Contents
1
2
3
Typical application circuit and block diagram . . . . . . . . . . . . . . . . . . . . 4
1.1
Application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
1.2
Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Pins description and connection diagrams . . . . . . . . . . . . . . . . . . . . . . 5
2.1
Pin descriptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.2
Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Electrical specifications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3.1
Absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3.2
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
4
Device description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
5
Driver section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
5.1
6
7
Power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Soft start and disable . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
6.1
Low-side-less start up (LSLess) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
6.2
Enable / disable . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Protections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
7.1
Overcurrent protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
7.1.1
8
2/35
Overcurrent threshold setting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
7.2
Feedback disconnection protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
7.3
Undervoltage lock out . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Application details . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
8.1
Output voltage selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
8.2
Compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
8.3
Soft-start time calculation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
8.4
Layout guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
8.5
Embedding L6726A-based VRs… . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Doc ID 12754 Rev 4
L6726A
9
10
Contents
Application Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
9.1
Output inductor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
9.2
Output capacitors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
9.3
Input capacitors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
20 A demonstration board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
10.1
11
Board description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
10.1.1
Power input (VIN) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
10.1.2
Power output (VOUT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
10.1.3
IC additional supply (VCC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
10.1.4
Test points . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
10.1.5
Demonstration board efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
5 A demonstration board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
11.1
Board description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
11.1.1
Power input (VIN) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
11.1.2
Power output (VOUT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
11.1.3
IC additional supply (VCC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
11.1.4
Test points . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
11.1.5
Demonstration board efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
12
Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
13
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Doc ID 12754 Rev 4
3/35
Typical application circuit and block diagram
L6726A
1
Typical application circuit and block diagram
1.1
Application circuit
Figure 1.
Typical application circuit
VIN = 1.5V to 19V (1)
VCC = 5V to 12V
CDEC
D
5
7
VCC
BOOT
FB
COMP
/ DIS
CF
RFB
ROS
CP
RF
3
RD
1
CBOOT
L6726A
6
GND
UGATE
PHASE
LGATE
/ OC
2
CHF
CBULK
HS
RgHS
8
4
L
RgLS
LS
Vout
RSN
COUT
LOAD
CSN
ROCSET
L6726A Reference Schematic
(1) Up to 12V with Vcc > 5V
1.2
Block diagram
Block diagram
VCC
Figure 2.
VOCTH
OCP
CONTROL LOGIC
& PROTECTIONS
ISS
BOOT
CLOCK
S
PWM
Q
R
OSCILLATOR
TRANSCONDUCTANCE
ERROR AMPLIFIER
ADAPTIVE ANTI
CROSS CONDUCTION
DISABLE
HS
UGATE
PHASE
VCC
LS
LGATE
/ OC
GND
+
-
L6726A
FB
IOCSET
COMP
/ DIS
4/35
0.8V
Doc ID 12754 Rev 4
L6726A
2
Pins description and connection diagrams
Pins description and connection diagrams
Figure 3.
Pins connection (top view)
BOOT
UGATE
GND
LGATE / OC
2.1
1
8
2
7
3
L6726A
4
6
5
PHASE
COMP / DIS
FB
VCC
Pin descriptions
Table 2.
Pins descriptions
Pin n
Name
1
BOOT
HS driver supply.
Connect through a capacitor (100 nF) to the floating node (LS-drain) pin
and provide necessary bootstrap diode from VCC.
2
UGATE
HS driver output. Connect to HS MOSFET gate.
3
GND
4
5
6
Function
All internal references, logic and drivers are connected to this pin.
Connect to the PCB ground plane.
LGATE. LS driver output. Connect to LS MOSFET gate.
OC. Over current threshold set. During a short period of time following VCC
rising over UVLO threshold, a 10μA current is sourced from this pin.
Connect to GND with an ROCSET resistor greater than 5kΩ to program OC
LGATE / OC Threshold. The resulting voltage at this pin is sampled and held internally
as the OC set point. Maximum programmable OC threshold is 0.55 V. A
voltage greater than 0.75V (max) activates an internal clamp and causes
OC threshold to be set at 400 mV. ROCSET not connected sets the 400 mV
default threshold.
VCC
FB
Device and LS driver power supply.
Operative range from 4.1 V to 13.2 V. Filter with at least 1μF MLCC to GND.
Error amplifier inverting input.
Connect with a resistor RFB to the output regulated voltage. Additional
resistor ROS to GND may be used to regulate voltages higher than the
reference.
7
COMP. Error amplifier output. Connect with an RF - CF // CP to GND to
compensate the device control loop in conjunction to the FB pin.
During the soft-start phase, a 10 μA current is sourced from this pin so the
COMP / DIS
compensation capacitors also act to program the SS time.
DIS. The device can be disabled by pulling this pin lower than 0.4 V (min).
Setting free the pin, the device enables again.
8
HS driver return path, current-reading and adaptive-dead-time monitor.
Connect to the LS drain to sense RdsON drop to measure the output current.
This pin is also used by the adaptive-dead-time control circuitry to monitor
when HS MOSFET is OFF.
PHASE
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5/35
Pins description and connection diagrams
2.2
L6726A
Thermal data
Table 3.
Symbol
Thermal data
Parameter
Value
Unit
RthJA
Thermal resistance junction to ambient(1)
85
°C/W
TMAX
Maximum junction temperature
150
°C
TSTG
Storage temperature range
-40 to 150
°C
TJ
Junction temperature range
-20 to 150
°C
1. Measured with the component mounted on a 2S2P board in free air (6.7 cm x 6.7 cm, 35 μm (P) and
17.5 μm (S) copper thickness).
6/35
Doc ID 12754 Rev 4
L6726A
Electrical specifications
3
Electrical specifications
3.1
Absolute maximum ratings
Table 4.
Absolute maximum ratings
Symbol
VCC
Parameter
Value
Unit
-0.3 to 15
V
15
45
V
-0.3 to (VBOOT - VPHASE) + 0.3
-1
VBOOT + 0.3
V
-8 to 30
V
-0.3 to VCC + 0.3
-2.5
V
-0.3 to 3.6
V
to GND
VBOOT
to PHASE
to GND
VUGATE
to PHASE
to PHASE; t < 50ns
to GND
VPHASE
to GND
VLGATE
to GND
to GND; t < 50ns
FB, COMP to GND
3.2
Electrical characteristics
VCC = 12 V; TA = -20 °C to +85 °C unless otherwise specified.
Table 5.
Symbol
Electrical characteristics
Parameter
Test conditions
Min.
Typ.
Max.
Unit
13.2
V
13.2
V
19.0
V
Recommended operating conditions
VCC
VIN
Device supply voltage
4.1
See Figure 1
Conversion input voltage
VCC < 7.0 V
Supply current and power-ON
ICC
VCC supply current
UGATE and LGATE = OPEN
IBOOT
BOOT supply current
UGATE = OPEN; PHASE to GND
UVLO
VCC Turn-ON
VCC rising
6
mA
0.5
mA
4.1
Hysteresis
0.2
V
V
Oscillator
FSW
Main oscillator accuracy
ΔVOSC
PWM ramp amplitude
dMAX
Maximum duty cycle
TA = 0 °C to +70 °C
243
270
297
225
270
315
kHz
1.1
80
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V
%
7/35
Electrical specifications
Table 5.
Symbol
L6726A
Electrical characteristics (continued)
Parameter
Test conditions
Min.
Typ.
Max.
-1
-
1
-1.5
-
1.5
Unit
Reference
Output voltage accuracy
VOUT = 0.8 V, TA = 0 °C to 70 °C
VOUT = 0.8 V
%
Transconductance error amplifier
gm
Transconductance(1)
IFB
Input bias current
A0
F0
ICOMP
5
Sourced from FB
mS
100
nA
Open loop gain(1)
70
dB
Unity gain(1)
4
MHz
Source current
360
μA
Sink current
-360
μA
10
μA
0.5
V
Current capability
Soft-Start and disable
ISS
Soft-start current
From COMP pin
DIS
Disable threshold
COMP falling
IUGATE
HS source current
BOOT - PHASE = 5 V to 12 V
1.5
A
RUGATE
HS sink resistance
BOOT - PHASE = 5 V to 12 V
1.1
Ω
ILGATE
LS source current
VCC = 5 V to 12 V
1.5
A
RLGATE
LS sink resistance
VCC = 5 V to 12 V
0.65
Ω
10
μA
0.4
Gate drivers
Over-current protection
IOCSET
OCSET current source
Sourced from LGATE pin.
See Section 7.1.1
VOC_SW
OC switch-over threshold
VLGATE/OC rising
VOCTH_FIXED Fixed OC threshold
VPHASE to GND
1. Guaranteed by design, not subject to test.
8/35
Doc ID 12754 Rev 4
780
-400
mV
mV
L6726A
4
Device description
Device description
L6726A is a single-phase PWM controller with embedded high-current drivers that provides
complete control logic and protections to realize in an easy and simple way a general DCDC step-down converter. Designed to drive N-channel MOSFETs in a synchronous buck
topology, with its high level of integration this 8-pin device allows reducing cost and size of
the power supply solution.
L6726A is designed to operate from a 5 V or 12 V supply bus. Thanks to the high precision
0.8V internal reference, the output voltage can be precisely regulated to as low as 0.8 V with
±1% accuracy over line and temperature variations (between 0 °C and +70 °C). The
switching frequency is internally set to 270 kHz.
This device provides a simple control loop with externally compensated transconductance
error-amplifier and programmable soft start. Low-side-less feature allows the device to
perform soft-start over pre-charged output avoiding negative spikes at the load side.
In order to avoid load damages, L6726A provides programmable threshold over current
protection. Output current is monitored across low-side MOSFET RdsON, saving the use of
expensive and space-consuming sense resistor. L6726A also features FB disconnection
protection, preventing dangerous uncontrolled output voltages in case of floating FB pin.
Doc ID 12754 Rev 4
9/35
Driver section
5
L6726A
Driver section
The integrated high-current drivers allow using different types of power MOSFET (also
multiple MOSFETs to reduce the equivalent RdsON), maintaining fast switching transition.
The driver for high-side MOSFET uses BOOT pin for supply and PHASE pin for return. The
driver for low-side MOSFET uses the VCC pin for supply and GND pin for return.
The controller embodies an anti-shoot-through and adaptive dead-time control to minimize
low side body diode conduction time, maintaining good efficiency while saving the use of
Schottky diode:
●
to check for high-side MOSFET turn off, PHASE pin is sensed. When the voltage at
PHASE pin drops down, the low-side MOSFET gate drive is suddenly applied;
●
to check for low-side MOSFET turn off, LGATE pin is sensed. When the voltage at
LGATE has fallen, the high-side MOSFET gate drive is suddenly applied.
If the current flowing in the inductor is negative, voltage on PHASE pin will never drop. To
allow the low-side MOSFET to turn-on even in this case, a watchdog controller is enabled: if
the source of the high-side MOSFET doesn't drop, the low side MOSFET is switched on so
allowing the negative current of the inductor to recirculate. This mechanism allows the
system to regulate even if the current is negative.
Power conversion input is flexible: 5 V, 12 V bus or any bus that allows the conversion (See
maximum duty cycle limitation and recommended operating conditions) can be chosen
freely.
10/35
Doc ID 12754 Rev 4
L6726A
5.1
Driver section
Power dissipation
L6726A embeds high current MOSFET drivers for both high side and low side MOSFETs: it
is then important to consider the power that the device is going to dissipate in driving them
in order to avoid overcoming the maximum junction operative temperature.
Two main terms contribute in the device power dissipation: bias power and drivers power.
●
Device bias power (PDC) depends on the static consumption of the device through the
supply pins and it is simply quantifiable as follow (assuming to supply HS and LS
drivers with the same VCC of the device):
P DC = V CC ⋅ ( I CC + I BOOT )
●
Drivers power is the power needed by the driver to continuously switch on and off the
external MOSFETs; it is a function of the switching frequency, the voltage supply of the
driver and total gate charge of the selected MOSFETs. It can be quantified considering
that the total power PSW dissipated to switch the MOSFETs (easy calculable) is
dissipated by three main factors: external gate resistance (when present), intrinsic
MOSFET resistance and intrinsic driver resistance. This last term is the important one
to be determined to calculate the device power dissipation. The total power dissipated
to switch the MOSFETs results:
P SW = F SW ⋅ [ Q gHS ⋅ ( V BOOT – V PHASE ) + Q gLS ⋅ V CC ]
where VBOOT - VPHASE is the voltage across the bootstrap capacitor.
External gate resistors helps the device to dissipate the switching power since the same
power PSW will be shared between the internal driver impedance and the external resistor
resulting in a general cooling of the device.
Figure 4.
Soft start (left) and disable (right)
Doc ID 12754 Rev 4
11/35
Soft start and disable
6
L6726A
Soft start and disable
L6726A implements a soft start to smoothly charge the output filter avoiding high in-rush
currents to be required from the input power supply. The device sources a 10 µA soft start
current from COMP, linearly charging the compensation network capacitors. The ramping
COMP voltage is compared to the oscillator triangular waveform generating PWM pulses of
increasing width that charge the output capacitors.
When the FB voltage crosses 800 mV, the output voltage is in regulation: soft start phase
will end and the transconductance error amplifier output will be enabled closing the control
loop.
In the event of an over current during soft start, the over current logic will override the soft
start sequence and will shut down the PWM logic and both the high side and low side gates.
This condition is latched, cycle VCC to recover.
The device sources soft start current only when VCC power supply is above UVLO
threshold and over current threshold setting phase has been completed.
6.1
Low-side-less start up (LSLess)
L6726A performs a special sequence in enabling LS driver to switch: during the soft-start
phase, the LS driver results disabled (LS = OFF) until the HS starts to switch. This avoids
the dangerous negative spike on the output voltage that can happen if starting over a precharged output and limits the output discharge (amount of output discharge depends on
programmed SS time length: the shorter the programmed SS, the more limited the output
discharge).
If the output voltage is pre-charged to a voltage higher than the final one, the HS would
never start to switch. In this case, LS is enabled and discharges the output to the final
regulation value.
Figure 5.
6.2
LSLess startup (left) vs non-LSLess startup (right)
Enable / disable
The device can be disabled by pushing COMP / DIS pin under 0.4 V (min). In this condition
HS and LS MOSFETs are turned off, and the 10 μA SS current is sourced from COMP / DIS
pin. Setting free the pin, the device enables again performing a new SS.
12/35
Doc ID 12754 Rev 4
L6726A
Protections
7
Protections
7.1
Overcurrent protection
The overcurrent feature protects the converter from a shorted output or overload, by sensing
the output current information across the low side MOSFET drain-source on-resistance,
RdsON. This method reduces cost and enhances converter efficiency by avoiding the use of
expensive and space-consuming sense resistors.
The low side RdsON current sense is implemented by comparing the voltage at the PHASE
node when LS MOSFET is turned on with the programmed OCP threshold voltage,
internally held. If the monitored voltage drop (GND to PHASE) exceeds this threshold, an
Overcurrent event is detected. If two overcurrent events are detected in two consecutive
switching cycles, the protection will be triggered and the device will turn off both LS and HS
MOSFETs in a latched condition.
To recover from over current protection triggered, VCC power supply must be cycled.
7.1.1
Overcurrent threshold setting
L6726A allows to easily program an overcurrent threshold ranging from 50 mV to 550 mV,
simply by adding a resistor (ROCSET) between LGATE and GND.
During a short time following VCC rising over UVLO threshold, an internal 10µA current
(IOCSET) is sourced from LGATE pin, determining a voltage drop across ROCSET. This
voltage drop will be sampled and internally held by the device as overcurrent Threshold. The
OC setting procedure overall time length ranges from 5.5 ms to 6.5 ms, proportionally to the
threshold being set.
Connecting a ROCSET resistor between LGATE and GND, the programmed threshold will be:
I OCSET ⋅ R OCSET
I OCth = ------------------------------------------R dsON
ROCSET values range from 5 kΩ to 55 kΩ.
If the voltage drop across ROCSET is too low, the system will be very sensitive to start-up
inrush current and noise. This can result in undesired OCP triggering. In this case, consider
increasing ROCSET value.
In case ROCSET is not connected, the device switches the OCP threshold to a 400 mV
default value: an internal safety clamp on LGATE is triggered as soon as LGATE voltage
reaches 700 mV (typ), enabling the 400 mV default threshold and suddenly ending OC
setting phase.
See Figure 6 for OC threshold setting procedure timings picture and oscilloscope sample
waveforms.
7.2
Feedback disconnection protection
In order to provide load protection even if FB pin is not connected, a 100 nA bias current is
always sourced from this pin. If FB pin is not connected, bias current will permanently pull
up FB: this forces COMP pin low, avoiding output voltage rising to dangerous levels.
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13/35
Protections
L6726A
Figure 6.
OC threshold setting procedure timings (top) and waveforms (bottom)
ROCSET connected
ROCSET not connected
UVLO Th
UVLO Th
VCC
VCC
PWM ramp
bottom edge
PWM ramp
bottom edge
Enable Th
Enable Th
COMP
COMP
700mV
700mV
LGATE
LGATE
5.5ms - 6.5ms
tDELAY
Setting Procedure
7.3
Setting Procedure
tDELAY
Undervoltage lock out
In order to avoid anomalous behaviors of the device when the supply voltage is too low to
support its internal rails, UVLO is provided: the device will start up when VCC reaches
UVLO upper threshold and will shutdown when VCC drops below UVLO lower threshold.
The 4.1 V maximum UVLO upper threshold allows L6726A to be supplied from 5 V and 12 V
busses in or-ing diode configuration.
Figure 7.
14/35
OCP trip, default threshold, LS: STD38NH02L (left) UVLO turn off (right)
Doc ID 12754 Rev 4
L6726A
Application details
8
Application details
8.1
Output voltage selection
L6726A is capable to precisely regulate an output voltage as low as 0.8 V. In fact, the device
comes with a fixed 0.8 V internal reference that guarantees the output regulated voltage to
be within ±1% tolerance over line and temperature variations between 0 °C and 70 °C
(excluding output resistor divider tolerance, when present).
Output voltage higher than 0.8 V can be achieved by adding a resistor ROS between FB pin
and ground. Referring to Figure 1, the steady state DC output voltage will be:
R FB ⎞
V OUT = V REF ⋅ ⎛⎝ 1 + ---------R OS⎠
where VREF is 0.8 V.
8.2
Compensation network
The control loop shown in Figure 8 is a voltage mode control loop. The error amplifier is a
transconductance type with fixed gain (3.3 ms typ.). The FB voltage is regulated to the
internal reference, thus the output voltage is fixed accordingly to the output resistor divider
(when present).
Transconductance error amplifier output current generates a voltage across ZF, which is
compared to oscillator saw-tooth waveform to provide PWM signal to the driver section.
PWM signal is then transferred to the switching node with VIN amplitude. This waveform is
filtered by the output filter.
Figure 8.
PWM control loop
VIN
OSC
ΔV OSC
_
L
+
COUT
PWM
COMPARATOR
ESR
OTA
+
COMP
_
VREF
FB
CF
RF
V OUT
R
CP
R FB
R OS
OUTPUT
DIVIDER
ZF
The converter transfer function is the small signal transfer function between the voltage at
the output node of the EA (COMP) and VOUT. This function has a double pole (complex
conjugate) at frequency FLC depending on the L-COUT resonance and a zero at FESR
Doc ID 12754 Rev 4
15/35
Application details
L6726A
depending on the output capacitor ESR. The DC gain of the modulator is simply the input
voltage VIN divided by the peak-to-peak oscillator voltage ΔVOSC.
VOUT is scaled and transferred to FB node by the output resistor divider.
The compensation network closes the loop joining FB and COMP node with transfer
function ideally equal to -gm·ZF.
Compensation goal is to close the control loop assuring high DC regulation accuracy, good
dynamic performances and stability. To achieve this, the overall loop needs high DC gain,
high bandwidth and good phase margin.
High DC gain is achieved giving an integrator shape to compensation network transfer
function. Loop bandwidth (F0dB) can be fixed choosing the right RF; however, for stability, it
should not exceed FSW/2π. To achieve a good phase margin, the control loop gain has to
cross 0 dB axis with -20 dB/decade slope.
As an example, Figure 9 shows an asymptotic bode plot of a type II compensation.
Figure 9.
Example of type II compensation.
Gain
[dB]
OTA
open loop
gain
closed loop
gain
compensation
gain
FZ
FP
20log (gm·RF )
converter
open loop
gain
0dB
20log [VIN/ΔVOSC·ROS/(RFB+ROS)]
F0dB
FLC
●
Open loop converter singularities:
a)
b)
●
1
F LC = --------------------------------2π L ⋅ C OUT
1
F ESR = ------------------------------------------2π ⋅ C OUT ⋅ ESR
Compensation Network singularities frequencies:
a)
b)
16/35
FESR
1
F Z = -----------------------------2π ⋅ R F ⋅ C F
1
F P = ------------------------------------------------CF ⋅ CP ⎞
⎛
2π ⋅ R F ⋅ --------------------⎝ C F + C P⎠
Doc ID 12754 Rev 4
Log (Freq)
L6726A
Application details
Type II compensation relies on the zero introduced by the output capacitors bank to achieve
stability. Thus, a needed condition to successfully apply type II compensation is F ESR < F 0dB
(usually true when output capacitor is based on electrolytic, aluminium electrolytic or
tantalum capacitor).
To define compensation network components values, the below suggestions may be
followed:
a)
Set the output resistor divider in order to obtain the desired output voltage:
V OUT
R FB
---------- = -------------–1
R OS
V REF
Usual values of RFB and ROS ranges from some hundreds of Ω to some kΩ
(consider trade-off between power dissipation on output resistor divider and offset
introduced by FB bias current).
If the desired output voltage is equal to internal reference, ROS has to be NC and
FB pin can be directly connected to VOUT.
b)
Set RF in order to obtain the desired closed loop regulator bandwidth according to
the approximated formula:
F 0dB ⋅ F ESR ΔV OSC 1 R FB + R OS
- ⋅ ------------------- ⋅ -------- ⋅ ---------------------------R F = -----------------------------2
R OS
V IN
gm
F LC
If VOUT = VREF, just consider (RFB+ROS)/ROS factor equal to 1.
c)
Place FZ below FLC (typically 0.2·FLC):
5
C F = -------------------------------2π ⋅ R F ⋅ F LC
d)
Place FP at 0.5·FSW:
CF
1
C P = ---------------------------------------------------- ≅ ------------------------------π ⋅ R F ⋅ C F ⋅ F SW – 1 π ⋅ R F ⋅ F SW
e)
Check that compensation network gain is lower than open loop transconductance
EA gain.
f)
Estimate phase margin obtained (it should be greater than 45°) and repeat,
modifying parameters, if necessary.
Doc ID 12754 Rev 4
17/35
Application details
8.3
L6726A
Soft-start time calculation
To calculate SS time (tSS), the following approximated equation can be used (CP<<CF):
t SS
V OUT
C F ⋅ -------------- ⋅ ΔV OSC
V IN
= ------------------------------------------------I SS
The previous equation refers only to VOUT ramp up time. The time elapsed from the end of
OC setting phase or COMP set free to the beginning of VOUT ramp up (see Figure 6) can be
approximately estimated as follow:
C F ⋅ 0.8V
t delay = ----------------------I SS
Once calculated tSS, also the current delivered by the converter during SS to charge the
output capacitor bank can be estimated:
C OUT ⋅ V OUT
I startup = --------------------------------t SS
8.4
Layout guidelines
L6726A provides control functions and high current integrated drivers to implement highcurrent step-down DC-DC converters. In this kind of application, a good layout is very
important.
The first priority when placing components for these applications has to be reserved to the
power section, minimizing the length of each connection and loop as much as possible. To
minimize noise and voltage spikes (EMI and losses) power connections (highlighted in
Figure 10) must be a part of a power plane and anyway realized by wide and thick copper
traces: loop must be anyway minimized. The critical components, i.e. the power MOSFETs,
must be close one to the other. The use of multi-layer printed circuit board is recommended.
Figure 10. Power connections (heavy lines)
VIN
UGATE
CIN
PHASE
L
L6726A
COUT
LGATE
LOAD
GND
The input capacitance (CIN), or at least a portion of the total capacitance needed, has to be
placed close to the power section in order to eliminate the stray inductance generated by the
copper traces. Low ESR and ESL capacitors are preferred, MLCC are suggested to be
connected near the HS drain.
Use proper number of vias when power traces have to move between different planes on the
PCB in order to reduce both parasitic resistance and inductance. Moreover, reproducing the
same high-current trace on more than one PCB layer will reduce the parasitic resistance
associated to that connection.
18/35
Doc ID 12754 Rev 4
L6726A
Application details
Connect output bulk capacitors (COUT) as near as possible to the load, minimizing parasitic
inductance and resistance associated to the copper trace, also adding extra decoupling
capacitors along the way to the load when this results in being far from the bulk capacitors
bank.
Gate traces and phase trace must be sized according to the driver RMS current delivered to
the power MOSFET. The device robustness allows managing applications with the power
section far from the controller without losing performances. Anyway, when possible, it is
recommended to minimize the distance between controller and power section. See
Figure 11 for drivers current paths.
Small signal components and connections to critical nodes of the application, as well as
bypass capacitors for the device supply, are also important. Locate bypass capacitor (VCC
and Bootstrap capacitor) and loop compensation components as close to the device as
practical. For over current programmability, place ROCSET close to the device and avoid
leakage current paths on LGATE / OC pin, since the internal current source is only 10 μA
Systems that do not use Schottky diode in parallel to the Low-Side MOSFET might show big
negative spikes on the PHASE pin. This spike must be limited within the absolute maximum
ratings (for example, adding a gate resistor in series to HS MOSFET gate, or a phase
resistor in series to PHASE pin), as well as the positive spike, but has an additional
consequence: it causes the bootstrap capacitor to be over-charged. This extra-charge can
cause, in the worst case condition of maximum input voltage and during particular
transients, that boot-to-phase voltage overcomes the absolute maximum ratings also
causing device failures. It is then suggested in this case to limit this extra-charge by adding a
small resistor in series to the bootstrap diode (RD in Figure 1).
Figure 11. Drivers turn-on and turn-off paths
LS DRIVER
LS MOSFET
HS DRIVER
VCC
HS MOSFET
BOOT
CGD
RGATE
CGD
RINT
RGATE
LGATE
RINT
UGATE
CGS
CDS
GND
RPHASE
CGS
CDS
PHASE
Doc ID 12754 Rev 4
19/35
Application details
8.5
L6726A
Embedding L6726A-based VRs…
When embedding the VR into the application, additional care must be taken since the whole
VR is a switching DC/DC regulator and the most common system in which it has to work is a
digital system such as MB or similar. In fact, latest MBs have become faster and more
powerful: high speed data busses are more and more common and switching-induced noise
produced by the VR can affect data integrity if additional layout guidelines are not followed.
Few easy points must be considered mainly when routing traces in which switching high
currents flow (switching high currents cause voltage spikes across the stray inductance of
the traces causing noise that can affect the near traces):
When reproducing high current path on internal layers, keep all layers the same size in order
to avoid “surrounding” effects that increase noise coupling.
Keep safe guard distance between high current switching VR traces and data busses,
especially if high-speed data busses, to minimize noise coupling.
Keep safe guard distance or filter properly when routing bias traces for I/O sub-systems that
must walk near the VR.
Possible causes of noise can be located in the PHASE connections, MOSFETs gate drive
and Input voltage path (from input bulk capacitors and HS drain). Also GND connection
must be considered if not insisting on a power ground plane. These connections must be
carefully kept far away from noise-sensitive data busses.
Since the generated noise is mainly due to the switching activity of the VR, noise emissions
depend on how fast the current switches. To reduce noise emission levels, it is also possible,
in addition to the previous guidelines, to reduce the current slope and thus to increase the
switching times: this will cause, as a consequence of the higher switching time, an increase
in switching losses that must be considered in the thermal design of the system.
20/35
Doc ID 12754 Rev 4
L6726A
Application Information
9
Application Information
9.1
Output inductor
Inductor value is defined by a compromise between dynamic response, ripple, efficiency,
cost and size. Usually, inductance is calculated to maintain inductor ripple current (ΔIL)
between 20% and 30% of maximum output current. Given the switching frequency (FSW),
the input voltage (VIN), the output voltage (VOUT) and the desired ripple current (ΔIL),
inductance can be calculated as follows:
V IN – V OUT V OUT
L = ------------------------------ ⋅ -------------F SW ⋅ ΔI L
V IN
Figure 12 shows the ripple current vs. the output voltage for different inductance, with
VIN = 5 V and VIN = 12 V.
Increasing inductance reduces inductor ripple current (and output voltage ripple
accordingly) but, at the same time, increases the converter response time to load transients.
Higher inductance means that the inductor needs more time to change its current from initial
to final value. Until the inductor has not finished its charging, the additional output current is
supplied by output capacitors. Minimizing the response time lead to minimize the output
capacitance required. If the compensation network is designed with high bandwidth, during
an heavy load transient the device is able to saturate duty cycle (0% or 80%). When this
condition is reached, the response time is limited only by the time required to charge the
inductor.
Figure 12. Inductor current ripple vs output voltage
12
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ir 8
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rr 6
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Vin =1 2 V, L =1 u H
Vin =1 2 V, L =2 u H
Vin =5 V, L =5 0 0 n H
Vin =5 V, L =1 .5 u H
0
0
1
2
3
Ou tp u t vo ltag e [V ]
Doc ID 12754 Rev 4
4
5
21/35
Application Information
9.2
L6726A
Output capacitors
Output capacitors choice depends on the application constraints in point of output voltage
ripple and output voltage deviation during a load transient.
During steady-state conditions, the output voltage ripple is influenced by ESR and
capacitance of the output capacitors as follows:
ΔV OUT_ESR = ΔI L ⋅ ESR
1
ΔV OUT_C = ΔI L ⋅ --------------------------------------8 ⋅ C OUT ⋅ F SW
Where ΔIL is the inductor current ripple. These contribution are not in phase, so total ripple
will be lower than the sum of their moduli. Even ESL and board parasitic inductance can
contribute significantly to output ripple.
During a load variation, the output capacitors supply to the load the additional current or
absorb the current in excess delivered by the inductor until converter reaction is completed.
In fact, even if the controller react immediately to the load transient saturating the duty cycle
to 80% or 0%, the current slew rate is limited by the inductance. At first approximation,
output voltage drop, based on ESR and capacitor charge/discharge and considering an
ideal load-step, can be estimated as follows:
ΔV OUT_ESR = ΔI OUT ⋅ ESR
2
L ⋅ ΔI OUT
ΔV OUT_C = ------------------------------------2 ⋅ C OUT ⋅ ΔV L
Where ΔVL is the voltage applied to the inductor during the transient ( D MAX ⋅ VIN – V OUT for
the load appliance or VOUT for the load removal).
MLCC capacitors typically have low ESR to minimize the ripple but also have low
capacitance that do not minimize the capacitive voltage deviation during load transient. On
the contrary, electrolytic capacitors usually have higher capacitance to minimize capacitive
voltage deviation during load transient, but also higher ESR value resulting in higher ripple
voltage and resistive voltage drop. For these reasons, a mix between electrolytic and MLCC
capacitor is usually suggested to minimize ripple as well as reducing voltage deviation in
dynamic conditions.
9.3
Input capacitors
The input capacitor bank is designed mainly to stand input rms current, which depends on
output current (IOUT) and duty-cycle (D) for the regulation as follows:
I rms = I OUT ⋅ D ⋅ ( 1 – D )
The equation reaches its maximum value, IOUT/2, when D = 0.5. Losses depend on input
capacitor ESR:
P = ESR ⋅ Irms
22/35
2
Doc ID 12754 Rev 4
L6726A
10
20 A demonstration board
20 A demonstration board
L6726A demonstration board realizes on a four-layer PCB a step-down DC/DC converter
and shows the operation of the device in a general purpose application. Input voltage can
range from 5 V to 12 V bus. Output voltage is programmed to 1.25 V. The voltage regulator
can deliver up to 20 A output current. The switching frequency is 270 kHz.
Figure 13. 20 A demonstration board (left) and components placement (right)
Figure 14. 20 A demonstration board top (left) and bottom (right) layers
Doc ID 12754 Rev 4
23/35
20 A demonstration board
L6726A
Figure 15. 20 A demonstration board inner layers
24/35
Doc ID 12754 Rev 4
Doc ID 12754 Rev 4
R6
NC
R18
20k
LGATE
GND
UGATE
BOOT
0
GNDCC
VCC
0
4
3
2
1
C35
2k
470pF
68nF
0
R12
0
NC
R8
VCC
FB
COMP
PHASE
L6726A/27
GND
VCC
C24
LGATE
GND
UGATE
BOOT
U1
R7
COMP
GND
VCC
R16
0
5
6
7
8
FB
3.3
FB
0
R13
3.9k
0
UGATE
3.3
R2
C2
1800uF
NC
R14
2.2k
R9
NC
C36
LGATE
C3
NC
R15
R5
R3
C10
BOOT
100nF
0
LGATE
PHASE
UGATE
0
R17
PHASE
COMP
0
C38
D1
1N4148
VCC_PIN
VCC
3.3
1uF
R1
C14
1uF
0
C1
1800uF
0
1
HSG 1
LSG1
NC
0
0
NC
C4
NC
0
C5
PHASE
GNDIN_POWER
GNDIN1
C37
R10
NC
0
NC
0
R11
LSG2
Q5
STD95NH02LT4
1
Q4
STD70NH02LT4
0
Q6
NC
0
1
L2
2
T60-18 6Ts
C23
6.8nF
R4
1.8
0
ᤡ
C15
0
2200uF
C25
0
NC
C13
4.7uF
HSG 4
Q1
NC
0
C12
C11
0
4.7uF
4.7uF
0
0
NC
C9
NC
C8
VIN_POWER
0
0
NC
C7
NC
C6
2
VIN1
C16
C27
0
0
NC
C26
NC
0
0
C18
0
NC
C29
OUT
0
2200uF
C28
NC
C17
NC
0
NC
OUT
LSG1 4
Q2
NC
ᤢ
C19
0
NC
C30
0
NC
0
NC
C31
0
NC
C20
PHASE 1
Q3
NC
2
0
NC
C32
0
NC
C21
0
NC
C33
0
NC
C22
LSG2 4
NC
L1
OUT
5
6
7
8
VIN_POWER
2
3
2
5
6
7
8
1
2
3
PHASE
VCC_PIN
PHASE
COMP
HSD
3
GND
3
5
6
7
8
1
2
3
HSD
1
2
3
U
R
W
F
X
G
Q
,
G
Q\
D
W
L
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W
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I
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R
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RX
I
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U
L
R
G
GI
$
C34
0
NC
0
GNDOUT1
GNDOUT
VOUT1
VOUT
GND
OUT
PHASE
FB
L6726A
20 A demonstration board
Figure 16. 20 A demonstration board schematic
V
Z
R
O
O
D
&
5
5
Q
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L
W
F
Q
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3
2
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25/35
20 A demonstration board
Table 6.
L6726A
20 A demonstration board - bill of material
Qty
Reference
Description
Package
Capacitors
2
C1, C2
Electrolytic Cap 1800 μF 16 V
Nippon Chemi-Con KZJ or KZG
1
C10
MLCC, 100 nF, 25 V, X7R
SMD0603
3
C11 to C13
MLCC, 4.7 μF, 16 V, X5R
Murata GRM31CR61C475MA01
SMD1206
2
C14, C38
MLCC, 1 μF, 16 V, X7R
SMD0805
2
C18, C25
Electrolytic Cap 2200 μF 6.3 V
Nippon Chemi-Con KZJ or KZG
1
C23
MLCC, 6.8 nF, X7R
1
C24
MLCC, 68 nF, X7R
1
C35
MLCC, 470 pF, X7R
Radial 10 x 25 mm
Radial 10 x 20mm
SMD0603
Resistors
3
R1, R2, R17
Resistor, 3R3, 1/16 W, 1%
3
R3, R5, R16
Resistor, 0R, 1/8 W, 1%
1
R4
Resistor, 1R8, 1/8 W, 1%
2
R11, R12
Resistor, 0R, 1/16 W, 1%
1
R9
Resistor, 2K2, 1/16 W, 1%
1
R13
Resistor, 3K9, 1/16 W, 1%
1
R7
Resistor, 2K, 1/16 W, 1%
1
R18
Resistor, 20K, 1/16 W, 1%
L1
Inductor, 1.25 μH, T60-18, 6Turns
Easymagnet AP106019006P-1R1M
SMD0603
SMD0805
SMD0603
Inductor
1
na
Active components
1
D1
Diode, 1N4148 or BAT54
1
Q5
STD70NH02LT4
1
Q6
STD95NH02LT4
1
U1
Controller, L6726A
SOT23
DPACK
26/35
Doc ID 12754 Rev 4
SO8
L6726A
20 A demonstration board
10.1
Board description
10.1.1
Power input (VIN)
This is the input voltage for the power conversion. The high-side MOSFET drain is
connected to this input. Supply must be compliant with VIN recommended operating
conditions and capacitors rating.
If VIN voltage is compliant also to VCC range listed in recommended operating conditions, it
can supply also the device through R16 resistor.
10.1.2
Power output (VOUT)
This is the output voltage of the power conversion. The output voltage is programmed to
1.25 V. It can be changed by replacing R13 (adjusting of compensation network may be
needed). R18 allows to adjust OCP threshold.
10.1.3
IC additional supply (VCC)
The controller can be supplied separately from the power conversion through VCC input. In
this case, to separate VCC from VIN, R16 resistor must be removed.
10.1.4
Test points
The following test points are provided to allow easy probing of important signals:
COMP: Output of the error amplifier;
–
FB: Inverting input of the error amplifier;
–
PH: Phase pin of the device;
–
LG: Low-side gate pin of the device;
–
UG: High-side gate pin of the device.
Demonstration board efficiency
Figure 17. 20 A demonstration board efficiency
Efficiency [%]
10.1.5
–
100
95
90
85
80
75
70
65
60
55
50
Vin=12V, Vout=1.25V
Vin=5V, Vout=1.25V
Vin=12V, Vout=2.5V
Vin=5V, Vout=2.5V
0
2
4
6
8
10
12
14
16
18
20
22
Output Current [A]
Doc ID 12754 Rev 4
27/35
5 A demonstration board
11
L6726A
5 A demonstration board
L6726A demonstration board realizes on a two-layer PCB a step-down DC/DC converter
and shows the operation of the device in a general-purpose low-current application. Input
voltage can range from 5 V to 12 V bus. Output voltage is programmed at 1.25 V. The
application can deliver an output current in excess of 5 A. The switching frequency is 270
kHz.
Figure 18. 5 A demonstration board (left) and components placement (right)
Figure 19. 5 A demonstration board top (left) and bottom (right) layers
28/35
Doc ID 12754 Rev 4
R18
NC
LGATE
0
ᤡ
4
3
2
UGATE
GNDIN1
ᤢ
5
6
7
8
Doc ID 12754 Rev 4
0
R8
R10
0
C35
2.2nF
0
R13
3.9K
R14
NC
2.2k
C36
NC
LGATE
R17
3.3
0
R5
0
PHASE
R3
C10
100nF
BOOT
LSG1
HSG1
2
4
1
0
7 8
3
VIN_POWER
U5A
STS9D8NH3LL
U5B
STS9D8NH3LL
HSD
5 6
0
L3
NC
L2
2.2uH
C23
6.8nF
R4
1.8
2
2
1
1
0
ᤡ
C12
10uF
ᤡ
ᤡ
C29
NC
0
C39
22uF
OUT
0
ᤡ
0
ᤢ
C18
NC
OUT
0
OUT
C40
NC
V
U
R
W
L
F
D
S
D
F
F
L
P
D
U
H
F
NC
LGATE
R9
UGATE
PHASE PIN
PHASE
UGATE
3.3
C51
10uF
ᤢ
0
ᤢ
0
C29A
NC
ᤢ
C30
330uF
OUT
W
Q
L
U
S
W
R
R
I
O
D
X
G
820
R7
FB
U
H
O
O
N
UR
U
RW
Q
ZR
WF
H
1
H
QK
W
R
LU
D
W
D
H
VQ
Q
HH
F
SD
PO
RS
&
COMP
R1
3.3
VCC
0
R2
V
U
R
W
L
F
D
S
D
F
P
X
O
D
W
Q
D
7
C24
220nF
VCC_PIN
FB
COMP
PHASE PIN
C38
1uF
D1
BAT54
U
R
W
L
F
D
S
D
F
F
L
W
\
O
R
U
W
F
H
O
(
L6726A
VCC
FB
COMP
PHASE
U1
GND
VCC
R16
0
W
Q
L
U
S
W
R
R
I
O
D
X
G
C14
1uF
LGATE/OC
GND
UGATE
BOOT
VCC
VIN1
GND
1
0
GNDCC
VCC
0
GNDIN_POWER
VIN_POWER
BOOT
GND
V
U
R
W
L
F
D
S
D
F
F
L
P
D
U
H
F
L6726A
W
Q
L
U
S
W
R
R
I
O
D
X
G
COMP
V
R
P
6
+
U
D
H
Q
H
F
D
O
S
FB
VOUT
0
GNDOUT1
GNDOUT
VOUT1
L6726A
5 A demonstration board
Figure 20. 5 A demonstration board schematic
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OUT
5 A demonstration board
Table 7.
L6726A
5 A demonstration board - bill of material
Qty
Reference
Description
Package
Capacitors
2
C12, C51
10 μF, 25 V, X5R
Murata GRM31CR61E106KA12
SMD1206
1
C10
MLCC, 100nF, 25V, X7R
SMD0603
2
C14, C38
MLCC, 1 μF, 16 V, X7R
SMD0805
1
C39
MLCC, 22 μF, 6.3 V, X5R
Murata GRM31CR60J226ME19
SMD1206
1
C30
330 μF, 6.3 V, 40 mΩ
Sanyo 6TPB330M
SMD7343
1
C23
MLCC, 6.8 nF, X7R
1
C24
MLCC, 220 nF, X7R
1
C35
MLCC, 2.2 nF, X7R
1
R4
Resistor, 1R8, 1/8 W, 1%
4
R3, R5, R10, R16
Resistor, 0R, 1/16 W, 1%
3
R1, R2, R17
Resistor, 3R3, 1/16 W, 1%
1
R9
Resistor, 2K2, 1/16 W, 1%
1
R13
Resistor, 3K9, 1/16 W, 1%
1
R7
Resistor, 820R, 1/16 W, 1%
L1
Inductor, 2.20 μH,
Wurth 744324220LF
SMD0603
Resistors
SMD0805
SMD0603
SMD0603
Inductor
1
na
Active Components
1
D1
Diode, BAT54
1
Q5
Mosfet, STS9D8NH3LL
SO8
1
U1
Controller, L6726A
SO8
11.1
Board description
11.1.1
Power input (VIN)
SOT23
This is the input voltage for the power conversion. The high-side MOSFET drain is
connected to this input. Supply must be compliant with VIN recommended operating
conditions and capacitors rating.
If VIN voltage is compliant also to VCC range listed in recommended operating conditions, it
can supply also the device through R16 resistor.
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Doc ID 12754 Rev 4
L6726A
11.1.2
5 A demonstration board
Power output (VOUT)
This is the output voltage of the power conversion. The output voltage is programmed to
1.25 V. It can be changed by replacing R13 (adjusting of compensation network may be
needed). Adding R18 allows to adjust OCP threshold.
11.1.3
IC additional supply (VCC)
The controller can be supplied separately from the power conversion through VCC input. In
this case, to separate VCC from VIN, R16 resistor must be removed.
11.1.4
Test points
The following test points are provided to allow easy probing of important signals:
COMP: output of the error amplifier;
–
FB: inverting input of the error amplifier;
–
PH: Phase pin of the device;
–
LG: Low-side gate pin of the device;
–
UG: High-side gate pin of the device.
Demonstration board efficiency
Figure 21. 5 A demonstration board efficiency
Efficiency [%]
11.1.5
–
100
95
90
85
80
75
70
65
60
55
50
Vin = 12V, Vout = 1.25V
Vin = 5V, Vout = 1.25V
Vin = 12V, Vout = 2.5V
Vin = 5V, Vout = 2.5V
0
1
2
3
4
5
6
Output Current [A]
Doc ID 12754 Rev 4
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Package mechanical data
12
L6726A
Package mechanical data
In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK® packages, depending on their level of environmental compliance. ECOPACK®
specifications, grade definitions and product status are available at: www.st.com.
ECOPACK is an ST trademark.
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Doc ID 12754 Rev 4
L6726A
Package mechanical data
Table 8.
SO-8 mechanical data
mm.
inch
Dim.
Min
Typ
Max
Min
Typ
Max
A
1.35
1.75
0.053
0.069
A1
0.10
0.25
0.004
0.010
A2
1.10
1.65
0.043
0.065
B
0.33
0.51
0.013
0.020
C
0.19
0.25
0.007
0.010
D (1)
4.80
5.00
0.189
0.197
E
3.80
4.00
0.15
0.157
e
1.27
0.050
H
5.80
6.20
0.228
0.244
h
0.25
0.50
0.010
0.020
L
0.40
1.27
0.016
0.050
k
0° (min.), 8° (max.)
ddd
0.10
0.004
1. D and F does not include mold flash or protrusions. Mold flash or potrusions shall not exceed 0.15mm
(.006inch) per side.
Figure 22. Package dimensions
Doc ID 12754 Rev 4
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Revision history
13
L6726A
Revision history
Table 9.
34/35
Document revision history
Date
Revision
Changes
16-Oct-2006
1
Initial release.
26-Oct-2006
2
Mechanical data dimensions updated
30-Jul-2007
3
Updated Figure 1 on page 4, tables 2, 3, 4, 5
23-Mar-2010
4
Added:
Section 9 on page 21, Section 10 on page 23, Section 11 on page 28
Updated: Table 5 on page 7
Doc ID 12754 Rev 4
L6726A
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Doc ID 12754 Rev 4
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