LTC3708 Fast 2-Phase, No RSENSE Buck Controller with Output Tracking DESCRIPTION FEATURES n n n n n n n n n n n n n n n The LTC®3708 is a dual, 2-phase synchronous step-down switching regulator with output voltage up/down tracking capability. The IC allows either coincident or ratiometric tracking. Multiple LTC3708s can be daisy-chained in applications requiring more than two voltages to be tracked. Power supply sequencing is accomplished using an external soft-start timing capacitor. Very Low Duty Factor Operation (tON(MIN) < 85ns) No RSENSE™ Option for Maximum Efficiency Very Fast Transient Response Programmable Output Voltage Up/Down Tracking 2-Phase Operation Reduces Input Capacitance 0.6V ±1% Output Voltage Reference External Frequency Synchronization Monotonic Soft-Start Onboard High Current MOSFET Drivers Wide VIN Range: Up to 36V Adjustable Cycle-by-Cycle Current Limit Instant Output Overvoltage Protection Optional Short-Circuit Shutdown Timer Power Good Output with 100μs Masking Available in 5mm × 5mm QFN Package The LTC3708 uses a constant on-time, valley current mode control architecture to deliver very low duty factors without requiring a sense resistor. Operating frequency is selected by an external resistor and is compensated for variations in input supply voltage. An internal phase-locked loop allows the IC to be synchronized to an external clock. Fault protection is provided by an output overvoltage comparator and an optional short-circuit shutdown timer. The regulator current limit level is user programmable. A wide supply range allows voltages as high as 36V to be stepped down to 0.6V output. APPLICATIONS n n Digital Signal Processors Network Servers L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATION High Efficiency Dual Output Step-Down Converter 5V 10Ω 1μF 4.7μF VOUT1 (0.5V/DIV) VIN 3.3V TO 28V 10μF 50V s4 100k VOUT2 (0.5V/DIV) + VOUT1 2.5V 15A POSCAP + 330μF 4V s2 M3 BG1 SENSE1– PGND1 19.1k 1.5M 6.04k 33k 10k LTC3708 180pF L2 1.2μH + B340A M4 SENSE2– PGND2 VFB1 VFB2 TRACK2 FCB ION1 ION2 ITH1 ITH2 INTLPF EXTLPF RUN/SS TRACK1 VRNG2 SGND VRNG1 0.1μF 0.22μF BG2 0.01μF 0.01μF M2 SW2 SENSE2+ SW1 SENSE1+ B340A VIN 6.04k 0.22μF DRVCC PGOOD VCC TG1 TG2 BOOST2 BOOST1 25k 3708 TA01b 100 9.0 95 7.5 90 6.0 12.1k fIN 1M VIN 1k 100k 5V 6.04k 33k 0.1μF 80 20VIN TO 2.5VOUT 5VIN TO 2.5VOUT 20VIN TO 1.8VOUT 5VIN TO 1.8VOUT 3.0 1.5 75 70 0.01 180pF M1, M2: RENESAS HAT2168 M3, M4: RENESAS HAT2165 4.5 85 0.01μF 3708 TA01 L1: PANASONIC ETQP3HIR4BF L2: PANASONIC ETQP2HIR2BF 2ms/DIV VOUT2 1.8V POSCAP 15A 470μF 2.5V s2 POWER LOSS (W) 12.1k M1 EFFICIENCY (%) L1 1.4μH 0 1 0.1 LOAD CURRENT (A) 10 3708 TA01c 3708fb 1 LTC3708 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) Input Supply Voltage (VCC, DRVCC) ............. 7V to –0.3V Boosted Topside Driver Supply Voltage BOOST1, 2 ............................................ 42V to –0.3V Switch Voltage (SW1, 2) .............................. 36V to –5V SENSE1+, SENSE2+ Voltages ....................... 36V to –5V SENSE1–, SENSE2– Voltages .................... 10V to –0.3V ION1, ION2 Voltages .................................... 21V to –0.3V (BOOST – SW) Voltages .............................. 7V to –0.3V RUN/SS, PGOOD Voltages .......................... 7V to –0.3V PGOOD DC Current ................................................. 5mA TRACK1, TRACK2 Voltages ..............VCC + 0.3V to –0.3V VRNG1, VRNG2 Voltages .................... VCC + 0.3V to –0.3V ITH1, ITH2 Voltages.................................... 2.7V to –0.3V VFB1, VFB2 Voltages .................................. 2.7V to –0.3V INTLPF, EXTLPF Voltages ......................... 2.7V to –0.3V FCB Voltages ............................................... 7V to –0.3V Operating Temperature Range (Note 5).... –40°C to 85°C Junction Temperature (Note 2) ........................... 125°C Storage Temperature Range................... –65°C to 125°C Reflow Peak Body Temperature ........................... 260°C SENSE1+ SW1 TG1 BOOST1 ION1 PGOOD FCB VRNG1 TOP VIEW 32 31 30 29 28 27 26 25 24 SENSE1– RUN/SS 1 ITH1 2 23 PGND1 VFB1 3 22 BG1 TRACK1 4 21 DRVCC 33 SGND 5 20 BG2 19 PGND2 TRACK2 6 18 SENSE2– VFB2 7 ITH2 8 17 VCC SENSE2+ SW2 TG2 BOOST2 ION2 VRNG2 INTLPF EXTLPF 9 10 11 12 13 14 15 16 UH PACKAGE 32-LEAD (5mm s 5mm) PLASTIC QFN TJMAX = 125°C, θJA = 34°C/W EXPOSED PAD (PIN 33) IS SGND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3708EUH#PBF LTC3708EUH#TRPBF 3708 32-Lead (5mm × 5mm) Plastic QFN –40°C to 85°C LEAD BASED FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3708EUH LTC3708EUH#TR 3708 32-Lead (5mm × 5mm) Plastic QFN –40°C to 85°C Consult LTC Marketing for parts specified with wider operating temperature ranges. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 5V, DRVCC = 5V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 2.4 250 3 400 mA μA –50 –100 nA 0.594 0.591 0.600 0.600 0.606 0.609 V V 0.594 0.600 0.606 Main Control Loop IQ Input DC Supply Current Normal Shutdown IFB1,2 Feedback Pin Input Current ITH = 1.2V (Notes 3, 4) VREF Internal Reference Voltage ITH = 1.2V, 0°C to 85°C (Notes 3, 4) ITH = 1.2V (Notes 3, 4) VFB1,2 Feedback Voltage ITH = 1.2V (Note 3) ΔVFB(LINEREG)1,2 Feedback Voltage Line Regulation VCC = 4.5V to 6.5V (Note 3) l 0.02 V %/V 3708fb 2 LTC3708 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 5V, DRVCC = 5V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS –0.05 –0.2 % 1.2 1.45 1.7 mS 94 186 116 233 138 280 ns ns 50 85 ns 270 350 ns 160 110 220 mV mV mV ΔVFB(LOADREG)1,2 Feedback Voltage Load Regulation ITH = 0.5V to 1.9V (Note 3) gm(EA)1,2 Error Amplifier Transconductance ITH = 1.2V (Note 3) tON1,2 On-Time ION = 60μA, VFCB = 0V ION = 30μA, VFCB = 0V tON(MIN)1,2 Minimum On-Time ION = 180μA tOFF(MIN)1,2 Minimum Off-Time ION = 30μA VSENSE(MAX)1,2 Maximum Current Sense Threshold VRNG = 1V, VFB = 0.565V VRNG = 0V, VFB = 0.565V VRNG = VCC, VFB = 0.565V 143 100 200 VSENSE(MIN)1,2 Minimum Current Sense Threshold VRNG = 1V, VFB = 0.635V VRNG = 0V, VFB = 0.635V VRNG = VCC, VFB = 0.635V ΔVFB(OV)1,2 Overvoltage Fault Threshold 8.5 10 11.5 % ΔVFB(UV)1,2 Undervoltage Fault Threshold –380 –420 –460 mV VRUN/SS(ON) RUN Pin Start Threshold 0.8 1.3 1.8 V VRUN/SS(LE) RUN Pin Latchoff Enable Threshold RUN/SS Pin Rising 2.6 3 3.3 V VRUN/SS(LT) RUN Pin Latchoff Threshold RUN/SS Pin Falling 2.2 2.5 2.8 V l 125 90 180 –62 –42 –88 l mV mV mV IRUN/SS(C) Soft-Start Charge Current VRUN/SS = 0V –0.5 –1.2 –2 μA IRUN/SS(D) Soft-Start Discharge Current VRUN/SS = VRUN/SS(LE), VFB1,2 = 0V 0.8 2 3 μA VCC(UVLO) Undervoltage Lockout VCC Falling 3.2 3.6 V VCC(UVLOR) Undervoltage Lockout Release VCC Rising 3.5 3.8 V TG RUP1,2 TG Driver Pull-Up On-Resistance TG High (Note 6) TG RDOWN1,2 TG Driver Pull-Down On-Resistance TG Low (Note 6) 2 Ω BG RUP1,2 BG Driver Pull-Up On-Resistance BG High (Note 6) 3 Ω BG RDOWN1,2 BG Driver Pull-Down On-Resistance BG Low (Note 6) 1 Ω ITRACK1,2 TRACK Pin Input Current ITH = 1.2V, VTRACK = 0.2V (Note 3) VFB(TRACK1,2) Feedback Voltage at Tracking VTRACK = 0V, ITH = 1.2V (Note 3) VTRACK = 0.2V, ITH = 1.2V (Note 3) VTRACK = 0.4V, ITH = 1.2V (Note 3) ΔVFBH1,2 PGOOD Upper Threshold ΔVFBL1,2 2 Ω Tracking –100 –150 nA –10 190 390 0 200 400 –10 210 410 mV mV mV Either VFB Rising 8.5 10 11.5 % PGOOD Lower Threshold Either VFB Falling –8.5 –10 –11.5 % ΔVFB(HYS)1,2 PGOOD Hysteresis VFB Returning 3 5 % VPGL PGOOD Low Voltage IPGOOD = 5mA 0.1 0.4 V ±1 μA PGOOD Output IPGOOD PGOOD Leakage Current VPGOOD = 7V PG Delay PGOOD Delay VFB Falling 100 1.9 2.1 2.3 V 1 1.5 2 V μs Phase-Locked Loops VFCB(DC) Forced Continuous Threshold Measured with a DC Voltage at FCB Pin VFCB(AC) Clock Input Threshold Measured with a AC Pulse at FCB Pin IEXTLPF External Phase Detector Output Current Sourcing Capability Sinking Capability fFCB < fSW1, VEXTLPF = 0V fFCB > fSW1, VEXTLPF = 2.4 20 –20 μA μA 3708fb 3 LTC3708 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 5V, DRVCC = 5V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS IINTLPF Internal Phase Detector Output Current Sourcing Capability Sinking Capability fSW1 < fSW2, VINTLPF = 0V fSW1 > fSW2, VINTLPF = 2.4 tON1 Modulation Range by External PLL Up Modulation Down Modulation ION1 = 60μA, VEXTLPF = 1.8V ION1 = 60μA, VEXTLPF = 0.6V 186 tON2 Modulation Range by Internal PLL Up Modulation Down Modulation ION1 = 60μA, VEXTLPF = 1.8V ION1 = 60μA, VEXTLPF = 0.6V 186 tON(PLL)1 tON(PLL)2 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliabilty and lifetime. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD as follows: TJ = TA + (PD • 34°C/W) Note 3: The LTC3708 is tested in a feedback loop that adjusts VFB to achieve a specified error amplifier output voltage (ITH). MIN TYP MAX UNITS 20 –20 μA μA 233 58 80 ns ns 233 58 80 ns ns Note 4: Internal reference voltage is tested indirectly by extracting error amplifier offset from the feedback voltage. Note 5: The LTC3708E is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 6: RDS(ON) limit is guaranteed by design and/or correlation to static test. TYPICAL PERFORMANCE CHARACTERISTICS Load Transient on Channel 1 Load Transient on Channel 2 IOUT1 10A/DIV IOUT2 10A/DIV VOUT1 100mV/DIV VOUT1 100mV/DIV VOUT2 100mV/DIV VOUT2 100mV/DIV 20μs/DIV 3708 G01 20μs/DIV Coincident Tracking 3708 G02 Ratiometric Tracking VOUT1 0.5V/DIV VOUT1 0.5V/DIV VOUT2 0.5V/DIV VOUT2 0.5V/DIV 2ms/DIV 3708 G03 2ms/DIV 3708 G04 3708fb 4 LTC3708 TYPICAL PERFORMANCE CHARACTERISTICS Soft-Start Power Loss vs Input Voltage 6 VOUT1 2V/DIV VOUT2 2V/DIV POWER LOSS (W) 5 RUN/SS 5V/DIV IL1 5A/DIV VOUT = 2.5V 4 3 VOUT = 1.8V 2 1 3708 G05 50ms/DIV IOUT = 15A 0 5 10 15 20 INPUT VOLTAGE (V) 25 3708 G06 Power Loss vs Load Current VIN = 5V 3.0 VOUT = 2.5V Frequency vs Load Current 250 IOUT = 15A 240 2.5 FREQUENCY (kHz) POWER LOSS (W) Frequency vs Input Voltage 260 VOUT = 1.8V 2.0 1.5 1.0 220 200 FREQUENCY (kHz) 3.5 EXTERNAL SYNCHRONIZATION (ANY IOUT) IOUT = 0A 200 180 0.5 0 10 100 1000 10000 160 100000 5 10 15 20 0 25 10 5 LOAD CURRENT (A) 0 3708 G09 Current Sense Threshold vs ITH Voltage On-Time vs Temperature 10000 300 300 CURRENT SENSE THRESHOLD (mV) ION = 30μA 250 ON-TIME (ns) 1000 200 150 ION = 60μA 100 100 50 10 1000 3708 G10 15 3708 G08 On-Time vs ION Current ON-TIME (ns) FORCED CONTINUOUS MODE EXTERNAL SYNCHRONIZATION DISCONTINUOUS MODE INPUT VOLTAGE (V) 3707 G07 10 100 ION CURRENT (μA) 100 50 LOAD (mA) 1 150 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3708 G11 VRNG = 250 2V 1.4V 200 1V 150 0.7V 100 0.5V 50 0 –50 –100 –150 –200 0 0.5 1.5 1 ITH VOLTAGE (V) 2 2.5 3708 G12 3708fb 5 LTC3708 TYPICAL PERFORMANCE CHARACTERISTICS Maximum Current Sense Threshold vs Temperature 300 250 200 150 100 50 0 0.5 1.5 1.25 1 VRNG VOLTAGE (V) 0.75 1.75 2 160 Load Regulation (Figure 13 Circuit) 0 VRNG = 1V –0.1 FORCED CONTINUOUS MODE 150 –0.2 $VOUT (%) 350 MAXIMUM CURRENT SENSE THRESHOLD (mV) MAXIMUM CURRENT SENSE THRESHOLD (mV) Maximum Current Sense Threshold vs VRNG Voltage 140 130 –0.3 –0.4 DISCONTINUOUS MODE –0.5 120 –0.6 110 –50 –25 –0.7 0 3708 G13 3 0 25 50 75 100 125 150 TEMPERATURE (°C) 6 9 3708 G15 3708 G14 Error Amplifier gm vs Temperature SENSE Pin Input Current vs Temperature RUN/SS Pin Current vs Temperature 150 1.6 15 12 LOAD CURRENT (A) 3 140 1.5 120 ISENSE (μA) 1.4 gm (mS) RUN/SS PIN CURRENT (μA) 130 1.3 1.2 ISENSE– 110 100 ISENSE+ 90 80 70 1.1 2 PULL-DOWN CURRENT 1 0 PULL-UP CURRENT –1 60 1.0 –50 –25 0 50 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 3708 G16 700 Undervoltage Lockout Threshold vs Temperature UNDERVOLTAGE LOCKOUT THRESHOLD (V) 4.0 RUN/SS THRESHOLD (V) 600 VFB (mV) 400 300 200 3.5 LATCHOFF ENABLE 3.0 2.5 LATCHOFF THRESHOLD 100 1 1.25 2 1.75 1.5 RUN/SS VOLTAGE (V) 2.25 2.5 3708 G19 2.0 –50 –25 0 125 100 3708 G18 RUN/SS Latch-Off Thresholds vs Temperature 500 50 25 0 75 TEMPERATURE (°C) –25 3708 G17 Feedback Voltage vs RUN/SS (Soft-Start) 0 –2 –50 25 50 75 100 125 150 TEMPERATURE (°C) 25 50 75 100 125 150 TEMPERATURE (°C) 3708 G20 4.5 4.0 3.5 3.0 2.5 2.0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3708 G21 3708fb 6 LTC3708 TYPICAL PERFORMANCE CHARACTERISTICS On-Time vs EXTLPF Voltage 500 450 450 400 400 ION1 = 30μA 350 300 250 200 ION1 = 60μA 150 VSW1 10V/DIV 250 200 100 50 0 1.2 1.1 1.3 EXTLPF VOLTAGE (V) 1.4 VSW2 10V/DIV ION2 = 60μA 150 50 1 IIN 2A/DIV VIN 200mV/DIV 300 100 0 2-Phase Operation ION2 = 30μA 350 tON2 (ns) tON1 (ns) On-Time vs INTLPF Voltage 500 0.6 0.8 1.2 1.4 1 1.6 INTLPF VOLTAGE (V) 1.8 2.0 VIN = 15V 1μs/DIV VOUT1 = 5V VOUT2 = 3.3V IOUT5 = IOUT3 = 2A 3708 G024 3708 G23 3708 G22 Load Transient Response Without External Synchronization IOUT1 10A/DIV Load Transient Response with External Synchronization IOUT1 10A/DIV fS = 200kHz fS = 240kHz fS = 220kHz SW1 10V/DIV SW1 10V/DIV VOUT1 50mV/DIV VOUT1 50mV/DIV 10μs/DIV 3708 G25 10μs/DIV Discontinuous Mode Operation VOUT 20mV/DIV fS = 220kHz 3708 G26 Power Good Mask VFB 0.2V/DIV PGOOD 2V/DIV IL 0.5A/DIV VIN = 15V VOUT = 5V VFCB = 5V IOUT = 20mA 2μs/DIV 3708 G027 100μs/DIV 3708 G28 3708fb 7 LTC3708 PIN FUNCTIONS RUN/SS (Pin 1): Run Control and Soft-Start Input. A capacitor to ground at this pin sets the ramp rate of the output voltage (approximately 0.5s/μF) and the time delay for overcurrent latchoff (see Applications Information). Forcing this pin below 0.8V shuts down the LTC3708. ITH1, ITH2 (Pins 2, 8): Error Amplifier Compensation Point and Current Control Threshold. The current comparator threshold increases with this control voltage. The voltage ranges from 0V to 2.4V with 0.8V corresponding to zero sense voltage (zero current). VFB1, VFB2 (Pins 3, 7): Error Amplifier Feedback Input. This pin connects the error amplifier input to an external resistive divider from VOUT . Additional compensation can be implemented, if desired, using this pin. TRACK1, TRACK2 (Pins 4, 6): Tie TRACK2 pin to a resistive divider connected to the output of channel 1 for either coincident or ratiometric output tracking. TRACK1 is used in the same manner between multiple LTC3708s (see Applications Information). To disable this feature, tie the pins to VCC. Do Not Float These Pins. SGND (Pins 5, 33): Signal Ground. All small-signal components and compensation components should connect to this ground and eventually connect to PGND at one point. The Exposed Pad of the LTC3708EUH must be soldered to the PCB. EXTLPF (Pin 9): Filter Connection for the External PLL. This PLL is used to synchronize the LTC3708 to an external clock. If external clock is not used, leave this pin floating. INTLPF (Pin 10): Filter Connection for the Internal PLL. This PLL is used to phase shift the second channel to the first channel by 180°. VCC (Pin 17): Main Input Supply. Decouple this pin to SGND with an RC filter (10Ω, 1μF for example). DRVCC (Pin 21): Driver Supply. Provides supply to the drivers for the bottom gates. Also used for charging the bootstrap capacitors. BG1, BG2 (Pins 22, 20): Bottom Gate Drive. Drives the gate of the bottom N-channel MOSFET between ground and DRVCC. PGND1, PGND2 (Pins 23, 19): Power Ground. Connect this pin closely to the source of the bottom N-channel MOSFET, the (–) terminal of CDRVCC and the (–) terminal of CIN. SENSE1–, SENSE2– (Pins 24, 18): Current Sense Comparator Input. The (–) input to the current comparator is used to accurately Kelvin sense the bottom side of the sense resistor or MOSFET. SENSE1+, SENSE2+ (Pins 25, 16): Current Sense Comparator Input. The (+) input to the current comparator is normally connected to the SW node unless using a sense resistor (See Applications Information). SW1, SW2 (Pins 26, 15): Switch Node. The (–) terminal of the bootstrap capacitor CB connects here. This pin swings from a Schottky diode voltage drop below ground up to VIN. TG1, TG2 (Pins 27, 14): Top Gate Drive. Drives the top N-channel MOSFET with a voltage swing equal to DRVCC superimposed on the switch node voltage SW. BOOST1, BOOST2 (Pins 28, 13): Boosted Floating Driver Supply. The (+) terminal of the bootstrap capacitor CB connects here. This pin swings from a diode voltage drop below DRVCC up to VIN + DRVCC. ION1, ION2 (Pins 29, 12): On-Time Current Input. Tie a resistor from VIN to this pin to set the one-shot timer current and thereby set the switching frequency. PGOOD (Pin 30): Power Good Output. Open-drain logic output that is pulled to ground when either or both output voltages are not within ±10% of the regulation point. The output voltage must be out of regulation for at least 100μs before the power good output is pulled to ground. FCB (Pin 31): Forced Continuous and External Clock Input. Tie this pin to ground to force continuous synchronous operation or to VCC to enable discontinuous mode operation at light load. Feeding an external clock signal into this pin will synchronize the LTC3708 to the external clock and enable forced continuous mode. VRNG1, VRNG2 (Pins 32, 11): Sense Voltage Range Input. The voltage at this pin is ten times the nominal sense voltage at maximum output current and can be programmed from 0.5V to 2V. The sense voltage defaults to 70mV when this pin is tied to ground, 140mV when tied to VCC. 3708fb 8 LTC3708 FUNCTIONAL DIAGRAM RIPLL CIPLL INTLPF FCB ION CLOCK DETECTOR RON FROM CHANNEL 2 TG ENABLE PHASE DETECTOR (PD2) EXTLPF 0.6V REF PHASE DETECTOR (PD1) VIN + VCC CIN CVCC REPLL CEPLL BOOST TO CHANNEL 2 OST OST tON = 0.7 (10pF) IION TG FCNT R M1 ON Q SW S 20k + SENSE+ SWITCH LOGIC + ICMP L1 VOUT + DB COUT DRVCC IREV – CB – SHDN BG OV CDRVCC M2 1.4V PGND VRNG SENSE– s 0.7V – OV 3.3μA – EA VFB R2 + 1 240k 0.66V SGND Q4 R1 + – ITH UV CC 0.54V + RC PGOOD Q1 Q2 VREF 0.6V Q3 + – TRACK ENABLE >100μs BLANKING 1.3V RUN SHDN FROM CHANNEL 2 OV AND UV COMPARATORS 1.2μA DUPLICATE FOR SECOND CHANNEL CONTROLLER 6V –+ 1.3V RUN/SS CSS NOTE: THE RUN/SS PIN ONLY CLAMPS VREF FOR PHASE 1 NOT PHASE 2. 3708 FD 3708fb 9 LTC3708 OPERATION (Refer to Functional Diagram) Main Control Loop The LTC3708 uses a constant on-time, current mode stepdown architecture with two control channels operating at 180 degrees out of phase. In normal operation, each top MOSFET is turned on for a fixed interval determined by its own one-shot timer OST. When the top MOSFET is turned off, the bottom MOSFET is turned on until the current comparator ICMP trips, restarting the one-shot timer and repeating the cycle. The trip level of the current comparator is set by the ITH voltage which is the output of each error amplifier, EA. Inductor current is determined by sensing the voltage between the SENSE– and SENSE+ pins using either the bottom MOSFET on-resistance or a separate sense resistor. At low load currents, the inductor current can drop to zero and become negative. This is detected by current reversal comparator IREV, which then shuts off M2 resulting in discontinuous operation. Both switches will remain off with the output capacitor supplying the load current until the ITH voltage rises above the zero current level (0.8V) to initiate another cycle. Discontinuous mode operation is disabled when the FCB pin is brought below 1.9V, forcing continuous synchronous operation. The main control loop is shut down by pulling the RUN/SS pin low, turning off both M1 and M2. Releasing the pin allows an internal 1.2μA current source to charge an external soft-start capacitor, CSS. When this voltage reaches 1.3V, the controller turns on and begins switching, but with the effective reference voltage clamped at 0V. As CSS continues to charge, the effective reference ramps up at the same rate and controls the rise rate of the output voltage. Operating Frequency The operating frequency is determined implicitly by the top MOSFET on-time and the duty cycle required to maintain regulation. The one-shot timer generates an on-time that is proportional to the ideal duty cycle, thus holding frequency approximately constant with changes in VIN. The nominal frequency can be adjusted with an external resistor RON. When the LTC3708 is synchronized to an external clock, the operating frequency will then be solely determined by the external clock. Output Overvoltage Protection An overvoltage comparator OV guards against transient overshoots (>10%) as well as other more serious conditions that may overvoltage the output. In this condition, M1 is turned off and M2 is turned on and held on until the condition is cleared. Short-Circuit Detection and Protection After the controller has been started and given adequate time to charge the output capacitors, the RUN/SS capacitor is used as the short-circuit time-out capacitor. If either one of the output voltages falls to less than 70% of its nominal output voltage, the RUN/SS capacitor begins discharging on the assumption that the output is in an overcurrent and/or short-circuit condition. If the condition lasts for a long enough period, as determined by the size of the RUN/SS capacitor, both controllers will be shut down until the RUN/SS pin voltage is recycled. This built-in latchoff can be overridden by providing >5μA pull-up at a compliance of 5V to the RUN/SS pin. This current shortens the soft-start period but also prevents net discharge of the RUN/SS capacitor during an overcurrent and/or shortcircuit condition. Power Good (PGOOD) Pin Overvoltage and undervoltage comparators OV and UV pull the PGOOD output low if the output feedback voltage exceeds a ±10% window around the regulation point. In addition, the output feedback voltage must be out of this window for a continuous duration of at least 100μs before PGOOD is pulled low. This is to prevent any glitch on the feedback voltage from creating a false power bad signal. The PGOOD will indicate high immediately when the feedback voltage is in regulation. 3708fb 10 LTC3708 OPERATION (Refer to Functional Diagram) DRVCC Power for the top and bottom MOSFET drivers is derived from the DRVCC pin. The top MOSFET driver is powered from a floating bootstrap capacitor, CB. This capacitor is normally recharged from DRVCC through an external Schottky diode, DB, when the top MOSFET is turned off. 2-Phase Operation For the LTC3708 to operate optimally as a 2-phase controller, the resistors connected to the ION pins must be selected such that the free-running frequency of each channel is close to that of the other. An internal phase-locked loop (PLL) will then ensure that channel 2 operates at the same frequency as channel 1, but phase shifted by 180°. The loop filter connected to the INTLPF pin provides stability to the PLL. For external clock synchronization, a second PLL is incorporated to adjust the on-time of channel 1 until its frequency is the same as the external clock. Compensation for the external PLL is through the EXTLPF pin. The loop filter components tied to the INTLPF and EXTLPF pins are used to compensate the internal PPL and external PLL respectively. The typical value ranges are: INTLPF: RIPLL = 2kΩ to 10kΩ, CIPLL = 10nF to 100nF EXTLPF: REPLL ≤ 1kΩ, CEPLL = 10nF to 100nF For noise suppression, a capacitor with a value of 1nF or less should be placed from INTLPF to ground and EXTLPF to ground. The LTC3708’s 2-phase operation brings considerable benefits to portable applications and automatic electronics. It lowers the input filtering requirement, reduces electromagnetic interference (EMI) and increases the power conversion efficiency. Until the introduction of the 2-phase operation, dual switching regulators operated both channels in phase (i.e., single phase operation). This means that both controlling switches turned on at the same time, causing current pulses of up to twice the amplitude of those for one regulator to be drawn from the input capacitor or battery. Such operation results in higher input RMS current, larger and/or more expensive input capacitors, more power loss and worse EMI in the input source (whether a wall adapter or a battery). In contrast to single phase operation, the two channels of a 2-phase switching regulator are operated 180 degrees out of phase. This effectively interleaves the current pulses drawn by the switches, greatly reducing the overlap time where they add together. The result is a significant reduction in total RMS input current, which in turn allows less expensive input capacitors to be used, reduces shielding requirements for EMI and improves real world operating efficiency. Figure 1 compares the input waveforms for a representative single phase dual switching regulator to the 2-phase dual switching regulator. An actual measurement of the RMS input current under these conditions shows that 2-phase dropped the input current from 2.53ARMS to 1.55ARMS. 5V SWITCH 20V/DIV 3.3V SWITCH 20V/DIV INPUT CURRENT 5A/DIV INPUT VOLTAGE 500mV/DIV IIN(MEAS) = 2.53ARMS (1a) IIN(MEAS) = 1.55ARMS 3708 F01 (1b) Figure 1. Input Waveforms Comparing Single Phase (1a) and 2-Phase (1b) Operation for Dual Switching Regulators Converting 12V to 5V and 3.3V at 3A Each 3708fb 11 LTC3708 (Refer to Functional Diagram) While this is an impressive reduction in itself, remember that the power losses are proportional to I2RMS, meaning that the actual power wasted is reduced by a factor of 2.66. The reduced input ripple current also means that less power is lost in the input power path, which could include batteries, switches, trace/connector resistances and protection circuitry. Improvements in both conducted and radiated EMI also directly accrue as a result of the reduced RMS input current and voltage. Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative duty cycles which, in turn, are dependent upon the input voltage, VIN. Figure 2 shows how the RMS input current varies for single phase and 2-phase operation for 3.3V and 5V regulators over a wide input voltage range. It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range, but in fact extend over a wide region. A good rule of thumb for most applications is that 2-phase operation will reduce the input capacitance requirement to that for just one channel operating at maximum current and 50% duty cycle. 3.0 SINGLE PHASE DUAL CONTROLLER 2.5 INPUT RMS CURRENT (A) OPERATION 2.0 1.5 2-PHASE DUAL CONTROLLER 1.0 0.5 0 VO1 = 5V/3A VO2 = 3.3V/3A 0 10 20 30 INPUT VOLTAGE (V) 40 3708 F02 Figure 2. RMS Input Current Comparison 3708fb 12 LTC3708 APPLICATIONS INFORMATION The basic LTC3708 application circuit is shown on the first page of this data sheet. External component selection is primarily determined by the maximum load current and begins with the selection of the power MOSFET switches and/or sense resistor. For the LTC3708, the inductor current is determined by the RDS(ON) of the synchronous MOSFET or by a sense resistor when the user opts for more accurate current sensing. The desired amount of ripple current and operating frequency largely determines the inductor value. Finally, CIN is selected for its ability to handle the large RMS current into the converter and COUT is chosen with low enough ESR to meet the output voltage ripple specification. across this resistor, connect the SENSE+ pin to the source of the synchronous MOSFET and the SENSE– pin to the other end of the resistor. The SENSE+ and SENSE– pins provide the Kelvin connections, ensuring accurate voltage measurement across the resistor. Using a sense resistor provides a well-defined current limit, but adds cost and reduces efficiency. Alternatively, one can use the synchronous MOSFET as the current sense element by simply connecting the SENSE+ pin to the switch node SW and the SENSE– pin to the source of the synchronous MOSFET, eliminating the sense resistor. This improves efficiency, but one must carefully choose the MOSFET on-resistance as discussed below. Maximum Sense Voltage and VRNG Pin Power MOSFET Selection Inductor current is determined by measuring the voltage across the RDS(ON) of the synchronous MOSFET or through a sense resistor that appears between the SENSE+ and SENSE– pins. The maximum sense voltage is set by the voltage applied to the VRNG pin and is equal to approximately VRNG/7. The current mode control loop will not allow the inductor current valleys to exceed VRNG/(7 • RSENSE). In practice, one should allow some margin for variations in the LTC3708 and external component values. A good guide for selecting the sense resistance is: Each output stage of the LTC3708 requires two external N-channel power MOSFETs, one for the top (main) switch and one for the bottom (synchronous) switch. Important parameters for the power MOSFETs are the breakdown voltage V(BR)DSS, threshold voltage VGS(TH), on-resistance RDS(ON), reverse transfer capacitance, CRSS, and maximum current, IDS(MAX). RSENSE = VRNG 10 • IOUT (MAX) The voltage of the VRNG pin can be set using an external resistive divider from VCC between 0.5V and 2V, resulting in nominal sense voltages of 50mV to 200mV. Additionally, the VRNG pin can be tied to ground or VCC, in which case the nominal sense voltage defaults to 70mV or 140mV, respectively. The maximum allowed sense voltage is about 1.4 times this nominal value. Connecting the SENSE+ and SENSE– Pins The LTC3708 provides the user with an optional method to sense current through a sense resistor instead of using the RDS(ON) of the synchronous MOSFET. When using a sense resistor, it is placed between the source of the synchronous MOSFET and ground. To measure the voltage The gate drive voltage is set by the 5V DRVCC supply. Consequently, logic-level threshold MOSFETs must be used in LTC3708 applications. If the driver’s voltage is expected to drop below 5V, then sub-logic level threshold MOSFETs should be considered. When the bottom MOSFET is used as the current sense element, particular attention must be paid to its on-resistance. MOSFET on-resistance is typically specified with a maximum value RDS(ON)(MAX) at 25°C. Additional margin is required to accommodate the rise in MOSFET on-resistance with temperature: RDS(ON)(MAX) = RSENSE ρT The ρT term is a normalization factor (unity at 25°C) accounting for the significant variation in on-resistance with temperature, typically about 0.4%/°C. For a maximum junction temperature of 100°C, using a value ρ100°C = 1.3 is reasonable (see Figure 3). 3708fb 13 LTC3708 APPLICATIONS INFORMATION Operating Frequency The choice of operating frequency is a trade-off between efficiency and component size. Low frequency operation improves efficiency by reducing MOSFET switching and driving losses but requires larger inductance and/or capacitance to maintain low output ripple voltage. 1.5 1.0 0.5 0 –50 50 100 0 JUNCTION TEMPERATURE (°C) 150 3708 F03 The operating frequency of LTC3708 applications is determined implicitly by the one-shot timer that controls the on time, tON, of the top MOSFET switch. The on time is set by the current into the ION pin according to: tON = Figure 3. RDS(ON) vs Temperature The power dissipated by the top and bottom MOSFETs strongly depends upon their respective duty cycles and the load current. When the LTC3708 is operating in continuous mode, the duty cycles for the MOSFETs are: D TOP DBOT V = OUT VIN V –V = IN OUT VIN PTOP = DTOP • IOUT(MAX)2 • ρT(TOP) • RDS(ON) + • IOUT(MAX) • CRSS • f • ⎛ 1 1 ⎞ RDR • ⎜ + ⎟ VGS(TH) ⎠ ⎝ DRVCC – VGS(TH) ( ) Tying a resistor, RON, from VIN to the ION pin yields an on time inversely proportional to VIN. For a step-down converter, this results in approximately constant frequency operation as the input supply varies: f= VOUT 0.7 • RON (10pF ) Figure 4 shows how RON relates to switching frequency for several common output voltages. The resulting power dissipation in the MOSFETs at maximum output current are: (0.5) • VIN2 ( 0.7 10pF IION ) PBOT = DBOT • IOUT(MAX)2 • ρT(BOT) • RDS(ON) Both MOSFETs have I2R losses and the top MOSFET includes an additional term for transition losses, which are largest at high input voltages. The bottom MOSFET losses are greatest when the bottom duty cycle is near 100%, during a short circuit or at high input voltage. 1000 SWITCHING FREQUENCY (kHz) RT NORMALIZED ON-RESISTANCE 2.0 VOUT = 3.3V VOUT = 1.5V VOUT = 2.5V 100 100 1000 RON (kΩ) 10000 3708 F04 Figure 4. Switching Frequency vs RON 3708fb 14 LTC3708 APPLICATIONS INFORMATION PLL and Frequency Synchronization In the LTC3708, there are two onboard phase-locked loops (PLL). One PLL is used to achieve frequency locking and 180° phase shift between the two channels while the second PLL locks onto the rising edge of an external clock. Since the LTC3708 uses a constant on-time architecture, the error signal generated by the phase detector of the PLL is used to vary the on time to achieve frequency locking and phase separation. The variable on-time range is from 0.5 • tON to 2 • tON, where tON is the initial on time set by the RON resistor. To fully utilize the frequency synchronization range of the PLL, it is advisable to set the initial on time properly so that the two channels have close free-running frequencies. Frequencies far apart may exceed the synchronization capability of the PLL. If the two output voltages are VOUT1 and VOUT2, for example, RON resistors should then be selected proportionally: RON1 VOUT1 = RON2 VOUT2 Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and ripples in the output voltage. Highest efficiency operation is obtained at low frequency with small ripple current. However, achieving this requires a large inductor. There is a trade-off between component size and efficiency. A reasonable starting point is to choose a ripple current that is about 40% of IOUT(MAX). Note that the largest ripple current occurs at the highest VIN. To guarantee that ripple current does not exceed a specified maximum, the inductance should be chosen according to: ⎛ VOUT ⎞ ⎛ ⎞ V L=⎜ 1 – OUT ⎟ ⎟ ⎜ ⎝ f • ΔIL(MAX) ⎠ ⎝ VIN(MAX) ⎠ Once the value for L is known, the type of inductor must be selected. A variety of inductors designed for high current, low voltage applications are available from manufacturers such as Sumida and Panasonic. Schottky Diode Selection In this case, channel 1 will first be synchronized to the external frequency and channel 2 will then be synchronized to channel 1 with 180° phase separation. The Schottky diodes in parallel with both bottom MOSFETs conduct during the dead time between the conduction of the power MOSFET switches. They are intended to prevent the body diode of the bottom MOSFET from turning on and storing charge during the dead time, which causes a modest (about 1%) efficiency loss. The diodes can be rated for about one-half to one-fifth of the full load current since they are on for only a fraction of the duty cycle. In order for the diodes to be effective, the inductance between them and the bottom MOSFETs must be as small as possible, mandating that these components be placed as close as possible in the circuit board layout. The diodes can be omitted if the efficiency loss is tolerable. Inductor Selection CIN and COUT Selection Given the desired input and output voltages, the inductor value and operating frequency determine the ripple current: The selection of CIN is simplified by the 2-phase architecture and its impact on the worst-case RMS current drawn through the input network (battery/fuse/capacitor). It can be shown that the worst-case RMS current occurs when only one controller is operating. The controller with the Similarly, if the external PLL is engaged to synchronize to an external frequency of fEXT, RON1 should be selected close to: RON1 = VOUT1 0.7 • fEXT • 10pF ⎛ ⎞ VOUT2 hence, ⎜ RON2 = ⎟ ⎝ 0.7 • fEXT • 10pF ⎠ ⎛V ⎞⎛ V ⎞ ΔIL = ⎜ OUT ⎟ ⎜ 1 – OUT ⎟ VIN ⎠ ⎝ f •L ⎠ ⎝ 3708fb 15 LTC3708 APPLICATIONS INFORMATION highest (VOUT)(IOUT) product needs to be used in the formula below to determine the maximum RMS current requirement. Increasing the output current, drawn from the other out-of-phase controller, will actually decrease the input RMS ripple current from this maximum value (see Figure 2). The type of input capacitor, value and ESR rating have efficiency effects that need to be considered in the selection process. The capacitance value chosen should be sufficient to store adequate charge to keep pulsating input currents down. 20μF to 40μF is usually sufficient for a 25W output supply operating at 200kHz. The ESR of the capacitor is important for capacitor power dissipation as well as overall efficiency. All of the power (RMS ripple current2 • ESR) not only heats up the capacitor but wastes power from the battery. Medium voltage (20V to 35V) ceramic, tantalum, OS-CON and switcher-rated electrolytic capacitors can be used as input capacitors, but each has drawbacks: ceramic voltage coefficients are very high and may have audible piezoelectric effects; tantalums need to be surge-rated; OS-CONs suffer from higher inductance, larger case size and limited surface-mount applicability; electrolytics’ higher ESR and dryout possibility require several to be used. 2-phase systems allow the lowest amount of capacitance overall. As little as one 22μF or two to three 10μF ceramic capacitors are an ideal choice in a 20W to 35W power supply due to their extremely low ESR. Even though the capacitance at 20V is substantially below their rating at zero-bias, very low ESR loss makes ceramics an ideal candidate for highest efficiency battery operated systems. Also consider parallel ceramic and high quality electrolytic capacitors as an effective means of achieving ESR and bulk capacitance goals. In continuous mode, the current of the top N-channel MOSFET is approximately a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current of one channel must be used. The maximum RMS capacitor current is given by: IRMS ≈ IMAX ( ) 1/ 2 ⎡ VOUT VIN − VOUT ⎤ ⎣ ⎦ VIN This formula has a maximum at VIN = 2VOUT , where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. The benefit of the LTC3708 2-phase operation can be calculated by using the equation above for the higher power channel and then calculating the loss that would have resulted if both controller channels switch on at the same time. The total RMS power lost is lower when both controllers are operating due to the interleaving of current pulses through the input capacitor’s ESR. This is why the input capacitor’s requirement calculated above for the worst-case controller is adequate for the dual controller design. Remember that input protection fuse resistance, battery resistance and PC board trace resistance losses are also reduced due to the reduced peak currents in a 2-phase system. The overall benefit of a 2-phase design will only be fully realized when the source impedance of the power supply/battery is included in the efficiency testing. The drains of the two top MOSFETS should be placed within 1cm of each other and share a common CIN(s). Separating the drains and CIN may produce undesirable voltage and current resonances at VIN. The selection of COUT is driven by the effective series resistance (ESR) required to minimize voltage ripple and load step transients. The output ripple (ΔVOUT) is determined by: ⎛ 1 ⎞ ΔVOUT ≈ ΔIL ⎜ ESR + ⎟ ⎝ 8fC OUT ⎠ where f = operating frequency, COUT = output capacitance, and ΔIL = ripple current in the inductor. The output ripple is highest at maximum input voltage since ΔIL increases with input voltage. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the necessary RMS current rating. 3708fb 16 LTC3708 APPLICATIONS INFORMATION Manufacturers such as Nichicon, United Chemi-Con and Sanyo can be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest (ESR)(size) product of any aluminum electrolytic at a somewhat higher price. An additional ceramic capacitor in parallel with OS-CON capacitors is recommended to reduce the inductance effects. In surface mount applications multiple capacitors may need to be used in parallel to meet the ESR, RMS current handling and load step requirements of the application. Aluminum electrolytic, dry tantalum and special polymer capacitors are available in surface mount packages. Special polymer capacitors offer very low ESR but have lower storage capacity per unit volume than other capacitor types. These capacitors offer a very cost-effective output capacitor solution and are an ideal choice when combined with a controller having high loop bandwidth. Tantalum capacitors offer the highest capacitance density and are often used as output capacitors for switching regulators having controlled soft-start. Several excellent surge-tested choices are the AVX TPS, AVX TPSV or the KEMET T510 series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Aluminum electrolytic capacitors can be used in cost-driven applications providing that consideration is given to ripple current ratings, temperature and long term reliability. A typical application will require several to many aluminum electrolytic capacitors in parallel. A combination of the above mentioned capacitors will often result in maximizing performance and minimizing overall cost. Other capacitor types include Nichicon PL series, Sanyo POSCAP, NEC Neocap, Cornell Dubilier ESRE and Sprague 595D series. Consult manufacturers for other specific recommendations. Top MOSFET Driver Supply (CB, DB in the Functional Diagram) An external bootstrap capacitor, CB, connected to the BOOST pin supplies the gate drive voltage for the topside MOSFET. This capacitor is charged through diode DB from DRVCC when the switch node is low. Note that the average voltage across CB is approximately DRVCC. When the top MOSFET turns on, the switch node rises to VIN and the BOOST pin rises to approximately VIN + DRVCC. The boost capacitor needs to store about 100 times the gate charge required by the top MOSFET. In most applications 0.1μF to 0.47μF is adequate. Discontinuous Mode Operation and FCB Pin The FCB pin determines whether the bottom MOSFET remains on when current reverses in the inductor. Tying this pin above its 2.3V threshold (typically to VCC) enables discontinuous operation where the bottom MOSFET turns off when inductor current reverses. The load current at which current reverses and discontinuous operation begins depends on the amplitude of the inductor ripple current and the ripple current depends on the choice of inductor value and operating frequency as well as the input and output voltages. Tying the FCB pin below 1.9V forces continuous synchronous operation, allowing current to reverse at light loads and maintaining high frequency operation. Besides providing a logic input to force continuous operation, the FCB pin acts as the input for external clock synchronization. Upon detecting the presence of an external clock signal, channel 1 will lock on to this external clock and this will be followed by channel 2 (see PLL and Frequency Synchronization). The LTC3708 defaults to forced continuous mode when sychronized to an external clock or when the PGOOD signal is low. Fault Conditions: Current Limit The maximum inductor current is inherently limited in a current mode controller by the maximum sense voltage. In the LTC3708, the maximum sense voltage is controlled by the voltage on the VRNG pin. With valley current control, the maximum sense voltage and the sense resistance determine the maximum allowed inductor valley current. The corresponding output current limit is: ILIMIT = VSNS(MAX ) RDS(ON) • ρT + 1 • ΔIL 2 3708fb 17 LTC3708 APPLICATIONS INFORMATION Caution should be used when setting the current limit based upon the RDS(ON) of the MOSFETs. The maximum current limit is determined by the minimum MOSFET on-resistance. Data sheets typically specify nominal and maximum values for RDS(ON), but not a minimum. A reasonable assumption is that the minimum RDS(ON) lies the same amount below the typical value as the maximum lies above it. Consult the MOSFET manufacturer for further guidelines. For a more accurate current limiting, a sense resistor can be used. Sense resistors in the 1W power range can be easily available in the 5%, 2% or 1% tolerance. The temperature coefficient of these resistors is very low, ranging from ±250ppm/°C to ±75ppm/°C. In this case, the (RDS(ON) • ρT) product in the above equation can simply be replaced by the RSENSE value. 2.0 SWITCHING FREQUENCY (MHz) The current limit value should be checked to ensure that ILIMIT(MIN) > IOUT(MAX). The minimum value of current limit generally occurs with the largest VIN at the highest ambient temperature, conditions which cause the largest power loss in the converter. Note that it is important to check for self-consistency between the assumed junction temperature and the resulting value of ILIMIT , which heats the junction. 1.5 DROPOUT REGION 1.0 0.5 0 0 3708 F05 Figure 5. Maximum Switching Frequency vs Duty Cycle Soft-Start and Latchoff with the RUN/SS Pin The RUN/SS pin provides a means to shut down the LTC3708 as well as a timer for soft-start and overcurrent latchoff. Pulling the RUN/SS pin below 0.8V shuts down the LTC3708. Releasing the pin allows an internal 1.2μA internal current source to charge the external capacitor, CSS. If RUN/SS has been pulled all the way to ground, there is a delay before starting of about: Minimum Off Time and Dropout Operation The minimum off time tOFF(MIN) is the smallest amount of time that the LTC3708 is capable of turning on the bottom MOSFET, tripping the current comparator and turning the MOSFET back off. This time is generally about 270ns. The minimum off time limit imposes a maximum duty cycle of tON/(tON + tOFF(MIN)). If the maximum duty cycle is reached, due to a dropping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is: VIN(MIN) = VOUT tON + tOFF(MIN) tON A plot of maximum frequency vs duty cycle is shown in Figure 5. 1.0 0.25 0.50 0.75 DUTY CYCLE (VOUT/VIN) tDELAY = ( ) 1.3V • C = 1.1s/μF CSS 1.2μA SS When the RUN/SS voltage reaches the ON threshold (typically 1.3V), the LTC3708 begins operating with a clamp on channel 1’s reference voltage. The clamp level is one threshold voltage below RUN/SS. As the voltage on RUN/SS continues to rise, channel 1’s reference is raised at the same rate, achieving monotonic output voltage soft-start (Figure 6). When RUN/SS rises 0.6V above the ON threshold, the reference clamp is invalidated and the internal precision reference takes over. When channel 2 is tracked to channel 1, soft-start on channel 2 is automatically achieved (see Output Voltage Tracking). 3708fb 18 LTC3708 APPLICATIONS INFORMATION VIN 3.3V OR 5V RUN/SS D1 $V = 0.6V RUN/SS RSS* ON ON THRESHOLD CSS TIME 3708 F07 VOUT1 *OPTIONAL TO OVERRIDE OVERCURRENT LATCHOFF Figure 7. RUN/SS Pin Interfacing with Latchoff Defeated TIME Output Voltage Tracking 3708 F06 Figure 6. Monotonic Soft-Start Waveforms Controlled soft-start requires that the timing capacitor, CSS, be made large enough to guarantee that the output can track the voltage rise on the RUN/SS pin. The minimum CSS capacitance can be calculated: CSS > R1 + R2 30μA • RSENSE • • COUT R1 VRNG where R1 and R2 are the feedback resistive dividers (Functional Diagram), COUT is the output capacitance and RSENSE is the current sense resistance. When bottom MOSFET RDS(ON) is used for current sensing, RSENSE should be replaced with the worst-case RDS(ON)(MAX). Generally, 0.1μF is more than sufficient for CSS. After the controller has been started and given adequate time to charge the output capacitor, CSS is used as a shortcircuit timer. After the RUN/SS pin charges above 3V and if either output voltage falls below 70% of its regulated value, a short-circuit fault is assumed. A 2μA current then begins discharging CSS. If the fault condition persists until the RUN/SS pin drops to 2.5V, the controller turns off all power MOSFETs, shutting down both channels. The RUN/SS pin must be actively pulled down to ground in order to restart operation. Overcurrent latchoff operation is not always needed or desired and can prove annoying during troubleshooting. This feature can be overridden by adding a pull-up current of >5μA to the RUN/SS pin (Figure 7). The additional current prevents the discharge of CSS during a fault and also shortens the soft-start period. The LTC3708 allows the user to program how the second channel output ramps up and down by means of the TRACK2 pin. Through this pin, the second channel output can be set up to either coincidently or ratiometrically track the channel 1 output, as shown in Figure 8. Similar to RUN/SS, the TRACK2 pin acts as a clamp on channel 2’s reference voltage. VOUT2 is referenced to the TRACK2 voltage when the TRACK2 < 0.6V and to the internal precision reference when TRACK2 > 0.6V. To implement the tracking in Figure 8a, connect an extra resistive divider to the output of channel 1 and connect its midpoint to the TRACK2 pin. The ratio of this divider should be selected the same as that of channel 2’s feedback divider (Figure 9a). In this tracking mode, VOUT1 must be set higher than VOUT2. To implement the ratiometric tracking in Figure 8b, no extra divider is needed; simply connect the TRACK2 pin to the VFB1 pin (Figure 9b). By selecting different resistors, the LTC3708 can achieve different modes of tracking including the two in Figure 8. So which mode should be programmed? While either mode in Figure 8 satisfies most practical applications, there does exist some trade-off. The ratiometric mode saves a pair of resistors but the coincident mode offers better output regulation. This can be better understood with the help of Figure 10. At the input stage of channel 2’s error amplifier, two common anode diodes are used to clamp the equivalent reference voltage and an additional diode is used to match the shifted common mode voltage. The top two current sources are of the same amplitude. In the coincident mode, the TRACK2 voltage is substantially higher than 0.6V at steady state and effectively turns off 3708fb 19 LTC3708 APPLICATIONS INFORMATION VOUT1 OUTPUT VOLTAGE OUTPUT VOLTAGE VOUT1 VOUT2 VOUT2 3708 F08 TIME TIME (8a) Coincident Tracking (8b) Ratiometric Tracking Figure 8. Two Different Modes of Output Voltage Tracking VOUT1 VOUT2 R3 R1 TO VFB1 PIN TO TRACK2 PIN R4 VOUT1 R3 VOUT2 R1 TO TRACK2 PIN TO VFB2 PIN R2 R4 R3 TO VFB1 PIN R2 TO VFB2 PIN R4 3708 F09 (9a) Coincident Tracking Setup (9b) Ratiometric Tracking Setup Figure 9. Setup for Coincident and Ratiometric Tracking ⎛ R1 VOUT1 R3 VOUT 2 ⎞ ⎜⎝ R2 = 0.6 − 1, R4 = 0.6 − 1⎟⎠ I I + D1 D2 EA2 TRACK2 0.6V – D3 VFB2 3708 F10 I-V characteristic of the diodes, it does impose a finite amount of output voltage deviation. Further, when channel 1’s output experiences dynamic excursions (under load transient, for example), channel 2 will be affected as well. For better output regulation, use the coincident tracking mode instead of ratiometric. Figure 10. Equivalent Input Circuit of Error Amplifier of Channel 2 The number of resistors in Figure 9a can be further reduced with the scheme in Figure 11. D1. D2 and D3 will therefore conduct the same current and offer tight matching between VFB2 and the internal precision 0.6V reference. In the ratiometric mode, however, TRACK2 equals 0.6V even at steady state. D1 will divert part of the bias current and make VFB2 slightly lower than 0.6V. Although this error is minimized by the exponential In a system that requires more than two tracked supplies, multiple LTC3708s can be daisy-chained through the TRACK1 pin. TRACK1 clamps channel 1’s reference in the same manner TRACK2 clamps channel 2. To eliminate the possibility of multiple LTC3708s coming on at different times, only the master LTC3708’s RUN/SS pin should be 3708fb 20 LTC3708 APPLICATIONS INFORMATION connected to a soft-start capacitor. All other LTC3708s should have their RUN/SS pins pulled up to VCC with a resistor between 50k and 300k. Figure 12 shows the circuit with four outputs. Three of them are programmed in the coincident mode while the fourth one tracks ratiometrically. If output tracking is not needed, connect the TRACK pins to VCC. Do Not Float These Pins. Although all dissipative elements in the circuit produce losses, four main sources account for most of the losses in LTC3708 circuits: VOUT2 R1 R4 R2 R5 TO VFB2 PIN TO TRACK2 PIN 1. DC I2R Losses. These arise from the resistances of the MOSFETs, inductor and PC board traces and cause the efficiency to drop at high output currents. In continuous mode, the average output current flows through L, but is chopped between the top and bottom MOSFETs. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L and the board traces to obtain the DC I2R loss. For example, if RDS(ON) = 0.01Ω and RL = 0.005Ω, the loss will range from 15mW up to 1.5W as the output current varies from 1A to 10A. TO VFB1 PIN R3 3708 F11 Figure 11. Alternative Setup for Coincident Tracking ⎛ R1+ R2 VOUT1 R1 R4 VOUT 2 ⎞ ⎜⎝ R3 = 0.6 – 1, R2 + R3 = R5 = 0.6 − 1⎟⎠ TRACK1 TRACK2 TO VCC VOUT1 R4 R5 LTC3708 “MASTER” R1 VFB1 R2 R2 R2 VOUT2 R3 VFB2 RUN/SS VOUT1 R2 CSS TO VCC VOUT3 R4 100k TRACK1 TRACK2 LTC3708 “SLAVE” VFB1 R2 RUN/SS The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. VOUT4 OUTPUT VOLTAGE VOUT1 Efficiency Considerations VOUT3 VOUT4 R5 VOUT2 VFB2 R2 3708 F12 TIME (12a) Circuit Setup (12b) Output Voltage Figure 12. Four Outputs with Tracking and Ratiometric Sequencing ⎛ R1 VOUT1 R5 VOUT 4 ⎞ R3 VOUT 2 R4 VOUT 3 ⎜⎝ R2 = 0.6 − 1, R2 = 0.6 − 1, R2 = 0.6 − 1, R2 = 0.6 − 1⎟⎠ 3708fb 21 LTC3708 APPLICATIONS INFORMATION 2. Transition Loss. This loss arises from the brief amount of time the top MOSFET spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V and can be estimated from: Transition Loss ≈ (0.5) • VIN2 • IOUT • CRSS • f • ⎛ 1 1 ⎞ RDS(ON)_ DRV ⎜ + ⎟ ⎝ DRVCC − VGS(TH) VGS(TH) ⎠ 3. DRVCC and VCC Current. This is the sum of the MOSFET driver and control currents. The driver current supplies the gate charge QG required to switch the power MOSFETs. This current is typically much larger than the control circuit current. In continuous mode operation: IGATECHG = f(QG(TOP) + QG(BOT)) 4. CIN Loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator. It must have a very low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries. The LTC3708 2-phase architecture typically halves this CIN loss over the single phase solutions. Other losses, including COUT ESR loss, Schottky conduction loss during dead time and inductor core loss generally account for less than 2% additional loss. When making any adjustments to improve efficiency, the final arbiter is the total input current for the regulator at your operating point. If you make a change and the input current decreases, then you improve the efficiency. If there is no change in input current, then there is no change in efficiency. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ΔILOAD (ESR), where ESR is the effective series resistance of COUT . ΔILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problems. The ITH pin external components shown in Figure 13 will provide adequate compensation for most applications. For a detailed explanation of switching control loop theory see Linear Technology Application Note 76. Design Example As a design example, take a supply with the following specifications: VIN = 7V to 28V (15V nominal), VOUT1 = 2.5V, VOUT2 = 1.8V, IOUT1(MAX) = IOUT2(MAX) = 10A, f = 500kHz and VOUT2 to track VOUT1. First calculate the timing resistor: RON1 = 2.5V = 714k (0.7V)(500kHz)(10pF ) Select a standard value of 715k. RON2 = 1.8 V = 514k (0.7V)(500kHz)(10pF ) Select a standard value of 511k. Next, choose the feedback resistors: R1 2.5V = – 1 = 3.17 R2 0.6 V Select R1 = 31.6k, R2 = 10k. R3 1.8 V = – 1= 2 R4 0.6 V Select R3 = 20k, R4 = 10k. For VOUT2 to coincidently track VOUT1 at start-up, connect an extra pair of R3 and R4 across VOUT1 with its midpoint tied to the TRACK2 pin. 3708fb 22 LTC3708 APPLICATIONS INFORMATION Third, design the inductors for about 40% ripple current at the maximum VIN: L1 = 2.5V ⎛ 2.5V ⎞ ⎜ 1– ⎟ = 1.1μH (500kHz)(0.4)(10A) ⎝ 28V ⎠ A standard 1μH inductor will result in 45% of ripple current (4.5A) at worst case. L2 = 1.8 V ⎛ 1.8 V ⎞ ⎜ 1– ⎟ = 0.8 μH (500kHz)(0.4)(10A) ⎝ 28V ⎠ L2 can also use 1μH to save some BOM (Bill of Material) cost; the resulting ripple current is 3.4A. The selection of MOSFETs is simplified by the fact that both channels have the same maximum output current. Select the top and bottom MOSFETs for one channel and the same MOSFETs can be used for the other. Take channel 1 for calculation and begin with the bottom synchronous MOSFET. As stated previously in the Power MOSFET Selection section, the major criterion in selecting the bottom MOSFET is low RDS(ON). Choose an Si4874 for example: RDS(ON) = 0.0083Ω (nom) 0.010Ω (max), θJA = 40°C/W. The nominal sense voltage is: VSNS(NOM) = (10A)(1.3)(0.0083) = 108mV Tying VRNG1 to 1.1V will set the current sense voltage range for a nominal value of 110mV with the current limit occurring at 146mV. To check if the current limit is acceptable, assume a junction temperature of about 80°C above a 70°C ambient with ρ150°C = 1.5: ILIMIT ≥ 146mV 1 + (4.1A ) = 11.8 A (1.5)(0.010Ω) 2 and double check the assumed TJ in the MOSFET: PBOT = 28 V – 2.5V 2 11.8 A ) (1.5)(0.010Ω) = 1.9W ( 28 V TJ = 70°C + (1.90W)(40°C/W) = 146° Because the top MOSFET is on for only a short time, an Si4884 will be sufficient: RDS(ON) = 0.0165Ω (max), CRSS = 190pF, VGS(TH) = 1V, θJA = 42°C/W. Checking its power dissipation at current limit with ρ130°C = 1.6: 2.5V 2 2 11.8 A ) (1.6)(0.0165Ω) + (0.5)(28 V ) ( 28 V (11.8A)(190pF )(500kHz)(2Ω)⎛⎜⎝ 5V 1– 1V + 11V ⎞⎟⎠ = 0.33W + 1.10W = 1.43W PTOP = TJ = 70°C + (1.43W)(42°C/W) = 130° The junction temperatures for both top and bottom MOSFETs will be significantly less at nominal current, but the above analysis shows that careful attention to PCB layout and heat sinking will be necessary in this circuit. The same MOSFETs (Si4874 and Si4884) can be used for channel 2. Finally, an input capacitor is chosen for an RMS current rating of about 5A at 85°C and the output capacitors are chosen for a low ESR of 0.013Ω to minimize output voltage changes due to inductor ripple current and load steps. The ripple voltage will be only: ⎛ ⎞ 1 ΔVOUT1(RIPPLE) = ΔIL1 • ⎜ ESR + ⎟ ⎝ 8 • f • COUT ⎠ ⎛ ⎞ 1 = 4.5A • ⎜ 0.013Ω + ⎟ 8 • 500kHz • 470μF ⎠ ⎝ = 60mV ⎛ ⎞ 1 ΔVOUT2(RIPPLE) = ΔIL2 • ⎜ ESR + ⎟ ⎝ 8 • f • COUT ⎠ ⎛ ⎞ 1 = 3.4A • ⎜ 0.013Ω + ⎟ 8 • 500kHz • 470μF ⎠ ⎝ = 46mV However, a 0A to 10A load step will cause an output change of up to: ΔVOUT(STEP) = ΔILOAD(ESR) = (10A)(0.013Ω) = 130mV An optional 22μF ceramic output capacitor is included to minimize the effect of ESL in the output ripple. The complete circuit is shown in Figure 13. 3708fb 23 LTC3708 APPLICATIONS INFORMATION VIN 7V TO 28V CIN 10μF 35V s4 BAT54A 5V + 10Ω 4.7μF 1μF BOOST2 VCC PGND1 17 4 31 21 VCC TRACK1 FCB DRVCC 27 TG1 TG2 28 BOOST2 BOOST1 0.22μF 26 SW2 SW1 25 SENSE2+ SENSE1+ 22 BG1 LTC3708EUH BG2 24 SENSE2– SENSE1– 23 PGND1 PGND2 32 VRNG2 VRNG1 VRNG2 29 ION1 ION2 3 VFB2 VFB1 6 TRACK2 PWRGD 9 EXTLPF INTLPF 0.01μF 2 ITH1 ITH2 1μF 1μF VOUT1 2.5V 10A 22μF 6.3V X7R L1 1μH + COUT1 470μF 4V B340A M1 M3 PGND1 20k 1% 56pF 10k 1% 31.6k 1% VIN R2 10k 1% 715k RUN/SS 100pF 20k 680pF 1 SGND 5 0.1μF 14 13 BOOST2 M2 L2 1μH 0.22μF 15 16 + 20 B340A M4 VOUT1 1.8V 10A COUT2 470μF 4V 22μF 6.3V X7R 18 19 11 511k 12 7 30 10 8 39k VIN VCC 20k 1% 56pF 5V 100k PGOOD 10k 11k 20k 100pF 680pF 10k 1% 0.022μF 0.01μF 1nF 3708 F13 CIN: UNITED CHEMI-CON THCR60EIH106ZT COUT1, COUT2: SANYO POSCAP 4TPD470M L1, L2: SUMIDA CEP125-1R0M M1, M2: VISHAY Si4884 M3, M4: VISHAY Si4874 Figure 13. Design Example: 2.5V/10A and 1.8V/10A at 500kHz with Output Tracking PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3708. These items are also illustrated graphically in Figure 14. Figure 15 further shows the current waveforms present in the various branches of the 2-phase synchronous Buck regulators operating in the continuous mode. • Place the loop of M1, M3 and CIN1 in a compact area. This loop conducts high pulsating current and its area needs to be minimized. Place M2, M4 and CIN2 in the same way. • Place CIN1 and CIN2 within the distance of 1cm. Longer distance may cause a large resonant loop. • Connect the negative plates of COUT1 and CDR1 to PGND1 before it joins PGND2 at the ground plane. Connect COUT2 and CDR2 in the same way so that power grounds are separated before they meet at a single point. • Cover the board area under the LTC3708 with a SGND plane. For the LTC3708EUH, solder the back of the IC to this plane. Separate SGND from the power ground and connect all signal components (ITH, VFB, ION, VCC, EXTLPF, INTLPF, VRNG, TRACK and RUN/SS) to the SGND plane before it joins PGND. Connect SGND to the gound plane at a single point. • Run SENSE+ and SENSE– across the bottom MOSFET (or RSENSE when a separate current sensing resistor is used) with Kelvin connection (Figure 16). Route SENSE+ and SENSE– together with minimum PC trace separation. The filter capacitor (when used) between SENSE+ and SENSE– should be as close to the LTC3708 as possible. • Keep the high dV/dt nodes SW, TG and BOOST away from sensitive small-signal nodes. 3708fb 24 LTC3708 APPLICATIONS INFORMATION RON1 FCB TG1 R1 PGOOD VFB1 R3 SW1 ITH1 BOOST ION1 CB1 COUT1 SENSE1+ EXTLPF D1 L1 SENSE1– TRACK1 PGND1 VRNG1 M3 CDR1 CIN1 VIN CIN2 CVCC LTC3708 DRVCC 5V SGND CDR2 BG2 M2 5V VCC BG1 M1 M4 PGND2 CSS RUN/SS VRNG2 SENSE2– TRACK2 L2 D2 SENSE2+ COUT2 INTLPF BOOST2 ION2 SW2 ITH2 TG2 VFB2 CB2 R4 R2 RON2 3708 F15 Figure 14. LTC3708 Layout Diagram • Connect the decoupling capacitors CDR1 and CDR2 close to the DRVCC and PGND pins. Connect CB1 and CB2 close to the BOOST and SW pins. • Connect the decoupling capacitor CVCC right across the VCC pin and SGND plane. Connect the EA compensation components close to the ITH pins. Connect the PLL loop filter close to the EXTLPF and INTLPF pins. Connect the ION decoupling capacitor close to the ION pins. • Flood all unused areas on all layers with copper. Flooding will reduce the temperature rise of the power components. You can connect the copper area to any DC net (VIN, VOUT , GND or to any other DC rail in your system). 3708fb 25 LTC3708 APPLICATIONS INFORMATION SW1 L1 D1 VOUT1 COUT1 + RL1 CERAMIC VIN RIN CIN + SW2 L2 D2 BOLD LINES INDICATE HIGH, SWITCHING CURRENT LINES. KEEP LINES TO A MINIMUM LENGTH. VOUT2 COUT2 + RL2 CERAMIC 3708 F15 Figure 15. Branch Current Waveforms D G D S D S D S RSENSE MOSFET SENSE+ SENSE– SENSE+ SENSE– (16a) Sensing the Bottom MOSFET 3708 F16 (16b) Sensing a Resistor Figure 16. Kelvin Sensing 3708fb 26 LTC3708 APPLICATIONS INFORMATION 5V 10Ω 1μF 4.7μF DB1 B340LA M3 19.1k 1% VIN 6.04k 1% CB1 0.1μF 6.04k 1% 15k 470pF 1000pF 3.32k 470pF LTC3708 150pF CSS 0.1μF L2 1.22μH + B340LA M4 BG2 SENSE2– PGND2 VFB1 VFB2 TRACK2 FCB ION1 ION2 ITH1 ITH2 INTLPF EXTLPF RUN/SS TRACK1 VRNG2 SGND VRNG1 0.01μF M2 CB2 0.1μF SW2 SENSE2+ SW1 SENSE1+ SENSE1– PGND1 RON1 1.5M DB2 VCC DRVCC PGOOD TG1 TG2 BOOST2 BOOST1 BG1 56pF 100k + VOUT1 2.5V 15A COUT1 + 330μF 4V s2 12.1k 1% M1 L1 1.43μH VIN 7V TO 24V CIN 10μF 25V s6 COUT2 470μF 2.5V s2 VOUT2 1.8V 15A 12.1k 1% FREQ = 220kHz fIN VIN RON2 1.1M 475Ω 130k 24.9k 1000pF 5V 0.047μF 6.04k 1% 15k 150pF 1000pF 470pF 3708 F17 COUT1: SANYO POSCAP 4TPD330M COUT2: SANYO POSCAP 2R5TPD470M CIN: TAIYO YUDEN: TMK325BJ106KM DB1, DB2: CMDSH-3 L1: PANASONIC ETQP3H1R4BF L2: PANASONIC ETQP2H1R2BF M1, M2: RENESAS HAT2168H M3, M4: RENESAS HAT2165H Figure 17. High Efficiency, Dual Output Power Supply with External Frequency Synchronization 3708fb 27 28 + VIN RUN/SS SGND GND 1.8V 6A GND 9V TO 20V 10k 1% 30.1k 1% COUT1 470μF 2.5V 100pF 6.04k 1% 100pF B240A L1 2.2μH C1 10μF 25V 12.1k 1% C2 10μF 25V 10k 59k VIN VCC Si4860DY Si4860DY 1.2M 0Ω 0Ω BOOST1 1nF 15k 150pF 100pF 0.22μF 0Ω BOOST2 BAT54A 1nF * 1μF VCC 10Ω VCC 5V 1M VIN SW2 SENSE2+ * VFB2 PWRGD INTLPF ITH2 SGND ION2 SENSE2– BG2 PGND2 VRNG2 0.1μF VFB1 TRACK2 EXTLPF ITH1 RUN/SS ION1 SENSE1– BG1 PGND1 VRNG1 TG2 BOOST2 LTC3708EUH SENSE1+ SW1 BOOST1 TG1 VCC TRACK1 FCB DRVCC 2.2μF 6.3V X5R 5V 15k 1nF 100pF VIN VCC 10nF 9.09k 604k 0Ω 0Ω BOOST2 470pF 10k 30.9k Si4860DY B240A Si4860DY 10k 1% L2 2.2μH C3 10μF 25V 3708 TA03 10k 1% 100pF B240A *SIGNAL GROUND, ROUTED SEPARATELY COUT1, COUT2: SANYO 2R5TPD470M C1 TO C4: TAIYO YUDEN TMK325BJ106MM L1, L2: SUMIDA CDEP105-2R2MC-88 1nF 150pF 0.22μF 0Ω 2.2μF 2.2μF DDR II Supplies with Transient Coupling + 5V 0.9V ±4A 100k PGOOD COUT2 470μF 2.5V GND C4 10μF 25V LTC3708 TYPICAL APPLICATIONS 3708fb LTC3708 TYPICAL APPLICATIONS Dual-Phase, 30A Power Supply with 10mV Output Ripple VIN 5V + CIN1 100μF 6.3V CIN2-CIN7 4.7μF 6.3V s6 TRACK BAT54A 1μF PGND1 10Ω 1μF BOOST2 VCC 1μF VOUT 1V 30A COUT1 1μF 6.3V M1 L1 0.19μH + 0.22μF PGND2 VCC TRACK1 FCB DRVCC TG1 TG2 BOOST2 BOOST1 SW1 B340A M3 100k R1 VCC 10k 0.1% 274k VIN LTC3708 BG2 PGND1 PGND2 VRNG1 VRNG2 RUN/SS 0.01μF 1000pF 220pF CIN1: SANYO OS-CON 6SVP100M COUT3, COUT4: SANYO POSCAP 2R5TPD470M L1, L2: PANASONIC ETQP4LR19 M1 TO M4: RENESAS HAT2165 22.1k VRNG1 B340A M4 COUT4 470μF 2.5V s2 + COUT2 1μF 6.3V VIN 274k VIN ION2 VFB2 PWRGD INTLPF ITH2 100k PGOOD 1nF SGND 220pF 100k VCC 22k 100pF SENSE2– SENSE1– ION1 VFB1 TRACK2 EXTLPF ITH1 R2 15k 0.1% L2 0.19μH 0.22μF SW2 BG1 100pF M2 SENSE2+ SENSE1+ COUT3 470μF 2.5V s2 BOOST2 10k 0.01μF 0.01μF 3708 TA05 3708fb 29 LTC3708 TYPICAL APPLICATIONS 12V/12A and 5V/12A at 300kHz Application VIN 20V TO 28V + 100μF 50V C1 3.3μF 50V X5R C2 3.3μF 50V X5R 5V BAT54A BOOST1 BOOST2 10Ω 5V VCC C3 3.3μF 50V X5R C4 3.3μF 50V X5R C6 1μF V TRACK1 FCB DRV CC CC Q1 HAT2167H 0Ω TG2 TG1 BOOST1 VOUT1 12V 12A L1 3.5μH 0.1μF COUT1 150μF 16V s2 B340LA 22pF 75k 10k 191k 10k 22pF Q2 HAT2167H 0Ω 100pF 24.9k SENSE1+ SENSE2+ 100pF BG2 PGND2 VRNG2 ION1 1nF 1nF 88.7k + 75k SGND 5V 47pF 100k 5.11k 24.9k 10k 150pF 470pF 1.5nF 0.1μF COUT2 220μF 6.3V s2 VOUT2 5V 12A PGOOD ITH2 RUN/SS 2.2nF 33.2k VIN VCC D3 B340LA VFB2 PGOOD INTLPF ITH1 150pF Q4 HAT2167H 2.2M ION2 VFB1 TRACK2 EXTLPF 0Ω 0Ω SENSE2– PGND1 VRNG1 5.6M L2 2.4μH 0.1μF SW2 SENSE1– 88.7k VIN VCC C6 3.3μF 50V X5R BOOST2 LTC3708EUH BG1 0Ω Q3 HAT2167H 0Ω BOOST2 SW1 + 2.2μF C5 3.3μF 50V X5R 15k 47pF 22nF 3708 TA04 COUT1: SANYO 16SVP150M COUT2: SANYO 6TPD220M C1 TO C6: TDK C4532X5R1H335M L1: SUMIDA CDEP147-3R5MC-H L2: SUMIDA CDEP147-2R4MC 3708fb 30 LTC3708 PACKAGE DESCRIPTION UH Package 32-Lead Plastic QFN (5mm × 5mm) (Reference LTC DWG # 05-08-1693 Rev D) 0.70 ±0.05 5.50 ±0.05 4.10 ±0.05 3.45 ± 0.05 3.50 REF (4 SIDES) 3.45 ± 0.05 PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC RECOMMENDED SOLDER PAD LAYOUT APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 5.00 ± 0.10 (4 SIDES) BOTTOM VIEW—EXPOSED PAD 0.75 ± 0.05 R = 0.05 TYP 0.00 – 0.05 PIN 1 NOTCH R = 0.30 TYP OR 0.35 s 45° CHAMFER R = 0.115 TYP 31 32 0.40 ± 0.10 PIN 1 TOP MARK (NOTE 6) 1 2 3.50 REF (4-SIDES) 3.45 ± 0.10 3.45 ± 0.10 (UH32) QFN 0406 REV D 0.200 REF NOTE: 1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE M0-220 VARIATION WHHD-(X) (TO BE APPROVED) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 0.25 ± 0.05 0.50 BSC 3708fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 31 LTC3708 TYPICAL APPLICATION Area = 650mm2, Height = 3mm VIN 7V TO 24V 5V BAT54A 2.2μF 6.3V BOOST1 BOOST2 5V 10Ω 1μF VCC C9 10μF 25V 1μF Q1A Si4816BDY VCC TRACK1 FCB DRVCC TG1 C16 100μF 6.3V L1 1.8μH 0.1μF BOOST2 Q1B Si4816BDY 20k 31.6k 100k SENSE2+ SENSE1– VIN VCC SENSE2– PGND1 VRNG1 PGND2 VRNG2 ION1 10k 10k 150pF 220pF VIN VCC 1000pF 20k 220pF 560pF 100k 5.11k 20k 10k 150pF 470pF 560pF 0.1μF 5V PGOOD ITH2 RUN/SS SGND SGND 20k 100k VFB2 PGOOD INTLPF ITH1 20k VOUT2 1.8V 5A C15 100μF 6.3V + 750k ION2 1000pF VFB1 TRACK2 EXTLPF C13 150μF 4V Q2B Si4816BDY BG2 BG1 1M 220pF L2 1.8μH 0.1μF SW2 SENSE1+ C1 150μF 4V BOOST2 LTC3708EUH SW1 + Q2A Si4816BDY TG2 BOOST1 VOUT1 2.5V 5A C11 10μF 25V 1μF 220pF 22nF 15k 3708 TA06 SGND VIN RUN/SS BAT54W C1, C13: SANYO 4TPE150MAZB C9, C11: TAIYO YUDEN TMK325BJ106KM C15, C16: TDK C3225X5R0J107M L1, L2: TOKO FDV0630-1R8M RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1778 Wide Operating Range, No RSENSE Step-Down Controller Single Channel, GN16 Package LTC3709 2-Phase, No RSENSE Step-Down Controller with Tracking/Sequencing Single Output, Remote Sensing LTC3728 Dual, 550kHz, 2-Phase Synchronous Step-Down Switching Regulator Fixed Frequency, Dual Output LTC3729 550kHz, PolyPhase®, High Efficiency, Synchronous Step-Down Switching Regulator Fixed Frequency, Single Output, Up to 12-Phase Operation LTC3731 3-Phase, 600kHz, Synchronous Buck Switching Regulator Controller 3-Phase, Single Output LTC3778 Wide Operating Range, No RSENSE Step-Down Controller Single Channel, Separate VON Programming PolyPhase is a registered trademark of Linear Technology Corporation 3708fb 32 Linear Technology Corporation LT 1207 REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2006