NCP1631 Interleaved, 2-Phase Power Factor Controller The NCP1631 integrates a dual MOSFET driver for interleaved PFC applications. Interleaving consists of paralleling two small stages in lieu of a bigger one, more difficult to design. This approach has several merits like the ease of implementation, the use of smaller components or a better distribution of the heating. Also, Interleaving extends the power range of Critical Conduction Mode that is an efficient and cost−effective technique (no need for low trr diodes). In addition, the NCP1631 drivers are 180° phase shift for a significantly reduced current ripple. Housed in a SOIC16 package, the circuit incorporates all the features necessary for building robust and compact interleaved PFC stages, with a minimum of external components. http://onsemi.com MARKING DIAGRAM A WL Y WW G General Features • Near−Unity Power Factor • Substantial 180° Phase Shift in All Conditions Including Transient • • • • • • • • • Phases Frequency Clamped Critical Conduction Mode (FCCrM) i.e., Fixed Frequency, Discontinuous Conduction Mode Operation with Critical Conduction Achievable in Most Stressful Conditions FCCrM Operation Optimizes the PFC Stage Efficiency Over the Load Range Out−of−phase Control for Low EMI and a Reduced rms Current in the Bulk Capacitor Frequency Fold−back at Low Power to Further Improve the Light Load Efficiency Accurate Zero Current Detection by Auxiliary Winding for Valley Turn On Fast Line / Load Transient Compensation High Drive Capability: −500 mA / +800 mA Signal to Indicate that the PFC is Ready for Operation (“pfcOK” Pin) VCC Range: from 10 V to 20 V Safety Features = Assembly Location = Wafer Lot = Year = Work Week = Pb−Free Package PIN ASSIGNMENT ZCD2 FB ZCD1 1 REF5V/pfcOK Rt DRV1 GND OSC Vcc Vcontrol FFOLD DRV2 BO Latch OVP / UVP CS (Top View) ORDERING INFORMATION Device Package Shipping† NCP1631DR2G SOIC−16 (Pb−Free) 2500 / Tape & Reel †For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging Specification Brochure, BRD8011/D. Typical Applications • Output Over and Under Voltage Protection • Brown−Out Detection with a 50−ms Delay to Help • • • • NCP1631G AWLYWWG SOIC−16 D SUFFIX CASE 751B • Computer Power Supplies • LCD / Plasma Flat Panels • All Off Line Appliances Requiring Power Factor Meet Hold−up Time Specifications Soft−Start for Smooth Start−up Operation Programmable Adjustment of the Maximum Power Over Current Limitation Detection of Inrush Currents Correction *For additional information on our Pb−Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting Techniques Reference Manual, SOLDERRM/D. © Semiconductor Components Industries, LLC, 2009 August, 2009 − Rev. 0 1 Publication Order Number: NCP1631/D NCP1631 Vin Vaux2 Vout Rbo1 Rovp1 Rzcd2 Rout1 1 2 Rout2 Ac lin e EMI Filter Rbo2 Cin OVPin Cosc Rovp2 Ccomp2 Rcomp1 Icoil1 Rt RFF Cbo2 Ccomp1 16 Rzcd1 15 3 14 4 13 5 12 6 11 7 8 10 L1 L2 Vaux2 Icoil2 D1 Vout pfcOK D2 M1 Vcc LOAD M2 9 Cbulk Rocp OVPin RCS Iin Figure 1. Typical Application Schematic Table 1. MAXIMUM RATINGS TABLE Symbol Rating Pin Value Unit VCC(MAX) Maximum Power Supply Voltage Continuous 11 −0.3, +20 V VMAX Maximum Input Voltage on Low Power Pins 1, 2, 3, 4, 6, 7, 8, 9, 10, 15, and 16 −0.3, +9.0 V 5 −0.3, VControl(clamp) (Note 1) V 550 145 mW °C/W −40 to +125 °C 150 °C −65 to +150 °C 300 °C 2 kV 200 V VControl(MAX) PD RqJ−A TJ VControl Pin Maximum Input Voltage Power Dissipation and Thermal Characteristics Maximum Power Dissipation @ TA = 70°C Thermal Resistance Junction−to−Air Operating Junction Temperature Range TJ(MAX) Maximum Junction Temperature TS(MAX) Storage Temperature Range TL(MAX) Lead Temperature (Soldering, 10s) ESD Capability, HBM model (Note 2) ESD Capability, Machine Model (Note 2) Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect device reliability. 1. “VControl(clamp)” is the pin5 clamp voltage. 2. This device(s) contains ESD protection and exceeds the following tests: Human Body Model 2000 V per JEDEC Standard JESD22−A114E Machine Model Method 200 V per JEDEC Standard JESD22−A115−A 3. This device contains latch−up protection and exceeds 100 mA per JEDEC Standard JESD78. http://onsemi.com 2 NCP1631 Table 2. TYPICAL ELECTRICAL CHARACTERISTICS TABLE (Conditions: VCC = 15 V, Vpin7 = 2 V, Vpin10 = 0 V, TJ from −25°C to +125°C, unless otherwise specified) Characteristics Test Conditions Symbol Min Typ Max Unit VCC increasing VCC decreasing VCC decreasing VCC(on) VCC(off) VCC(hyst) VCC(reset) 11 9.5 1.5 4.0 11.85 10 2.0 6.0 12.7 10.5 − 7.5 VCC = 9.4 V ICC(start) − 50 100 Fsw = 130 kHz (Note 4) VCC = 15 V, Vpin10 = 5 V VCC = 15 V, pin 7 grounded ICC1 ICC(latch) ICC(off) − – − 3.5 0.4 0.4 7.0 0.8 0.8 Pin 6 open IOSC(clamp) 31.5 35 38.5 mA Pin 6 grounded IOSC(CH1) 126 140 154 mA Ipin6 = 50 mA IOSC(CH2) 76.5 85 93.5 mA Pin 6 grounded IOSC(DISCH1) 94.5 105 115.5 mA Ipin6 = 50 mA IOSC(DISCH2) 45 50 55 mA Ipin6 = 50 mA, Vpin5 = 2.5 V VFF 0.9 1.0 1.3 V STARTUP AND SUPPLY CIRCUITS Supply Voltage Startup Threshold Minimum Operating Voltage Hysteresis VCC(on) – VCC(off) Internal Logic Reset V Startup current Supply Current Device Enabled/No output load on pin6 Current that discharges VCC in latch mode Current that discharges VCC in OFF mode mA mA OSCILLATOR AND FREQUENCY FOLDBACK Clamping Charging Current Charge Current with no frequency foldback Charge Current @ Ipin6 = 50 mA Maximum Discharge Current with no frequency foldback Discharge Current @ Ipin6 = 50 mA Voltage on pin 6 Oscillator Upper Threshold VOSC(high) − 5 − V Oscillator Lower Threshold VOSC(low) 3.6 4.0 4.4 V VOSC(swing) 0.93 0.98 1.03 V Ipin9 = 100 mA Ipin9 = 10 mA VCS(TH100) VCS(TH10) −20 −10 0 0 20 10 mV Tj = 25°C Tj = −25°C to 125°C IILIM1 IILIM2 202 194 210 210 226 226 mA Iin−rush 11 14 17 mA Ipin14 = 100 mA Ipin14 = −100 mA Ipin11 = 100 mA Ipin11 = −100 mA RSNK1 RSRC1 RSNK2 RSRC2 – – – – 7 15 7 15 15 25 15 25 VDRV1 = 10 V VDRV1 = 0 V VDRV2 = 10 V VDRV2 = 0 V ISNK1 ISRC1 ISNK1 ISRC1 − − − − 800 500 800 500 − − − − CDRV1 = 1 nF, VDRV1 = 1 to 10 V CDRV2 = 1 nF, VDRV2 = 1 to 10 V tr1 tr2 − − 40 40 − − Oscillator Swing (Note 5) CURRENT SENSE Current Sense Voltage Offset Current Sense Protection Threshold Threshold for In−rush Current Detection (Note 5) GATE DRIVE Drive Resistance DRV1 Sink DRV1 Source DRV2 Sink DRV2 Source Drive Current Capability (Note 5) DRV1 Sink DRV1 Source DRV2 Sink DRV2 Source Rise Time DRV1 DRV2 Ω mA ns 4. DRV1 and DRV2 pulsating at half this frequency, that is, 65 kHz. 5. Not tested. Guaranteed by design and characterization. http://onsemi.com 3 NCP1631 Table 2. TYPICAL ELECTRICAL CHARACTERISTICS TABLE (Conditions: VCC = 15 V, Vpin7 = 2 V, Vpin10 = 0 V, TJ from −25°C to +125°C, unless otherwise specified) Characteristics Test Conditions Symbol Min Typ Max CDRV1 = 1 nF, VDRV1 = 10 to 1 V CDRV2 = 1 nF, VDRV2 = 10 to 1 V tf1 tf2 – – 20 20 – – VREF 2.44 2.500 2.56 Unit GATE DRIVE Fall Time DRV1 DRV2 ns REGULATION BLOCK Feedback Voltage Reference Error Amplifier Source Current Capability @ Vpin2 = 2.4 V IEA(SRC) −20 Error Amplifier Sink Current Capability @ Vpin2 = 2.6 V IEA(SNK) +20 Error Amplifier Gain V mA GEA 110 200 290 mS IControl(boost) 184 230 276 mA Vpin2 = 2.5 V IFB(bias) −500 500 nA @ Vpin2 = 2.4 V @ Vpin2 = 2.6 V VControl(clamp) VControl(MIN) VControl(range) − − 2.7 3.6 0.6 3 − − 3.3 V @ Vpin2 = 2.6 V, Ipin6 = 90 mA @ Vpin2 = 2.4 V, Ipin6 = 90 mA VREGUL(MIN) VREGUL(Clamp) − − − 1.66 0.1 − V Ratio (Vout(low) Detect Threshold / VREF) (Note 5) FB falling Vout(low)/VREF 95.0 95.5 96.0 % Ratio (Vout(low) Detect Hysteresis / VREF) (Note 5) FB rising Hout(low)/VREF − − 0.5 % Vpin2 = 3 V DMIN − − 0 % Vpin7 = 1.1 V, Ipin3 = 50 mA Vpin7 = 1.1 V, Ipin3 = 200 mA (Note 5) Vpin7 = 2.2 V, Ipin3 = 100 mA (Note 5) Vpin7 = 2.2 V, Ipin3 = 400 mA (Note 5) ton1 ton2 ton3 ton4 14.5 1.10 4.00 0.35 19.5 1.35 5.00 0.41 22.5 1.60 6.00 0.48 ms VBO = Vpin7 = 1.1 V, Ipin3 = 50 mA VBO = Vpin7 = 1.1 V, Ipin3 = 200 mA VBO = Vpin7 = 2.2 V, Ipin3 = 50 mA VBO = Vpin7 = 2.2 V, Ipin3 = 200 mA VRt1 VRt2 VRt3 VRt4 1.071 1.071 2.169 2.169 1.096 1.096 2.196 2.196 1.121 1.121 2.223 2.223 V Not tested Vton(MAX) Pin 5 Source Current when (Vout(low) Detect) is activated Pin2 Bias Current Pin 5 Voltage: Internal VREGUL Voltage (measured on pin 6): SKIP MODE Duty Cycle RAMP CONTROL (valid for the two phases) Maximum DRV1 and DRV2 On−Time (FB pin grounded) Pin 3 voltage Maximum Vton Voltage Pin 3 Current Capability IRt(MAX) Pin 3 sourced current below which the controller is OFF Pin 3 Current Range 5 1 IRt(off) Not tested − V − 7 IRt(range) 20 0.40 0.20 mA mA 1000 mA 0.60 0.30 V ZERO VOLTAGE DETECTION CIRCUIT (valid for ZCD1 and ZCD2) ZCD Threshold Voltage VZCD increasing VZCD falling VZCD(TH),H VZCD(TH),L ZCD Hysteresis VZCD decreasing VZCD(HYS) 0.25 Ipin1 = 5.0 mA Ipin1 = −5.0 mA VZCD(high) VZCD(low) 10 −0.65 Input Clamp Voltage High State Low State 0.50 0.25 V V 4. DRV1 and DRV2 pulsating at half this frequency, that is, 65 kHz. 5. Not tested. Guaranteed by design and characterization. http://onsemi.com 4 NCP1631 Table 2. TYPICAL ELECTRICAL CHARACTERISTICS TABLE (Conditions: VCC = 15 V, Vpin7 = 2 V, Vpin10 = 0 V, TJ from −25°C to +125°C, unless otherwise specified) Characteristics Test Conditions Symbol Min Typ Max Unit Internal Input Capacitance (Note 5) CZCD − 10 − pF ZCD Watchdog Delay tZCD 80 200 320 ms VBO(TH) 0.97 1.00 1.03 V IBO 6 7 8 mA Brown−Out Blanking Time (Note 5) tBO(BLANK) 38 50 62 ms Brown−Out Monitoring Window (Note 5) tBO(window) 38 50 62 ms VBO(clamp) − 965 − mV IBO(clamp) 100 − − mA VBO(HYS) 10 35 60 mV IBO(PNP) 100 − − mA VBO(PNP) 0.35 0.70 0.90 V VOVP 2.425 2.500 2.575 V Ratio (VOVP / VREF) (Note 5) VOVP/VREF 99.2 99.7 100.2 % Ratio UVP Threshold over VREF VUVP/VREF 8 12 16 % Vpin8 = 2.5 V Vpin8 = 0.3 V IOVP(bias) −500 − 500 nA VLatch 2.375 2.500 2.625 V Vpin10 = 2.3 V ILatch(bias) −500 − 500 nA Pin 15 Voltage Low State Vpin7 = 0 V, Ipin15 = 250 mA VREF5V(low) − 60 120 mV Pin 15 Voltage High State Vpin7 = 0 V, Ipin15 = 5 mA VREF5V(high) 4.7 5.0 5.3 V IREF5V 5 10 − mA Thermal Shutdown Threshold TSHDN 130 140 150 °C Thermal Shutdown Hysteresis TSHDN(HYS) − 50 − °C ZERO VOLTAGE DETECTION CIRCUIT (valid for ZCD1 and ZCD2) BROWN−OUT DETECTION Brown−Out Comparator Threshold Brown−Out Current Source Pin 7 clamped voltage if VBO < VBO(TH) during tBO(BLANK) Ipin7 = −100 mA Current Capability of the BO Clamp Hysteresis VBO(TH) – VBO(clamp) Ipin7 = − 100 mA Current Capability of the BO pin Clamp PNP Transistor Pin BO voltage when clamped by the PNP Ipin7 = − 100 mA OVER AND UNDER VOLTAGE PROTECTIONS Over−Voltage Protection Threshold Pin 8 Bias Current LATCH INPUT Pin Latch Threshold for Shutdown Pin Latch Bias Current pfcOK / REF5V Current Capability THERMAL SHUTDOWN 4. DRV1 and DRV2 pulsating at half this frequency, that is, 65 kHz. 5. Not tested. Guaranteed by design and characterization. http://onsemi.com 5 NCP1631 Table 3. DETAILED PIN DESCRIPTION Pin Number Name Function 1 ZCD2 2 FB This pin receives a portion of the pre−converter output voltage. This information is used for the regulation and the “output low” detection (VOUTL) that drastically speed−up the loop response when the output voltage drops below 95.5% of the wished level. 3 RT The resistor placed between pin 3 and ground adjusts the maximum on−time of our system for both phases, and hence the maximum power that can be delivered by the PFC stage. 4 OSC Connect a capacitor to set the clamp frequency of the PFC stage. If wished, this frequency can be reduced in light load as a function of the resistor placed between pin 6 and ground (frequency fold−back). If the coil current cycle is longer than the selected switching period, the circuit delays the next cycle until the core is reset. Hence, the PFC stage can operate in Critical Conduction Mode in the most stressful conditions. 5 VControl The error amplifier output is available on this pin. The capacitor connected between this pin and ground adjusts the regulation loop bandwidth that is typically set below 20 Hz to achieve high Power Factor ratios. Pin5 is grounded when the circuit is off so that when it starts operation, the power increases slowly (soft−start). 6 Freq. Foldback Apply a resistor between pin 6 and ground to adjust the oscillator charge current. Clamped not to exceed 100 mA, this charge current is made proportional to the power level for a reduced switching frequency at light load and an optimum efficiency over the load range. 7 BO (Brown−out Protection) Apply an averaged portion of the input voltage to detect brown−out conditions when Vpin2 drops below 1 V. A 100−ms internal delay blanks short mains interruptions to help meet hold−up time requirements. When it detects a brown−out condition, the circuit stops pulsing and grounds the “pfcOK” pin to disable the downstream converter. Also an internal 7−mA current source is activated to offer a programmable hysteresis. The pin2 voltage is internally re−used for feed−forward. Ground pin 2 to disable the part. 8 OVP / UVP The circuit turns off when Vpin9 goes below 480 mV (UVP) and disables the drive as long as the pin voltage exceeds 2.5 V (OVP). 9 CS This pin monitors a negative voltage proportional to the coil current. This signal is sensed to limit the maximum coil current and protect the PFC stage in presence of in−rush currents. 10 Latch Apply a voltage higher than 2.5 V to latch−off the circuit. The device is reset by unplugging the PFC stage (practically when the circuit detects a brown−out detection) or by forcing the circuit VCC below VCCRST (4 V typically). Operation can then resume when the line is applied back. 11 DRV2 This is the gate drive pin for phase 2 of the interleaved PFC stage. The high current capability of the totem pole gate drive (+0.5/−0.8 A) makes it suitable to effectively drive high gate charge power MOSFETs. 12 VCC This pin is the positive supply of the IC. The circuit starts to operate when VCC exceeds 12 V and turns off when VCC goes below 10 V (typical values). After start−up, the operating range is 9.5 V up to 20 V. 13 GND Connect this pin to the pre−converter ground. 14 DRV1 This is the gate drive pin for phase 1 of the interleaved PFC stage. The high current capability of the totem pole gate drive (+0.5/−0.8 A) makes it suitable to effectively drive high gate charge power MOSFETs. 15 REF5V / pfcOK The pin15 voltage is high (5 V) when the PFC stage is in a normal, steady state situation and low otherwise. This signal serves to “inform” the downstream converter that the PFC stage is ready and that hence, it can start operation. 16 ZCD1 This is the zero current detection pin for phase 2 of the interleaved PFC stage. Apply the voltage from an auxiliary winding to detect the core reset of the inductor and the valley of the MOSFET drain source voltage This is the zero current detection pin for phase 1 of the interleaved PFC stage. Apply the voltage from an auxiliary winding to detect the core reset of the inductor and the valley of the MOSFET drain source voltage. http://onsemi.com 6 NCP1631 pfcOK Vout low detect VD D − + TS D Error Amplifier ±20 mA Vref + SK I P ( 0.6 V cl amp vo ltag e Fault management i s acti vated) 5R OC P STOP VREGUL OF F 4R 3V pfcOK OVP IFF FFOLD DT VBOcomp pfcOK IRt_low Generation of the charge current for the internal timing capacitors (max on−time setting for the two phases) DRV1 ICH On−time control for the two phases Brown−out detection with 50−ms delay Vton CLK1 In−rush Vpwm1 ZCD1 Zero current detection for phase 1 Stup DT OFF DRV1 − + Vovp = Vref In−rush OSC IFF pfcOK OFF pfcOK/ REF5V Current Sense Block (Building of ICS proportional to ICOIL) UVP OC P OVP − In−rush ICS > 210 mA ICS > 14 mA QZCD1 QZCD2 Vcc < Vcc(r eset) − + Latch Ou tpu t Buffer 1 S Lstup Q R BO_NOK Vcc_OK Vr ef VZCD2 DRV1 REF5V VZCD1 + 12% Vr ef Oscillator block with interleaving and frequency foldback VZCD1 All the RS latches are RESET dom inant OVLflag1 OVP Q Lpwm1 R STOP DRV2 Ou tpu t Buffer 2 Vcc Vpwm2 CLK2 DRV2 S Vpwm1 VZCD2 VDMG1 Q Lpwm2 R R L SHDN Q S Ics − ZCD2 VBO S STOP VDMG2 Zero current detection for phase 2 BO Vcc Vpwm2 CLK1 VBO BO_NOK VBOcomp CLK2 In−rush Vton processing circuitry DRV2 OF F In−rush SKIP OFF Generation of the oscillator charge current IFF as a function of VREGUL (frequency fold−back) IRt < 7 mA Rt Vcc(on) Vcc(off) UVLO Vcontrol Vcc Vcc_OK BO_NOK IRt_low OVLflag1 Regul Iref SHDN UVP − FB VD D SHDN CS + 0.955*Vref Vref Internal Thermal Shutdown 230 mA DRV1 DRV2 GND Figure 2. Functional Block Diagram http://onsemi.com 7 NCP1631 Detailed Operating Description The NCP1631 integrates a dual MOSFET driver for interleaved, 2−phase PFC applications. It drives the two branches in so−called Frequency Clamped Critical conduction Mode (FCCrM) where each phase operates in Critical conduction Mode (CrM) in the most stressful conditions and in Discontinuous Conduction Mode (DCM) otherwise, acting as a CrM controller with a frequency clamp (given by the oscillator). According to the conditions, the PFC stage actually jumps from DCM to CrM (and vice versa) with no discontinuity in operation and without degradation of the current shape. Furthermore, the circuit incorporates protection features for a rugged operation together with some special circuitry to lower the power consumed by the PFC stage in no−load conditions. More generally, the NCP1631 is ideal in systems where cost−effectiveness, reliability, low stand−by power and high power factor are the key parameters: A “pfcOK” signal. The circuit detects when the PFC stage is in steady state or if on the contrary, it is in a start−up or fault condition. In the first case, the “pfcOK” pin (pin15) is in high state and low otherwise. This signal is to disable the downstream converter unless the bulk capacitor is charged and no fault is detected. Finally, the downstream converter can be optimally designed for the narrow voltage provided by the PFC stage in normal operation. Safety Protections. The NCP1631 permanently monitors the input and output voltages, the input current and the die temperature to protect the system from possible over−stresses and make the PFC stage extremely robust and reliable. In addition to the aforementioned OVP protection, one can list: Maximum Current Limit: the circuit permanently senses the total input current and prevents it from exceeding the preset current limit, still maintaining the out−of−phase operation. In−rush Detection: the NCP1631 prevents the power switches turn on for the large in−rush currents sequence that occurs during the start−up phase. Fully Stable FCCrM and Out−Of−Phase Operation. Unlike master/slave controllers, the NCP1631 utilizes an interactive−phase approach where the two branches operate independently. Hence, the two phases necessarily operate in FCCrM, preventing risks of undesired dead−times or continuous conduction mode sequences. In addition, the circuit makes them interact so that they run out−of−phase. The NCP1631 unique interleaving technique substantially maintains the wished 180° phase shift between the 2 branches, in all conditions including start−up, fault or transient sequences. Under−Voltage Protection: this feature is mainly to prevent operation in case of a failure in the OVP monitoring network (e.g., bad connection). Brown−Out Detection: the circuit stops operating if the line magnitude is too low to protect the PFC stage from the excessive stress that could damage it in such conditions. Thermal Shutdown: the circuit stops pulsing when its junction temperature exceeds 150°C typically and resumes operation once it drops below about 100°C (50°C hysteresis). Optimized Efficiency Over The Full Power Range. The NCP1631 optimizes the efficiency of your PFC stage in the whole line/load range. Its clamp frequency is a major contributor at nominal load. For medium and light load, the clamp frequency linearly decays as a function of the power to maintain high efficiency levels even in very light load. The power threshold under which frequency reduces is programmed by the resistor placed between pin 6 and ground. To prevent any risk of regulation loss at no load, the circuit further skips cycles when the error amplifier reaches its low clamp level. NCP1631 Operating Modes The NCP1631 drives the two branches of the interleaved in FCCrM where each phase operates in Critical conduction Mode (CrM) in the most stressful conditions and in Discontinuous Conduction Mode (DCM) otherwise, acting as a CrM controller with a frequency clamp (given by the oscillator). According to the conditions, the PFC stage actually jumps from DCM to CrM (and vice versa) with no discontinuity in operation and without degradation of the current shape. The circuit can also transition within an ac line cycle so that: • CrM reduces the current stress around the sinusoid top. • DCM limits the frequency around the line zero crossing. This capability offers the best of each mode without the drawbacks. The way the circuit modulates the MOSFET on−time allows this facility. Fast Line / Load Transient Compensation. Characterized by the low bandwidth of their regulation loop, PFC stages exhibit large over and under−shoots when abrupt load or line transients occur (e.g. at start−up). The NCP1631 dramatically narrows the output voltage range. First, the controller dedicates one pin to set an accurate Over−Voltage Protection level and interrupts the power delivery as long as the output voltage exceeds this threshold. Also, the NCP1631 dynamic response enhancer drastically speeds−up the regulation loop when the output voltage is 4.5% below its desired level. As a matter of fact, a PFC stage provides the downstream converter with a very narrow voltage range. http://onsemi.com 8 NCP1631 Figure 3. DCM and CRM Operation Within a Sinusoid Cycle for One Branch NCP1631 On−time Modulation The NCP1631 operates in voltage mode. As portrayed by Figure 6, the MOSFET on time t1 is controlled by the signal Vton generated by the regulation block as follows: Let’s study the ac line current absorbed by one phase of the interleaved PFC converter. The current waveform of the inductor (L) during one switching period (Tsw) is portrayed by Figure 5. The ac line current is the averaged value of the coil current as the result of the EMI filter “polishing” action. Hence, the line current produced by one of the phase is: ǒ Ǔǒt T) t ǓV t Iin + 1 1 2 L 1 2 sw in t1 + (eq. 1) ǒt (tT) t )Ǔ 1 1 Equation 1 shows that Iin is proportional to Vin if t1(t 1 ) t2) is a constant. Tsw Forcing 2 sw Ǔ to be a constant for proper power factor correction can be changed into: ǒt (tT) t )Ǔ 1 1 (eq. 2) Where: • Ct is the internal timing capacitor • It is the internal current source for the timing capacitor. The It charge current is constant for a given resistor placed on the Rt pin. Ct is also a constant. Hence, the condition Where (Tsw = t1 + t2 + t3) is the switching period and Vin is the ac line rectified voltage. ǒ C tVTON It ǒ 2 sw constant is what the NCP1631 does to perform FCCrM operation that is, to operate in discontinuous or critical conduction mode according to the conditions, without degradation of the power factor. Ǔ is constant. VTON(t 1 ) t2) T sw The output of the regulation block (VCONTROL) is linearly changed into a signal (VREGUL) varying between 0 and 1.66 V. (VREGUL) is the voltage that is injected into the PWM section to modulate the MOSFET duty−cycle. However, the NCP1631 inserts some circuitry that processes (VREGUL) to form the signal (VTON) that is used in the PWM section instead of (VREGUL) (see Figure 7). (VTON) is modulated in response to the dead−time sensed during the precedent current cycles, that is, for a proper shaping of the ac line current. This modulation leads to: VTON + Figure 4. Boost Converter T swVREGUL t1 ) t2 or: VTON t1 ) t2 + VREGUL (eq. 3) Tsw Substitution of Equation 3 into Equation 2 leads to the following on−time expression: ǒ Ct t1 + Ǔ TswVREGUL t1)t2 It (eq. 4) Replacing “t1” by its expression of Equation 4, Equation 1 simplifies as follows: Iin(phase1) + I in(phase2) + Figure 5. Inductor Current in DCM http://onsemi.com 9 V in C tVREGUL 2L It (eq. 5) NCP1631 Given the regulation low bandwidth of the PFC systems, (VCONTROL) and then (VREGUL) are slow varying signals. Hence, the line current absorbed by each phase is: Iin(phase1) + I in(phase2) + k V in where: k + constant + ƪ ƫ From this equation, we can check that if Vpin7 (BO voltage) is 1 V and Rt is 20 kW (Ipin3 = 50 mA) that the on−time is 20 ms as given by parameter Ton1. Since: (eq. 6) VREGUL(max) + 1.66 V C tVREGUL 2 L It Ton + Hence, the input current is then proportional to the input voltage and the ac line current is properly shaped. One can note that this analysis is also valid for CrM operation that is just a particular case of this functioning where (t3=0), which leads to (t1+t2=Tsw) and (VTON=VREGUL). That is why the NCP1631 automatically adapts to the conditions and jumps from DCM and CrM (and vice versa) without power factor degradation and without discontinuity in the power delivery. The charging current It is internally processed to be proportional to the square of the line magnitude. Its value is however programmed by the pin 3 resistor to adjust the available on−time as defined by the Ton1 to Ton4 parameters of the data sheet. From these data, we can deduce: Rt 2 t1 + T on(ms) + 50 n Vpin7 2 C tVREGUL It Vpin7 + 2 Ǹ2 Vin(rms) k BO p where kBO is the scale down factor of the BO sensing network ǒ k BO + Ǔ R bo2 R bo1 ) R bo2 (see Brown−out section) We can deduce the total input current value and the average input power: Iin(rms) ^ (R t)2V REGUL 26.9 @ 10 12 L k BO 2V in,rms (eq. 8) (Rt)2V REGUL 26.9 @ 10 12 L k BO 2 (eq. 9) Pin,avg ^ (eq. 7) timing capacitor s aw −too th PWM comparator + to PWM latch − VREGUL R1 + OA1 Vton SKIP − C1 S3 OC P 0.5* (I se nse − 210 m) S1 IN 1 S2 −> V ton d u ring (t1+t2) −> 0 V d u ring t3 (d e a d −time) −> V ton *(t1+t2)/T in average OV P OF F VBOcomp (from BO block) pfcOK DT (high during dead−time) In−rus h The integrator OA1 amplifies the error between VREGUL and IN1 so that in average, (VTON*(t1+t2)/Tsw) equates VREGUL. Figure 6. PWM Circuit and Timing Diagram Figure 7. VTON Processing Circuit The “VTON processing circuit” is “informed” when there is an OVP condition or a skip sequence, not to over−dimension VTON in that conditions. Otherwise, an OVP sequence or a skipped cycle would be viewed as a “normal” dead−time phase by the circuit and VTON would inappropriately increase to compensate it. (Refer to Figure 7). The output of the “VTON processing circuit” is also grounded when the circuit is in OFF state to discharge the capacitor C1 and initialize it for the next active phase. Finally, the “VTON” is not allowed to be further increased compared to VREGUL when the circuit has not completed the start−up phase (pfcOK low) and if VBOcomp from the brown−out block is high (refer to brown−out section for more information). http://onsemi.com 10 350,00 3,50 300,00 3,00 250,00 2,50 200,00 2,00 150,00 1,50 100,00 1,00 Vin ton 50,00 0,00 Ton (ms) Vin (V) NCP1631 0 2 4 6 8 10 time (ms) 12 14 16 0,50 18 20 0,00 Figure 8. Input Voltage and On−time vs. Time (example with FSW = 100 kHz, Pin = 150 W, VAC = 230 V, L = 200 mH) Regulation Block and Low Output Voltage Detection The swing of the error amplifier output is limited within an accurate range: • It is forced above a voltage drop (VF) by the “low clamp” circuitry. When this circuitry is activated, the power demand is minimum and the NCP1631 enters skip mode (the controller stops pulsating) until the clamp is no more active. • It is clamped not to exceed 3.0 V + the same VF voltage drop. A trans−conductance error amplifier with access to the inverting input and output is provided. It features a typical trans−conductance gain of 200 mS and a typical capability of ±20 mA. The output voltage of the PFC stage is typically scaled down by a resistors divider and monitored by the inverting input (feed−back pin – pin2). The bias current is minimized (less than 500 nA) to allow the use of a high impedance feed−back network. The output of the error amplifier is pinned out for external loop compensation (pin5). Typically a type−2 compensator is applied between pin5 and ground, to set the regulation bandwidth below 20 Hz, as need in PFC applications (refer to application note AND8407). FB 230 m A VDD − 0.955*Vref pfcOK + Vout low detect Hence, Vpin5 features a 3 V voltage swing. Vpin5 is then offset down by (VF) and further divided before it connects to the “Vton processing block” and the PWM section. Finally, the output of the regulation is a signal (“VREGUL” of the block diagram) that varies between 0 and 1.66 V. E rr o r Am p lifier ±20 m A − V control + Vref OVLflag1 OFF 5R SKIP (0 .6 V c lamp vo ltage is activated) VREGUL 3V 4R Figure 9. Regulation Block Figure 10. Correspondence Between VCONTROL and VREGUL http://onsemi.com 11 NCP1631 Zero Current Detection Provided the low bandwidth of the regulation loop, sharp variations of the load, may result in excessive over and under−shoots. Over−shoots are limited by the Over− Voltage Protection (see OVP section). To contain the under−shoots, an internal comparator monitors the feed−back signal (Vpin2) and when Vpin2 is lower than 95.5% of its nominal value, it connects a 230 mA current source to speed−up the charge of the compensation capacitor (Cpin5). Finally, it is like if the comparator multiplied the error amplifier gain by 10. One must note that this circuitry for under−shoots limitation, is not enabled during the start−up sequence of the PFC stage but only once the converter has stabilized (that is when the “pfcOK” signal of the block diagram, is high). This is because, at the beginning of operation, the pin5 capacitor must charge slowly and gradually for a soft start−up. While the on time is constant, the core reset time varies with the instantaneous input voltage. The NCP1631 determines the demagnetization completion by sensing the inductor voltage, more specifically, by detecting when the inductor voltage drops to zero. Practically, an auxiliary winding in flyback configuration is taken off of the boost inductor and gives a scaled down version of the inductor voltage that is usable by the controller (Figure 12). In that way, the ZCD voltage (“VAUX”) falls and starts to ring around zero volts when the inductor current drops to zero. The NCP1631 detects this falling edge and allows the next driver on time. Figure 1 shows how it is implemented. For each phase, a comparator detects when the voltage of the ZCD winding exceeds 0.5 V. When this is the case, the coil is in demagnetization phase and the latch LZCD is set. This latch is reset when the next driver pulse occurs. Rzcd2 1 Rzcd1 ZCD2 16 ZCD1 Vin VDMG1 Negative and positive clamp Negative and positive clamp + − 0.5 V AND1 S LZCD Qzcd1 Q S QZCD R CLK1 R (from phase In−rush management block) 200−ms delay S OFF (from Fault management VDMG2 block) + − 0.5 V PWM latch PH1 Vzcd1 SET1 Q S R Q Q L1 DRV1 14 M1 output buffe r 1 reset signal (from PH1 PWM comparator) DT R Vzcd2 Vcc D1 D2 Vout Vin L2 SET2 PWM latch PH2 output buffe r 2 Vcc Qzcd2 S Q CLK2 R (from phase management In−rush reset signal block) (from PH2 PWM comparator) DRV2 11 M2 Cbulk Cbulk Figure 11. Zero Current Detection To prevent negative voltages on the ZCD pins (ZCD1 for phase 1 and ZCD2 for phase 2), these pins are internally clamped to about 0 V when the voltage applied by the corresponding ZCD winding is negative. Similarly, the ZCD pins are clamped to VZCD(high) (10 V typical), when the ZCD voltage rises too high. Because of these clamps, a resistor (RZCD of Figure 11) is necessary to limit the current from the ZCD winding to the ZCD pin. The clamps are designed to respectively source and sink 5 mA minimum. It is recommended not to exceed this 5 mA level within the ZCD clamps for a proper operation. At startup or after an inactive period (because of a protection that has tripped for instance), there is no energy in the ZCD winding and therefore no voltage signal to activate the ZCD comparator. This means that the driver will never turn on. To avoid this, an internal watchdog timer is integrated into the controller. If the driver remains low for more than 200 ms (typical), the timer sets the LZCD latch as the ZCD winding signal would do. Obviously, this 200−ms delay acts as a minimum off−time if there is no demagnetization winding while it has no action if there is a ZCD voltage provided by the auxiliary winding. http://onsemi.com 12 NCP1631 Figure 12. Zero Current Detection Timing Diagram (VAUX is the Voltage Provided by the ZCD Winding) Current Sense not to sink more than 5 mA from the CS pin for a proper operation. Two functions use ICS: the over current protection and the in−rush current detection. The NCP1631 is designed to monitor a negative voltage proportional to total input current, i.e., the current drawn by the two interleaved branches (Iin). As portrayed by Figure 13, a current sense resistor (RCS) is practically inserted within the return path to generate a negative voltage (VCS) proportional to Iin. The circuit uses VCS to detect when Iin exceeds its maximum permissible level. To do so, the circuit incorporates an operational amplifier that sources the current necessary to maintain the CS pin voltage null (refer to Figure 13). By inserting a resistor ROCP between the CS pin and RCS, we adjust the current that is sourced by the CS pin (ICS) as follows: * [R CSICOIL] ) [ROCPI CS] + 0 Over−Current Protection (OCP) If ICS exceeds IILIM1 (210 mA typical), an over−current is detected and the on−time is decreased proportionally to the difference between the sensed current IIN and the 210 mA OCP threshold. The on−time reduction is done by injecting a current Ineg in the negative input of the “VTON processing circuit” OPAMP. (See Figure 7) Ineg + 0.5(I CS * 210 m) (eq. 10) This current is injected each time the OCP signal is high. Which leads to: ICS + RCS I R OCP COIL (eq. 12) The maximum coil current is: (eq. 11) ICOIL(max) + In other words, the pin 9 current (ICS) is proportional to the coil current. A negative clamp protects the circuit from the possible negative voltage that can be applied to the pin. This protection is permanently active (even if the circuit off). The clamp is designed to sustain 5 mA. It is recommended R OCP I RCS ILIM1 (eq. 13) In−rush Current Detection When the PFC stage is plugged to the mains, the bulk capacitor is abruptly charged to the line voltage. The charge current (named in−rush current) can be very huge http://onsemi.com 13 NCP1631 MOSFETs from the possible excessive stress it could suffer from if it was allowed to turn on while a huge current flowed through the coil as it can be the case at start−up or during an over−load transient. The propagation delay (ICS < Iin−rush) to (drive outputs high) is in the range of few ms. However when the circuit starts to operate, the NCP1631 disables this protection to avoid that the current produced by one phase and sensed by the circuit prevents the other branch from operating. Practically, some logic grounds the In−rush protection output when it detects the presence of current cycles with a zero current detection signal provided by the auxiliary winding (Figure 13). EMI Filter VIN Vaux2 IIN Vaux1 Curr e nt Mirror CIN The pin voltage is maintained to 0 V CS 9 ICS ICS ICS OC P (ICS is proportional to the coil current) ICS In−rush Iin−rush = 14 mA Negative clamp M2 (from ZCD block) QZCD1 QZCD2 ROCP RCS L2 L1 IILIM1 = 210 mA ICS D1 VOUT D2 M1 DRV 2 DRV 1 DRV 1 DRV 2 CBULK LOAD Ac lin e depending on the presence or absence of an effective in−rush limiting circuitry. If the MOSFET turns on during this severe transient, it may be over−stressed and finally damaged. That is why, the NCP1631 permanently monitors the input current and delays the MOSFET turn on until the in−rush current has vanished. This is the function of the ICS comparison to the Iin−rush threshold (14 mA typical). When ICS exceeds Iin−rush, the comparator output (“In−rush”) is high and prevents the PWM latches from setting (see block diagram). Hence, the two drivers cannot turn high and the MOSFETs cannot switch on. This is to guarantee that the MOSFETs remain off as long as if the input current exceeds 10% of its maximum value. This feature protects the IIN The CS block performs the over−current protection and the in−rush current detection. Figure 13. Current Sense Block Over−Voltage Protection While PFC circuits often use one single pin for both the Over−Voltage Protection (OVP) and the feed−back, the NCP1631 dedicates one specific pin for the under−voltage and over−voltage protections. The NCP1631 configuration allows the implementation of two separate feed−back networks (see Figure 15): 1. One for regulation applied to pin 2. 2. Another one for the OVP function (pin 8). Vout (bulk voltage) Vout (bulk voltage) Rout1 FB 1 2 3 Rout3 Rout2 OVP FB 15 14 4 13 5 12 6 11 7 10 8 Rout1 16 Rovp1 Rout2 Rovp2 OVP 9 1 2 16 3 14 4 13 5 12 6 11 7 10 8 Figure 14. Configuration with One Feed−back Network for Both OVP and Regulation Figure 15. Configuration with Two Separate Feed−back Networks http://onsemi.com 14 15 9 NCP1631 should be used to allow operation of the downstream converter. The double feed−back configuration offers some up−graded safety level as it protects the PFC stage even if there is a failure of one of the two feed−back arrangements. However, if wished, one single feed−back arrangement is possible as portrayed by Figure 14. The regulation and OVP blocks having the same reference voltage, the resistance ratio Rout2 over Rout3 adjusts the OVP threshold. More specifically, Oscillator Section – Phase Management The oscillator generates the clock signal that dictates the maximum switching frequency for the global system (fosc). In other words, each of the two interleaved branches cannot operate above the clamp frequency that is half the oscillator frequency (fosc/2). The oscillator frequency (fosc) is adjusted by the capacitor applied to pin 4. Typically, a 440 pF capacitor approximately leads to a 120−kHz operating frequency, meaning a 60−kHz clamp frequency for each branch. The oscillator frequency should be kept below 500 kHz (which corresponds to a pin4 capacitor in the range of 100 pF). As shown by Figure 16, two current sources IOSC(clamp) (35 mA typical) and IOSC(CH) (105 mA typical) charge the pin 4 capacitor until its voltage exceeds VOSC(high) (5 V typically). At that moment, the output of the COMP_OSC comparator (“SYNC” of Figure 16) turns high and changes the COMP_OSC reference threshold that drops from VOSC(high) down to VOSC(low) (hysteresis). The system enters a discharge phase where the ICH current source is disabled and instead a sink current IOSC(DISCH) (105 mA typ.) discharges the pin 4 capacitor. This sequence lasts until Vpin4 goes below VOSC(low) when the “SYNC” signal turns low and a new charging phase starts. A divider by two uses the “SYNC” information to manage the phases of the interleaved PFC: the first SYNC pulse sets “phase 1”, the second one, “phase 2”, the third one phase 1 again... etc... According to the selected phase, the “SYNC” signal sets the relevant “Clock generator latch” that will generate the clock signal (“CLK1” for phase 1, “CLK2” for phase 2) when SYNC drops to zero (falling edge detector). So, the drivers are synchronized to SYNC falling edge. Actually, the drivers cannot turn on at this very moment if the demagnetization of the coil is not yet complete (CrM operation). In this case, the clock signal is maintained high until the driver turns high (the clock generator latches are reset by the corresponding driver is high − reset on rising edge detector). Also, the discharge time can be prolonged if when Vpin4 drops below VOSC(low), the driver of the phase cannot turn on because the core is not reset yet (CrM operation). In this case, Vpin4 decreases until the driver turns high. The further discharge of Vpin4 below VOSC(low) helps maintain a substantial 180° phase shift in CrM that is in essence, guaranteed in DCM. In the two conditions (CrM or DCM), operation is stable and robust. Figure 17 portrays the clock signal waveforms in different cases: − In fixed frequency operation (DCM), the cycle time of the coil current is shorter than an oscillator period. Hence, as soon as the clock signal goes high, the driver can turn on and reset the clock generator latch. The clock signal is then a short pulse. The bulk regulation voltage (“Vout(nom)”) is: Vout(nom) + R out1 ) R out2 ) R out3 @ V ref R out2 ) R out3 (eq. 14) The OVP level (“Vout(ovp)”) is: Vout(ovp) + R out1 ) R out2 ) R out3 @ V ref R out2 (eq. 15) The ratio OVP level over regulation level is: Vout(ovp) Vout(nom) +1) Rout3 Rout2 (eq. 16) For instance, (Vout(nom) = 105% x Vout(nom)) leads to: (Rout3 = 5% x Rout2). When the circuit detects that the output voltage exceeds the OVP level, it maintains the power switch open to stop the power delivery. As mentioned previously, the “VTON processing circuit” is “informed” when there is an OVP condition, not to over−dimension VTON in that conditions. Otherwise, an OVP sequence would be viewed as a dead−time phase by the circuit and VTON would inappropriately increase to compensate it (refer to Figure 7). PfcOK / REF5V Signal The NCP1631 can communicate with the downstream converter. The signal “pfcOK/REF5V” is high (5 V) when the PFC stage is in normal operation (its output voltage is stabilized at the nominal level) and low otherwise. More specifically, “pfcOK/REF5V” is low: − During the PFC stage start−up, that is, as long as the output voltage has not yet stabilized at the right level. The start−up phase is detected by the latch “LSTUP” of the block diagram in Figure 2. “LSTUP” is set during each “off” phase so that its output (“STUP“) is high when the circuit enters an active phase. The latch is reset when the error amplifier stops charging its output capacitor, that is, when the output voltage of the PFC stage has reached its desired regulation level. At that moment, “STUP” falls down to indicate the end of the start−up phase. − Any time, the circuit is off or a fault condition is detected as described by the “Fault management and OFF mode” section Finally, “pfcOK/REF5V” is high when the PFC output voltage is properly and safely regulated. “pfcOK/REF5V” http://onsemi.com 15 NCP1631 current has totally vanished. In other words, critical conduction mode (CrM) operation is obtained. The clamp frequency can be computed using the following equation: − However, the coil current can possibly be non zero when the clock signal turns high. The circuit would enter Continuous Conduction Mode (CCM) if the MOSFET turned on in that moment. In order to avoid CCM operation, the clock is prevented from setting the PWM latch until the core is reset (that is as low as “VZCD” of Figure 8 is low). The clock signal remains high during this waiting phase (refer to Figure 12). Hence the next MOSFET conduction time occurs as soon as the coil FFOLD (eq. 17) where COSC is the pin 4 external capacitor and Cpin the pin 4 parasitic capacitance (about 10 pF). 105 mA Current Mirror IFF 60 m C OSC ) 10 p fosc ^ VREGUL RFF VREGUL IFF pfcOK Circuitry for Frequency Foldback SYNCbar IOSC(CH) = IFF R CLK1 Ge nera tion latch IOSC(clamp) Co mp _OSC OS C CL K1 SYNCbar DRV 1 S SYNC Q Q_ph1 P h ase1 COSC VOSC(high)/ VOSC(low) DRV 2 divider by tw o P hase2 IOSC(DISCH) = IFF Q_ph2 Q_ph2 R Q_ph1 CLK2 Ge nera tion latch S Q SYNCbar Figure 16. Oscillator Block http://onsemi.com 16 CL K2 NCP1631 Figure 17. Typical Waveforms (Tdelay not shown here for the sake of simplicity) Frequency Foldback “VREGUL” is the signal derived from Vcontrol that is effectively used to modulate the MOSFET on−time. VREGUL is buffered and applied to pin 6 (“Frequency fold−back” pin). A resistor RFF is to be connected to pin 6 to sink a current proportional to VREGUL In addition, the circuit features the frequency fold−back function to improve the light load efficiency. Practically, the oscillator charge and discharge currents (IOSC(CH) and IOSC(DISCH) of Figure 16) are not constant but dependent on the power level. More specifically, IOSC(CH) and IOSC(DISCH) linearly vary as a function of Vcontrol output of the regulation block that thanks to the feed−forward featured by the NCP1631, is representative of the load. The practical implementation is portrayed by Figure 16. ǒ Ipin6 + I FF + Ǔ V REGUL . R FF This current is clamped not to exceed 105 mA and copied by a current mirror to form IOSC(CH) and IOSC(DISCH). As a matter of fact, the oscillator charge current is: IOSC(CH) + I OSC(clamp) ) V REGUL R FF if ǒ Ǔ VREGUL v 105 mA RFF IOSC(CH) + I OSC(clamp) ) I OSC(CH1) + I OSC(CHT1) + 140 mA The oscillator charge current is then an increasing function of VREGUL and is clamped to 140 mA. The oscillator discharge current is: IOSC(DISCH) + V REGUL R FF IOSC(DISCH) + I OSC(DISCH1) + 105 mA if ǒ (eq. 18) otherwise Ǔ VREGUL v 105 mA RFF (eq. 19) otherwise value for (IFF = 105 mA). If we consider the clamp frequency fOSC computed by Equation 17 as the nominal value obtained at full load and if we name it “fOSC(nom)”: The oscillator discharge current is also an increasing function of VREGUL and is clamped to105 mA. As a consequence, the clamp frequency is also an increasing function of VREGUL until it reaches a maximum http://onsemi.com 17 NCP1631 if ǒVREGUL w R FF @ 105 mAǓ fOSC + f OSC(nom) fOSC + VREGUL(R FFIOSC(clamp) ) VREGUL) 60 m R FF(RFFI OSC(clamp) ) 2V REGUL) Let’s illustrate this operation on an example. VREGUL is the control signal that varies between 0 and 1.66 V, (VREGUL = 1.66 V) corresponding to the maximum power (Pin)HL that can virtually be delivered by the PFC stage as selected by the timing resistor (for more details, you can refer to the application note AND8407). @ f OSC(nom) if ǒVREGUL v R FF @ 105 mAǓ (eq. 20) If one decides to start to reduce the clamp frequency when the power goes below (Pin)HL/2, the oscillator charge current should start to decrease when VREGUL is 0.83 V. Hence, the pin 6 resistor (“RFF”) must be selected so that pin 6 sources 105 mA when VREGUL equates 0.83 V: RFF + 0.83 V + 7.9 kW 105 mA (eq. 21) Let’s take (RFF = 8.2 kW) which is a normalized value. This selection leads to: if ǒVREGUL w 8.2 k @ 105 m + 860 mVǓ fOSC + f OSC(nom) fOSC + V REGUL(RFFI OSC(clamp) ) V REGUL) 492 m(RFFI OSC(clamp) ) 2V REGUL) @ f OSC(nom) if ǒVREGUL v 860 mVǓ (eq. 22) For instance, if the nominal frequency (fOSC(nom)) is 120 kHz, the following characteristic is obtained. 150 100 Fosc (kHz) fOSC(nom) = 120 kHz 50 0 0 0.5 1 1.5 VREGUL (V) Figure 18. Fold−back Characteristic of the Clamp Frequency with RFF = 8.2 kW and fOSC(nom) = 120 kHz If pin6 is grounded (accidently or not), the circuit operates properly with a constant 140 mA oscillator charge current and a 105 mA discharge current. The clamp frequency equates its nominal value over the whole load range. If pin6 is open, the oscillator charge current is equal to IOSC(clamp) but the oscillator discharge current is null and hence the PFC stage cannot operate. A minimum discharge current and hence a minimum clamp frequency can be forced by placing a resistor between pin 4 and ground. For instance, a 1.5−MW resistor forces a 3.3−mA discharge current when the oscillator capacitor is fully charged and about 2.6 mA when it is near the oscillator low threshold (4 V). A transistor pulls the pin 6 down during startup to disable the frequency fold−back function. Skip Mode The circuit features the frequency fold−back that leads to a very efficient stand−by mode. In order to ensure a proper regulation in no load conditions even if this feature is not used (pin 6 grounded), the circuit skips cycles when the error amplifier output is at its minimum level. The error amplifier output is maintained between about 0.6 V and 3.6 V thanks to active clamps. A skip sequence occurs as long as the 0.6 V clamp circuitry is triggered and switching operation is recovered when the clamp is inactive. http://onsemi.com 18 NCP1631 Brown−Out Protection The brown−out pin receives a portion of the input voltage (VIN). As VIN is a rectified sinusoid, a capacitor must integrate the ac line ripple so that a voltage proportional to the average value of (VIN) is applied to the brown−out pin. IRt_ low IRt < 7 mA Current M irror Rt IRt VBO RRt Vin VBO IRt Feed−forward circuitry Ac line EM I Filter Cin RCS Rbo1 Cbo 1V BO Rbo2 S LBO Q R s2 Vdd This PNP transistor maintains the BO pin below the BO threshold when the circuit is not fed enough to control the state of the BO block VBOcomp s1 7 mA 980 mV Clamp BO_NOK Tdelay 50-ms delay reset 50-ms delay res et This voltage (“VBOcomp”) is high when Vpin7 is below 1 V Ci r c uitr y fo r brown−out detection Figure 19. Brown−out Block The main function of the BO block is to detect too low input voltage conditions. A 7−mA current source lowers the BO pin voltage when a brown−out condition is detected. This is for hysteresis purpose as required by this function. In nominal operation, the voltage applied to pin7 must be higher than the 1 V internal voltage reference. In this case, the output of the comparator BO_Comp (VBOcomp) is low (see Figure 19). Conversely, if Vpin7 goes below 1 V, the BO_Comp output turns high and a 965 mV voltage source is connected to the BO pin to maintain the pin level near 1 V. Then, a 50−ms blanking delay is activated during which no fault is detected. The main goal of the 50−ms lag is to help meet the hold−up requirements. In case of a short mains interruption, no fault is detected and hence, the “pfcOK” signal remains high and does not disable the downstream converter. In addition, pin7 being kept at 965 mV, there is almost no extra delay between the line recovery and the occurrence of a proper voltage applied to pin2, that otherwise would exist because of the large capacitor typically placed between pin7 and ground to filter the input voltage ripple. As a result, the NCP1631 effectively “blanks” any mains interruption that is shorter than 25 ms (minimum guaranteed value of the 50−ms timer). At the end of this 50−ms blanking delay, another timer is activated that sets a 50−ms window during which a fault can be detected. This is the role of the second 50−ms timer of Figure 19: • if the output of OPAMP is high at the end of the first delay (50−ms blanking time) and before the second 50−ms delay time is elapsed, a brownout condition is detected • if the output of OPAMP remains low for the duration of the second delay, no fault is detected. When the “BO_NOK” signal is high: − The drivers are disabled, the “Vcontrol” pin is grounded to recover operation with a soft−start when the fault has gone and the “pfcOK” voltage turns low to disable the downstream converter. − The OPAMP output is separated from pin7 (Figure 19) to prevent the operational amplifier from maintaining 1 V on pin7 (as done by the switches s1 and s2 in the representation of Figure 19). Instead, Vpin2 drops to the value that is externally forced (by Vin, Rbo1, Rbo2 and Cbo2 in Figure 19). As a consequence, the OPAMP output remains high and the “BO_NOK” signal stays high until the line recovers. http://onsemi.com 19 NCP1631 then enabled once the temperature drops below about 80°C (60°C hysteresis). The temperature shutdown keeps active as long as the circuit is not reset, that is, as long as VCC keeps higher than VCCRESET. The reset action forces the TSD threshold to be the upper one (140°C). This ensures that any cold start−up will be done with the right TSD level. − The 7−mA current source is enabled that lowers the pin7 voltage for hysteresis purpose. A short delay (Tdelay) is added to get sure that these three actions are properly done before the PFC driver is disabled and the “Vcontrol” and “pfcOK” pins are grounded. At startup, a pnp transistor ensures that the BO pin voltage remains below the 1 V threshold until VCC reaches VCC(on). This is to guarantee that the circuit starts operation in the right state, that is, “BONOK” high. When VCC exceeds VCC(on), the pnp transistor turns off and the circuit enables the 7−mA current source (IBO). Also, (IBO) is enabled whenever the part is in off−mode, but at startup, IBO is disabled until VCC reaches VCC(on). Under−Voltage Lockout Section The NCP1631 incorporates an Under−Voltage Lockout block to prevent the circuit from operating when the power supply is not high enough to ensure a proper operation. An UVLO comparator monitors the pin 12 voltage (VCC) to allow the NCP1631 operation when VCC exceeds 12 V typically. The comparator incorporates some hysteresis (2.0 V typically) to prevent erratic operation as the VCC crosses the threshold. When VCC goes below the UVLO comparator lower threshold, the circuit turns off. The circuit off state consumption is very low: < 50 mA. This low consumption enables to use resistors to charge the VCC capacitor during the start−up without the penalty of a too high dissipation. Brown−out Resistors Calculation The BO resistors can be calculated with the following equations (for more details, refer to the application note AND8407) (V in,avg)boH Rbo1 + Rbo2 + ȡ *ȧ(V Ȣ ȡ f10 ȣȣ ȧ1 * 3f ȧȧ Ȣ ȤȤ line in,avg) boL line IHYST ǒ Rbo1 (Vin,avg)boL VBO(th) ǒ 1* ǓǓ fBO 3fline (eq. 23) Output Drive Section The circuit embeds two drivers to control the two interleaved branches. Each output stage contains a totem pole optimized to minimize the cross conduction current during high frequency operation. The gate drive is kept in a sinking mode whenever the Under−Voltage Lockout (UVLO) is active or more generally whenever the circuit is off. Its high current capability (−500 mA/+800 mA) allows it to effectively drive high gate charge power MOSFET. (eq. 24) *1 Feed−forward As shown by Figure 19, The BO circuit also generates an internal current proportional to the input voltage average value (IRt). The pin7 voltage is buffered and made available on pin 3. Placing a resistor between pin 3 and ground, enables to adjust a current proportional to the average input voltage. This current (IRt) is internally copied and squared to form the charge current for the timing capacitor of each phase. Since this current is proportional to the square of the line magnitude, the conduction time is made inversely proportional to the line magnitude. This feed−forward feature makes the transfer function and the power delivery independent of the ac line level. Only the regulation output (VREGUL) controls the power amount. If the IRt current is too low ( below 7 mA), the controller goes in OFF mode to avoid damaging the MOSFETs with too long conduction time. Reference Section The circuit features an accurate internal reference voltage (VREF). VREF is optimized to be ±2.4% accurate over the temperature range (the typical value is 2.5 V). VREF is the voltage reference used for the regulation and the over−voltage protection. The circuit also incorporates a precise current reference (IREF) that allows the Over−Current Limitation to feature a ±6% accuracy over the temperature range. Fault Management and OFF Mode The circuit detects a fault if the Rt pin is open (Figure 20). Practically, if the pin sources less than 7 mA, the “IRt_Low” signal sets a latch that turns off the circuit if its output (Rt(open)) is high. A 30−ms blanking time avoids parasitic fault detections. The latch is reset when the circuit is in UVLO state (too low VCC levels for proper operation). Thermal Shutdown (TSD) An internal thermal circuitry disables the circuit gate drive and then keeps the power switch off when the junction temperature exceeds 140°C typically. The output stage is http://onsemi.com 20 NCP1631 Internal Thermal Shutdown Vref VDD TSD Vcc Regul Iref Stdwn UVP BO_NOK Vcc_OK UVLO 12 V / 10 V Rt(open) Q R S 30−ms blanking time IRt_Low (Ipin3 < 7 mA) OF F Fault management Figure 20. Fault Management Block When any of the following faults is detected: − brown−out (“BO_NOK”) − Under−Voltage Protection (“UVP”) − Latch−off condition (“Stdwn”) − Die over−temperature (“TSD”) − Too low current sourced by the Rt pin (“Rt(open)”) − “UVLO” (improper Vcc level for operation) The circuit turns off. In this mode, the controller stops operating. The major part of the circuit sleeps and its consumption is minimized (< 500 mA). More specifically, when the circuit is in OFF state: − The two drive outputs are kept low − The 7−mA current source of the brown−out block is enabled to set the proper start−up BO threshold. − The pin5 capacitor (Vcontrol) is discharged and kept grounded along the OFF time, to initialize it for the next operating sequence, where it must be slowly and gradually charged to offer some soft−start. − The “pfcOK” pin is grounded. − The output of the “VTON processing block” is grounded When the circuit recovers after a fault, the first watchdog time is around 20 ms instead of 200 ms to allow a faster re−start. In OFF mode at startup, the consumption is very low (< 50 mA). The brown−out block is initialized not to allow operation (“BO_NOK” high) by default. The PNP clamp is active and maintains the BO pin level below 1 V. The 7−mA current source is enabled only when VCC reaches VCC(on) threshold. http://onsemi.com 21 NCP1631 Figure 21. Start−up and Brown Out Conditions http://onsemi.com 22 NCP1631 PACKAGE DIMENSIONS SOIC−16 CASE 751B−05 ISSUE K −A− 16 9 −B− P 1 8 PL 0.25 (0.010) 8 B M S G R K F X 45 _ C −T− SEATING PLANE J M D NOTES: 1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. 2. CONTROLLING DIMENSION: MILLIMETER. 3. DIMENSIONS A AND B DO NOT INCLUDE MOLD PROTRUSION. 4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER SIDE. 5. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.127 (0.005) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. MILLIMETERS INCHES DIM MIN MAX MIN MAX A 9.80 10.00 0.386 0.393 B 3.80 4.00 0.150 0.157 C 1.35 1.75 0.054 0.068 D 0.35 0.49 0.014 0.019 F 0.40 1.25 0.016 0.049 G 1.27 BSC 0.050 BSC J 0.19 0.25 0.008 0.009 K 0.10 0.25 0.004 0.009 M 0_ 7_ 0_ 7_ P 5.80 6.20 0.229 0.244 R 0.25 0.50 0.010 0.019 16 PL 0.25 (0.010) M T B S A S SOLDERING FOOTPRINT 8X 6.40 16X 1 1.12 16 16X 0.58 1.27 PITCH 8 9 DIMENSIONS: MILLIMETERS ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. 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