PM6675AS High efficiency step-down controller with embedded 2 A LDO regulator Features ■ ■ Switching section – 4.5 V to 36 V input voltage range – 0.6 V, ±1 % voltage reference – Selectable 1.5 V fixed output voltage – Adjustable 0.6 V to 3.3 V output voltage – 1.237 V ±1 % reference voltage available – Very fast load transient response using constant-on-time control loop – No RSENSE current sensing using low side MOSFETs' RDS(ON) – Negative current limit – Latched OVP and UVP – Soft-start internally fixed at 3 ms – Selectable pulse skipping at light load – Selectable No-audible (33 kHz) pulse skip mode – Ceramic output capacitors supported – Output voltage ripple compensation – Output soft-end LDO regulator section – Adjustable 0.6 V to 3.3 V output voltage – Selectable ±1 Apk or ±2 Apk current limit – Dedicated power-good signal – Ceramic output capacitors supported – Output soft-end Applications ■ Industrial application on 24 V ■ Graphic cards VFQFPN-24 4x4 Description The PM6675AS device consists of a single high efficiency step-down controller and an independent low drop-out (LDO) linear regulator. The constant on-time (COT) architecture assures fast transient response supporting both electrolytic and ceramic output capacitors. An embedded integrator control loop compensates the DC voltage error due to the output ripple. Selectable low-consumption mode allows the highest efficiency over a wide range of load conditions. The low-noise mode sets the minimum switching frequency to 33 kHz for audio-sensitive applications. The LDO linear regulator can sink and source up to 2 Apk. Two fixed current limit (±1 A- ±2 A) can be chosen. An active soft-end is independently performed on both the switching and the linear regulators outputs when disabled. ■ Embedded computer systems Table 1. Device summary Order codes Package PM6675AS Packaging Tube VFQFPN-24 4x4 (exposed pad) PM6675ASTR February 2008 Tape and reel Rev 1 1/48 www.st.com Contents PM6675AS Contents 1 Typical application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 2 Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 3 2.1 Connections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.2 Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Electrical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 3.1 Maximum rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 3.2 Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 3.3 Recommended operating conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 4 Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9 5 Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 6 Typical operating characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 7 Device description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 7.1 2/48 Switching section - constant on-time pwm controller . . . . . . . . . . . . . . . . 16 7.1.1 Constant-on-time architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 7.1.2 Output ripple compensation and loop stability . . . . . . . . . . . . . . . . . . . . 19 7.1.3 Pulse-skip and no-audible pulse-skip modes . . . . . . . . . . . . . . . . . . . . . 24 7.1.4 Mode-of-operation selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26 7.1.5 Current sensing and current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27 7.1.6 POR, UVLO and soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28 7.1.7 Switching section power-good signal . . . . . . . . . . . . . . . . . . . . . . . . . . . 29 7.1.8 Switching section output discharge . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29 7.1.9 Gate drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 7.1.10 Reference voltage and bandgap . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 7.1.11 Switching section OV and UV protections . . . . . . . . . . . . . . . . . . . . . . . 30 PM6675AS Contents 7.1.12 7.2 8 Device thermal protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 LDO linear regulator section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31 7.2.1 LDO section current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32 7.2.2 LDO section soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 7.2.3 LDO section power-good signal . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 7.2.4 LDO section output discharge . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33 Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 8.1 External components selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 8.1.1 Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35 8.1.2 Input capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37 8.1.3 Output capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38 8.1.4 MOSFETs selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39 8.1.5 Diode selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41 8.1.6 VOUT current limit setting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41 8.1.7 All ceramic capacitors application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42 9 Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45 10 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47 3/48 Typical application circuit PM6675AS 1 Typical application circuit Figure 1. Application circuit +5V R LP VIN C IN3 C IN2 R1 C IN R2 23 LDO PG VLDO 8 22 VOSC VCC 18 B O O HGATE 21 T PHASE LIN 4 LPG 24 6 AVCC C IN4 10 VSEL LILIM VLDOIN 12 NOSKIP 3 CSNS 2 LFB 5 15 14 13 7 COMP VREF SWEN LEN LGND SPG PGND SGND 1 VSNS 11 C OUT2 SMPS PG 4/48 L 20 VSMPS LGATE 17 PM6675AS PM6675A LOUT C BOOT C BYP C OUT 19 16 R LIM 9 C INT PM6675AS Pin settings 2 Pin settings 2.1 Connections CSNS PHASE HGATE BOOT LIN LOUT Pin connection (through top view) 19 24 1 18 LGND VCC LFB LGATE NOSKIP PGND PM6675AS PM6675A LPG SPG LEN SGND 6 13 SWEN LILIM COMP VSEL 12 VSNS 7 VOSC AVCC VREF Figure 2. 5/48 Pin settings 2.2 PM6675AS Pin description Table 2. 6/48 Pin functions N° Pin Function 1 LGND 2 LFB 3 NOSKIP 4 LPG 5 SGND Ground Reference for analog circuitry, control logic and VTTREF buffer. Connect together with the thermal pad and VTTGND to a low impedance ground plane. See the Application Note for details. 6 AVCC +5 V supply for internal logic. Connect to +5 V rail through a simple RC filtering network. 7 VREF High accuracy output voltage reference (1.237 V) for multilevel pins setting. It can deliver up to 50 uA. Connect a 100 nF capacitor between VREF and SGND in order to enhance noise rejection. 8 VOSC Frequency Selection. Connect to the central tap of a resistor divider to set the desired switching frequency. The pin cannot be left floating. See Section 7: Device description for details. 9 VSNS Switching section output remote sensing and discharge path during output soft-end. Connect as close as possible to the load via a low noise PCB trace. 10 VSEL Fixed output selector and feedback input for the switching controller. If VSEL pin voltage is higher than 4 V, the fixed 1.5 V output is selected. If VSEL pin voltage is lower than 4 V, it is used as negative input of the error amplifier. See Section 7.1.4: Mode-of-operation selection for details. 11 COMP DC voltage error compensation input pin for the switching section. Refer to Mode of Operation Selection section for more details. 12 LILIM Current limit selector for the LDO. Connect to SGND for ±1 A current limit or to +5 V for ±2 A current limit. 13 SWEN Switching Controller Enable. When tied to ground, the switching output is turned off and a soft-end is performed. 14 LEN Linear Regulator Enable. When tied to ground, the LDO output is turned off and a soft-end is performed. 15 SPG Switching Section power-good signal (open drain output). High when the switching regulator output voltage is within ±10 % of nominal value. 16 PGND Power ground for the switching section. 17 LGATE Low-side gate driver output. 18 VCC LDO power ground. Connect to negative terminal of VTT output capacitor. LDO remote sensing. Connect as close as possible to the load via a low noise PCB trace. Pulse-skip/no-audible pulse-skip modes selector. See Section 7.1.4: Mode-of-operation selection LDO section power-good signal (open drain output). High when LDO output voltage is within ±10 % of nominal value. +5 V low-side gate driver supply. Bypass with a 100 nF capacitor to PGND. PM6675AS Pin settings Table 2. Pin functions (continued) N° Pin Function 19 CSNS Current sense input for the switching section. This pin must be connected through a resistor to the drain of the synchronous rectifier (RDSon sensing) to set the current limit threshold. 20 PHASE Switch node connection and return path for the high side gate driver. 21 HGATE High-Side Gate Driver Output 22 BOOT Bootstrap capacitor connection. Input for the supply voltage of the high-side gate driver. 23 LIN 24 LOUT Linear Regulator Input. Bypass to LGND by a 10 µF ceramic capacitor for noise rejection enhancement. LDO linear regulator output. Bypass with a 20 µF (2x10 µF MLCC) filter capacitor. 7/48 Electrical data PM6675AS 3 Electrical data 3.1 Maximum rating Table 3. Absolute maximum ratings (1) Symbol Parameter Value VAVCC AVCC to SGND -0.3 to 6 VVCC VCC to SGND -0.3 to 6 PGND, LGND to SGND VPHASE PTOT Unit -0.3 to 0.3 HGATE and BOOT to PHASE -0.3 to 6 HGATE and BOOT to PGND -0.3 to 44 PHASE to SGND -0.3 to 38 LGATE to PGND -0.3 to VVCC +0.3 V CSNS, SPG, LEN, SWEN, LILIM, COMP, VSEL, VSNS, VOSC, VREF, NOSKIP to SGND -0.3 to VAVCC + 0.3 LPG,VREF, LOUT, LFB to SGND -0.3 to VAVCC + 0.3 LIN, LOUT, LPG, LIN to LGND -0.3 to VAVCC + 0.3 Power dissipation @TA = 25°C 2.3 W 1. Free air operating conditions unless otherwise specified. Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. Exposure to absolute maximum rated conditions for extended periods may affect device reliability. 3.2 Thermal data Table 4. Thermal data Symbol 3.3 Parameter Unit 42 °C/W RthJA Thermal resistance junction to ambient TSTG Storage temperature range -50 to 150 TA Operating ambient temperature range -40 to 85 TJ Junction operating temperature range -40 to 125 °C Recommended operating conditions Table 5. Recommended operating conditions Symbol VIN 8/48 Value Parameter Min Typ Max Input voltage range 4.5 36 VAVCC IC supply voltage 4.5 5.5 VVCC IC supply voltage 4.5 5.5 Unit V PM6675AS Electrical characteristics 4 Electrical characteristics Table 6. Electrical characteristics TA = - 25 °C to 85 °C , VCC = AVCC = +5 V, LIN = 1.5 V and LOUT= 0.6 V if not otherwise specified (1). Symbol Parameter Test condition Min Typ Max Unit Supply section Operating current (Switching + LDO) SWEN, LEN, VSEL and NOSKIP connected to AVCC, No load on LOUT output. 2 ISW Operating current (switching) SWEN, VSEL and NOSKIP connected to AVCC, LEN coneected to SGND. 1 ISHDN Shutdown operating current SWEN and LEN tied to SGND. 10 IIN AVCC Under Voltage Lockout upper threshold mA 4.1 4.25 µA 4.4 V UVLO AVCC Under Voltage Lockout lower threshold 3.85 UVLO hysteresis 4.0 4.1 70 mV On-time (SMPS) tON On-time duration VSEL low, NOSKIP low, VOSC = 300 mV 530 630 730 VVSNS = 2 V VOSC = 500 mV 320 380 440 300 350 ns 1.237 1.249 V ns OFF-TIME (SMPS) tOFFMIN Minimum Off-Time Voltage reference Voltage accuracy 4.5 V< VIN < 36 V Load regulation -50 µA < IVREF < 50 µA 1.224 -4 4 mV Undervoltage Lockout Fault Threshold 800 SMPS output VOUT SMPS fixed output voltage Feedback output voltage accuracy 1.5 VSEL connected to AVCC, NOSKIP tied to SGND, No Load -1.5 V 1.5 % 9/48 Electrical characteristics Table 6. PM6675AS Electrical characteristics (continued) TA = -25 °C to 85 °C , VCC = AVCC = +5 V, LIN = 1.5 V and LOUT= 0.6 V if not otherwise specified. (1) Symbol Parameter Test condition Min Typ Max Unit CSNS input bias current 90 100 110 µA Comparator offset -6 Current limit and zero crossing comparator ICSNS Positive current limit threshold VPGND - VCSNS 100 mV fixed negative current limit threshold VZC,OFFS 6 110 Zero crossing comparator offset -11 -5 1 HGATE high state (pullup) 2.0 3 HGATE low state (pulldown) 1.8 2.7 LGATE high state (pullup) 1.4 2.1 LGATE low state (pulldown) 0.6 0.9 High and low side gate drivers HGATE driver on-resistance Ω LGATE driver on-resistance UVP/OVP protections and PGOOD signals OVP Over voltage threshold 112 115 118 UVP Under voltage threshold 67 70 73 SMPS upper threshold 107 110 113 SMPS lower threshold 86 90 93 LDO upper threshold 107 110 113 LDO lower threshold 86 90 93 % PGOOD IPG,LEAK SPG and LPG leakeage current SPG and LPG forced to 5.5 V VPG,LOW SPG and LPG low level voltage ILPG,SINK = ISPG,SINK = 4 mA 1 µA 150 250 mV 3 4 ms Soft-start section (SMPS) Soft-start ramp time (4 steps current limit) 2 Soft-start current limit step µA 25 Soft end section Switching section discharge resistance 15 25 35 LDO section discharge resistance 15 25 35 Ω LDO section LDO reference voltage VLREF 10/48 LDO output accuracy respect to VREF 600 -1 mA < ILDO < 1 mA -20 20 -1 A < ILDO < 1 A -25 25 mV PM6675AS Table 6. Electrical characteristics Electrical characteristics (continued) TA = -25 °C to 85 °C , VCC = AVCC = +5 V, LIN = 1.5 V and LOUT= 0.6 V if not otherwise specified. (1) Symbol Parameter LDO sink current limit ILDO,CL LDO source current limit ILIN,BIAS Test condition Min Typ Max VLFB > VLREF , LILIM = 5 V -3 -2.3 -2 VLFB > VLREF, LILIM = 0 V -1.6 -1.3 -1 0.9 ⋅ VLREF < VLFB < VLREF, LILIM=5V 2 2.4 3 0.9 ⋅ VLREF < VLFB < VLREF, LILIM = 0 V 1 1.3 1.6 VLFB < 0.9 ⋅ VLREF, LILIM = 5 V 1 1.3 1.6 VLFB < 0.9 ⋅ VLREF, LILIM = 0 V 0.5 0.8 1.1 1 10 LDO input bias current, on LEN connected to AVCC, no load LDO input bias current, off LEN = 0 V, no load ILFB,BIAS LFB input bias current ILFB,LEAK LFB leakage current LEN connected to AVCC VLFB = 0.6 V LEN=0V, VLFB = 0.6V Unit A 1 µA -1 1 -1 1 Power management section Fixed mode VVTHVSEL VAVCC -0.7 VSEL pin thresholds VAVCC -1.3 Adjustable mode Forced-PWM mode VVTHNOSKIP NOSKIP pin thresholds No-audible mode VAVCC -0.8 VAVCC -1.5 1.0 Pulse-skip mode VVTHLEN, VVTHSWEN LEN, SWEN turn off level VVTHLILIM LILIM pin thresholds V 0.5 0.4 LEN, SWEN turn on level 1.6 ±2A LDO current limit VAVCC -0.8 ±1A LDO current limit 0.5 IIN,LEAK Logic input leakage current LEN, SWEN and LILIM = 5 V 10 IIN3,LEAK Multilevel input leakage current VSEL and NOSKIP = 5 V 10 IOSC,LEAK VOSC pin leakage current VOSC = 1 V 1 µA Thermal shutdown TSHDN Shutdown temperature (2) 150 °C 1. Specifications referred to TJ = TA. All the parameters at operating temperatures extremes are guaranteed by design and statistical correlation (not production tested). 2. Guaranteed by design. Not production tested. 11/48 Block diagram PM6675AS 5 Block diagram Figure 3. Functional and block diagram VREF VOSC Vr = 0.6V 1.236V Bandgap BOOT LFB Level shifter Ton HGATE 1-shot LIN PHASE Ton min 1-shot _ LOUT Anti Cross Conduction VCC Toff min 1-shot LILIM LGATE + PGND LDS LEN 0.6V Zero Crossing & Current Limit LGND _ Vr +10% LPG VREF COMP + + - Vr +10% SWEN + _ gm + Vr -10% SGND AVCC UVP/OVP SWEN UVLO Vr LDS LDS CSNS + - Vr SPG + Vr -10% LEN VSNS LILIM NOSKIP CONTROL LOGIC Thermal Shutdown LEN Table 7. SWEN 12/48 SWEN SDS adj VSEL Legend Switching controller enable LEN LDO regulator enable LDS LDO output discharge enable SDS Switching output discharge enable LILIM LDO regulator current limit fix PM6675AS Typical operating characteristics 6 Typical operating characteristics Figure 4. Efficiency vs output load FSW = 330 kHz VOUT=1.5 V, VIN = 24 V VOUT - Efficiency Figure 5. Switching frequency vs output current, VOUT = 1.5 V, VIN = 24 V SW Frequency VS VOUT Load Forced PWM Pulse Skip Non Audible PS 100 500 90 400 70 Frequency [kHz] Efficiency [%] 80 60 50 40 30 20 300 Forced PWM Pulse Skip 200 Non Audible PS 100 10 0 0.001 0.010 0.100 1.000 0 0.010 10.000 Current [A] 0.100 1.000 10.000 Current [A] Figure 6. Switching frequency vs input Figure 7. voltage, VOUT = 1.5 V, IVOUT = 2 A, forced PWM mode VOUT load regulation, VIN = 24 V VOUT - Load Regulation SW Frequency VS Input Voltage 1.540 550 Forced PWM 1.535 Pulse Skip 450 Voltage [V] Frequency [kHz] 1.530 350 Non Audible PS 1.525 1.520 1.515 1.510 1.505 1.500 0.001 250 4 14 24 34 Figure 8. 0.010 0.100 1.000 10.000 Current [A] Voltage [V] LOUT load regulation LDOIN = VOUT, VOUT in forced PWM mode Figure 9. VOUT = 1.5 V, VIN = 24 V, IVOUT = 0 A, pulse-skip mode LOUT - Load Regulation 1.090 1.080 Voltage [V] 1.070 1.060 1.050 1.040 1.030 1.020 -1.500 -1.000 -0.500 0.000 0.500 1.000 1.500 Current [A] 13/48 Typical operating characteristics PM6675AS Figure 10. VOUT = 1.5V , VIN = 24V, IVOUT = 0 A, forced-PWM mode Figure 11. VOUT = 1.5 V, VIN = 24 V, no load, Non-audible pulse-skip mode (33 kHz) Figure 12. VOUT Soft-start @150mΩ load, pulse-skip mode Figure 13. LOUT turn on, VOUT in pulse-skip mode Figure 14. VOUT Load Transient (VIN = 24 V, LOAD = 0 A -> 7 A @2.5 A/µs). pulse-skip mode Figure 15. LOUT load transient (VIN = 24 V, LOAD = -1.5 A -> 1.5 A @2.5 A/µs). pulse-skip mode 14/48 PM6675AS Typical operating characteristics Figure 16. VOUT and LOUT output voltages. VOUT soft-end. LOUT powered by an auxiliary rail Figure 17. VOUT and LOUT output voltages LOUT soft-end Figure 18. UV protection, pulse-skip mode LOUT powered by an auxiliary rail Figure 19. OV protection, pulse-skip mode Figure 20. VOUT current limit protection during a load transient (0 A to 9 A @2.5A/µs) Figure 21. LOUT current limit during an output short 15/48 Device description 7 PM6675AS Device description The PM6675AS combines a single high efficiency step-down controller and an independent Low Drop-Out (LDO) linear regulator in the same package. The switching controller section is a high-performance, pseudo-fixed frequency, ConstantOn-Time (COT) based regulator specifically designed for handling fast load transient over a wide range of input voltage. The switching section output can be easily set to a fixed 1.5 V voltage without additional components or adjusted in the 0.6 V to 3.3 V range using an external resistor divider. The Switching Mode Power Supply (SMPS) can handle different modes of operation in order to minimize noise or power consumption, depending on the application needs. Selectable lowconsumption and low-noise modes allow the highest efficiency and a 33 kHz minimum switching frequency respectively at light loads. The current sensing is lossless, based on the Low-Side MOSFET turn-on resistance. The input of the LDO can be either the switching section output or a lower voltage rail in order to reduce the total power dissipation. Linear regulator stability is achieved by filtering its output with a ceramic capacitor (20 µF or greater). The LDO linear regulator can sink and source up to 2 Apk. Two fixed current limit (±1A-±2A) can be chosen. An active soft-end is independently performed on both the switching and the linear regulators outputs when disabled. 7.1 Switching section - constant on-time pwm controller The PM6675AS employes a pseudo-fixed frequency, Constant On-Time (COT) controller as the core of the switching section. As well known, the COT controller concerns of a relatively simple algorithm and uses the ripple voltage derived across the output capacitor ESR to trigger the On-Time one-shot generator. In this way, the output capacitor ESR acts as a current sense resistor providing the appropriate ramp signal to the PWM comparator. Nearly constant switching frequency is achieved by the system loop in steady-state operating conditions by varying the On-Time duration, avoiding thus the need for a clock generator. The On-Time one shot duration is directly proportional to the output voltage, sensed at VSNS pin, and inversely proportional to the input voltage, sensed at the VOSC pin, as follows: Equation 1 TON = K OSC VSNS +τ VOSC where KOSC is a constant value (130 ns typ.) and τ is the internal propagation delay (40ns typ.). The one-shot generator directly drives the high-side MOSFET at the beginning of each switching cycle allowing the inductor current to increase; after the On-Time has expired, an Off-Time phase, in which the low-side MOSFET is turned on, follows. 16/48 PM6675AS Device description The Off-Time duration is solely determined by the output voltage: when lower than the set value (i.e. the voltage at VSNS pin is lower than the internal reference = 0.6 V), the synchronous rectifier is turned off and a new cycle begins (Figure 22). Figure 22. Inductor current and output voltage in steady state conditions Inductor current Output voltage Vreg Ton t Toff The duty-cycle of the buck converter is, in steady-state conditions, given by Equation 2 D= VOUT VIN The switching frequency is thus calculated as Equation 3 fSW D = = TON VOUT α VIN 1 = OSC ⋅ V α OUT K OSC K OSC SNS VOSC where Equation 4 a α OSC = VOSC VIN α OUT = VSNS VOUT Equation 4 b 17/48 Device description PM6675AS Referring to the typical application schematic (fig. 1 and 23), the final expression is then: Figure 23. Switching frequency selection and VOSC pin VIN PM6675AS R1 VOSC R2 Equation 5 fSW = α OSC R2 1 = ⋅ K OSC R1 + R 2 K OSC Even if the switching frequency is theoretically independent from input and output voltages, parasitic parameters involved in power path (like MOSFET on-resistance and inductor DCR) introduce voltage drops responsible of a slight dependence on load current. In addition, the internal delay is cause of a light dependence from input voltage. The PM6675AS switching frequency can be set by an external divider connected to the VOSC pin. The voltage seen at this pin must be greater than 0.8 V and lower than 2 V in order to ensure system's linearity. 7.1.1 Constant-on-time architecture Figure 24 shows the simplified block diagram of the Constant-On-Time controller. The switching regulator of the PM6675AS owns a one-shot generator that turns on the highside MOSFET when the following conditions are simultaneously satisfied: the PWM comparator is high (i.e. output voltage is lower than Vr = 0.6 V), the synchronous rectifier current is below the current limit threshold and the minimum off-time has expired. A minimum off-time constrain (300 ns typ.) is introduced to assure the boot capacitor charge and allow inductor valley current sensing on low-side MOSFET. A minimum On-Time is also introduced to assure the start-up switching sequence. Once the on-time has timed out, the high side switch is turned off, while the synchronous rectifier is ignited according to the anti-cross conduction management circuitry. When the output voltage reaches the valley limit (determined by internal reference Vr=0.6 V), the low-side MOSFET is turned off according to the anti-cross conduction logic once again, and a new cycle begins. 18/48 PM6675AS Device description Figure 24. Switching section simplified block diagram VOSC Positive Current Limit comparator CSNS Toff-min + Level shifter - 100uA S Q PWM Comparator Integrator R Q gm Anti crossconduction circuitry Ton-min 0.6V + + - + 2.5V 500mV VCC VSEL<4V VSEL + - T on 2.5V S VSNS S Q VOSC LS Q VSNS Zero-crossing Comparator R + _ Q LGATE PGND R Min fsw counter 0.6V VBG bandgap 1.236V PULSE - SKIP SGND 7.1.2 HGATE PHASE - COMP HS 1-Shot generator + VBG BOOT VOSC VREF Output ripple compensation and loop stability The loop is closed connecting the center tap of the output divider (internally, when the fixed output voltage is chosen, or externally, using the VSEL pin in the adjustable output voltage mode). The feedback node is the negative input of the error comparator, while the positive input is internally connected to the reference voltage (Vr = 0.6 V). When the feedback voltage becomes lower than the reference voltage, the PWM comparator goes high and sets the control logic, turning on the high-side MOSFET. After the On-Time (calculated as previously described) the system releases the high-side MOSFET and turns-on the synchronous rectifier. The voltage drop along ground and supply PCB paths, used to connect the output capacitor to the load, is a source of DC error. Further the system regulates the output voltage valley, not the average, as shown in Figure 22. Thus, the voltage ripple on the output capacitor is an additional source of DC error. To compensate this error, an integrative network is introduced in the control loop, by connecting the output voltage to the COMP pin through a capacitor (CINT) as shown in Figure 25. 19/48 Device description PM6675AS Figure 25. Circuitry for output ripple compensation COMP PIN VOLTAGE ∆V Vr t COMP OUTPUT VOLTAGE VREF I=gm(V1-Vr) - PWM Comparator CFILT ∆V gm CINT VCINT RINT t + Vr + RFb1 V1 RFb2 ESR VSNS COUT The additional capacitor is used to reduce the voltage on the COMP pin when higher than 300 mVpp and is unnecessary for most of applications. The transconductance amplifier (gm) generates a current, proportional to the DC error, used to charge the CINT capacitor. The voltage across the CINT capacitor feeds the negative input of the PWM comparator, forcing the loop to compensate the total static error. An internal voltage clamp forces the COMP pin voltage range to ±150 mV respect to VREF. This is useful to avoid or smooth output voltage overshoot during a load transient. When the Pulse-Skip Mode is entered, the clamping range is automatically reduced to 60 mV in order to enhance the recovering capability. In case the ripple amplitude is larger than 150 mV, an additional capacitor CFILT can be connected between the COMP pin and ground to reduce ripple amplitude, otherwise the integrator will operate out of its linearity range. This capacitor is unnecessary for most of applications and can be omitted. The design of the external feedback network depends on the output voltage ripple. If the ripple is higher than approximately 20 mV, the correct CINT capacitor is usually enough to keep the loop stable. The stability of the system depends firstly on the output capacitor zero frequency. The following condition must be satisfied: Equation 6 fSW > k × fZout = 20/48 k 2π × C out × ESR PM6675AS Device description where k is a fixed design parameter (k > 3). It determinates the minimum integrator capacitor value: Equation 7 CINT > gm Vr ⎛f ⎞ Vout 2π ⋅ ⎜ SW − fZout ⎟ k ⎝ ⎠ ⋅ where gm = 50 µs is the integrator transconductance. If the ripple on the COMP pin is greater than the integrator 150 mV, the auxiliary capacitor CFILT can be added. If q is the desired attenuation factor of the output ripple, CFILT is given by: Equation 8 CFILT = CINT ⋅ (1 − q) q In order to reduce the noise on the COMP pin, it is possible to add a resistor RINT that, together with CINT and CFILT, realizes a low pass filter. The cutoff frequency fCUT must be greater (10 or more times) than the switching frequency: Equation 9 RINT = 2π ⋅ fCUT 1 CINT ⋅ CFILT ⋅ CINT + CFILT If the ripple is very small (lower than approximately 20 mV), a different compensation network, called "Virtual-ESR" Network, is needed. This additional circuit generates a triangular ripple that is added to the output voltage ripple at the input of the integrator. The complete control scheme is shown in Figure 26. 21/48 Device description PM6675AS Figure 26. "Virtual-ESR" network COMP PIN VOLTAGE T NODE VOLTAGE ΔV2 VREF ΔV1 t t COMP RINT CINT T gm Vr C VSNS OUTPUT VOLTAGE + - CFILT R1 R VREF I=gm(V1-Vr) PWM Comparator + RFb1 V1 RFb2 ESR ΔV COUT t The ripple on the COMP pin is the sum of the output voltage ripple and the triangular ripple generated by the Virtual-ESR Network. In fact the Virtual-ESR Network behaves like a further equivalent series resistor RVESR. A good trade-off is to design the network in order to achieve an RVESR given by: Equation 10 R VESR = VRIPPLE − ESR ∆IL where ∆IL is the inductor current ripple and VRIPPLE is the total ripple at the T node, chosen greater than approximately 20 mV. The new closed-loop gain depends on CINT. In order to ensure stability it must be verified that: Equation 11 CINT > gm Vr ⋅ 2π ⋅ fZ Vout where: Equation 12 fZ = 22/48 1 2π ⋅ C out ⋅ R TOT PM6675AS Device description and Equation 13 R TOT = ESR + R VERS Moreover, the CINT capacitor must meet the following condition: Equation 14 fSW > k ⋅ fZ = k 2π ⋅ C out ⋅ R TOT where RTOT is the sum of the ESR of the output capacitor and the equivalent ESR given by the Virtual-ESR Network (RVESR). The k parameter must be greater than unity (k > 3) and determines the minimum integrator capacitor value CINT: Equation 15 CINT > gm Vr ⋅ ⎛ fSW ⎞ Vout 2π ⋅ ⎜ − fZ ⎟ ⎝ k ⎠ The capacitor of the Virtual-ESR Network, C, is chosen as follow Equation 16 C > 5 ⋅ CINT and R is calculated to provide the desired triangular ripple voltage: Equation 17 R= L R VESR ⋅ C Finally, the R1 resistor can be selected according to expression 18: Equation 18 ⎛ 1 ⎞ ⎟ R ⋅ ⎜⎜ π ⋅ fZ ⋅ C ⎟⎠ ⎝ R1 = 1 R− π ⋅ fZ ⋅ C 23/48 Device description 7.1.3 PM6675AS Pulse-skip and no-audible pulse-skip modes High efficiency at light load conditions is achieved by PM6675AS entering the Pulse-Skip Mode (if enabled). At light load conditions the zero-crossing comparator truncates the lowside switch On-Time as soon as the inductor current becomes negative; in this way the comparator determines the On-Time duration instead of the output ripple (see Figure 27). Figure 27. Inductor current and output voltage at light load with Pulse-Skip Inductor current Output voltage Vreg TON TOFF As a consequence, the output capacitor is left floating and its discharge depends solely on the current drained from the load. When the output ripple on the pin COMP falls under the reference, a new shot is triggered and the next cycle begins. The Pulse-Skip mode is naturally obtained enabling the zero-crossing comparator and automatically takes part in the COT algorithm when the inductor current is about half the ripple current amount, i.e. migrating from continuous conduction mode (C.C.M.) to discontinuous conduction mode (D.C.M.). The output current threshold related to the transition between PWM Mode and Pulse-Skip Mode can be approximately calculated as: Equation 19 ILOAD (PWM2Skip) = VIN − VOUT ⋅ TON 2 ⋅L At higher loads, the inductor current never crosses the zero and the device works in pure PWM mode with a switching frequency around the nominal value. A physiological consequence of Pulse-Skip Mode is a more noisy and asynchronous (than normal conditions) output, mainly due to very low load. If the Pulse-Skip is not compatible with the application, the PM6675AS allows the user to choose also between forced-PWM and No-Audible Pulse-Skip alternative modes (see Chapter 7.1.4 for details). 24/48 PM6675AS Device description No-audible pulse-skip mode Some audio-noise sensitive applications cannot accept the switching frequency to enter the audible range as is possible in Pulse-Skip mode with very light loads. For this reason, the PM6675AS implements an additional feature to maintain a minimum switching frequency of 33 kHz despite of a slight efficiency loss. At very light load conditions, if any switching cycle has taken place within 30 µs (typ.) since the last one (because of the output voltage is still higher than the reference), a No-audible pulse-skip cycle begins. The low-side MOSFET is turned on and the output is driven to fall until the reference has been crossed. Then, the high-side switch is turned on for a Ton period and, once it has expired, the synchronous rectifier is enabled until the inductor current reaches the zero-crossing threshold (see Figure 28). Figure 28. Inductor current and output voltage at light load with non-audible pulse-skip Inductor current Output voltage Vreg TMAX TON TOFF TIDLE t For frequencies higher than 33 kHz (due to heavier loads) the device works in the same way as in Pulse-Skip mode. It is important to notice that in both pulse-skip and no-audible PulseSkip modes the switching frequency changes not only with the load but also with the input voltage. 25/48 Device description 7.1.4 PM6675AS Mode-of-operation selection Figure 29. VSEL and NOSKIP multifunction pin configurations VOUT +5V PM6675AS R9 R8 VSEL VREF NOSKIP The PM6675AS has been designed to satisfy the widest range of applications. The device is provided of some multilevel pins which allow the user to choose the appropriate configuration. The VSEL pin is used to firstly decide between fixed preset or adjustable (user defined) output voltages. When the VSEL pin is connected to +5 V, the PM6675AS set the switching section output voltage to 1.5 V without the need of an external divider. Applications requiring different output voltages can be managed by PM6675AS simply setting the adjustable mode. If the VSEL pin voltage is higher than 4 V, the fixed output mode is selected. Connecting an external divider to the VSEL pin, it is used as negative input of the error amplifier and the output voltage is given by expression (20). Equation 20 VOUTADJ = 0.6 ⋅ R8 + R9 R8 The output voltage can be set in the range from 0.6 V to 3.3 V. The NOSKIP is the power saving algorithm selector: if tied to +5 V, the forced-PWM (fixed frequency) control is performed. If grounded or connected to VREF pin (1.237 V reference voltage), the Pulse-Skip or Non-Audible Pulse-Skip Modes are respectively selected. Table 8. Mode-of-operation settings summary VSEL NOSKIP VOUT VNOSKI P > 4.2 V VVSEL > 4.3V VVSEL < 3.7V 1V < VNOSKIP < 3.5 V Forced-PWM 1.5 V Non-Audible Pulse-skip < 0.5 V Pulse-Skip VNOSKIP > 4.2 V Forced-PWM 1 V < VNOSKIP < 3.5 V VNOSKIP < 0.5 V 26/48 Operating mode ADJ Non-Audible Pulse-skip Pulse-Skip PM6675AS 7.1.5 Device description Current sensing and current limit The PM6675AS switching controller employes a valley current sensing algorithm to properly handle the current limit protection and the inductor current zero-crossing information. The current is sensed during the conduction time of the low-side MOSFET. The current sensing element is the low-side MOSFET on-resistance. The sensing scheme is visible in Figure 30. Figure 30. Current sensing scheme HGATE PHASE RILIM CSNS LGATE PGND An internal 100 µA current source is connected to CSNS pin that is also the non-inverting input of the positive current limit comparator. When the voltage drop developed across the sensing parameter equals the voltage drop across the programming resistor RILIM, the controller skips subsequent cycles until the overcurrent is detected or the output UV protection latches off the device (see par. Chapter 7.1.4 UV and OV Protections). Referring to Figure 30, the RDSon sensing technique is tailored to all low cost, high efficiency applications. It must be taken into account that the current limit circuit actually regulates the inductor valley current. This means that RILIM must be calculated to set a limit threshold given by the maximum DC output current plus half of the inductor ripple current: Equation 21 ICL = 100µA ⋅ RILIM R SENSE where RSENSE is the sensing device (RDSon). The PM6675AS provides also a fixed negative current limit to prevent excessive reverse inductor current when the switching section sinks current from the load in forced-PWM (3rd quadrant working conditions). This negative current limit threshold is measured between PHASE and PGND pins, comparing the drop magnitude on PHASE pin with an internal 110mV fixed voltage. 27/48 Device description 7.1.6 PM6675AS POR, UVLO and soft-start The PM6675AS automatically performs an internal startup sequence during the rising phase of the analog supply of the device (AVCC). The switching controller remains in a stand-by state until AVCC crosses the upper UVLO threshold (4.25 V typ.), keeping active the internal discharge MOSFETs (only if AVCC > 1 V). The soft-start allows a gradual increase of the internal current limit threshold during startup reducing the input/output surge currents. At the beginning of start-up, the PM6675AS current limit is set to 25 % of nominal value and the under voltage protection is disabled. Then, the current limit threshold is sequentially brought to 100 % in four steps of approximately 750 µs (figure 13). Figure 31. Soft-start waveforms Switching output Current limit threshold SWEN Time After a fixed 3 ms total time, the soft-start finishes and UVP is released: if the output voltage doesn't reach the under voltage lower threshold within soft-start duration, the UVP condition is detected; the device performs a soft end and latches off. Depending on the load conditions, the inductor current may or may not reach the nominal value of the current limit (Figure 32 on page 29 shows two examples). 28/48 PM6675AS Device description Figure 32. Soft-start at heavy load (a) and short-circuit (b) condition, pulse-skip enabled (b) (a) 7.1.7 Switching section power-good signal The SPG pin is an open drain output used to monitor output voltage through VSNS (in fixed output voltage mode) or VSEL (in adjustable output voltage mode) pins and is enabled after the soft-start timer has expired. The SPG signal is held low if the output voltage drops 10 % below or rises 10 % above the nominal regulated value. The SPG output can sink current up to 4 mA. 7.1.8 Switching section output discharge Active soft-end of the output occurs when the SWEN (SWitching ENable) is forced low. When the switching section is turned off, an internal 25 Ω resistor discharges the output through the VSNS pin. Figure 33. Switching section soft-end VOUT Resistive Discharge SWEN 29/48 Device description 7.1.9 PM6675AS Gate drivers The integrated high-current gate drivers allow using different power MOSFETs. The highside driver employes a bootstrap circuit which is supplied by the +5 V rail. The BOOT and PHASE pins work respectively as supply and return path for the high-side driver, while the low-side driver is directly feed through VCC and PGND pins. An important feature of the PM6675AS gate drivers is the Adaptive Anti-Cross-Conduction circuitry, which prevents high-side and low-side MOSFETs from being turned on at the same time. When the high-side MOSFET is turned off, the voltage at the PHASE node begins to fall. The low-side MOSFET is turned on only when the voltage at the PHASE node reaches an internal threshold (2.5 V typ.). Similarly, when the low-side MOSFET is turned off, the high-side one remains off until the LGATE pin voltage is above 1 V. The power dissipation of the drivers is a function of the total gate charge of the external power MOSFETs and the switching frequency, as shown in the following equation: Equation 22 PD (driver ) = VDRV ⋅ Q g ⋅ fSW The low-side driver has been designed to have a low-resistance pull-down transistor (0.6 Ω typ.) in order to prevent undesired ignition of the low-side MOSFET due to the Miller effect. 7.1.10 Reference voltage and bandgap The 1.237 V internal bandgap reference has a granted accuracy of ±1 % over the -25 °C to 85 °C temperature range. The VREF pin is a buffered replica of the bandgap voltage. It can supply up to ±100 µA and is suitable to set the intermediate level of NOSKIP multifunction pin. A 100 nF (min.) bypass capacitor toward SGND is required to enhance noise rejection. If VREF falls below 0.8 V (typ.), the system detects a fault condition and all the circuitry is turned off. An internal divider derives a 0.6 V±1 % voltage (Vr) from the bandgap. This voltage is used as reference for both the switching and the linear sections. The Over-Voltage Protection, the Under-Voltage Protection and the power-good signals are also referred to Vr. 7.1.11 Switching section OV and UV protections When the switching output voltage is about 115 % of its nominal value, a latched OverVoltage Protection (OVP) occurs. In this case the synchronous rectifier immediately turns on while the high-side MOSFET turns off. The output capacitor is rapidly discharged and the load is preserved from being damaged. The OVP is also active during the soft-start. Once an OVP has taken part, a toggle on SWEN pin or a power-on-reset is necessary to exit from the latched state. When the switching output voltage is below 70 % of its nominal value, a latched UnderVoltage Protection occurs. This event causes the switching section to be immediately disabled and both switches to be opened. The controller performs a soft-end and the output is eventually kept to ground, turning the low side MOSFET on when the voltage is lower than 400 mV. 30/48 PM6675AS Device description The Under-Voltage Protection circuit is enabled only at the end of the soft-start. Once an UVP has taken part, a toggle on SWEN pin or a Power-On-Reset is necessary to clear the fault state and restart the section. 7.1.12 Device thermal protection The internal control circuitry of the PM6675AS self-monitors the junction temperature and turns all outputs off when the 150 °C limit has been overran. This event causes the switching section to be immediately disabled and both switches to be opened. The controller performs a soft-end and both the outputs are eventually kept to ground, then the low side MOSFET is turned on when the voltage of the switching section is lower than 400 mV. The thermal fault is a latched protection and normal operating condition is restored by a Power-On Reset or toggling SWEN and LEN pins at the same time. Table 9. Switching section OV, UV and OT faults management Fault Conditions Over voltage VOUT > 115 % of the nominal value Under voltage Junction over temperature 7.2 Action LGATE pin is forced high and the device latches off. Exit by a Power-On Reset or toggling SWEN LGATE pin is forced high after the soft-end, then the VOUT < 70 % of the nominal device latches off. Exit by a Power-On Reset or value toggling SWEN. TJ > +150 °C LGATE pin is forced high after the soft-end, then the device latches off. Exit by a Power-On Reset or toggling SWEN and LEN after temperature drop. LDO linear regulator section The independent Low-Drop-Out (LDO) linear regulator has been designed to sink and source up to 2 A peak current and 1 A continuously. The LDO output voltage can be adjusted in the range 0.6 V to 3.3 V simply connecting a resistor divider as shown in Figure 34 on page 32. Equation 23 VLDO ADJ = 0.6 ⋅ R19 + R20 R20 31/48 Device description PM6675AS Figure 34. LDO output voltage selection PM6675AS VLOUT LOUT Cc COUT R19 LFB R20 LGND A compensation capacitor Cc must be added to adjust the dynamic response of the loop. The value of Cc is calculated according to the desired bandwidth of the LDO regulator and depends on the value of the feedback resistors. In most of applications the pole due to the compensation capacitor is placed at 100-200 kHz (equation 24). Equation 24 fp = 1 = 200kHz 2π(R19 ⊕ R20) ⋅ C C The LIN input can be connected to the switching section output for compact solutions or to a lower supply, if available in the system, in order to reduce the power dissipation of the LDO. A minimum output capacitance of 20 µF (2x10 µF MLCC capacitors) is enough to assure stability and fast load transient response. 7.2.1 LDO section current limit The LDO regulator can handle up to ±2 Apk, depending on the LDO input voltage and the LILIM pin setting. The output current is limited to ±1 A or ±2 A if the LILIM pin is connected to SGND or AVCC respectively (Figure 35). Figure 35. LDO current limit setting +5V PM6675AS ±2A CL ±1A CL LILIM The maximum current that the LDO can source depends also on the input and output voltages. Due to the high side MOSFET of the output stage, the LDO cannot source the limit current at high output voltages. Figure 36 shows the maximum current that the LDO can source as function of the input and output voltages. For output voltages higher than 2 V, the maximum output current is limited as reported. 32/48 PM6675AS Device description Figure 36. LDO current limit setting 2.2 ILOUT [A] 2.0 VOUT=1.05V 1.8 VOUT=1.2V 1.6 VOUT=1.5V VOUT=1.8V 1.4 VOUT=2.0V 1.2 VOUT=2.2V VOUT=2.5V 1.0 VOUT=3.3V 0.8 0.6 0.4 0.2 0.0 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 VLIN [V] 7.2.2 LDO section soft-start The LDO section soft-start is performed by clamping the current limit. During startup, the LDO current limit voltage is set to 1 A and the output voltage increases linearly. When the output voltage rises above 90 % of the nominal value, the current limit is released to 2 A according to the LILIM pin setting. 7.2.3 LDO section power-good signal The LPG pin is an open drain output used to monitor the LDO output voltage through LFB pin. The LPG signal is held low if the output voltage drops 10 % below or rises 10 % above the nominal regulated value. The LPG output can sink current up to 4 mA. 7.2.4 LDO section output discharge Active soft-end of the LDO output occurs when the LEN (Linear ENable) is forced low. When the LDO section is turned off, an internal 25 Ω resistor, directly connected to the LOUT pin, discharges the output. Figure 37. LDO section soft-end VLDO Resistive Discharge LEN 33/48 Application information 8 PM6675AS Application information The purpose of this chapter is to show the design procedure of the switching section. The design starts from three main specifications: ● The input voltage range, provided by the battery or the AC adapter. The two extreme values (VINmax and VINmin) are important for the design. ● The maximum load current, indicated with ILOAD,MAX . ● The maximum allowed output voltage ripple VRIPPLE,MAX. It’s also possible that specific designs should involve other specifications. The following paragraphs will guide the user into a step-by-step design. 8.1 External components selection The PM6675AS employes a pseudo-fixed frequency, Constant On-Time (COT) controller as the core of the switching section. The switching frequency can be set by connecting an external divider to the VOSC pin. The voltage seen at this pin must be greater than 0.8 V and lower than 2 V in order to take advantage of the internal block linearity. Nearly constant switching frequency is achieved by the system loop in steady-state operating conditions by varying the On-Time duration, avoiding thus the need for a clock generator. The On-Time one shot duration is directly proportional to the output voltage, sensed at VSNS pin, and inversely proportional to the input voltage, sensed at the VOSC pin, as follows: Equation 25 TON = K OSC VSNS +τ VOSC where KOSC is a constant value (130 ns typ.) and τ is the internal propagation delay (40 ns typ.). The duty-cycle of the buck converter is, in steady-state conditions, given by Equation 26 D= VOUT VIN The switching frequency is thus calculated as Equation 27 fSW 34/48 VOUT α VIN D 1 = = = OSC ⋅ VSNS TON α OUT K OSC K OSC ⋅ VOSC PM6675AS Application information Equation 28 a α OSC = VOSC VIN α OUT = VSNS VOUT Equation 28 b Referring to the typical application schematic (figs. 1 and 23), the final expression is then: Equation 29 fSW = α OSC R2 1 = ⋅ K OSC R1 + R 2 K OSC Even if the switching frequency is theoretically independent from battery and output voltages, parasitic parameters involved in power path (like MOSFETs on-resistance and inductor DCR) introduce voltage drops responsible of a slight dependence on load current. In addition, the internal delay is cause of a light dependence from input voltage. Table 10. 8.1.1 Typical values for switching frequency selection R1 (kΩ) R2 (kΩ) Approx switching frequency (kHz) 330 11 250 330 13 300 330 15 350 330 18 400 330 20 450 330 22 500 Inductor selection Once the switching frequency has been defined, the inductance value depends on the desired inductor ripple current. Low inductance value means great ripple current that brings to poor efficiency and great output noise. On the other hand a great current ripple is desirable for fast transient response when a load step is applied. Otherwise, great inductance brings to good efficiency but the transient response is critical, especially if VINmin - Vout is little. Moreover a minimum output ripple voltage is necessary to assure system stability and jitter-free operations (see Output capacitor selection paragraph). The product of the output capacitor ESR multiplied by the inductor ripple current must be taken into consideration. A good trade-off between the transient response time, the efficiency, the cost and the size is to choose the inductance value in order to maintain the inductor ripple current between 20 % and 50 % (usually 40 %) of the maximum output current. The maximum inductor ripple current, ∆ILMAX, occurs at the maximum input voltage. 35/48 Application information PM6675AS With these considerations, the inductance value can be calculated with the following expression: Equation 30 L= VIN − VOUT VOUT ⋅ fsw ⋅ ∆IL VIN where fSW is the switching frequency, VIN is the input voltage, VOUT is the output voltage and is the inductor current ripple. Once the inductor value is determined, the inductor current ripple is then recalculated: Equation 31 ∆IL,MAX = VIN,MAX − VOUT fsw ⋅ L ⋅ VOUT VIN,MAX The next step is the calculation of the maximum r.m.s. inductor current: Equation 32 IL,RMS = (ILOAD,MAX ) 2 + (∆IL,MAX ) 2 12 The inductor must have an r.m.s. current greater than IL,RMS in order to assure thermal stability. Then the calculation of the maximum inductor peak current follows: Equation 33 IL,PEAK = ILOAD,MAX + ∆IL,MAX 2 ILPEAK is important in inductor selection in term of its saturation current. The saturation current of the inductor should be greater than ILPEAK not only in case of hard saturation core inductors. Using soft-ferrite cores it is possible (but not advisable) to push the inductor working near its saturation current. In Table 11 some inductors suitable for typical working conditions are listed. Table 11. 36/48 Evaluated inductors (@ fsw = 400 kHz) Manufacturer Series Inductance (µH) +40°C rms current (A) -30% saturation current (A) COILCRAFT MLC1538-102 1 13.4 21.0 COILCRAFT MVR1261C-112 1.1 20 20 WURTH 7443552100 1 16 20 COILTRONICS HC8-1R2 1.2 16.0 25.4 PM6675AS Application information In Pulse-Skip Mode, low inductance values produce a better efficiency versus load curve, while higher values result in higher full-load efficiency because of the smaller current ripple. 8.1.2 Input capacitor selection In a buck topology converter the current that flows through the input capacitor is pulsed and with zero average value. The RMS input current can be calculated as follows: Equation 34 2 ICinRMS = ILOAD ⋅ D ⋅ (1 − D) + 1 D ⋅ (∆IL ) 2 12 Neglecting the second term, the equation 34 is reduced to: Equation 35 ICinRMS = ILOAD D ⋅ (1 − D) The losses due to the input capacitor are thus maximized when the duty-cycle is 0.5: Equation 36 Ploss = ESR Cin ⋅ ICinRMS (max) 2 = ESR Cin ⋅ (0.5 ⋅ ILOAD (max)) 2 The input capacitor should be selected with a RMS rated current higher than ICinRMS. Tantalum capacitors are good in term of low ESR and small size, but they occasionally can burn out if subjected to very high current during operation. Multi-Layers-Ceramic-Capacitors (MLCC) have usually a higher RMS current rating with smaller size and they remain the best choice. The drawback is their quite high cost. It must be taken in account that MLCC capacitance decreases when the operating voltage is near the rated voltage. In table 12 some MLCC suitable for most of applications are listed. Table 12. Evaluated MLCC for input filtering Manufacturer Series Capacitance (µF) Rated voltage (V) Maximum Irms @100 kHz (A) TAIYO YUDEN UMK325BJ106KM-T 10 50 2 TAIYO YUDEN GMK316F106ZL-T 10 35 2.2 TAIYO YUDEN GMK325F106ZH-T 10 35 2.2 TAIYO YUDEN GMK325BJ106KN 10 35 2.5 TDK C3225X5R1E106M 10 25 37/48 Application information 8.1.3 PM6675AS Output capacitor selection Using tantalum or electrolytic capacitors, the selection is made referring to ESR and voltage rating rather than by a specific capacitance value. The output capacitor has to satisfy the output voltage ripple requirements. At a given switching frequency, small inductor values are useful to reduce the size of the choke but increase the inductor current ripple. Thus, to reduce the output voltage ripple a low ESR capacitor is required. To reduce jitter noise between different switching regulators in the system, it is preferable to work with an output voltage ripple greater than 25 mV. As far as it concerns the load transient requirements, the Equivalent Series Resistance (ESR) of the output capacitor must satisfy this relationship: Equation 37 ESR ≤ VRIPPLE,MAX ∆IL,MAX where VRIPPLE is the maximum tolerable ripple voltage. In addition, the ESR must be enough high to meet stability requirements. The output capacitor zero must be lower than the switching frequency: Equation 38 fSW > fZ = 1 2π ⋅ ESR ⋅ C out If ceramic capacitors are used, the output voltage ripple due to inductor current ripple is negligible; then the inductance could be smaller, reducing the size of the choke. In this case it is important that the output capacitor can adsorb the inductor energy without generating an over-voltage condition when the system changes from a full load to a no load condition. The minimum output capacitance can be chosen by the following equation: Equation 39 C OUT,min = 2 L ⋅ ILOAD ,MAX Vf 2 − Vi 2 where Vf is the output capacitor voltage after the load transient and Vi is the output capacitor voltage before the load transient. 38/48 PM6675AS Application information In Table 13 some tested polymer capacitors are listed. Table 13. Evaluated output capacitors Manufacturer Series Capacitance (µF) Rated voltage (V) ESR max @100kHz (mΩ) SANYO 4TPE220MF 220 4V 15 to 25 4TPE150MI 150 4V 18 4TPC220M 220 4V 40 TNCB OE227MTRYF 220 2.5 V 25 HITACHI 8.1.4 MOSFETs selection In SMPS converters, power management efficiency is a high level requirement, so the power dissipation on the power switches becomes an important factor in switches selection. Losses of high-side and low-side MOSFETs depend on their working condition. Considering the high-side MOSFET, the power dissipation is calculated as: Equation 40 PDHighSide = Pconduction + Pswitching Maximum conduction losses are approximately given by: Equation 41 Pconduction = R DSon ⋅ VOUT 2 ⋅ ILOAD,MAX VIN. min where RDSon is the MOSFET drain-source on-resistance. Switching losses are approximately given by: Equation 42 Pswitching = VIN ⋅ (ILOAD (max) − 2 ∆IL ∆I ) ⋅ t on ⋅ fsw VIN ⋅ (ILOAD (max) + L ) ⋅ t off ⋅ fsw 2 2 + 2 where tON and tOFF are the turn-on and turn-off times of the MOSFET and depend on the gate-driver current capability and the gate charge Qgate. A greater efficiency is achieved with low RDSon. Unfortunately low RDSon MOSFETs have a great gate charge. As general rule, the RDSon . Qgate product should be minimized to find out the suitable MOSFET. Logic-level MOSFETs are recommended, as long as low-side and high-side gate drivers are powered by VVCC = +5 V. The breakdown voltage of the MOSFETs (VBRDSS) must be greater than the maximum input voltage VINmax. Below some tested high-side MOSFETs are listed. 39/48 Application information Table 14. PM6675AS Evaluated high-side MOSFETs Manufacturer Type RDSon (mΩ) Gate charge (Nc) Rated reverse voltage (V) ST STS12NH3LL 10.5 12 30 ST STS7NF60L 17 25 60 IR IRF7811 9 18 30 In buck converters the power dissipation of the synchronous MOSFET is mainly due to conduction losses: Equation 43 PDLowSide ≅ Pconduction Maximum conduction losses occur at the maximum input voltage: Equation 44 ⎛ V Pconduction = R DSon ⋅ ⎜1 − OUT ⎜ VIN,MAX ⎝ ⎞ ⎟ ⋅ ILOAD,MAX 2 ⎟ ⎠ The synchronous rectifier should have the lowest RDSon as possible. When the high-side MOSFET turns on, high dV/dt of the phase node can bring up even the low-side gate through its gate-drain capacitance CRES, causing cross-conduction problem. Once again, the choice of the low-side MOSFET is a trade-off between on resistance and gate charge; a good selection should minimizes the ratio CRSS / CGS where CGS = CISS - CRSS. Below some tested low-side MOSFETs are listed. Table 15. Evaluated low-side MOSFETs Manufacturer Type RDSon (mΩ) C GD C GS Rated reverse voltage (V) ST STS12NH3LL 13.5 0.069 30 ST STS25NH3LL 40 0.011 30 IR IRF7811 24 0.054 30 Dual N-MOS can be used in applications with low output current. Figure 16 shows some suitable dual MOSFETs for applications requiring about 3 A. Table 16. 40/48 Suitable dual MOSFETs Manufacturer Type RDSon (mΩ) Gate charge (nC) Rated reverse voltage (V) ST STS8DNH3LL 25 10 30 IR IRF7313 46 33 30 PM6675AS 8.1.5 Application information Diode selection A rectifier across the synchronous switch is recommended. The rectifier works as a voltage clamp across the synchronous rectifier and reduces the negative inductor swing during the dead time between turning the high-side MOSFET off and the synchronous rectifier on. Moreover it increases the efficiency of the system. The reverse voltage should be greater than the maximum input voltage VINmax and a minimum recovery reverse charge is preferable. Table 17 shows some evaluated diodes. Table 17. 8.1.6 Evaluated free-wheeling rectifiers Manufacturer Type Forward voltage (V) Rated reverse voltage (V) ST STPS1L30M 0.34 30 ST STPS1L30A 0.34 30 ST STPS1L60A 0.56 60 VOUT current limit setting The valley current limit is set by RCSNS and must be chosen to support the maximum load current. The valley of the inductor current ILvalley is: Equation 45 ILvalley = ILOAD (max) − ∆IL 2 The output current limit depends on the current ripple as shown in Figure 38: Figure 38. Valley current limit waveforms Inductor current Current Inductor current MAX LOAD 2 MAX LOAD 1 Valley current limit Time Being fixed the valley threshold, the more the current ripple is greater, the more the DC output current is greater. If an output current limit greater than over all the input voltage range is required, the minimum current ripple must be considered in the previous formula. Then the resistor RCSNS is: Equation 46 R CSNS = R DSon ⋅ ILvalley 100uA 41/48 Application information PM6675AS where RDSon is the drain-source on-resistance low-side switch. Consider the temperature effect and the worst case value in RDSon calculation (typically +0.4 %/°C). The accuracy of the valley current also depends on the offset of the internal comparator (±6 mV). The negative valley-current limit (if the device works in forced-PWM mode) is given by: Equation 47 110mV RDSon INEG = 8.1.7 All ceramic capacitors application Design of external feedback network depends on the output voltage ripple across the output capacitors ESR. If the ripple is great enough (at least 20 mV), the compensation network simply consist of a CINT capacitor. Figure 39. Integrative compensation Ton One-shot generator VSNS VOUT + PWM Comparator - VREF + COMP gm Integrator CFILT RINT - Vr=0.6 CINT The stability of the system depends firstly on the output capacitor zero frequency. It must be verified that: Equation 48 fSW > k ⋅ fZout = 42/48 k 2π ⋅ R out C out PM6675AS Application information where k is a free design parameter greater than unity (k > 3) . It determinates the minimum integrator capacitor value CINT: Equation 49 CINT > gm Vref ⎞ Vo ⎛f 2π ⋅ ⎜⎜ SW − fZout ⎟⎟ k ⎠ ⎝ ⋅ If the ripple on pin COMP is greater than the integrator output dynamic (150 mV), an additional capacitor Cfilt could be added in order to reduce its amplitude. If q is the desired attenuation factor of the output ripple, select: Equation 50 C filt = CINT ⋅ (1 − q) q In order to reduce noise on pin COMP, it’s possible to introduce a resistor RINT that, together with CINT and Cfilt, realizes a low pass filter. The cutoff frequency must be much greater (10 or more times) than the switching frequency of the section: Equation 51 RINT = 2π ⋅ fCUT 1 CINT ⋅ CFILT CINT + CFILT For most of applications both RINT and Cfilt are unnecessary. If the ripple is very small (e.g. such as with ceramic capacitors), a further compensation network, called “Virtual ESR” network, is needed. This additional part generates a triangular ripple that substitutes the ESR output voltage ripple. The complete compensation scheme is represented in Figure 40. Figure 40. Virtual ESR network L R R1 CINT VOUT C RINT PWM Comparator Ton Generation Block + gm + - VREF - CFILT 0.6V Integrator 43/48 Application information PM6675AS Select C as shown: Equation 52 C > 5 ⋅ CINT Then calculate R in order to have enough ripple voltage on the integrator input: Equation 53 R= L R VESR ⋅ C Where RVERS is the new virtual output capacitor ESR. A good trade-off is to consider an equivalent ESR of 30-50 mΩ, even though the choice depends on inductor current ripple. Then choose R1 as follows: Equation 54 ⎛ 1 ⎞ ⎟ R ⋅ ⎜⎜ CπfZ ⎟⎠ ⎝ R1 = 1 R− CπfZ 44/48 PM6675AS 9 Package mechanical data Package mechanical data In order to meet environmental requirements, ST offers these devices in ECOPACK® packages. These packages have a Lead-free second level interconnect. The category of second Level Interconnect is marked on the package and on the inner box label, in compliance with JEDEC Standard JESD97. The maximum ratings related to soldering conditions are also marked on the inner box label. ECOPACK is an ST trademark. ECOPACK specifications are available at: www.st.com. Table 18. VFQFPN-24 4mm x 4mm mechanical data mm. Dim. Min Typ Max 0.80 0.90 1.00 A1 0.0 0.05 A2 0.65 0.80 D 4.00 D1 3.75 E 4.00 E1 3.75 A θ P 12° 0.24 0.42 e 0.50 N 24.00 Nd 6.00 Ne 6.00 0.40 0.60 L 0.30 0.50 b 0.18 D2 1.95 2.10 2.25 E2 1.95 2.10 2.25 0.30 45/48 Package mechanical data Figure 41. Package dimensions 46/48 PM6675AS PM6675AS 10 Revision history Revision history Table 19. Document revision history Date Revision 19-Feb-2008 1 Changes Initial release. 47/48 PM6675AS Please Read Carefully: Information in this document is provided solely in connection with ST products. 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