STMICROELECTRONICS L6566BTR

L6566B
Multi-mode controller for SMPS
Features
■
Selectable multi-mode operation:
fixed frequency or quasi-resonant
■
On-board 700 V high-voltage start-up
■
Advanced light load management
■
Low quiescent current (< 3 mA)
■
Adaptive UVLO
■
Line feedforward for constant power capability
vs mains voltage
■
Pulse-by-pulse OCP, shutdown on overload
(latched or autorestart)
■
Hi-end AC-DC adapter/charger
■
Transformer saturation detection
■
LCD TV/monitor, PDP
■
Programmable frequency modulation for EMI
reduction
■
digital consumer, IT equipment
■
single-stage PFC
■
Latched or autorestart OVP
■
Brownout protection
■
-600/+800 mA totem pole gate driver with
active pull-down during UVLO
■
SO16N package
Applications
Block diagram
VREF
1
HV
5
TIME
OUT
15
6.4V
7.7V
Q
CS
LEB
7
1.5 V
UVLO
VCC
UVLO_SHF
400 uA
6
+
PWM
Vth
+
OCP
-
VCC
Hiccup-mode
OCP logic
5.7V
+
+
FMOD
+
OVPL
Icharge
LINE VOLTAGE
FEEDFORWARD
Reference
voltages
Internal supply
VOLTAGE
REGULATOR
&
ADAPTIVE UVLO
VCC
OVP
LOW CLAMP OFF2
& DISABLE
+
VCC
VFF
9
SOFT-START
&
FAULT MNGT
I HV
COMP
SS
14
10
-
Figure 1.
SO16N
BURST-MODE
14V
OCP2
4
GD
13
OSCILLATOR
R
Q
50 mV
100 mV
ZCD
-
DRIVER
S
MODE SELECTION
&
TURN-ON LOGIC
12
TIME
OUT OVPL
ZERO CURRENT
DETECTOR
+
11
4.5V
OVERVOLTAGE
PROTECTION
OFF2
OVP
LATCH
+
MODE/SC
-
OSC
DIS
8
IC_LATCH
16
AC_OK
3V
15 µA
0.450V
0.485V
-
AC_FAIL
+
UVLO
DISABLE
3
May 2008
Rev 2
1/51
www.st.com
51
Contents
L6566B
Contents
1
Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
2
Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
3
2.1
Connections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
2.2
Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Electrical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
3.1
Maximum rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
3.2
Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
4
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
5
Application information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
2/51
5.1
High-voltage start-up generator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
5.2
Zero current detection and triggering block; oscillator block . . . . . . . . . . 21
5.3
Burst-mode operation at no load or very light load . . . . . . . . . . . . . . . . . . 24
5.4
Adaptive UVLO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
5.5
PWM control block . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
5.6
PWM comparator, PWM latch and voltage feedforward blocks . . . . . . . . 27
5.7
Hiccup-mode OCP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
5.8
Frequency modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
5.9
Latched disable function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
5.10
Soft-start and delayed latched shutdown upon overcurrent . . . . . . . . . . . 33
5.11
OVP block . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
5.12
Brownout protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
5.13
Slope compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
5.14
Summary of L6566B power management functions . . . . . . . . . . . . . . . . 41
L6566B
Contents
6
Application examples and ideas . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
7
Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
8
Order codes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49
9
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50
3/51
List of tables
L6566B
List of tables
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.
Table 7.
Table 8.
Table 9.
Table 10.
Table 11.
4/51
Pin functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Thermal data. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
L6566B light load management features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
L6566B protections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
External circuits that determine IC behavior upon OVP and OCP . . . . . . . . . . . . . . . . . . . 45
SO16N mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
Order codes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50
L6566B
List of figures
List of figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.
Figure 9.
Figure 10.
Figure 11.
Figure 12.
Figure 13.
Figure 14.
Figure 15.
Figure 16.
Figure 17.
Figure 18.
Figure 19.
Figure 20.
Figure 21.
Figure 22.
Figure 23.
Figure 24.
Figure 25.
Figure 26.
Figure 27.
Figure 28.
Figure 29.
Figure 30.
Figure 31.
Figure 32.
Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Typical system block diagram. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Pin connection (through top view) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Multi-mode operation with QR option active . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
High-voltage start-up generator: internal schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Timing diagram: normal power-up and power-down sequences . . . . . . . . . . . . . . . . . . . . 19
Timing diagram showing short-circuit behavior (SS pin clamped at 5V). . . . . . . . . . . . . . . 20
Zero current detection block, triggering block, oscillator block and related logic . . . . . . . . 20
Drain ringing cycle skipping as the load is gradually reduced . . . . . . . . . . . . . . . . . . . . . . 22
Operation of ZCD, triggering and oscillator blocks (QR option active) . . . . . . . . . . . . . . . . 23
Load-dependent operating modes: timing diagrams . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Addition of an offset to the current sense lowers the burst-mode operation threshold . . . . 25
Adaptive UVLO block. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Possible feedback configurations that can be used with the L6566B . . . . . . . . . . . . . . . . . 26
Externally controlled burst-mode operation by driving pin COMP: timing diagram. . . . . . . 27
Typical power capability change vs. input voltage in QR flyback converters . . . . . . . . . . . 28
Left: Overcurrent setpoint vs. VFF voltage; right: Line Feedforward function block . . . . . . 29
Hiccup-mode OCP: timing diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Frequency modulation circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Operation after latched disable activation: timing diagram . . . . . . . . . . . . . . . . . . . . . . . . . 33
Soft-start pin operation under different operating conditions and settings . . . . . . . . . . . . . 34
OVP Function: internal block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
OVP function: timing diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Maximum allowed duty cycle vs. switching frequency for correct OVP detection. . . . . . . . 37
Brownout protection: internal block diagram and timing diagram . . . . . . . . . . . . . . . . . . . . 38
Voltage sensing techniques to implement brownout protection with the L6566B . . . . . . . . 39
Slope compensation waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
Typical low-cost application schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
Typical full-feature application schematic (QR operation) . . . . . . . . . . . . . . . . . . . . . . . . . 44
Typical full-feature application schematic (FF operation) . . . . . . . . . . . . . . . . . . . . . . . . . . 45
Frequency foldback at light load (FF operation) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Latched shutdown upon mains overvoltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
5/51
Description
1
L6566B
Description
The L6566B is an extremely versatile current-mode primary controller ICs, specifically
designed for high-performance offline flyback converters. It is also suited for single-stage
single-switch input-current-shaping converters (single-stage PFC) for applications supposed
to comply with EN61000-3-2 or JEITA-MITI regulations.
Both fixed-frequency (FF) and quasi-resonant (QR) operation are supported. The user can
pick either of the two depending on application needs. The device features an externally
programmable oscillator: it defines converter’s switching frequency in FF mode and the
maximum allowed switching frequency in QR mode.
When FF operation is selected, the ICs work like a standard current-mode controller with a
maximum duty cycle limited at 70 % min. The oscillator frequency can be modulated to
mitigate EMI emissions.
QR operation, when selected, occurs at heavy load and is achieved through a transformer
demagnetization sensing input that triggers MOSFET’s turn-on. Under some conditions,
ZVS (zero-voltage switching) can be achieved. Converter’s power capability rise with the
mains voltage is compensated by line voltage feedforward. At medium and light load, as the
QR operating frequency equals the oscillator frequency, a function (valley skipping) is
activated to prevent further frequency rise and keep the operation as close to ZVS as
possible.
With either FF or QR operation, at very light load the ICs enter a controlled burst-mode
operation that, along with the built-in non-dissipative high-voltage start-up circuit and the low
quiescent current, helps keep low the consumption from the mains and meet energy saving
recommendations.
An innovative adaptive UVLO helps minimize the issues related to the fluctuations of the
self-supply voltage due to transformer’s parasites.
The protection functions included in this device are: not-latched input undervoltage
(brownout), output OVP (auto-restart or latch-mode selectable), a first-level OCP with
delayed shutdown to protect the system during overload or short circuit conditions (autorestart or latch-mode selectable) and a second-level OCP that is invoked when the
transformer saturates or the secondary diode fails short. A latched disable input allows easy
implementation of OTP with an external NTC, while an internal thermal shutdown prevents
IC overheating.
Programmable soft-start, leading-edge blanking on the current sense input for greater noise
immunity, slope compensation (in FF mode only), and a shutdown function for externally
controlled burst-mode operation or remote ON/OFF control complete the equipment of this
device.
6/51
L6566B
Description
Figure 2.
Typical system block diagram
FLYBACK DC-DC CONVERTER
Rectified
& Filtered
Mains
Voltage
Voutdc
L6566B
7/51
Pin settings
L6566B
2
Pin settings
2.1
Connections
Figure 3.
2.2
HVS
1
16
AC_OK
N.C.
2
15
VFF
GND
3
14
SS
GD
4
13
OSC
Vcc
5
12
MODE/SC
FMOD
6
11
ZCD
CS
7
10
VREF
DIS
8
9
COMP
Pin description
Table 1.
N°
8/51
Pin connection (through top view)
Pin functions
Pin
Function
1
HVS
High-voltage start-up. The pin, able to withstand 700 V, is to be tied directly to the
rectified mains voltage. A 1 mA internal current source charges the capacitor
connected between Vcc pin (5) and GND pin (3) until the voltage on the Vcc pin
reaches the turn-on threshold, then it is shut down. Normally, the generator is reenabled when the Vcc voltage falls below 5 V to ensure a low power throughput
during short circuit. Otherwise, when a latched protection is tripped the generator is
re-enabled 0.5 V below the turn-on threshold, to keep the latch supplied; or, when
the IC is turned off by pin COMP (9) pulled low the generator is active just below
the UVLO threshold to allow a faster restart.
2
N.C.
Not internally connected. Provision for clearance on the PCB to meet safety
requirements.
3
GND
Ground. Current return for both the signal part of the IC and the gate drive. All of
the ground connections of the bias components should be tied to a track going to
this pin and kept separate from any pulsed current return.
4
GD
Gate driver output. The totem pole output stage is able to drive power MOSFET’s
and IGBT’s with a peak current capability of 800 mA source/sink.
L6566B
Pin settings
Table 1.
N°
5
6
7
8
9
10
Pin functions (continued)
Pin
Function
Vcc
Supply voltage of both the signal part of the IC and the gate driver. The internal
high voltage generator charges an electrolytic capacitor connected between this
pin and GND (pin 3) as long as the voltage on the pin is below the turn-on threshold
of the IC, after that it is disabled and the chip is turned on. The IC is disabled as the
voltage on the pin falls below the UVLO threshold. This threshold is reduced at light
load to counteract the natural reduction of the self-supply voltage. Sometimes a
small bypass capacitor (0.1 µF typ.) to GND might be useful to get a clean bias
voltage for the signal part of the IC.
FMOD
Frequency modulation input. When FF mode operation is selected, a capacitor
connected from this pin to GND (pin 3) is alternately charged and discharged by
internal current sources. As a result, the voltage on the pin is a symmetrical
triangular waveform with the frequency related to the capacitance value. By
connecting a resistor from this pin to pin 13 (OSC) it is possible to modulate the
current sourced by the OSC pin and then the oscillator frequency. This modulation
is to reduce the peak value of EMI emissions by means of a spread-spectrum
action. If the function is not used, the pin will be left open.
CS
Input to the PWM comparator. The current flowing in the MOSFET is sensed
through a resistor, the resulting voltage is applied to this pin and compared with an
internal reference to determine MOSFET’s turn-off. The pin is equipped with 150 ns
min. blanking time after the gate-drive output goes high for improved noise
immunity. A second comparison level located at 1.5 V latches the device off and
reduces its consumption in case of transformer saturation or secondary diode short
circuit. The information is latched until the voltage on the Vcc pin (5) goes below
the UVLO threshold, hence resulting in intermittent operation. A logic circuit
improves sensitivity to temporary disturbances.
DIS
IC’s latched disable input. Internally the pin connects a comparator that, when the
voltage on the pin exceeds 4.5 V, latches off the IC and brings its consumption to a
lower value. The latch is cleared as the voltage on the Vcc pin (5) goes below the
UVLO threshold, but the HV generator keeps the Vcc voltage high (see pin 1
description). It is then necessary to recycle the input power to restart the IC. For a
quick restart pull pin 16 (AC_OK) below the disable threshold (see pin 16
description). Bypass the pin with a capacitor to GND (pin 3) to reduce noise pickup. Ground the pin if the function is not used.
COMP
Control input for loop regulation. The pin will be driven by the phototransistor
(emitter-grounded) of an optocoupler to modulate its voltage by modulating the
current sunk. A capacitor placed between the pin and GND (3), as close to the IC
as possible to reduce noise pick-up, sets a pole in the output-to-control transfer
function. The dynamics of the pin is in the 2.5 to 5 V range. A voltage below an
internally defined threshold activates burst-mode operation. The voltage at the pin
is bottom-clamped at about 2 V. If the clamp is externally overridden and the
voltage is pulled below 1.4 V the IC will shut down.
VREF
An internal generator furnishes an accurate voltage reference (5 V ± 2 %) that can
be used to supply few mA to an external circuit. A small film capacitor (0.1 µF typ.),
connected between this pin and GND (3), is recommended to ensure the stability of
the generator and to prevent noise from affecting the reference. This reference is
internally monitored by a separate auxiliary reference and any failure or drift will
cause the IC to latch off.
9/51
Pin settings
L6566B
Table 1.
N°
11
Pin functions (continued)
Pin
Function
ZCD
Transformer demagnetization sensing input for quasi-resonant operation and OVP
input. The pin is externally connected to the transformer’s auxiliary winding through
a resistor divider. A negative-going edge triggers MOSFET’s turn-on if QR mode is
selected.
A voltage exceeding 5 V shuts the IC down and brings its consumption to a lower
value (OVP). Latch-off or auto-restart mode is selectable externally. This function is
strobed and digitally filtered to increase noise immunity.
Operating mode selection. If the pin is connected to the VREF pin (7)
quasi-resonant operation is selected, the oscillator (pin 13, OSC) determines the
maximum allowed operating frequency.
Fixed-frequency operation is selected if the pin is not tied to VREF, in which case
12 MODE/SC the oscillator determines the actual operating frequency, the maximum allowed
duty cycle is set at 70 % min. and the pin delivers a voltage ramp synchronized to
the oscillator when the gate-drive output is high; the voltage delivered is zero while
the gate-drive output is low. The pin is to be connected to pin CS (7) via a resistor
for slope compensation.
13
14
15
16
10/51
OSC
Oscillator pin. The pin is an accurate 1 V voltage source, and a resistor connected
from the pin to GND (pin 3) defines a current. This current is internally used to set
the oscillator frequency that defines the maximum allowed switching frequency of
the L6566B, if working in QR mode, or the operating switching frequency if working
in FF mode.
SS
Soft-start current source. At start-up a capacitor Css between this pin and GND
(pin 3) is charged with an internal current generator. During the ramp, the internal
reference clamp on the current sense pin (7, CS) rises linearly starting from zero to
its final value, thus causing the duty cycle to increase progressively starting from
zero as well. During soft-start the adaptive UVLO function and all functions
monitoring pin COMP are disabled. The soft-start capacitor is discharged whenever
the supply voltage of the IC falls below the UVLO threshold. The same capacitor is
used to delay IC’s shutdown (latch-off or auto-restart mode selectable) after
detecting an overload condition (OLP).
VFF
Line voltage feedforward input. The information on the converter’s input voltage is
fed into the pin through a resistor divider and is used to change the setpoint of the
pulse-by-pulse current limitation (the higher the voltage, the lower the setpoint).
The linear dynamics of the pin ranges from 0 to 3 V. A voltage higher than 3 V
makes the IC stop switching. If feedforward is not desired, tie the pin to GND (pin 3)
directly if a latch-mode OVP is not required (see pin 11, ZCD) or through a 10 kΩ
min. resistor if a latch-mode OVP is required. Bypass the pin with a capacitor to
GND (pin 3) to reduce noise pick-up.
AC_OK
Brownout protection input. A voltage below 0.45 V shuts down (not latched) the IC,
lowers its consumption and clears the latch set by latched protections (DIS > 4.5 V,
SS > 6.4 V, VFF > 6.4 V). IC’s operation is re-enabled as the voltage exceeds
0.45 V. The comparator is provided with current hysteresis: an internal 15 µA
current generator is ON as long as the voltage on the pin is below 0.45 V and is
OFF if this value is exceeded. Bypass the pin with a capacitor to GND (pin 3) to
reduce noise pick-up. Tie to Vcc with a 220 to 680 kΩ resistor if the function is not
used.
L6566B
Electrical data
3
Electrical data
3.1
Maximum rating
Table 2.
Absolute maximum ratings
Symbol
Pin
VHVS
1
IHVS
Value
Unit
Voltage range (referred to ground)
-0.3 to 700
V
1
Output current
Self-limited
VCC
5
IC supply voltage (Icc = 20 mA)
Self-limited
VFMOD
6
Voltage range
-0.3 to 2
V
Vmax
7, 8, 10, 14
Analog inputs and outputs
-0.3 to 7
V
Vmax
9, 15, 16
Maximum pin voltage (Ipin ≤ 1 mA)
Self-limited
IZCD
11
Zero current detector max. current
±5
mA
VMODE/SC
12
Voltage range
-0.3 to 5.3
V
VOSC
13
Voltage range
-0.3 to 3.3
V
0.75
W
PTOT
Power dissipation @TA = 50 °C
TSTG
Storage temperature
-55 to 150
°C
Junction operating temperature range
-40 to 150
°C
Value
Unit
120
°C/W
TJ
3.2
Parameter
Thermal data
Table 3.
Symbol
RthJA
Thermal data
Parameter
Thermal resistance junction to ambient
11/51
Electrical characteristics
4
L6566B
Electrical characteristics
(TJ = -25 to 125°C, VCC = 12, CO = 1 nF; MODE/SC = VREF, RT = 20 kΩ from OSC to GND,
unless otherwise specified).
Table 4.
Electrical characteristics
Symbol
Parameter
Test condition
Min
Typ
Max
Unit
Supply voltage
Vcc
VccOn
VccOff
Operating range after turn-on
Turn-on threshold
Turn-off threshold
VCOMP > VCOMPL
10.6
23
VCOMP = VCOMPO
8
23
(1)
(1)
(1)
V
13
14
15
VCOMP > VCOMPL
9.4
10
10.6
VCOMP = VCOMPO
7.2
7.6
8.0
Hys
Hysteresis
VCOMP > VCOMPL
VZ
Zener voltage
Icc = 20 mA, IC disabled
V
V
4
23
V
25
27
V
Supply current
Istart-up
Start-up current
Before turn-on, Vcc = 13 V
200
250
µA
Iq
Quiescent current
After turn-on, VZCD = VCS = 1 V
2.6
2.8
mA
Icc
Operating supply current
MODE/SC open
4
4.6
mA
IC disabled
Iqdis
(2)
330
2500
Quiescent current
µA
IC latched off
440
500
High-voltage start-up generator
Breakdown voltage
IHV < 100 µA
700
Start voltage
IVcc < 100 µA
65
80
100
V
Icharge
Vcc charge current
VHV > VHvstart, Vcc > 3 V
0.55
0.85
1
mA
IHV, ON
ON-state current
IHV, OFF
OFF-state leakage current
VHV
VHVstart
VHV > VHvstart, Vcc > 3 V
1.6
VHV > VHvstart, Vcc = 0
0.8
VHV = 400 V
40
Vcc falling
12/51
Vcc restart voltage
mA
4.4
5
5.6
IC latched off
12.5
13.5
14.5
Disabled by
VCOMP < VCOMPOFF
9.4
10
10.6
(1)
VCCrestart
V
(1)
µA
V
L6566B
Table 4.
Electrical characteristics
Electrical characteristics (continued)
Symbol
Parameter
Test condition
Min
Typ
Max
Unit
4.95
5
5.05
V
Reference voltage
VREF
Output voltage
(1)
VREF
Total variation
IREF = 1 to 5 mA,
Vcc = 10.6 to 23 V
4.9
5.1
V
IREF
Short circuit current
VREF = 0
10
30
mA
Sink capability in UVLO
Vcc = 6 V; Isink = 0.5 mA
0.5
V
VOV
TJ = 25 °C; IREF = 1 mA
Overvoltage threshold
0.2
5.3
5.7
V
Internal oscillator
fsw
Oscillation frequency
Operating range
10
TJ = 25 °C, VZCD = 0,
MODE/SC = open
95
100
105
Vcc =12 to 23 V, VZCD = 0,
MODE/SC = open
93
100
107
0.97
1
1.03
V
75
%
VOSC
Voltage reference
(3)
Dmax
Maximum duty cycle
MODE/SC = open,
VCOMP = 5 V
300
70
kHz
Brownout protection
Vth
IHys
Voltage falling (turn-off)
0.432
0.450
0.468
V
Voltage rising (turn-on)
0.452
0.485
0.518
V
Vcc > 5 V, VVFF = 0.3 V
12
15
18
µA
3
3.15
3.3
V
-1
µA
Threshold voltage
Current hysteresis
(1)
VAC_OK_CL Clamp level
IAC_OK = 100 µA
Line voltage feedforward
VVFF = 0 to 3 V, VZCD < VZCDth
IVFF
Input bias current
VVFF
Linear operation range
VOFF
IC disable voltage
VVFFlatch
Kc
KFF
VZCD > VZCDth
3
Latch-off/clamp level
Control voltage gain
Feedforward gain
-0.7
(3)
(3)
-1
mA
0 to 3
V
3.15
3.3
V
VZCD > VZCDth
6.4
V
VVFF = 1 V, VCOMP = 4 V
0.4
V/V
VVFF = 1 V, VCOMP = 4 V
0.04
V/V
13/51
Electrical characteristics
Table 4.
L6566B
Electrical characteristics (continued)
Symbol
Parameter
Test condition
Min
Typ
Max
Unit
-1
µA
300
ns
100
ns
Current sense comparator
ICS
Input bias current
tLEB
Leading edge blanking
td(H-L)
VCSx
VCS = 0
150
Delay to output
VCOMP = VCOMPHI, VVFF = 0 V
0.92
1
1.08
Overcurrent setpoint
VCOMP = VCOMPHI, VVFF = 1.5 V
0.45
0.5
0.55
0
0.1
Hiccup-mode OCP level
(1)
1.5
1.6
VCOMP = VCOMPHI, VVFF = 3.0 V
VCSdis
250
1.4
V
V
PWM control
VCOMPHI
Upper clamp voltage
ICOMP = 0
5.7
V
VCOMPLO
Lower clamp voltage
ISOURCE = -1 mA
2.0
V
Linear dynamics upper limit
(1)
ICOMP
Max. source current
VCOMP = 3.3 V
RCOMP
Dynamic resistance
VCOMP = 2.6 to 4.8 V
VCOMPSH
VVFF = 0 V
(1)
VCOMPBM
Burst-mode threshold
Hys
Burst-mode hysteresis
ICLAMPL
Lower clamp capability
VCOMPOFF
Disable threshold
VCOMPO
Level for lower UVLO off
threshold (voltage falling)
VCOMPL
Level for higher UVLO off
threshold (voltage rising)
14/51
(1) MODE/SC
4.8
5
5.2
V
320
400
480
µA
25
kΩ
2.52
2.65
2.78
2.7
2.85
3
V
= open
20
VCOMP = 2 V
-3.5
Voltage falling
(3)
(3) MODE/SC
-1.5
1.4
mA
V
2.61
2.75
2.89
3.02
3.15
3.28
2.9
3.05
3.2
3.41
3.55
3.69
V
= open
(3)
(3)
mV
V
MODE/SC = open
L6566B
Table 4.
Electrical characteristics
Electrical characteristics (continued)
Symbol
Parameter
Test condition
Min
Typ
Max
Unit
5.4
5.7
6
V
Zero current detector/ overvoltage protection
VZCDH
Upper clamp voltage
IZCD = 3 mA
VZCDL
Lower clamp voltage
IZCD = - 3 mA
Arming voltage
(1)
positive-going edge
85
100
115
mV
Triggering voltage
(1)
negative-going edge
30
50
70
mV
VZCDA
VZCDT
IZCD
Internal pull-up
-0.4
VCOMP < VCOMPSH
VZCD < 2 V, VCOMP = VCOMPHI
V
-1
µA
-130
-100
-70
IZCDsrc
Source current capability
VZCD = VZCDL
-3
mA
IZCDsnk
Sink current capability
VZCD = VZCDH
3
mA
TBLANK1
Turn-on inhibit time
After gate-drive going low
VZCDth
TBLANK2
OVP threshold
OVP strobe delay
2.5
4.85
After gate-drive going low
5
µs
5.15
2
V
µs
Latched shutdown function
IOTP
VOTP
Input bias current
VDIS = 0 to VOTP
Disable threshold
(1)
4.32
4.5
-1
µA
4.68
V
Thermal shutdown
Vth
Shutdown threshold
160
°C
Hys
Hysteresis
50
°C
External oscillator (frequency modulation)
fFMOD
Oscillation frequency
---
Usable frequency range
Vpk
Peak voltage
Vvy
IFMOD
CMOD = 0.1 µF
600
750
0.05
(3)
900
Hz
15
kHz
1.5
V
Valley voltage
0.5
V
Charge/discharge current
150
µA
3
V
Mode selection / slope compensation
MODEth
Threshold for QR operation
SCpk
Ramp peak
(MODE/SC = open)
RS-COMP = 3 kΩ to GND, GD
pin high, VCOMP = 5 V
1.7
V
SCvy
Ramp starting value
(MODE/SC = open)
RS-COMP = 3 kΩ to GND,
GD pin high
0.3
V
Ramp voltage
(MODE/SC = open)
GD pin low
0
V
Source capability
(MODE/SC = open)
VS-COMP = VS-COMPpk
0.8
mA
15/51
Electrical characteristics
Table 4.
L6566B
Electrical characteristics (continued)
Symbol
Parameter
Test condition
Min.
Typ.
Max.
TJ = 25 °C, VSS < 2 V,
VCOMP = 4 V
14
20
26
TJ = 25 °C, VSS > 2 V,
VCOMP =VCOMPHi
3.5
5
6.5
Discharge current
VSS > 2 V
3.5
5
6.5
High saturation voltage
VCOMP = 4 V
VSSDIS
Disable level
(1)
VSSLAT
Latch-off level
VCOMP = VCOMPHi
VGDH
Output high voltage
IGDsource = 5 mA, Vcc = 12 V
VGDL
Output low voltage
IGDsink = 100 mA
Unit
Soft-start
ISS1
Charge current
ISS2
ISSdis
VSSclamp
VCOMP = VCOMPHi
µA
2
4.85
5
µA
V
5.15
V
6.4
V
11
V
0.75
V
Gate driver
Isourcepk
Isinkpk
9.8
Output source peak current
-0.6
A
Output sink peak current
0.8
A
tf
Fall time
40
ns
tr
Rise time
50
ns
VGDclamp
Output clamp voltage
IGDsource = 5 mA; Vcc = 20 V
UVLO saturation
Vcc = 0 to Vccon, Isink = 1 mA
10
1. Parameters tracking one another.
2. See Table 6 on page 41 and Table 7 on page 42
3. The voltage feedforward block output is given by:
16/51
Vcs = Kc (VCOMP − 2.5 ) − K FF VVFF
11.3
15
V
0.9
1.1
V
L6566B
5
Application information
Application information
The L6566B is a versatile peak-current-mode PWM controller specific for offline flyback
converters. The device allows either fixed-frequency (FF) or quasi-resonant (QR) operation,
selectable with the pin MODE/SC (12): forcing the voltage on the pin over 3 V (e.g. by tying
it to the 5 V reference externally available at pin VREF, 10) will activate QR operation,
otherwise the device will be FF-operated.
Irrespective of the operating option selected by pin 12, the device is able to work in different
modes, depending on the converter’s load conditions. If QR operation is selected (see
Figure 4):
1. QR mode at heavy load. Quasi-resonant operation lies in synchronizing MOSFET's
turn-on to the transformer’s demagnetization by detecting the resulting negative-going
edge of the voltage across any winding of the transformer. Then the system works
close to the boundary between discontinuous (DCM) and continuous conduction
(CCM) of the transformer. As a result, the switching frequency will be different for
different line/load conditions (see the hyperbolic-like portion of the curves in Figure 4).
Minimum turn-on losses, low EMI emission and safe behavior in short circuit are the
main benefits of this kind of operation.
2. Valley-skipping mode at medium/ light load. The externally programmable oscillator of
the L6566B, synchronized to MOSFET’s turn-on, enables the designer to define the
maximum operating frequency of the converter. As the load is reduced MOSFET’s turnon will not any more occur on the first valley but on the second one, the third one and
so on. In this way the switching frequency will no longer increase (piecewise linear
portion in Figure 4).
3. Burst-mode with no or very light load. When the load is extremely light or disconnected,
the converter will enter a controlled on/off operation with constant peak current.
Decreasing the load will then result in frequency reduction, which can go down even to
few hundred hertz, thus minimizing all frequency-related losses and making it easier to
comply with energy saving regulations or recommendations. Being the peak current
very low, no issue of audible noise arises.
Figure 4.
Multi-mode operation with QR option active
fosc
Input voltage
Valley-skipping
mode
f sw
Burst-mode
Quasi-resonant mode
0
0
P in
Pinmax
17/51
Application information
L6566B
If FF operation is selected:
1. FF mode from heavy to light load. The system operates exactly like a standard current
mode control, at a frequency fsw determined by the externally programmable oscillator:
both DCM and CCM transformer operation are possible, depending on whether the
power that it processes is greater or less than:
Equation 1
⎛ Vin VR ⎞
⎟⎟
⎜⎜
⎝ Vin + VR ⎠
Pin T =
2 fsw Lp
2.
2
where Vin is the input voltage to the converter, VR the reflected voltage (i.e. the
regulated output voltage times the primary-to-secondary turn ratio) and Lp the
inductance of the primary winding. PinT is the power level that marks the transition from
continuous to discontinuous operation mode of the transformer.
Burst-mode with no or very light load. This kind of operation is activated in the same
way and results in the same behavior as previously described for QR operation.
The L6566B is specifically designed for applications with no PFC front-end; pin 6 (FMOD)
features an auxiliary oscillator that can modulate the switching frequency (when FF
operation is selected) in order to mitigate EMI emissions by a spread-spectrum action.
5.1
High-voltage start-up generator
Figure 5 shows the internal schematic of the high-voltage start-up generator (HV generator).
It is made up of a high-voltage N-channel FET, whose gate is biased by a 15 MΩ resistor,
with a temperature-compensated current generator connected to its source.
Figure 5.
High-voltage start-up generator: internal schematic
HV
L6566B
1
15 M
Vcc_OK
HV_EN
I HV
5
CONTROL
I charge
3
GND
18/51
Vcc
L6566B
Application information
With reference to the timing diagram of Figure 6, when power is first applied to the converter
the voltage on the bulk capacitor (Vin) builds up and, at about 80 V, the HV generator is
enabled to operate (HV_EN is pulled high) so that it draws about 1 mA. This current, minus
the device’s consumption, charges the bypass capacitor connected from pin Vcc (5) to
ground and makes its voltage rise almost linearly.
Figure 6.
Timing diagram: normal power-up and power-down sequences
Vin
VHVstart
regulation is lost here
Vcc
(pin 5)
t
VccON
VccOFF
Vccrestart
t
GD
(pin 4)
t
HV_EN
t
Vcc_OK
Icharge
t
0.85 mA
Power-on
Normal
operation
Power-off
t
As the Vcc voltage reaches the turn-on threshold (14 V typ.) the device starts operating and
the HV generator is cut off by the Vcc_OK signal asserted high. The device is powered by
the energy stored in the Vcc capacitor until the self-supply circuit (typically an auxiliary
winding of the transformer and a steering diode) develops a voltage high enough to sustain
the operation. The residual consumption of this circuit is just the one on the 15 MΩ resistor
(≈10 mW at 400 Vdc), typically 50-70 times lower, under the same conditions, as compared
to a standard start-up circuit made with external dropping resistors.
At converter power-down the system will lose regulation as soon as the input voltage is so
low that either peak current or maximum duty cycle limitation is tripped. Vcc will then drop
and stop IC activity as it falls below the UVLO threshold (10 V typ.). The Vcc_OK signal is
de-asserted as the Vcc voltage goes below a threshold VCCrest located at about 5V. The HV
generator can now restart. However, if Vin < Vinstart, as illustrated in Figure 6, HV_EN is deasserted too and the HV generator is disabled. This prevents converter’s restart attempts
and ensures monotonic output voltage decay at power-down in systems where brownout
protection (see the relevant section) is not used.
The low restart threshold VCCrest ensures that, during short circuits, the restart attempts of
the device will have a very low repetition rate, as shown in the timing diagram of Figure 7 on
page 20, and that the converter will work safely with extremely low power throughput.
19/51
Application information
Figure 7.
L6566B
Timing diagram showing short-circuit behavior (SS pin clamped at 5 V)
Short circuit occurs here
Vcc
(pin 5)
VccON
Vcc OFF
Vccrestart
Trep
GD
(pin 4)
t
< 0.03Trep
Vcc_OK
t
Icharge
t
0.85 mA
t
Figure 8.
Zero current detection block, triggering block, oscillator block and
related logic
COMP
VFF
9
15
L6566B
ZCD
11
RZ1
+Vin
line
FFWD
BLANKING
TIME
5.7V
PWM
blanking
START
7
CS
4
GD
RZ2
R
100 mV
50 mV
+
TURN-ON
LOGIC
MONO
STABLE
Q
S
OSCILLATOR
Strobe
Rs
Reset
+
-
S/H
4:1
Counter
FAULT
AUXILIARY
OSCILLATOR
5V
6
13
FMOD
OSC
RMOD
CMOD
20/51
Q
DRIVER
RT
L6566B
5.2
Application information
Zero current detection and triggering block; oscillator block
The zero current detection (ZCD) and triggering blocks switch on the external MOSFET if a
negative-going edge falling below 50 mV is applied to the input (pin 11, ZCD). To do so the
triggering block must be previously armed by a positive-going edge exceeding 100 mV.
This feature is typically used to detect transformer demagnetization for QR operation, where
the signal for the ZCD input is obtained from the transformer’s auxiliary winding used also to
supply the L6566B. The triggering block is blanked for TBLANK = 2.5 µs after MOSFET’s
turn-off to prevent any negative-going edge that follows leakage inductance
demagnetization from triggering the ZCD circuit erroneously.
The voltage at the pin is both top and bottom limited by a double clamp, as illustrated in the
internal diagram of the ZCD block of Figure 8 on page 20. The upper clamp is typically
located at 5.7 V, while the lower clamp is located at -0.4 V. The interface between the pin
and the auxiliary winding will be a resistor divider. Its resistance ratio will be properly chosen
(see Section 5.11: OVP block on page 35) and the individual resistance values (RZ1, RZ2)
will be such that the current sourced and sunk by the pin be within the rated capability of the
internal clamps (± 3 mA).
At converter power-up, when no signal is coming from the ZCD pin, the oscillator starts up
the system. The oscillator is programmed externally by means of a resistor (RT) connected
from pin OSC (13) to ground. With good approximation the oscillation frequency fosc will be:
Equation 2
fosc ≈
2 ⋅ 10 3
RT
(with fosc in kHz and RT in kΩ). As the device is turned on, the oscillator starts immediately;
at the end of the first oscillator cycle, being zero the voltage on the ZCD pin, the MOSFET
will be turned on, thus starting the first switching cycle right at the beginning of the second
oscillator cycle. At any switching cycle, the MOSFET is turned off as the voltage on the
current sense pin (CS, 7) hits an internal reference set by the line feedforward block, and the
transformer starts demagnetization. If this completes (hence a negative-going edge appears
on the ZCD pin) after a time exceeding one oscillation period Tosc = 1/fosc from the previous
turn-on, the MOSFET will be turned on again - with some delay to ensure minimum voltage
at turn-on – and the oscillator ramp will be reset. If, instead, the negative-going edge
appears before Tosc has elapsed, it will be ignored and only the first negative-going edge
after Tosc will turn-on the MOSFET and synchronize the oscillator. In this way one or more
drain ringing cycles will be skipped (“valley-skipping mode”, Figure 9) and the switching
frequency will be prevented from exceeding fosc.
21/51
Application information
Figure 9.
L6566B
Drain ringing cycle skipping as the load is gradually reduced
VDS
VDS
VDS
t
TON
TFW
Tosc
Tosc
Pin = Pin'
(limit condition)
Note:
t
t
TV
Tosc
Pin = Pin'' < Pin'
Pin = Pin''' < P in''
When the system operates in valley skipping-mode, uneven switching cycles may be
observed under some line/load conditions, due to the fact that the OFF-time of the MOSFET
is allowed to change with discrete steps of one ringing cycle, while the OFF-time needed for
cycle-by-cycle energy balance may fall in between. Thus one or more longer switching
cycles will be compensated by one or more shorter cycles and vice versa. However, this
mechanism is absolutely normal and there is no appreciable effect on the performance of
the converter or on its output voltage.
If the MOSFET is enabled to turn on but the amplitude of the signal on the ZCD pin is
smaller than the arming threshold for some reason (e.g. a heavy damping of drain
oscillations, like in some single-stage PFC topologies, or when a turn-off snubber is used),
MOSFET’s turn-on cannot be triggered. This case is identical to what happens at start-up:
at the end of the next oscillator cycle the MOSFET will be turned on, and a new switching
cycle will take place after skipping no more than one oscillator cycle.
The operation described so far does not consider the blanking time TBLANK after MOSFET’s
turn off, and actually TBLANK does not come into play as long as the following condition is
met:
Equation 3
D ≤ 1−
TBLANK
Tosc
where D is the MOSFET duty cycle. If this condition is not met, things do not change
substantially: the time during which MOSFET’s turn-on is inhibited is extended beyond Tosc
by a fraction of TBLANK. As a consequence, the maximum switching frequency will be a little
lower than the programmed value fosc and valley-skipping mode may take place slightly
earlier than expected. However this is quite unusual: setting fosc = 150 kHz, the
phenomenon can be observed at duty cycles higher than 60 %. See Section 5.11: OVP
block on page 35 for further implications of TBLANK.
If the voltage on the COMP pin (9) saturates high, which reveals an open control loop, an
internal pull-up keeps the ZCD pin close to 2 V during MOSFET's OFF-time to prevent noise
from false triggering the detection block. When this pull-up is active, the ZCD pin might not
be able to go below the triggering threshold, which would stop the converter. To allow autorestart operation, however ensuring minimum operating frequency in these conditions, the
oscillator frequency that retriggers MOSFET's turn-on is that of the external oscillator
divided by 128. Additionally, to prevent malfunction at converter's start-up, the pull-up is
disabled during the initial soft-start (see the relevant section). However, to ensure a correct
22/51
L6566B
Application information
start-up, at the end of the soft-start phase the output voltage of the converter must meet the
condition:
Equation 4
Vout >
Ns
R Z1 I ZCD
Naux
where Ns is the turn number of the secondary winding, Naux the turn number of the
auxiliary winding and IZCD the maximum pull-up current (130 µA).
The operation described so far under different operating conditions for the converter is
illustrated in the timing diagrams of Figure 10.
If the FF option is selected the operation will be exactly equal to that of a standard currentmode PWM controller. It will work at a frequency fsw = fosc; both DCM and CCM
transformer's operation are possible, depending on the operating conditions (input voltage
and output load) and on the design of the power stage. The MOSFET is turned on at the
beginning of each oscillator cycle and is turned off as the voltage on the current sense pin
reaches an internal reference set by the line feedforward block. The maximum duty cycle is
limited at 70 % minimum. The signal on the ZCD pin in this case is used only for detecting
feedback loop failures (see Section 5.11: OVP block on page 35 ).
Figure 10. Operation of ZCD, triggering and oscillator blocks (QR option active)
ZCD
(pin 11)
ZCD
(pin 11)
100 mV
100 mV
100 mV
50 mV
50 mV
ZCD
(pin 11)
50 mV
Oscillator
ramp
Oscillator
ramp
Oscillator
ramp
ZCD
blanking
time
ZCD
blanking
time
ZCD
blanking
time
Arm/Trigger
Arm/Trigger
Arm/Trigger
ON-enable
ON-enable
ON-enable
PWM latch
Set
PWM latch
Set
PWM latch
Set
PWM latch
Reset
PWM latch
Reset
PWM latch
Reset
GD
(pin 4)
GD
(pin 4)
GD
(pin 4)
armed
trigger
a) full load
b) light load
c) start-up
23/51
Application information
5.3
L6566B
Burst-mode operation at no load or very light load
When the voltage at the COMP pin (9) falls 20 mV below a threshold fixed internally at a
value, VCOMPBM, depending on the selected operating mode, the L6566B is disabled with
the MOSFET kept in OFF state and its consumption reduced at a lower value to minimize
Vcc capacitor discharge.
The control voltage now will increase as a result of the feedback reaction to the energy
delivery stop (the output voltage will be slowly decaying), the threshold will be exceeded and
the device will restart switching again. In this way the converter will work in burst-mode with
a nearly constant peak current defined by the internal disable level. A load decrease will
then cause a frequency reduction, which can go down even to few hundred hertz, thus
minimizing all frequency-related losses and making it easier to comply with energy saving
regulations. This kind of operation, shown in the timing diagrams of Figure 11 along with the
others previously described, is noise-free since the peak current is low.
If it is necessary to decrease the intervention threshold of the burst-mode operation, this can
be done by adding a small DC offset on the current sense pin as shown in Figure 12 on
page 25.
Note:
The offset reduces the available dynamics of the current signal; thereby, the value of the
sense resistor must be determined taking this offset into account.
Figure 11. Load-dependent operating modes: timing diagrams
COMP
(pin 9)
20 mV
hyster.
VCOMPBM
t
fosc
MODE/SC=Open
fsw
MODE/SC=VREF
t
GD
(pin 4)
MODE/SC=Open
MODE/SC=VREF
FF Mode
QR Mode
Burst-mode
Burst-mode
Valley-skipping Mode
24/51
FF Mode
QR Mode
t
L6566B
Application information
Figure 12. Addition of an offset to the current sense lowers the burst-mode
operation threshold
Vcso = Vref
R
R + Rc
Vref
10
4
Rc
L6566B
R
7
3
Rs
5.4
Adaptive UVLO
A major problem when optimizing a converter for minimum no-load consumption is that the
voltage generated by the auxiliary winding under these conditions falls considerably as
compared even to a few mA load. This very often causes the supply voltage Vcc of the
control IC to drop and go below the UVLO threshold so that the operation becomes
intermittent, which is undesired. Furthermore, this must be traded off against the need of
generating a voltage not exceeding the maximum allowed by the control IC at full load.
To help the designer overcome this problem, the device, besides reducing its own
consumption during burst-mode operation, also features a proprietary adaptive UVLO
function. It consists of shifting the UVLO threshold downwards at light load, namely when
the voltage at pin COMP falls below a threshold VCOMPO internally fixed, so as to have more
headroom. To prevent any malfunction during transients from minimum to maximum load
the normal (higher) UVLO threshold is re-established when the voltage at pin COMP
exceeds VCOMPL and Vcc has exceeded the normal UVLO threshold (see Figure 13). The
normal UVLO threshold ensures that at full load the MOSFET will be driven with a proper
gate-to-source voltage.
Figure 13.
Adaptive UVLO block
VCOMP
(pin 9)
Vcc
V COMPL
V COMPO
5
+
-
Vcc
(pin 5)
R
9
-
S
t
UVLO
Q
+
VccOFF1
VccOFF2
+
SW
VCOMPL
VCOMPO
-
COMP
VccOFF1
VccOFF2
(*)
L6566B
(*) VccOFF2 < VccOFF1 is selected when Q is high
Q
t
t
25/51
Application information
5.5
L6566B
PWM control block
The device is specific for secondary feedback. Typically, there is a TL431 on the secondary
side and an optocoupler that transfers output voltage information to the PWM control on the
primary side, crossing the isolation barrier. The PWM control input (pin 9, COMP) is driven
directly by the phototransistor’s collector (the emitter is grounded to GND) to modulate the
duty cycle (Figure 14, left-hand side circuit).
In applications where a tight output regulation is not required, it is possible to use a primarysensing feedback technique. In this approach the voltage generated by the self-supply
winding is sensed and regulated. This solution, shown in Figure 14, right-hand side circuit,
is cheaper because no optocoupler or secondary reference is needed, but output voltage
regulation, especially as a result of load changes, is quite poor.
Figure 14. Possible feedback configurations that can be used with the L6566B
Vout
5
L6566B
Vcc
L6566B
9
COMP
Cs
9
Naux
COMP
TL431
Secondary feedback
Primary feedback
Ideally, the voltage generated by the self-supply winding and the output voltage should be
related by the Naux/Ns turn ratio only. Actually, numerous non-idealities, mainly
transformer's parasites, cause the actual ratio to deviate from the ideal one. Line regulation
is quite good, in the range of ± 2 %, whereas load regulation is about ± 5 % and output
voltage tolerance is in the range of ± 10 %.
The dynamics of the pin is in the 2.5 to 5 V range. The voltage at the pin is clamped
downwards at about 2 V. If the clamp is externally overridden and the voltage on the pin is
pulled below 1.4 V the L6566B will shut down. This condition is latched as long as the
device is supplied. While the device is disabled, however, no energy is coming from the selfsupply circuit, thus the voltage on the Vcc capacitor will decay and cross the UVLO
threshold after some time, which clears the latch and lets the HV generator restart. This
function is intended for an externally controlled burst-mode operation at light load with a
reduced output voltage, a technique typically used in multi-output SMPS, such as those for
TVs or monitors (see the timing diagram Figure 15 on page 27).
26/51
L6566B
Application information
Figure 15. Externally controlled burst-mode operation by driving pin COMP: timing
diagram
Vcc
(pin 5)
VccON
Standby is commanded here
VccOFF
Vccrestart
COMP
(pin 9)
t
GD
(pin 4)
t
Vcc_OK
Icharge
0.85 mA
Vout
t
t
t
t
5.6
PWM comparator, PWM latch and voltage feedforward blocks
The PWM comparator senses the voltage across the current sense resistor Rs and, by
comparing it to the programming signal delivered by the feedforward block, determines the
exact time when the external MOSFET is to be switched off. Its output resets the PWM
latch, previously set by the oscillator or the ZCD triggering block, which will assert the gate
driver output low. The use of PWM latch avoids spurious switching of the MOSFET that
might result from the noise generated (“double-pulse suppression”).
Cycle-by-cycle current limitation is realized with a second comparator (OCP comparator)
that senses the voltage across the current sense resistor Rs as well and compares this
voltage to a reference value Vcsx. Its output is or-ed with that of the PWM comparator (see
the circuit schematic in Figure 17 on page 29). In this way, if the programming signal
delivered by the feedforward block and sent to the PWM comparator exceeds Vcsx, it will be
the OCP comparator to reset first the PWM latch instead of the PWM comparator. The value
of Vcsx, thereby, determines the overcurrent setpoint along with the sense resistor Rs.
The power that QR flyback converters with a fixed overcurrent setpoint (like fixed-frequency
systems) are able to deliver changes with the input voltage considerably. With wide-range
mains, at maximum line it can be more than twice the value at minimum line, as shown by
the upper curve in the diagram of Figure 16 on page 28. The device has the line feedforward
function available to solve this issue.
It acts on the overcurrent setpoint Vcsx, so that it is a function of the converter’s input voltage
Vin sensed through a dedicated pin (15, VFF): the higher the input voltage, the lower the
27/51
Application information
L6566B
setpoint. This is illustrated in the diagram on the left-hand side of Figure 17 on page 29: it
shows the relationship between the voltage on the pin VFF and Vcsx (with the error amplifier
saturated high in the attempt of keeping output voltage regulation):
Equation 5
Vcsx = 1 −
VVFF
k
= 1 − Vin
3
3
Figure 16. Typical power capability change vs input voltage in QR flyback
converters
2.5
k=0
system not
compensated
2
[email protected] V inlim
P
k
1.5
1
system optimally
compensated
k = kopt
0.5
1
1.5
2
2.5
3
3.5
4
Vin
Vinmin
Note:
If the voltage on the pin exceeds 3 V switching ceases but the soft-start capacitor is not
discharged. The schematic in Figure 17 on page 29 shows also how the function is included
in the control loop.
With a proper selection of the external divider R1-R2, i.e. of the ratio k = R2 / (R1+R2), it is
possible to achieve the optimum compensation described by the lower curve in the diagram
of Figure 16.
The optimum value of k, kopt, which minimizes the power capability variation over the input
voltage range, is the one that provides equal power capability at the extremes of the range.
The exact calculation is complex, and non-idealities shift the real-world optimum value from
the theoretical one. It is therefore more practical to provide a first cut value, simple to be
calculated, and then to fine tune experimentally.
Assuming that the system operates exactly at the boundary between DCM and CCM, and
neglecting propagation delays, the following expression for kopt can be found:
Equation 6
k opt = 3 ⋅
28/51
Vin min ⋅ Vin max
VR
+ (Vin min + Vin max ) ⋅ VR
L6566B
Application information
Experience shows that this value is typically lower than the real one. Once the maximum
peak primary current, IPKpmax, occurring at minimum input voltage Vinmin has been found,
the value of Rs can be determined from (5):
Equation 7
Rs =
k opt
1−
3
Vin min
IPKp max
Figure 17. Left: overcurrent setpoint vs VFF voltage; right: line feedforward function block
Rectif ied Line Voltage
Vcsx [V]
1.2
Optional for
OVP settings
R1
VCOMP = Upper clamp
1
0.8
R2
Rs
VFF
CS
15
COMP
0.2
7
VOLTAGE
FEED
FORWARD
9
PWM
-
0.4
+
0.6
4
R
Q
DRIVER
+
S
OCP
GD
-
Vcsx
0.5
1
1.5
2
VVFF [V]
2.5
3
3.5
Hiccup
L6566B
1.5 V
DISABLE
-
0
Clock/ZCD
+
0
The converter is then tested on the bench to find the output power level Poutlim where
regulation is lost (because overcurrent is being tripped) both at Vin = Vinmin and
Vin = Vinmax.
If Poutlim @ Vinmax > Poutlim @ Vinmin the system is still undercompensated and k needs
increasing; if Poutlim @ Vinmax < Poutlim @ Vinmin the system is overcompensated and k
needs decreasing. This will go on until the difference between the two values is acceptably
low. Once found the true kopt in this way, it is possible that Poutlim turns out slightly different
from the target; to correct this, the sense resistor Rs needs adjusting and the above tuning
process will be repeated with the new Rs value. Typically a satisfactory setting is achieved in
no more than a couple of iterations.
In applications where this function is not wanted, e.g. because of a narrow input voltage
range, the VFF pin can be simply grounded, directly or through a resistor (see “Section 5.11:
OVP block on page 35”). The overcurrent setpoint will be then fixed at the maximum value of
1V. If a lower setpoint is desired to reduce the power dissipation on Rs, the pin can be also
biased at a fixed voltage using a divider from VREF (pin 10).
If the FF option is selected the line feedforward function can be still used to compensate for
the total propagation delay Td of the current sense chain (internal propagation delay td(H-L)
plus the turn-off delay of the external MOSFET), which in standard current mode PWM
controllers is done by adding an offset on the current sense pin proportional to the input
voltage. In that case the divider ratio k, which will be much smaller as compared to that used
with the QR option selected, can be calculated with the following equation:
29/51
Application information
L6566B
Equation 8
k opt = 3
Td
Rs Lp
where Lp is the inductance of the primary winding. In case a constant maximum power
capability vs. the input voltage is not required, the VFF pin can be grounded, directly or
through a resistor (see Section 5.11: OVP block on page 35), hence fixing the overcurrent
setpoint at 1 V, or biased at a fixed voltage through a divider from VREF to get a lower
setpoint.
It is possible to bypass the pin to ground with a small film capacitor (e.g. 1-10 nF) to ensure
a clean operation of the IC even in a noisy environment.
The pin is internally forced to ground during UVLO, after activating any latched protection
and when pin COMP is pulled below its low clamp voltage (see Section 5.5: PWM control
block on page 26).
5.7
Hiccup-mode OCP
A third comparator senses the voltage on the current sense input and shuts down the device
if the voltage on the pin exceeds 1.5 V, a level well above that of the maximum overcurrent
setpoint (1 V). Such an anomalous condition is typically generated by either a short circuit of
the secondary rectifier or a shorted secondary winding or a hard-saturated flyback
transformer.
Figure 18. Hiccup-mode OCP: timing diagram
Vcc
(pin 5)
Secondary diode is shorted here
VccON
Vcc OFF
Vcc restart
VCS
(pin 7)
GD
(pin 4)
1.5 V
t
t
OCP latch
t
Vcc_OK
t
t
To distinguish an actual malfunction from a disturbance (e.g. induced during ESD tests), the
first time the comparator is tripped the protection circuit enters a “warning state”. If in the
next switching cycle the comparator is not tripped, a temporary disturbance is assumed and
the protection logic will be reset in its idle state; if the comparator will be tripped again a real
malfunction is assumed and the L6566B will be stopped. Depending on the time relationship
30/51
L6566B
Application information
between the detected event and the oscillator, occasionally the device could stop after the
third detection.
This condition is latched as long as the device is supplied. While it is disabled, however, no
energy is coming from the self-supply circuit; hence the voltage on the Vcc capacitor will
decay and cross the UVLO threshold after some time, which clears the latch. The internal
start-up generator is still off, then the Vcc voltage still needs to go below its restart voltage
before the Vcc capacitor is charged again and the device restarted. Ultimately, this will result
in a low-frequency intermittent operation (Hiccup-mode operation), with very low stress on
the power circuit. This special condition is illustrated in the timing diagram of Figure 18 on
page 30.
5.8
Frequency modulation
To alleviate converter’s EMI emissions and reduce cost and size of the line filter, it is
advantageous to modulate its switching frequency, so that the resulting spread-spectrum
action distributes the energy of each harmonic of the switching frequency over a number of
side-band harmonics. Their overall energy will be unchanged but the individual amplitudes
will be smaller. This is what naturally occurs with QR operation, due to the twice-mainsfrequency ripple appearing on the input bulk capacitor, which translates into different DCMCCM boundary frequencies.
The L6566B is provided with a dedicated pin, FMOD (6), to perform this function if FF mode
is selected.
Figure 19. Frequency modulation circuit
L6566B
13
6
1V
1.5 V
OSC
FMOD
RMOD
0V
RT
0.5 V
CMOD
With reference to Figure 19, the capacitor CMOD is connected from FMOD to ground and is
alternately charged and discharged between 0.5 and 1.5 V by internal current generators
sourcing and sinking the same current (three times the current defined by the resistor RT on
pin OSC). Hence, the voltage across CMOD will be a symmetric triangle, whose frequency fm
is determined by CMOD. By connecting a resistor RMOD from FMOD to OSC, the current
sourced by pin OSC will be modulated according a triangular profile at a frequency fm. If
RMOD is considerably higher than RT, as normally is, both fm and the symmetry of the
triangle will be little affected.
With this arrangement it is possible to set, nearly independently, the frequency deviation
∆fsw and the modulating frequency fm, which define the modulation index:
31/51
Application information
L6566B
Equation 9
β=
∆fsw
fm
which is the parameter that the amplitude of the generated side-band harmonics depends
on.
The minimum frequency fsw_min (occurring on the peak of the triangle) and the maximum
frequency fsw_max (occurring on the valley of the triangle) will be symmetrically placed
around the centre value fsw, so that:
Equation 10
fsw _ min = fsw − 21 ∆fsw ;
fsw _ max = fsw + 21 ∆fsw
Then, RT will be found from (5) (see Section 5.2: Zero current detection and triggering block;
oscillator block on page 21), while RMOD and CMOD can be calculated as follows:
Equation 11
R MOD =
2 ⋅ 10 3
∆fsw
C MOD =
75
fm
where ∆fsw and fm (in kHz, with CMOD in nF and RMOD in kΩ) will be selected by the user so
to achieve the best compromise between attenuation of peak EMI emissions and clean
converter operation.
5.9
Latched disable function
The device is equipped with a comparator having the non-inverting input externally available
at the pin DIS (8) and with the inverting input internally referenced to 4.5 V. As the voltage
on the pin exceeds the internal threshold, the device is immediately shut down and its
consumption reduced to a low value.
The information is latched and it is necessary to let the voltage on the Vcc pin go below the
UVLO threshold to reset the latch and restart the device. To keep the latch supplied as long
as the converter is connected to the input source, the HV generator is activated periodically
so that Vcc oscillates between the start-up threshold VccON and VccON - 0.5 V. Activating the
HV generator in this way cuts its power dissipation approximately by three (as compared to
the case of continuous conduction) and keeps peak silicon temperature close to the average
value.
To let the L6566B restart it is then necessary to disconnect the converter from the input
source. Pulling pin 16 (AC_OK) below the disable threshold (see Section 5.12: Brownout
protection on page 37) will stop the HV generator until Vcc falls below Vccrestart, so that the
latch can be cleared and a quicker restart is allowed as the input source is removed. This
operation is shown in the timing diagram of Figure 20 on page 33.
32/51
L6566B
Application information
This function is useful to implement a latched overtemperature protection very easily by
biasing the pin with a divider from VREF, where the upper resistor is an NTC physically
located close to a heating element like the MOSFET, or the transformer. The DIS pin is a
high impedance input, thus it is prone to pick up noise, which might give origin to undesired
latch-off of the device. It is possible to bypass the pin to ground with a small film capacitor
(e.g. 1-10 nF) to prevent any malfunctioning of this kind.
Figure 20. Operation after latched disable activation: timing diagram
DIS
(pin 8)
4.5V
Vcc
(pin 5)
HV generator is turned on
Restart is quicker
t
Vcc ON
VccON -0.5
VccOFF
Disable latch is reset here
Vccrestart
GD
(pin 4)
HV generator turn-on is disabled here
Input source is removed here
Vin
t
t
VHVstart
t
AC_OK
(pin 16)
Vth
t
5.10
Soft-start and delayed latched shutdown upon overcurrent
At device start-up, a capacitor (CSS) connected between the SS pin (14) and ground is
charged by an internal current generator, ISS1, from zero up to about 2 V where it is
clamped. During this ramp, the overcurrent setpoint progressively rises from zero to the
value imposed by the voltage on the VFF pin (15, see Section 5.6: PWM comparator, PWM
latch and voltage feedforward blocks on page 27); MOSFET’s conduction time increases
gradually, hence controlling the start-up inrush current. The time needed for the overcurrent
setpoint to reach its steady state value, referred to as soft-start time, is approximately:
Equation 12
TSS =
V
⎞
Css
Css ⎛
⎜1 − VFF ⎟⎟
Vcsx (VVFF ) =
ISS1
ISS1 ⎜⎝
3 ⎠
During the ramp (i.e. until VSS = 2 V) all the functions that monitor the voltage on pin COMP
are disabled.
33/51
Application information
L6566B
The soft-start pin is also invoked whenever the control voltage (COMP) saturates high,
which reveals an open-loop condition for the feedback system. This condition very often
occurs at start-up, but may be also caused by either a control loop failure or a converter
overload/short circuit. A control loop failure results in an output overvoltage that is handled
by the OVP function of the L6566B (see next section). In case of QR operation, a short
circuit causes the converter to run at a very low frequency, then with very low power
capability. This makes the self-supply system that powers the device unable to keep it
operating, so that the converter will work intermittently, which is very safe. In case of
overload the system has a power capability lower than that at nominal load but the output
current may be quite high and overstress the output rectifier. In case of FF operation the
capability is almost unchanged and both short circuit and overload conditions are more
critical to handle.
The L6566B, regardless of the operating option selected, makes it easier to handle such
conditions: the 2 V clamp on the SS pin is removed and a second internal current generator
ISS2 = ISS1 /4 keeps on charging CSS. As the voltage reaches 5 V the device is disabled, if it
is allowed to reach 2 VBE over 5 V, the device will be latched off. In the former case the
resulting behavior will be identical to that under short circuit illustrated in Figure 7 on
page 20; in the latter case the result will be identical to that of Figure 20 on page 33. See
Section 5.9: Latched disable function on page 32 for additional details.
Figure 21. Soft-start pin operation under different operating conditions and settings
Vcc
(pin 5)
UVLO
Vcc falls below UVLO
before latching off
SS
(pin 14)
t
5V+2Vbe
5V
here the IC
shuts down
2V
COMP
(pin 9)
here the IC
latches off
t
GD
(pin 4)
t
t
START-UP
NORMAL
OPERATION
TEMPORARY
OVERLOAD
NORMAL
OPERATION
OVERLOAD
SHUTDOWN
RESTART
LATCHED
AUTORESTART
A diode, with the anode to the SS pin and the cathode connected to the VREF pin (10) is the
simplest way to select either auto-restart mode or latch-mode behavior upon overcurrent. If
the overload disappears before the Css voltage reaches 5 V the ISS2 generator will be
turned off and the voltage gradually brought back down to 2 V. Refer to the “Application
examples and Ideas” section (Table 7 on page 45) for additional hints.
If latch-mode behavior is desired also for converter’s short circuit, make sure that the supply
voltage of the device does not fall below the UVLO threshold before activating the latch.
Figure 21 shows soft-start pin behavior under different operating conditions and with
different settings (latch-mode or autorestart).
Note:
34/51
Unlike other PWM controllers provided with a soft-start pin, in the L6566B grounding the SS
pin does not guarantee that the gate driver is disabled.
L6566B
5.11
Application information
OVP block
The OVP function of the L6566B monitors the voltage on the ZCD pin (11) in MOSFET’s
OFF-time, during which the voltage generated by the auxiliary winding tracks converter’s
output voltage. If the voltage on the pin exceeds an internal 5 V reference, a comparator is
triggered, an overvoltage condition is assumed and the device is shut down. An internal
current generator is activated that sources 1 mA out of the VFF pin (15). If the VFF voltage
is allowed to reach 2 Vbe over 5 V, the L6566B will be latched off. See Section 5.9: Latched
disable function on page 32 for more details on IC’s behavior under these conditions. If the
impedance externally connected to pin 15 is so low that the 5+2 VBE threshold cannot be
reached or if some means is provided to prevent that, the device will be able to restart after
the Vcc has dropped below 5 V. Refer to the “Application examples and Ideas” section
(Table 7 on page 45) for additional hints.
Figure 22. OVP function: internal block diagram
ZCD
11
to triggering
block
L6566B
5V
40kΩ
+
PWM latch
R
Q
S
Q
COUT
5pF
OVP
Monostable
M1
Monostable
M2
2 µs
2-bit
counter
STROBE
0.5 µs
FF
R Q1
Fault
Counter
reset
S
The ZCD pin will be connected to the auxiliary winding through a resistor divider RZ1, RZ2
(see Figure 8 on page 20). The divider ratio kOVP = RZ2 / (RZ1 + RZ2) will be chosen equal
to:
Equation 13
k OVP =
5
Ns
Vout OVP Naux
where VoutOVP is the output voltage value that is to activate the protection, Ns the turn
number of the secondary winding and Naux the turn number of the auxiliary winding.
35/51
Application information
L6566B
Figure 23. OVP function: timing diagram
GD
(pin 4)
t
Vaux
0
ZCD
(pin 11)
t
5V
t
COUT
2 µs
STROBE
t
0.5 µs
t
OVP
t
COUNTER
RESET
COUNTER
STATUS
t
0
0
0
0 →1
1 →2
2 →0
0
0 →1
1 →2
2 →3
3 →4
FAULT
t
NORMAL OPERATION
TEMPORARY DISTURBANCE
FEEDBACK LOOP FAILURE
t
The value of RZ1 will be such that the current sourced by the ZCD pin be within the rated
capability of the internal clamp:
Equation 14
R Z1 ≥
1
3 ⋅ 10
−3
Naux
Vin max
Np
where Vinmax is the maximum dc input voltage and Ns the turn number of the primary
winding. See Section 5.2: Zero current detection and triggering block; oscillator block on
page 21 for additional details.
To reduce sensitivity to noise and prevent the latch from being erroneously activated, first
the OVP comparator is active only for a small time window (typically, 0.5 µs) starting 2 µs
after MOSFET’s turn-off, to reject the voltage spike associated to the positive-going edges
of the voltage across the auxiliary winding Vaux; second, to stop the L6566B the OVP
comparator must be triggered for four consecutive switching cycles. A counter, which is
reset every time the OVP comparator is not triggered in one switching cycle, is provided to
this purpose.
Figure 22 on page 35 shows the internal block diagram, while the timing diagrams in
Figure 23 illustrate the operation.
Note:
36/51
To use the OVP function effectively, i.e. to ensure that the OVP comparator will be always
interrogated during MOSFET’s OFF-time, the duty cycle D under open-loop conditions must
fulfill the following inequality:
L6566B
Application information
Equation 15
D + TBLANK 2 fsw ≤ 1
where TBLANK2 = 2 µs; this is also illustrated in the diagram of Figure 24.
Figure 24. Maximum allowed duty cycle vs switching frequency for correct OVP
detection
0.8
0.725
0.7
0.6
Dmax
0.5
0.4
0.3
0.2
5 .10
4
1 . 10
5
1.5 . 10
5
2 . 10
5
2.5 .10
5
3 . 10
5
3.5 .10
5
4 . 10
5
fsw [Hz]
5.12
Brownout protection
Brownout protection is basically a not-latched device shutdown function activated when a
condition of mains undervoltage is detected. There are several reasons why it may be
desirable to shut down a converter during a brownout condition, which occurs when the
mains voltage falls below the minimum specification of normal operation.
Firstly, a brownout condition may cause overheating of the primary power section due to an
excess of RMS current. Secondly, spurious restarts may occur during converter power
down, hence causing the output voltage not to decay to zero monotonically.
L6566B shutdown upon brownout is accomplished by means of an internal comparator, as
shown in the block diagram of Figure 25 on page 38, which shows the basic usage. The
inverting input of the comparator, available on the AC_OK pin (16), is supposed to sense a
voltage proportional to the RMS (peak) mains voltage; the non-inverting input is internally
referenced to 0.485 V with 35 mV hysteresis. If the voltage applied on the AC_OK pin before
the device starts operating does not exceed 0.485 V or if it falls below 0.45 V while the
device is running, the AC_FAIL signal goes high and the device shuts down, with the softstart capacitor discharged and the gate-drive output low. Additionally, if the device has been
latched off by some protection function (testified by Vcc oscillating between VccON and
VccON - 0.5 V) the AC_OK voltage falling below 0.45 V clears the latch. This may allow a
quicker restart as the input source is removed.
While the brownout protection is active the start-up generator keeps on working but, being
there no PWM activity, the Vcc voltage continuously oscillates between the start-up and the
HV generator restart thresholds, as shown in the timing diagram of Figure 25.
37/51
Application information
L6566B
Figure 25. Brownout protection: internal block diagram and timing diagram
Sensed voltage
VsenON
VsenOFF
VAC_OK
(pin 16)
t
0.485V
0.45V
Sensed
voltage
t
Vcc
AC_FAIL
5
L6566B
t
IHYS
RH
15 µA
AC_OK
16
15 µA
0.485V
0.45V
AC_FAIL
t
Vcc
(pin 5)
+
RL
t
GD
(pin 4)
t
Vout
t
The brownout comparator is provided with current hysteresis in addition to voltage
hysteresis: an internal 15 µA current sink is ON as long as the voltage applied on the
AC_OK pin is such that the AC_FAIL signal is high. This approach provides an additional
degree of freedom: it is possible to set the ON threshold and the OFF threshold separately
by properly choosing the resistors of the external divider (see below). With just voltage
hysteresis, instead, fixing one threshold automatically fixes the other one depending on the
built-in hysteresis of the comparator.
With reference to Figure 25, the following relationships can be established for the ON
(VsenON) and OFF (VsenOFF) thresholds of the sensed voltage:
Equation 16
Vsen ON − 0.485
0.485
= 15 ⋅ 10 − 6 +
RH
RL
Vsen OFF − 0.45 0.45
=
RH
RL
which, solved for RH and RL, yield:
Equation 17
RH =
38/51
Vsen ON − 1.078 ⋅ Vsen OFF
15 ⋅ 10
−6
;
RL = RH
0.45
Vsen OFF − 0.45
L6566B
Application information
Figure 26. Voltage sensing techniques to implement brownout protection with the
L6566B
AC mains (L/N)
HV Input bus
RH
AC mains (N/L)
AC_OK
RH
16
RL1
RL
16
RH
AC_OK
RL1
L6566B
VFF
15
RL2
15
RL2
L6566B
VFF
RL
CF
Optionalfor
OVPsettings
a)
Optionalfor
OVP settings
b)
It is typically convenient to use a single divider to bias both the AC_OK and the VFF pins, as
shown in Figure 26: this is possible because in all practical cases the voltage on the VFF pin
is lower than that on the AC_OK pin. Once RH and RL have been found as suggested above,
and kopt, either calculated from (6) or (8) or experimentally found, RL will be split as:
Equation 18
R L 2 = k opt ( R L + R H ) ; R L1 = R L − R L 2
Circuit a) senses the input voltage bus (across the bulk capacitor, downstream the bridge
rectifier); in this case, for a proper operation of the brownout function, VsenON must be lower
than the peak voltage at minimum mains and VsenOFF lower than the minimum voltage on
the input bulk capacitor at minimum mains and maximum load considering, in case, holdup
requirements during mains missing cycles as well. Brownout level will be load-dependent. In
case of latched shutdown, when the input source is removed it is necessary to wait until the
bulk capacitor voltage falls below the start voltage of the HV generator VHVstart in order for
the unit to restart, which may take even several seconds.
Circuit b) senses the mains voltage directly, upstream the bridge rectifier. It can be
configured either for half-wave sensing (only the line/neutral wire is sensed) or full-wave
sensing (both neutral and line are sensed); in the first case, assuming CF is large enough,
the sensed voltage will be equal to 1/π the peak mains voltage, while in the second case it
will be equal to 2/π the peak mains voltage. CF needs to be quite a big capacitor (in the uF)
to have small residual ripple superimposed on the dc level; as a rule-of-thumb, use a time
constant RL ·CF at least 4-5 times the maximum line cycle period in case of half-wave
sensing, 2-3 times in case of full-wave sensing. Then fine tune if needed, considering also
transient conditions such as mains missing cycles. Brownout level will not depend on the
load. When the input source is removed CF will be discharged after some ten ms then this
circuit is suitable to have a quick restart after a latched shutdown.
The AC_OK pin is a high impedance input connected to high value resistors, thus it is prone
to pick up noise, which might alter the OFF threshold when the converter is running or give
origin to undesired switch-off of the device during ESD tests. It is possible to bypass the pin
to ground with a small film capacitor (e.g. 1-10 nF) to prevent any malfunctioning of this kind.
The voltage on the pin is clamped upwards at about 3.15 V; then, if the function is not used
the pin has to be connected to Vcc through a resistor (220 to 680 kΩ).
39/51
Application information
5.13
L6566B
Slope compensation
The pin MODE/SC (12), when not connected to VREF, provides a voltage ramp during
MOSFET’s ON-time synchronous to that of the internal oscillator sawtooth, with 0.8 mA
minimum current capability. This ramp is intended for implementing additive slope
compensation on current sense. This is needed to avoid the sub-harmonic oscillation that
arises in all peak-current-mode-controlled converters working at fixed frequency in
continuous conduction mode with a duty cycle close to or exceeding 50 %.
Figure 27. Slope compensation waveforms
Internal
oscillator
GD
(pin 4)
t
MODE/SC
(pin 12)
t
t
The compensation will be realized by connecting a programming resistor between this pin
and the current sense input (pin 7, CS). The CS pin has to be connected to the sense
resistor with another resistor to make a summing node on the pin. Since no ramp is
delivered during MOSFET OFF-time (see Figure 27), no external component other than the
programming resistor is needed to ensure a clean operation at light loads.
Note:
The addition of the slope compensation ramp will reduce the available dynamics of the
current signal; thereby, the value of the sense resistor must be determined taking this into
account. Note also that the burst-mode threshold (in terms of power) will be slightly
changed.
If slope compensation is not required with FF operation, the pin shall be left floating.
40/51
L6566B
5.14
Application information
Summary of L6566B power management functions
It has been seen that the device is provided with a number of power management functions:
multiple operating mode upon loading conditions and protection functions. To help the
designer familiarize with these functions, in the following tables all of theme are summarized
with their respective activation mechanism and the resulting status of the most important
pins. This can be useful not only for a correct use of the IC but also for diagnostic purposes:
especially at prototyping/debugging stage, it is quite common to bump into unwanted
activation of some function, and these tables can be used as a sort of quick troubleshooting
guide.
Table 5.
L6566B light load management features
Feature Description
Burst
mode
Caused
by
IC
behavior
Controlled
ON-OFF
Pulse
operation for VCOMP <
low power VCOMPBM skipping
operation
consumptio
- Hys
n at light
load
Vcc_restart Consump. VREF
(V)
(Iqdis,mA)
(V)
N.A.
1.34 mA
5
SS
unchanged
VCOMP
OSC
(V)
(V)
VCOMPBM
-HYS to
VCOMPBM
0/1
FMOD
0
41/51
Application information
Table 6.
Protection
OVP
L6566B protections
Description
Caused by
Vcc
IC Iq VREF
IC
restart
behavior
(mA) (V)
(V)
SS
VCOMP
OSC
(V)
(V)
FMOD
VFF
VZCD>VZCDt
h for 4
Auto
consecutive
Output
restart(1)
overvoltage switching
protection cycles
5
2.2
5(6)
unchanged
(6)
0
0
0
unchanged
VFF >
VFFlatch
Latched
13.5
0.33
0
0
0
0
0
0
Auto
restart(2)
5
1.46
5(6)
0
0
unchanged
Latched
13.5
0.33
0
0
0
0
Auto
restart
5
1.46
0
0
unchanged
Latched
13.5
0.33
0
0
0
0
0
0
VCOMP
=VCOMPHi
OLP
L6566B
Output
overload
protection
VSS >
VSSDIS
VCOMP
=VCOMPHi
VSS >
VSSLAT
VCOMP
=VCOMPHi
VSS >
Output short VSSDIS(4)
Short circuit
circuit
protection
protection VCOMP
=VCOMPHi
VSS >
VSSLAT(6)
VSS
VCOMPHi
(6)
<VSSLAT(3)
0
0
VSS
VCOMPHi
(5)
<VSSLAT(6)
Transformer
saturation or
shorted
secondary
diode
protection
VCS >
VCSDIS
for 2-3
consecutive
switching
cycles
Latched
5
0.33
0
0
0
0
0
0
Externally
settable
overtempera
ture
protection
VDIS>VOTP
Latched
13.5
0.33
0
0
0
0
0
0
Internal
thermal
shutdown
Tj > 160oC
Auto
restart(5)
5
0.33
0
0
0
0
0
0
Brownout
Mains
undervoltag
e protection
VAC_OK <
Vth
Auto
restart
5
0.33
0
0
0
0
0
unchanged
Reference
drift
VREF drift
protection
VREF > Vov
Latched
13.5
0.33
0
0
0
0
0
0
Shutdown1
Gate driver
disable
VFF > Voff
Auto
restart
5
2.5
5
unchanged
unchang
ed
1
uncha
nged
unchanged
2nd OCP
OTP
42/51
L6566B
Table 6.
Application information
L6566B protections (continued)
Protection
Description
Caused by
Shutdown2
Shutdown
by VCOMP
low
VCOMP <
VCOMPOFF
ADAPTIVE
UVLO
Shutdown
by Vcc going
below Vccoff
(lowering of
Vccoff
threshold at
light load)
Vcc < 9.4V
(VCOMP >
VCOMPL)
Vcc < 7.2V
(VCOMP >
VCOMPO)
Vcc
IC Iq VREF
IC
restart
behavior
(mA) (V)
(V)
SS
VCOMP
OSC
(V)
(V)
FMOD
VFF
Latched
10
0.33
0
0
0
0
0
0
Auto
restart
5V
0.18
mA
0
0
0
0
0
0
1. Use One external diode from VFF (#15) to AC_OK (#16), cathode to AC_OK
2. Use one external diode from SS (#14) to VREF (#10), cathode to VREF
3. If Css and the Vcc capacitor are such that Vcc falls below UVLO before latch tripping (Figure 21 on page 34)
4. If Css and the Vcc capacitor are such that the latch is tripped before Vcc falls below UVLO (Figure 21 on page 34)
5. When TJ < 110 oC
6. Discharged to zero by Vcc going below UVLO
It is worth reminding that “Auto-restart” means that the device will work intermittently as long
as the condition that is activating the function is not removed; “Latched” means that the
device is stopped as long as the unit is connected to the input power source and the unit
must be disconnected for some time from the source in order for the device (and the unit) to
restart. Optionally, a restart can be forced by pulling the voltage of pin 16 (AC_OK) below
0.45 V.
43/51
Application examples and ideas
6
L6566B
Application examples and ideas
Figure 28. Typical low-cost application schematic
F1
fuse
Vin
88 to
264 Vac
NTC1
B1
CY1
CX1
T1
CX2
C1
Lx
R1
C2
CY2
D4
Vout
D1
C8A,B
R2
R3 470k
D2
C3
AC_OK
FMOD
DIS
6
1
C7 2.2 nF Y1
Vcc
HVS
16
5
8
VFF
4
R4
GD
D3
L6566B
12
10
MODE/SC
13
VREF
7
14
OSC
C4 R6
Optional f or
QR operation
R7
Q1
IC1
15
9
3
COMP
SS
C5
11
4
ZCD
R5
R9
GND
3
Optional f or
QR operation
C6
1
IC3
CS
2
C9
R8
TL431
R10
Figure 29. Typical full-feature application schematic (QR operation)
F1
fuse
Vin
88 to
264 Vac
NTC1
B1
CY1
CX1
T1
CX2
C1
R1
Lx
C2
D4
CY2
Vout
D1
C8A,B
R2
D2
1N4148
C3
R15
FMOD
AC_OK
HVS
6
1
11
5
16
4
R14
VFF
DIS
C7 2.2 nF Y1
Vcc
ZCD
R3
R4
GD
15
R7
Q1
IC1
D3 1N4148
L6566B
7
IC3 PC817A
CS
8
1
4
10
6
12
13
14
9
R5
3
R9
R18
R13
NTC2
R12
C4
VREF
MODE/SC
R6
OSC
C5
SS
COMP
C6
GND
3
2
C9
R8
TL431
R10
44/51
L6566B
Application examples and ideas
Figure 30. Typical full-feature application schematic (FF operation)
F1
fuse
Vin
88 to
264 Vac
NTC1
B1
CY1
CX1
T1
CX2
C1
R1
Lx
C2
D4
CY2
Vout
D1
C8A,B
R2
D2
1N4148
R16
C3
R15
MODE/SC
AC_OK
12
1
DIS
15
L6566B
6
VREF
13
FMOD
R11
R7
Q1
D3 1N4148
IC3 PC817A
CS
7
8
NTC2
R12
R4
GD
4
IC1
10
R13
R3
ZCD
11
5
16
R14
VFF
C7 2.2 nF Y1
Vcc
HVS
14
OSC
9
SS
R17
3
COMP
GND
1
4
R5
R9
R18
3
2
C9
R8
C10
C4
R6
C5
C6
TL431
R10
Table 7.
External circuits that determine IC behavior upon OVP and OCP
OVP latched
OVP auto-restart
SS
RH
AC_OK
OCP latched
VREF
14
RL2
L6566B
15
VFF
RL2
VREF
14
RL2
VFF
15
AC_OK
VREF
14
10
16
L6566B
RFF
SS
RH
10
16
RL1
3.3 + 1.3 ⋅ 10 −3 RL1
≥ 6. 4
R
1 + L1
RL 2
1N4148
SS
AC_OK
15
Diode needed if
1N4148
OCP auto-restart
10
L6566B
RL1
RFF ≈ 10 RL2
RH
VREF
14
16
RFF
VFF
AC_OK
10
16
RL1
SS
RH
L6566B
RL1
RL2
VFF
15
Diode needed if
3.3 + 1.3 ⋅ 10 −3 RL1
≥ 6.4
R
1 + L1
RL 2
45/51
Application examples and ideas
L6566B
Figure 31. Frequency foldback at light load (FF operation)
R1
MODE/SC
R2
Vref
10
12
COMP
L6566B
9
BC857C
13
OSC
RT
Figure 32. Latched shutdown upon mains overvoltage
Vin
Vin
BC857
Vcc
BC847
5
Vref
L6566B
8
DIS
15
VFF
DIS
8
>10 Rq
46/51
10
L6566B
Rq
15
VFF
L6566B
7
Package mechanical data
Package mechanical data
In order to meet environmental requirements, ST offers these devices in ECOPACK®
packages. These packages have a lead-free second level interconnect. The category of
second Level Interconnect is marked on the package and on the inner box label, in
compliance with JEDEC Standard JESD97. The maximum ratings related to soldering
conditions are also marked on the inner box label. ECOPACK is an ST trademark.
ECOPACK specifications are available at: www.st.com.
47/51
Package mechanical data
Table 8.
L6566B
SO16N mechanical data
mm.
inch
Dim.
Min
Typ
A
a1
Min
Typ
1.75
0.1
Max
0.069
0.25
a2
0.004
0.009
1.6
0.063
b
0.35
0.46
0.014
0.018
b1
0.19
0.25
0.007
0.010
C
0.5
c1
0.020
45°
(typ.)
D
9.8
10
0.386
0.394
E
5.8
6.2
0.228
0.244
e
1.27
0.050
e3
8.89
0.350
F
3.8
4.0
0.150
0.157
G
4.60
5.30
0.181
0.208
L
0.4
1.27
0.150
0.050
M
S
Figure 33. Package dimensions
48/51
Max
0.62
0.024
8°(max.)
L6566B
8
Order codes
Order codes
Table 9.
Order codes
Order codes
Package
Packaging
L6566B
SO16N
Tube
L6566BTR
SO16N
Tape and reel
49/51
Revision history
9
L6566B
Revision history
Table 10.
50/51
Document revision history
Date
Revision
Changes
20-Aug-2007
1
First release
29-May-2008
2
Updated Figure 29 on page 44, Table 2 on page 11
L6566B
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51/51