LTC3122 15V, 2.5A Synchronous Step-Up DC/DC Converter with Output Disconnect Description Features VIN Range: 1.8V to 5.5V, 500mV After Start-Up Output Voltage Range: 2.2V to 15V 800mA Output Current for VIN = 5V and VOUT = 12V Output Disconnects from Input When Shut Down Synchronous Rectification: Up to 95% Efficiency Inrush Current Limit Up to 3MHz Adjustable Switching Frequency Synchronizable to External Clock n Selectable Burst Mode® Operation: 25µA I Q n Output Overvoltage Protection nSoft-Start n <1µA I in Shutdown Q n 12-Lead, 3mm × 4mm × 0.75mm Thermally Enhanced DFN and MSOP Packages n n n n n n The LTC®3122 is a synchronous step-up DC/DC converter with true output disconnect and inrush current limiting. The 2.5A current limit along with the ability to program output voltages up to 15V makes the LTC3122 well suited for a variety of demanding applications. Once started, operation will continue with inputs down to 500mV, extending runtime in many applications. n Applications n n n n RF Power Piezo Actuators Small DC Motors 12V Analog Rail From Battery, 5V, or Backup Capacitor L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. The LTC3122 features output disconnect in shutdown, dramatically reducing input power drain and enabling VOUT to completely discharge. Adjustable PWM switching from 100kHz to 3MHz optimizes applications for highest efficiency or smallest solution footprint. The oscillator can also be synchronized to an external clock for noise sensitive applications. Selectable Burst Mode operation reduces quiescent current to 25µA, ensuring high efficiency across the entire load range. An internal soft-start limits inrush current during start-up. Other features include a <1µA shutdown current and robust protection under short-circuit, thermal overload, and output overvoltage conditions. The LTC3122 is offered in both a low profile 12-lead (3mm × 4mm × 0.75 mm) DFN package and a 12-lead thermally enhanced MSOP package. Typical Application 5V to 12V Synchronous Boost Converter with Output Disconnect 100 3.3µH 90 SW 4.7µF OFF ON BURST PWM SD LTC3122 PWM/SYNC 100nF CAP RT FB VCC VC SGND 57.6k 4.7µF VOUT 12V 800mA VOUT PGND 1.02M 22µF 113k 210k 80 10 Burst Mode OPERATION 70 1 PWM 60 50 40 30 PWM POWER LOSS 0.1 POWER LOSS (W) VIN EFFICIENCY (%) VIN 5V Efficiency Curve 20 10 10pF 390pF 0 0.01 3122 TA01a 0.1 10 1 100 LOAD CURRENT (mA) 0 1000 3122 TA01b 3122f 1 LTC3122 Absolute Maximum Ratings (Note 1) VIN Voltage ................................................... –0.3V to 6V VOUT Voltage ............................................. –0.3V to 18V SW Voltage (Note 2)................................... –0.3V to 18V SW Voltage (Pulsed < 100ns) (Note 2)........ –0.3V to 19V VC, RT Voltage ........................................... –0.3V to VCC CAP Voltage VOUT < 5.7V.............................–0.3V to (VOUT + 0.3V) 5.7V ≤ VOUT ≤ 11.7V...... (VOUT – 6V) to (VOUT + 0.3V) VOUT > 11.7V..................................(VOUT – 6V) to 12V All Other Pins................................................ –0.3V to 6V Operating Junction Temperature Range (Notes 3, 4)............................................. –40°C to 125°C Storage Temperature Range................... –65°C to 150°C MSE Lead Temperature (Soldering, 10sec)............ 300°C Pin Configuration TOP VIEW SW PGND 2 VIN 3 PWM/SYNC 4 VCC RT TOP VIEW 12 CAP 1 SW PGND VIN PWM/SYNC VCC RT 11 VOUT 13 PGND 10 SGND 9 SD 5 8 FB 6 7 VC 1 2 3 4 5 6 13 PGND 12 11 10 9 8 7 CAP VOUT SGND SD FB VC MSE PACKAGE 12-LEAD PLASTIC MSOP DE PACKAGE 12-LEAD (4mm × 3mm) PLASTIC DFN TJMAX = 125°C, θJA = 43°C/W (NOTE 5), θJC = 5°C/W EXPOSED PAD (PIN 13) IS PGND, MUST BE SOLDERED TO PCB FOR RATED THERMAL PERFORMANCE TJMAX = 125°C, θJA = 40°C/W (NOTE 5), θJC = 10°C/W EXPOSED PAD (PIN 13) IS PGND, MUST BE SOLDERED TO PCB FOR RATED THERMAL PERFORMANCE Order Information LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3122EDE#PBF LTC3122EDE#TRPBF 3122 12-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C LTC3122IDE#PBF LTC3122IDE#TRPBF 3122 12-Lead (4mm × 3mm) Plastic DFN –40°C to 125°C LTC3122EMSE#PBF LTC3122EMSE#TRPBF 3122 12-Lead Plastic MSOP –40°C to 125°C LTC3122IMSE#PBF LTC3122IMSE#TRPBF 3122 12-Lead Plastic MSOP –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3122f 2 LTC3122 Electrical Characteristics The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C (Note 3). VIN = 3.6V, VOUT = 12V, RT = 57.6k unless otherwise noted. PARAMETER CONDITIONS Minimum Start-Up Voltage VOUT = 0V l MIN. Input Voltage Range After VOUT ≥ 2.2V TYP MAX 1.7 1.8 UNITS V l 0.5 5.5 V Output Voltage Adjust Range l 2.2 15 V Feedback Voltage l 1.178 1.202 1.225 V Feedback Input Current VFB = 1.4V 1 50 nA Quiescent Current, Shutdown VSD = 0V, VOUT = 0V, Not Including Switch Leakage 0.01 1 µA Quiescent Current, Active VC = 0V, Measured On VIN, Non-Switching 500 700 µA Quiescent Current, Burst Measured on VIN, VFB > 1.4V Measured on VOUT, VFB > 1.4V 25 10 40 20 µA µA N-channel MOSFET Switch Leakage Current VSW = 15V, VOUT = 15V, VC = 0V 0.1 20 µA P-channel MOSFET Switch Leakage Current VSW = 0V, VOUT = 15V, VSD = 0V 0.1 20 µA N-channel MOSFET Switch On-Resistance 0.121 P-channel MOSFET Switch On-Resistance 0.188 N-channel MOSFET Current Limit l 2.5 3.5 90 94 1 Maximum Duty Cycle VFB = 1.0V l Minimum Duty Cycle VFB = 1.4V l l 0.85 SYNC Frequency Range l 0.1 PWM/SYNC Input High l 0.9 •VCC PWM/SYNC Input Low l PWM/SYNC Input Current VPWM/SYNC = 5.5V CAP Clamp Voltage VOUT > 6.1V, Referenced to VOUT VCC Regulation Voltage VIN < 2.8V, VOUT > 5V Error Amplifier Transconductance l SD Input Low l VSD = 5.5V Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Voltage transients on the SW pin beyond the DC limit specified in the Absolute Maximum Ratings are non-disruptive to normal operations when using good layout practices, as shown on the demo board or described in the data sheet or application notes. Note 3: The LTC3122 is tested under pulsed load conditions such that TA ≈ TJ. The LTC3122E is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3122I is guaranteed % % 1.15 MHz 3 MHz V 0.1•VCC V 0.01 1 µA –5.6 –6.0 V 4 4.25 4.5 V 95 120 µS 70 Soft-Start Time l A –5.2 Error Amplifier Output Current SD Input High Ω 4.5 0 Switching Frequency SD Input Current Ω ±25 µA 10 ms 1.6 V 1 0.25 V 2 µA to meet specifications over the full –40°C to 125°C operating junction temperature range. The junction temperature (TJ in °C) is calculated from the ambient temperature (TA in °C) and power dissipation (PD in Watts) according to the formula: TJ = TA + (PD • θJA) where θJA is the thermal impedance of the package. Note 4: The LTC3122 includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature shutdown is active. Continuous operation above the specified maximum operating junction temperature may result in device degradation or failure. Note 5: Failure to solder the exposed backside of the package to the PC board ground plane will result in a thermal impedance much higher than the rated package specifications. 3122f 3 LTC3122 Typical Performance Characteristics Configured as front page application unless otherwise specified. Efficiency vs Load Current, VOUT = 7.5V Efficiency vs Load Current, VOUT = 5V 100 100 90 90 80 80 EFFICIENCY (%) EFFICIENCY (%) 70 60 50 PWM 40 30 20 0 0.01 0.1 70 60 50 PWM 40 30 0 0.01 0.1 30 0 0.01 80mA ILOAD = 200mA 3122 G04 INPUT CURRENT 1A/DIV 3122 G05 2ms/DIV RDS(ON) vs Temperature, Both NMOS and PMOS 1.0 80 CHANGE IN RDS(ON) FROM 25°C (%) –0.1 –0.2 –0.3 –0.4 –0.5 140 3122 G07 60 40 20 0 –20 –40 –50 3122 G06 Oscillator Frequency vs Temperature CHANGE IN FREQUENCY FROM 25°C (%) 0.2 1000 VOUT 5V/DIV 80mA 500µs/DIV Feedback vs Temperature 0 1 100 10 LOAD CURRENT (mA) SD 5V/DIV IOUT 500mA/DIV 0.1 0.1 Inrush Current Control 800mA 1µs/DIV VIN = 5.4V VIN = 4.2V VIN = 2.6V 3122 G03 VOUT 500mV/DIV AC-COUPLED INDUCTOR CURRENT 1A/DIV CHANGE IN VFB FROM 25°C (%) PWM 40 Load Transient Response VOUT 20mV/DIV AC-COUPLED 40 90 TEMPERATURE (°C) 50 3122 G02 PWM Mode Operation –10 60 10 1 100 1000 10000 10 LOAD CURRENT (mA) 3122 G01 –0.6 –60 70 20 VIN = 5.4V VIN = 3.8V VIN = 2.3V 10 1 100 1000 10000 10 LOAD CURRENT (mA) BURST 80 20 VIN = 4.2V VIN = 3.3V VIN = 0.6V 10 90 BURST EFFICIENCY (%) BURST 100 Efficiency vs Load Current, VOUT = 12V –10 70 110 30 TEMPERATURE (°C) 150 3122 G08 0.5 0 –0.5 –1.0 –1.5 –2.0 –60 –10 90 40 TEMPERATURE (°C) 140 3122 G09 3122f 4 LTC3122 Typical Performance Characteristics Peak Current Limit Change vs Temperature 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 0.5 1.5 3.5 2.5 VIN (V) 4.5 5.5 70 1 60 0 –1 –2 250 –4 –50 –10 70 110 30 TEMPERATURE (°C) 120 100 50 4.5 100 80 60 40 0 0.5 5.5 1.5 3122 G13 SD Pin Threshold 3.5 2.5 VIN , RISING(V) FREQUENCY (MHz) 3122 G16 15 0 4 0.5 2 500 0 600 3122 G17 4 –10 70 110 30 TEMPERATURE (°C) 150 VOUT = 15V VOUT = 3.6V VOUT = 2.2V 3 10 1.0 400 300 RT (kΩ) 3122 G12 Frequency Accuracy 6 200 6 30 12 1.5 100 5 3122 G14 8 0 4 3122 G14 2.0 0 3 VIN (V) 45 –15 –50 5.5 PERIOD (µs) 400mV 1s/DIV 4.5 FREQUENCY PERIOD 2.5 900mV 2 60 Frequency vs RT 3.0 VOUT 5V/DIV 1 75 20 2.5 3.5 VIN , FALLING (V) 0 Burst Mode Quiescent Current Change vs Temperature VOUT = 2.2V VOUT = 5V VOUT = 7.5V VOUT = 12V 140 150 VSD 500mV/DIV 0 150 Burst Mode IZERO Current vs VIN VOUT = 2.2V VOUT = 5V VOUT = 7.5V VOUT = 12V 1.5 20 10 3122 G10 200 0 0.5 30 3122 G11 IZERO CURRENT (mA) OUTPUT CURRENT (mA) 300 40 –3 Burst Mode Maximum Output Current vs VIN 350 VOUT = 5V VOUT = 7.5V VOUT = 12V 50 CHANGE IN CURRENT FROM 25°C (%) OUTPUT CURRENT (A) 1.6 2 INPUT CURRENT (mA) VOUT = 5V VOUT = 7.5V VOUT = 12V 1.8 PWM Operation No Load Input Current vs VIN CHANGE IN FREQUENCY (%) 2.0 PEAK CURRENT LIMIT CHANGE FROM 25°C (%) PWM Mode Maximum Output Current vs VIN 2 1 0 –1 –2 –3 –4 0 1 3 2 4 VIN FALLING (V) 5 6 3122 G18 3122f 5 LTC3122 Typical Performance Characteristics Efficiency vs Frequency 0 100 90 VCC vs VIN 60 50 40 30 20 fOSC = 200kHz fOSC = 1MHz fOSC = 3MHz 10 10 100 LOAD CURRENT (mA) 4.0 –2 –3 VCC (V) VCAP, REFERRED TO VOUT (V) 70 EFFICIENCY (%) 4.5 –1 80 0 CAP Pin Voltage vs VOUT –4 –5 3.5 3.0 –6 1000 –7 0 2 4 8 6 10 VOUT (V) 12 14 2.5 VIN FALLING VIN RISING 0 Burst Mode Operation to PWM Mode VOUT 100mV/DIV AC-COUPLED 2 3122 G20 3122 G19 Burst Mode Operation 1 3 VIN (V) 4 5 6 3122 G21 PWM Mode to Burst Mode Operation VOUT 100mV/DIV AC-COUPLED VOUT 100mV/DIV AC-COUPLED VSW 10V/DIV VPWM/SYNC 2V/DIV INDUCTOR CURRENT 0.5A/DIV ILOAD = 50mA VPWM/SYNC 2V/DIV 3122 G22 5µs/DIV Burst Mode Transient ILOAD = 70mA ILOAD = 70mA 20µs/DIV 3122 G23 3122 G24 20µs/DIV Synchronized Operation Short-Circuit Response SHORT-CIRCUIT APPLIED VOUT 5V/DIV VOUT 200mV/DIV AC-COUPLED VSW 5V/DIV 100mA ILOAD 100mA/DIV 10mA VPWM/SYNC 5V/DIV 10mA 200µs/DIV 3122 G25 SHORT-CIRCUIT REMOVED SYNCHRONIZED TO 1.3MHz 1µs/DIV 3122 G26 INPUT CURRENT 2A/DIV 100µs/DIV 3122 G27 3122f 6 LTC3122 Pin Functions SW (Pin 1): Switch Pin. Connect an inductor from this pin to VIN. Keep PCB trace lengths as short and wide as possible to reduce EMI and voltage overshoot. An internal anti-ringing resistor is connected between SW and VIN after the inductor current has dropped to near zero, to minimize EMI. The anti-ringing resistor is also activated in shutdown and during the sleep periods of Burst Mode operation. VCC (Pin 5): VCC Regulator Output. Connect a low-ESR filter capacitor of at least 4.7µF from this pin to GND to provide a regulated rail approximately equal to the lower of VIN and 4.25V. When VOUT is higher than VIN, and VIN falls below 3V, VCC will regulate to the lower of approximately VOUT and 4.25V. A UVLO event occurs if VCC drops below 1.6V. Switching is inhibited, and a soft-start is initiated when VCC returns above 1.7V. PGND (Pins 2, 13): Power Ground. When laying out your PCB, provide a short, direct path between PGND and the output capacitor and tie directly to the ground plane. The exposed pad is ground and must be soldered to the PCB ground plane for rated thermal performance. RT (Pin 6): Frequency Adjust Pin. Connect an external resistor (RT) from this pin to SGND to program the oscillator frequency according to the formula: VIN (Pin 3): Input Supply Pin. The device is powered from VIN unless VOUT exceeds VIN and VIN is less than 3V. Place a low ESR ceramic bypass capacitor of at least 4.7µF from VIN to PGND. X5R and X7R dielectrics are preferred for their superior voltage and temperature characteristics. RT = 57.6/ƒOSC where ƒOSC is in MHz and RT is in kΩ. VC (Pin 7): Error Amplifier Output. A frequency compensation network is connected to this pin to compensate the control loop. See Compensating the Feedback Loop section for guidelines. • PWM/SYNC = High. Disable Burst Mode Operation and maintain low noise, constant frequency operation. FB (Pin 8): Feedback Input to the Error Amplifier. Connect the resistor divider tap to this pin. Connect the top of the divider to VOUT and the bottom of the divider to SGND. The output voltage can be adjusted from 2.2V to 15V according to this formula: • PWM/SYNC = Low. Enable Burst Mode operation. VOUT = 1.202V • (1 + R1/R2) • PWM/SYNC = External CLK. The internal oscillator is synchronized to the external CLK signal. Burst Mode operation is disabled. A clock pulse width between 100ns and 2µs is required to synchronize the oscillator. An external resistor must be connected between RT and GND to program the oscillator slightly below the desired synchronization frequency. SD (Pin 9): Logic Controlled Shutdown Input. Bringing this pin above 1.6V enables normal, free-running operation, forcing this pin below 0.25V shuts the LTC3122 down, with quiescent current below 1μA. Do not leave this pin floating. In non-synchronized applications, repeated clocking of the PWM/SYNC pin to affect an operating mode change is supported with these restrictions: VOUT (Pin 11): Output Voltage Sense and the Source of the Internal Synchronous Rectifier MOSFET. Driver bias is derived from VOUT. Connect the output filter capacitor from VOUT to PGND, as close to the IC as possible. A minimum value of 10µF ceramic is recommended. VOUT is disconnected from VIN when SD is low. PWM/SYNC (Pin 4): Burst Mode Operation Select and Oscillator Synchronization. Do not leave this pin floating. • Boost Mode (VOUT > VIN): IOUT <500µA: ƒPWM/SYNC ≤ 100Hz, IOUT ≥ 500µA: ƒPWM/SYNC ≤ 5kHz • Buck Mode (VOUT < VIN): IOUT <5mA: ƒPWM/SYNC ≤ 5Hz, IOUT ≥ 5mA: ƒPWM/SYNC ≤ 5kHz SGND (Pin 10): Signal Ground. When laying out a PC board, provide a short, direct path between SGND and the (–) side of the output capacitor. CAP (Pin 12): Serves as the Low Reference for the Synchronous Rectifier Gate Drive. Connect a low ESR filter capacitor (typically 100nF) from this pin to VOUT to provide an elevated ground rail, approximately 5.6V below VOUT, used to drive the synchronous rectifier. 3122f 7 LTC3122 Block Diagram 1 BULK CONTROL SIGNALS SW VIN ANTI-RING L1 VIN 1.8V TO 5.5V 3 COUT TSD VREF_UP OSC SD OVLO SD SHUTDOWN PWM LOGIC AND DRIVERS + – CURRENT SENSE SD PWM/SYNC PWM BURST SYNC CONTROL + – IZERO COMP OVLO –+ – – + VC 5 LDO 6 FB R1 8 1.202V VC SD TSD OVLO CVCC 4.7µF OSCILLATOR RPL 12 R2 gm ERROR AMPLIFIER ILIM REF ADAPTIVE SLOPE COMPENSATION VCC CAP CPL VIN VOUT VBEST C1 100nF 16.2V PGND 4 VOUT 2.2V TO 15V 11 VIN CIN 9 VOUT OSC REFERENCE UVLO CF SOFT-START VC CLAMP RT RC CC VREF_UP 1.202V RT THERMAL SD 7 TSD SGND 10 PGND 2 EXPOSED PAD 13 LTC3122 3122 BD THE VALUES OF RC, CC, AND CF ARE BASED UPON OPERATING CONDITIONS. PLEASE REFER TO COMPENSATING THE FEEDBACK LOOP SECTION FOR GUIDELINES TO DETERMINE OPTIMAL VALUES OF THESE COMPONENTS. 3122f 8 LTC3122 Operation With its low RDS(ON) and low gate charge internal N-channel MOSFET switch and P-channel MOSFET synchronous rectifier, the LTC3122 achieves high efficiency over a wide range of load current. High efficiency is achieved at light loads when Burst Mode operation is commanded. Operation can be best understood by referring to the Block Diagram. Low Voltage Operation The LTC3122 is designed to allow start-up from input voltages as low as 1.8V. When VOUT exceeds 2.2V, the LTC3122 continues to regulate its output, even when VIN falls to as low as 0.5V. The limiting factors for the application become the availability of the input source to supply sufficient power to the output at the low voltages, and the maximum duty cycle. Note that at low input voltages, small voltage drops due to series resistance become critical and greatly limit the power delivery capability of the converter. This feature extends operating times by maximizing the amount of energy that can be extracted from the input source. Low Noise Fixed Frequency Operation Soft-Start The LTC3122 contains internal circuitry to provide closedloop soft-start operation. The soft-start utilizes a linearly increasing ramp of the error amplifier reference voltage from zero to its nominal value of 1.202V in approximately 10ms, with the internal control loop driving VOUT from zero to its final programmed value. This limits the inrush current drawn from the input source. As a result, the duration of the soft-start is largely unaffected by the size of the output capacitor or the output regulation voltage. The closed loop nature of the soft-start allows the converter to respond to load transients that might occur during the soft-start interval. The soft-start period is reset by a shutdown command on SD, a UVLO event on VCC (VCC < 1.6V), an overvoltage event on VOUT (VOUT ≥ 16.2V), or an overtemperature event (thermal shutdown is invoked when the die temperature exceeds 170°C). Upon removal of these fault conditions, the LTC3122 will soft-start the output voltage. Error Amplifier The non-inverting input of the transconductance error amplifier is internally connected to the 1.202V reference and the inverting input is connected to FB. An external resistive voltage divider from VOUT to ground programs the output voltage from 2.2V to 15V via FB as shown in Figure 1. R1 VOUT = 1.202V 1+ R2 Selecting an R2 value of 121kΩ to have approximately 10µA of bias current in the VOUT resistor divider yields the formula: R1 = 100.67•(VOUT – 1.202V) where R1 is in kΩ. Power converter control loop compensation is set by a simple RC network between VC and ground. VOUT LTC3122 R1 + – The LTC3122 is an adjustable frequency, 100kHz to 3MHz synchronous boost converter housed in either a 12-lead 4mm × 3mm DFN or a thermally enhanced MSOP package. The LTC3122 offers the unique ability to start-up and regulate the output from inputs as low as 1.8V and continue to operate from inputs as low as 0.5V. Output voltages can be programmed between 2.2V and 15V. The device also features fixed frequency, current mode PWM control for exceptional line and load regulation. The current mode architecture with adaptive slope compensation provides excellent transient load response and requires minimal output filtering. An internal 10ms closed loop soft-start simplifies the design process while minimizing the number of external components. FB 1.202V R2 3122 F01 Figure 1. Programming the Output Voltage 3122f 9 LTC3122 Operation Internal Current Limit The current limit comparator shuts off the N-channel MOSFET switch once its threshold is reached. Peak switch current is limited to 3.5A, independent of input or output voltage, except when VOUT is below 1.5V, resulting in the current limit being approximately half of the nominal peak. Lossless current sensing converts the peak current signal of the N-channel MOSFET switch into a voltage that is summed with the internal slope compensation. The summed signal is compared to the error amplifier output to provide a peak current control command for the PWM. Zero Current Comparator The zero current comparator monitors the inductor current being delivered to the output and shuts off the synchronous rectifier when this current reduces to approximately 50mA. This prevents the inductor current from reversing in polarity, improving efficiency at light loads. Oscillator The internal oscillator is programmed to the desired switching frequency with an external resistor from the RT pin to GND according to the following formula: 57.6 ƒOSC (MHz) = RT (kΩ) The oscillator also can be synchronized to an external frequency by applying a pulse train to the PWM/SYNC pin. An external resistor must be connected between RT and GND to program the oscillator to a frequency approximately 25% below that of the externally applied pulse train used for synchronization. RT is selected in this case according to this formula: 73.2 RT (kΩ) = ƒSYNC (MHz) Output Disconnect The LTC3122’s output disconnect feature eliminates body diode conduction of the internal P-channel MOSFET rectifier. This allows for VOUT to discharge to 0V during shutdown, and draw no current from the input source. It also allows for inrush current limiting at turn-on, minimizing surge currents seen by the input supply. Note that to obtain the advantages of output disconnect, there must not be an external Schottky diode connected between SW and VOUT. The output disconnect feature also allows VOUT to be pulled high, without reverse current being backfed into the power source connected to VIN. Shutdown The boost converter is disabled by pulling SD below 0.25V and enabled by pulling SD above 1.6V. Note that SD can be driven above VIN or VOUT, as long as it is limited to less than the absolute maximum rating. Thermal Shutdown If the die temperature exceeds 170°C typical, the LTC3122 will go into thermal shutdown (TSD). All switches will be turned off until the die temperature drops by approximately 7°C, when the device re-initiates a soft-start and switching can resume. Boost Anti-Ringing Control The anti-ringing control connects a resistor across the inductor to damp high frequency ringing on the SW pin during discontinuous current mode operation when the inductor current has dropped to near zero. Although the ringing of the resonant circuit formed by L and CSW (capacitance on SW pin) is low energy, it can cause EMI radiation. VCC Regulator An internal low dropout regulator generates the 4.25V (nominal) VCC rail from VIN or VOUT, depending upon operating conditions. VCC is supplied from VIN when VIN is greater than 3.5V, otherwise the greater of VIN and VOUT is used. The VCC rail powers the internal control circuitry and power MOSFET gate drivers of the LTC3122. The VCC regulator is disabled in shutdown to reduce quiescent current and is enabled by forcing the SD pin above its threshold. A 4.7µF or larger capacitor must be connected between VCC and SGND. 3122f 10 LTC3122 Applications Information An overvoltage condition occurs when VOUT exceeds approximately 16.2V. Switching is disabled and the internal soft-start ramp is reset. Once VOUT drops below approximately 15.6V, a soft-start cycle is initiated and switching is enabled. If the boost converter output is lightly loaded so that the time constant product of the output capacitance, COUT, and the output load resistance, ROUT is near or greater than the soft-start time of approximately 10ms, the soft-start ramp may end before or soon after switching resumes, defeating the inrush current limiting of the closed loop soft-start following an overvoltage event. Short-Circuit Protection The LTC3122 output disconnect feature allows output short-circuit protection. To reduce power dissipation under overload and short-circuit conditions, the peak switch current limit is reduced to 1.6A. Once VOUT > 1.5V, the current limit is set to its nominal value of 3.5A. VIN > VOUT Operation The LTC3122 step-up converter will maintain voltage regulation even when the input voltage is above the desired output voltage. Note that operating in this mode will exhibit lower efficiency and a reduced output current capability. Refer to the Typical Performance Characteristics section for details. Burst Mode Operation When the PWM/SYNC pin is held low, the boost converter operates in Burst Mode operation to improve efficiency at light loads and reduce standby current at no load. The input thresholds for this pin are determined relative to VCC with a low being less than 10% of VCC and a high being greater than 90% of VCC. The LTC3122 will operate in fixed frequency PWM mode even if Burst Mode operation is commanded during soft-start. In Burst Mode operation, the LTC3122 switches asynchronously. The inductor current is first charged to 600mA by turning on the N-channel MOSFET switch. Once this current threshold is reached, the N-channel is turned off and the P-channel synchronous switch is turned on, delivering current to the output. When the inductor current discharges to approximately zero, the cycle repeats. In Burst Mode operation, energy is delivered to the output until the nominal regulation value is reached, at which point the LTC3122 transitions to sleep mode. In sleep, the output switches are turned off and the LTC3122 consumes only 25μA of quiescent current. When the output voltage droops approximately 1%, switching resumes. This maximizes efficiency at very light loads by minimizing switching and quiescent losses. Output voltage ripple in Burst Mode operation is typically 1% peak-to-peak. Additional output capacitance (10μF or greater), or the addition of a small feed-forward capacitor (10pF to 50pF) connected between VOUT and FB can help further reduce the output ripple. The maximum output current (IOUT) capability in Burst Mode operation varies with VIN and VOUT, as shown in Figure 2. 350 300 OUTPUT CURRENT (mA) Overvoltage Lockout 250 VOUT = 2.2V VOUT = 5V VOUT = 7.5V VOUT = 12V 200 150 100 50 0 0.5 1.5 2.5 3.5 VIN, FALLING (V) 4.5 5.5 3122 F02 Figure 2. Burst Mode Maximum Output Current vs VIN 3122f 11 LTC3122 Applications Information Pcb Layout Guidelines The high switching frequency of the LTC3122 demands careful attention to board layout. A careless layout will result in reduced performance. Maximizing the copper area for ground will help to minimize die temperature rise. A multilayer board with a separate ground plane is ideal, but not absolutely necessary. See Figure 3 for an example of a two-layer board layout. rent capability by reducing the inductor ripple current. The minimum inductance value, L, is inversely proportional to operating frequency and is given by the following equation: L> VIN • ( VOUT − VIN ) 3 µH and L > ƒ ƒ • Ripple • VOUT where: Ripple = Allowable inductor current ripple (amps peak-to-peak) ƒ = Switching Frequency in MHz PGND PGND CAP SW VIN 1 12 2 11 3 4 VCC 13 PGND 10 SGND 9 5 8 6 7 RT VOUT FB VC The inductor current ripple is typically set for 20% to 40% of the maximum inductor current. High frequency ferrite core inductor materials reduce frequency dependent power losses compared to cheaper powdered iron types, improving efficiency. The inductor should have low ESR (series resistance of the windings) to reduce the I2R power losses, and must be able to support the peak inductor current without saturating. Molded chokes and some chip inductors usually do not have enough core area to support the peak inductor currents of 3A to 4A seen on the LTC3122. To minimize radiated noise, use a shielded inductor. See Table 1 for suggested components and suppliers. 3122 F02 Figure 3. Traces Carrying High Current Are Direct (PGND, SW, VIN and VOUT). Trace Area at FB and VC Are Kept Low. Trace Length to Input Supply Should Be Kept Short. VIN and VOUT Ceramic Capacitors Should Be Placed as Close to the LTC3122 Pins as Possible Schottky Diode Although it is not required, adding a Schottky diode from SW to VOUT can improve the converter efficiency by about 4%. Note that this defeats the output disconnect and shortcircuit protection features of the LTC3122. Component Selection Inductor Selection The LTC3122 can utilize small surface mount inductors due to its high switching frequency (up to 3MHz). Larger values of inductance will allow slightly greater output cur- Table 1. Recommended Inductors PART NUMBER MAX DC VALUE DCR CURRENT (µH) (mΩ) (A) SIZE (mm) W×L×H Coilcraft LPS4018 Coilcraft MSS7341 Coilcraft MSS1260T 1 3.3 33 42 20 54.9 3.8 3.72 4.34 4 × 4 × 1.8 7.3 × 7.3 × 4.1 12.3 × 12.3 × 6.2 Coiltronics DRQ73 Coiltronics SD7030 Coiltronics DR125 0.992 3.3 33 24 24 59 3.99 3 3.84 7.6 × 7.6 × 3.55 7×7×3 12.5 × 12.5 × 6 Murata LQH6PP Murata LQH6PP 1 3.3 9 16 4.3 3.8 6 × 6 × 4.3 6 × 6 × 4.3 Sumida CDRH50D28RNP Sumida CDRH8D28NP Sumida CDRH129HF 1.2 3.3 33 13 18 53 4.8 4 4.25 5 × 5 × 2.8 8×8×3 12 × 12 × 10 3 31 3.2 6 × 6 × 4.5 TDK LTF5022T TDK SPM6530T TDK VLF12060T 1.2 3.3 33 25 20 53 4.2 4.1 3.4 5 × 5.2 × 2.2 7 × 7 × 3.2 11.7 × 12 × 6 Würth WE-PD 3.3 32.5 3.1 7.3 × 7.3 × 2 Taiyo-Yuden NR6045 3122f 12 LTC3122 Applications Information Output and Input Capacitor Selection Low ESR (equivalent series resistance) capacitors should be used to minimize the output voltage ripple. Multilayer ceramic capacitors are an excellent choice as they have extremely low ESR and are available in small footprints. X5R and X7R dielectric materials are preferred for their ability to maintain capacitance over wide voltage and temperature ranges. Y5V types should not be used. Although ceramic capacitors are recommended, low ESR tantalum capacitors may be used as well. When selecting output capacitors, the magnitude of the peak inductor current, together with the ripple voltage specification, determine the choice of the capacitor. Both the ESR (equivalent series resistance) of the capacitor and the charge stored in the capacitor each cycle contribute to the output voltage ripple. The ripple due to the charge is approximately: VRIPPLE(CHARGE) ≈ IP • VIN COUT • VOUT • ƒ where IP is the peak inductor current. of its rated capacitance when operated near its rated voltage. As a result it is sometimes necessary to use a larger capacitor value or a capacitor with a larger value and case size, such as 1812 rather than 1206, in order to actually realize the intended capacitance at the full operating voltage. Be sure to consult the vendor’s curve of capacitance vs DC bias voltage. Table 2 shows a sampling of capacitors suited for LTC3122 applications. Table 2. Representative Output Capacitors MANUFACTURER, PART NUMBER VALUE (µF) VOLTAGE (V) SIZE L × W × H (mm) TYPE, ESR (mΩ) AVX, 12103D226MAT2A 22 25 3.2 × 2.5 × 2.79, X5R Ceramic Kemet, C2220X226K3RACTU 22 25 5.7 × 5.0 × 2.4, X7R Ceramic Kemet, A700D226M016ATE030 22 16 7.3 × 4.3 × 2.8, Alum. Polymer, 30mΩ Murata, GRM32ER71E226KE15L 22 25 3.2 × 2.5 × 2.5, X7R Ceramic Nichicon, PLV1E121MDL1 82 25 8 × 8 × 12, Alum. Polymer, 25mΩ Panasonic, ECJ-4YB1E226M 22 25 3.2 × 2.5 × 2.5, X5R Ceramic Sanyo, 25TQC22MV 22 25 7.3 × 4.3 × 3.1, POSCAP, 50mΩ Sanyo, 16TQC100M 100 16 7.3 × 4.3 × 1.9, POSCAP, 45mΩ The ESR of COUT is usually the most dominant factor for ripple in most power converters. The ripple due to the capacitor ESR is: V VRIPPLE(ESR) = ILOAD • RESR • OUT VIN Sanyo, 25SVPF47M 47 25 6.6 × 6.6 × 5.9, OS-CON, 30mΩ Taiyo Yuden, TMK325BJ226MM-T 22 25 3.2 × 2.5 × 2.5, X5R Ceramic where RESR = capacitor equivalent series resistance. TDK, CKG57NX5R1E476M 47 25 6.5 × 5.5 × 5.5, X5R Ceramic The input filter capacitor reduces peak currents drawn from the input source and reduces input switching noise. A low ESR bypass capacitor with a value of at least 4.7µF should be located as close to the VIN pin as possible. Cap-XX GS230F 1.2Farads 4.5 39 × 17 × 3.8 28mΩ Cooper A1030-2R5155 1.5Farads 2.5 Ø = 10, L = 30 60mΩ Maxwell BCAP0050-P270 50Farads 2.5 Ø = 18, L = 40 20mΩ Low ESR and high capacitance are critical to maintain low output voltage ripple. Capacitors can be used in parallel for even larger capacitance values and lower effective ESR. Ceramic capacitors are often utilized in switching converter applications due to their small size, low ESR and low leakage currents. However, many ceramic capacitors experience significant loss in capacitance from their rated value with increased DC bias voltage. It is not uncommon for a small surface mount capacitor to lose more than 50% For applications requiring a very low profile and very large capacitance, the GS, GS2 and GW series from Cap-XX and PowerStor Aerogel Capacitors from Cooper all offer very high capacitance and low ESR in various low profile packages. A method for improving the converter’s transient response uses a small feed-forward series network of a capacitor and 3122f 13 LTC3122 Applications Information a resistor across the top resistor of the feedback divider (from VOUT to FB). This adds a phase-lead zero and pole to the transfer function of the converter as calculated in the Compensating the Feedback Loop section. possible. If the junction temperature rises above ~170°C, the part will go into thermal shutdown, and all switching will stop until the temperature drops approximately 7°C. Operating Frequency Selection The LTC3122 uses current mode control, with internal adaptive slope compensation. Current mode control eliminates the second order filter due to the inductor and output capacitor exhibited in voltage mode control, and simplifies the power loop to a single pole filter response. Because of this fast current control loop, the power stage of the IC combined with the external inductor can be modeled by a transconductance amplifier gmp and a current controlled current source. Figure 4 shows the key equivalent small signal elements of a boost converter. There are several considerations in selecting the operating frequency of the converter. Typically the first consideration is to stay clear of sensitive frequency bands, which cannot tolerate any spectral noise. For example, in products incorporating RF communications, the 455kHz IF frequency is sensitive to any noise, therefore switching above 600kHz is desired. Some communications have sensitivity to 1.1MHz and in that case a 1.5MHz switching converter frequency may be employed. A second consideration is the physical size of the converter. As the operating frequency is increased, the inductor and filter capacitors typically can be reduced in value, leading to smaller sized external components. The smaller solution size is typically traded for efficiency, since the switching losses due to gate charge increase with frequency. Another consideration is whether the application can allow pulse-skipping. When the boost converter pulse-skips, the minimum on-time of the converter is unable to support the duty cycle. This results in a low frequency component to the output ripple. In many applications where physical size is the main criterion, running the converter in this mode is acceptable. In applications where it is preferred not to enter this mode, the maximum operating frequency is given by: VOUT − VIN ƒMAX _ NOSKIP ≤ Hz VOUT • tON(MIN) where tON(MIN) = minimum on-time = 100ns. Thermal Considerations For the LTC3122 to deliver its full power, it is imperative that a good thermal path be provided to dissipate the heat generated within the package. This can be accomplished by taking advantage of the large thermal pad on the underside of the IC. It is recommended that multiple vias in the printed circuit board be used to conduct heat away from the IC and into a copper plane with as much area as Compensating the Feedback Loop The DC small-signal loop gain of the system shown in Figure 4 is given by the following equation: GBOOST = GEA • GMP • GPOWER • R2 R1+ R2 where GEA is the DC gain of the error amplifier, GMP is the modulator gain, and GPOWER is the inductor current to VOUT gain. – + gmp IL VOUT η • VIN •I 2 • VOUT L RPL 1.202V REFERENCE CF CPL + gma RC CC RO ERROR AMPLIFIER RL COUT MODULATOR VC RESR – R1 FB R2 3122 F04 CC: COMPENSATION CAPACITOR COUT: OUTPUT CAPACITOR CPL: PHASE LEAD CAPACITOR CF : HIGH FREQUENCY FILTER CAPACITOR gma: TRANSCONDUCTANCE AMPLIFIER INSIDE IC gmp: POWER STAGE TRANSCONDUCTANCE AMPLIFIER RC: COMPENSATION RESISTOR RL: OUTPUT RESISTANCE DEFINED AS VOUT/ILOADMAX RO: OUTPUT RESISTANCE OF gma RPL: PHASE LEAD RESISTOR R1, R2: FEEDBACK RESISTOR DIVIDER NETWORK RESR: OUTPUT CAPACITOR ESR η : CONVERTER EFFICIENCY (~90% AT HIGHER CURRENTS) Figure 4. Boost Converter Equivalent Model 3122f 14 LTC3122 Applications Information GEA = g ma • RO ≈ 950V/V (Not Adjustable; g ma = 95µS, RO ≈ 10MΩ) GMP = g mp = GPOWER = ∆IL ∆VC ≈ 3.4S (Not Adjustable) ∆VOUT η • VIN = ∆IL 2 •IOUT Combining the two equations above yields: GDC = GMP • GPOWER ≈ 1.7 • η • VIN IOUT V/V Converter efficiency η will vary with IOUT and switching frequency ƒOSC as shown in the typical performance characteristics curves. 2 Output Pole: P1 = Hz 2 • π • RL • COUT Error Amplifier Pole: P2 = 1 Hz 2 • π • RO • (CC + CF ) Error Amplifier Zero: Z1 = 1 Hz 2 • π • RC • CC ESR Zero: Z2 = RHP Zero: Z3 = 1 2 • π • RESR • COUT VIN2 • RL 2 • π • VOUT 2 • L Hz 1 Hz 2 • π • (R1+ RPL ) • CPL 1 Phase Lead Pole: P4 = Hz R1• R2 +R 2•π • •C R1+ R2 PL PL Error Amplifier Filter Pole: P5 = 1 Hz CC • CF 2 • π • RC • CC + CF GƒC ≈ GDC 2 ƒ 1+ C P1 V/V Decide how much phase margin (Φm) is desired. Greater phase margin can offer more stability while lower phase margin can yield faster transient response. Typically, Φm ≈ 60° is optimal for minimizing transient response time while allowing sufficient margin to account for component variability. Φ1 is the phase boost of Z1, P2, and P5 while Φ2 is the phase boost of Z5 and P4. Select Φ1 and Φ2 such that V Φ1 ≤ 74° ; Φ2 ≤ 2 • tan−1 OUT − 90° and 1.2V Hz ƒ High Frequency Pole: P3 > OSC 3 Phase Lead Zero: Z4 = The current mode zero (Z3) is a right half plane zero which can be an issue in feedback control design, but is manageable with proper external component selection. As a general rule, the frequency at which the open-loop gain of the converter is reduced to unity, known as the crossover frequency ƒC, should be set to less than one third of the right half plane zero (Z3), and under one eighth of the switching frequency ƒOSC. Once ƒC is selected, the values for the compensation components can be calculated using a bode plot of the power stage or two generally valid assumptions: P1 dominates the gain of the power stage for frequencies lower than ƒC and ƒC is much higher than P2. First calculate the power stage gain at ƒC, GƒC in V/V. Assuming the output pole P1 dominates GƒC for this range, it is expressed by: ƒ Φ1 + Φ2 = Φm + tan−1 C Z3 where VOUT is in V and ƒC and Z3 are in kHz. Setting Z1, P5, Z4, and P4 such that Z1= ƒC ƒ , P5 = ƒC a1, Z4 = C , P4 = ƒC a2 a1 a2 allows a1 and a2 to be determined using Φ1 and Φ2 Φ + 90° Φ + 90° a1 = tan2 1 , a2 = tan2 2 2 2 3122f 15 LTC3122 Applications Information The compensation will force the converter gain GBOOST to unity at ƒC by using the following expression for CC: 10 3 • g ma • R2 • G ƒC ( a 1 − 1) a 2 CC = pF 2π • ƒ C • (R1+ R2 ) a 1 (gma in µS, ƒ C in kHz, G ƒC in V/V) Once CC is calculated, RC and CF are determined by: RC = 6 10 • a1 kΩ (ƒC in kHz, C C in pF) 2π • ƒC • CC Once the compensation values have been calculated, obtaining a converter bode plot is strongly recommended to verify calculations and adjust values as required. Using the circuit in Figure 5 as an example, Table 3 shows the parameters used to generate the bode plot shown in Figure 6. Table 3. Bode Plot Parameters for Type II Compensation PARAMETER VALUE UNITS COMMENT VIN 5 V App Specific VOUT 12 V App Specific C CF = C a1 − 1 RL 15 Ω App Specific COUT 22 µF App Specific RESR 5 mΩ App Specific The values of the phase lead components are given by the expressions: 3.3 µH App Specific ƒOSC 1 MHz Adjustable R1 1020 kΩ Adjustable R2 113 kΩ Adjustable gma 95 µS Fixed RPL R1• R2 R1− a2 • R1+ R2 = kΩ and a2 − 1 CPL = 106 ( a2 − 1) (R1+ R2) 2π • ƒC • R12 a2 pF where R1, R2, and RPL are in kΩ and ƒC is in kHz. Note that selecting Φ2 = 0° forces a2 = 1, and so the converter will have Type II compensation and therefore no feedforward: RPL is open (infinite impedance) and CPL = 0pF. If a2 = 0.833 • VOUT (its maximum), feedforward is maximized; RPL = 0 and CPL is maximized for this compensation method. L RO 10 MΩ Fixed gmp 3.4 S Fixed η 80 % App Specific RC 210 kΩ Adjustable CC 390 pF Adjustable CF 10 pF Adjustable RPL 0 kΩ Optional CPL 0 pF Optional From Figure 6, the phase is 60° when the gain reaches 0dB, so the phase margin of the converter is 60°. The crossover frequency is 15kHz, which is more than three times lower than the 108.4kHz frequency of the RHP zero to achieve adequate phase margin. 3122f 16 LTC3122 Applications Information VIN 5V L1 3.3µH SW VIN CIN 4.7µF SD OFF ON LTC3122 PWM/SYNC BURST PWM C1 100nF FB VCC VC SGND R1 1.02M CAP RT RT 57.6k VOUT 12V 800mA VOUT COUT 22µF R2 113k RC 210k PGND CF 10pF CC 390pF CVCC 4.7µF 3122 F05a Transient Response with 400mA to 800mA Load Step Switching Waveforms with 800mA Load VOUT 100mV/DIV AC-COUPLED VOUT 500mV/DIV AC-COUPLED SW 10V/DIV INDUCTOR CURRENT 1A/DIV ILOAD 500mA/DIV 3122 F05b 100µs/DIV 3122 F05c Figure 5. 1MHz, 5V to 12V, 800mA Boost Converter 170 180 150 140 PHASE 130 100 110 60 90 20 70 –20 50 –60 30 –100 GAIN 10 –140 –180 –10 –30 0.01 PHASE (deg) GAIN (dB) 200ns/DIV 0.1 10 1 FREQUENCY (kHz) 100 –220 1000 3122 F06 Figure 6. Bode Plot for Example Converter 3122f 17 LTC3122 Applications Information VIN 5V L1 3.3µH SW VIN CIN 4.7µF SD OFF ON BURST PWM VOUT 12V 800mA VOUT C1 100nF LTC3122 CAP PWM/SYNC RPL 604k 1.02M CPL 10pF VCC RT 57.6k COUT 22µF FB RT SGND PGND VC RC 127k CF 33pF CC 220pF CVCC 4.7µF R2 113k 3122 F06 Figure 7. Boost Converter with Phase Lead The circuit in Figure 7 shows the same application as that in Figure 5 with Type III compensation. This is accomplished by adding CPL and RPL and adjusting CC, CF, and RC accordingly. Table 4 shows the parameters used to generate the bode plot shown in Figure 8. From Figure 8, the phase margin is still optimized at 60° and the crossover frequency remains 15kHz. Adding CPL and RPL provides some feedforward signal in Burst Mode operation, leading to lower output voltage ripple. Table 4. Bode Plot Parameters for Type III Compensation 170 PARAMETER 150 180 140 VALUE UNITS COMMENT VIN 5 V App Specific VOUT 12 V App Specific RL 15 Ω App Specific COUT 22 µF App Specific RESR 5 mΩ App Specific 30 3.3 µH App Specific 10 –140 1 MHz Adjustable –10 –180 R1 113 kΩ Adjustable R2 1020 kΩ Adjustable –30 0.01 gma 95 µS Fixed RO 10 MΩ Fixed gmp 3.4 S Fixed η 80 % App Specific ƒOSC RC 127 kΩ Adjustable CC 220 pF Adjustable CF 33 pF Adjustable RPL 604 kΩ Adjustable CPL 10 pF Adjustable GAIN (dB) 100 PHASE 110 60 90 20 70 –20 50 –60 GAIN 0.1 10 1 FREQUENCY (kHz) PHASE (deg) L 130 –100 100 –220 1000 3122 F08 Figure 8. Bode Plot Showing Phase Lead 3122f 18 LTC3122 Typical Applications Single Li-Cell to 6V, 5W Synchronous Boost Converter for RF Transmitter VIN 2.5V TO 4.2V L1 3.3µH SW VIN CIN 4.7µF OFF ON SD LTC3122 C1 100nF RT FB VCC VC SGND R1 487k CAP PWM/SYNC RT 57.6k VOUT 6V 833mA VOUT COUT 22µF RC 73.2k PGND CC 560pF CVCC 4.7µF VIN = 3.6V VOUT 500mV/DIV AC-COUPLED 833mA R2 121k OUTPUT CURRENT 500mA/DIV CF 47pF 80mA 80mA 3122 TA02b 100µs/DIV CIN, CVCC: 4.7µF, 16V, X7R, 1206 C1: 100nF, 16V, X7R, 1206 COUT: 22µF, 16V, X7R, 1812 L1: TDK SPM6530T-3R3M 3122 TA02a 2 AA Cell to 12V Synchronous Boost Converter, 180mA L1 3.3µH SW VIN OFF ON PWM/SYNC VOUT C1 100nF RT FB VCC VC RT 57.6k SGND CVCC 4.7µF CIN, CVCC: 4.7µF, 16V, X7R, 1206 C1: 100nF, 16V, X7R, 1206 COUT: 22µF, 25V, X7R, 1812 L1: TDK SPM6530T-3R3M R1 1.02M CAP PGND COUT 22µF RC 200k CC 560pF R2 113k CF 10pF 2.3 100 2.1 90 1.9 80 70 1.7 60 1.5 50 1.3 40 1.1 30 0.9 20 EFFICIENCY INPUT CURRENT 0.7 3122 TA03a 0.5 1.6 EFFICIENCY (%) CIN 4.7µF SD LTC3122 VOUT 12V 180mA INPUT CURRENT (A) VIN 1.8V TO 3V 1.8 2 2.2 2.4 2.6 VIN (V) 10 2.8 3 0 3.2 3122 TA03b 3122f 19 LTC3122 Typical Applications 3.3V to 12V Synchronous Boost Converter with Output Disconnect, 500mA VIN 3.3V L1 3.3µH SW VIN OFF ON CIN 4.7µF SD LTC3122 C1 100nF RT FB VCC VC SGND R1 1.02M CAP PWM/SYNC RT 57.6k VOUT 12V 500mA VOUT COUT 22µF RC 232k PGND CVCC 4.7µF CC 470pF R2 113k SW 5V/DIV INDUCTOR CURRENT 1A/DIV CF 10pF 500ns/DIV CIN, CVCC: 4.7µF, 16V, X7R, 1206 C1: 100nF, 16V, X7R, 1206 COUT: 22µF, 25V, X7R, 1812 L1: TDK SPM6530T-3R3M 3122 TA04b 3122 TA04a USB/Battery Powered Synchronous Boost Converter, 4.3V to 5V, 1A VIN 4.3V TO 5.5V L1 3.3µH SW VIN OFF ON CIN 4.7µF SD LTC3122 PWM/SYNC C1 100nF FB VCC VC SGND CVCC 4.7µF CIN, CVCC: 4.7µF, 16V, X7R, 1206 C1: 100nF, 16V, X7R, 1206 COUT: 100µF, 16V, X7R, 1812 L1: TDK SPM6530T-3R3M R1 383k CAP RT RT 57.6k VOUT 5V 1A VOUT PGND VIN = 4.3V VOUT 500mV/DIV AC-COUPLED COUT 100µF RC 43.2k CC 1000pF R2 121k 1A OUTPUT CURRENT 500mA/DIV CF 68pF 100mA 200µs/DIV 3122 TA05b 3122 TA05a 3122f 20 LTC3122 Typical Applications 5V to Dual Output Synchronous Boost Converter, ±15V C2 470nF L1 3.3µH VIN 5V –15.1 SW OFF ON SD C1 100nF CAP PWM/SYNC R1 1.3M COUT1 22µF FB RT R2 113k VC VCC RT 57.6k U1 SGND RC 365k PGND CVCC 4.7µF CC 150pF VOUT2 –15V CF 10pF COUT2 47µF Z1 14.9 –14.8 14.8 –14.7 14.7 –14.6 14.6 –14.5 14.5 VOUT2 –14.4 14.4 –14.3 14.3 –14.2 14.2 –14.1 CIN, CVCC: 4.7µF, 16V, X7R, 1206 COUT2: 47µF, 25V, X7R, 1206 C1: 100nF, 16V, X7R, 1206 COUT1: 22µF, 25V, X7R, 1812 C2: 470nF, 25V, X7R, 1206 L1: TDK SPM6530T-3R3M U1: CENTRAL SEMICONDUCTOR CBAT54S Z1: DIODES, INC. DDZ16ASF-7 15.0 VOUT1 –14.9 0 3122 TA06a 100 50 150 OUTPUT CURRENT (mA) VOUT1 (V) CIN 4.7µF LTC3122 VOUT1 15V VOUT VOUT2 (V) VIN 15.1 –15.0 14.1 200 3122 TA06b Single Li-Cell 3-LED Driver, 2.5V/4.2V to 350mA VIN 2.5V TO 4.2V L1 3.3µH VIN = 3.6V SW VIN OFF ON CIN 4.7µF SD LTC3122 PWM/SYNC VOUT FB VCC VC SGND VCC CAP RT RT 57.6k D1 C1 100nF PGND CVCC 4.7µF CIN, CVCC: 4.7µF, 6V, X7R, 1206 C1: 100nF, 6V, X7R, 1206 COUT: 22µF, 16V, X7R, 1812 L1: TDK SPM6530T-3R3M D1, D2, D3: CREE XPGWHT-L1-0000-00G51 LT1006 RC 2k CC 3.9nF + – R1 1.02M D2 COUT1 22µF D3 RS 0.1Ω SD 5V/DIV LED CURRENT 100mA/DIV R2 30.9k 2ms/DIV 3122 TA07b 3122 TA07a 3122f 21 LTC3122 Package Description DE/UE Package 12-Lead Plastic DFN (4mm × 3mm) (Reference LTC DWG # 05-08-1695 Rev D) 0.70 ±0.05 3.60 ±0.05 2.20 ±0.05 3.30 ±0.05 1.70 ± 0.05 PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC 2.50 REF RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 4.00 ±0.10 (2 SIDES) 7 R = 0.115 TYP 0.40 ± 0.10 12 R = 0.05 TYP PIN 1 TOP MARK (NOTE 6) 0.200 REF 3.30 ±0.10 3.00 ±0.10 (2 SIDES) 1.70 ± 0.10 0.75 ±0.05 6 0.25 ± 0.05 1 PIN 1 NOTCH R = 0.20 OR 0.35 × 45° CHAMFER (UE12/DE12) DFN 0806 REV D 0.50 BSC 2.50 REF 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING PROPOSED TO BE A VARIATION OF VERSION (WGED) IN JEDEC PACKAGE OUTLINE M0-229 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 3122f 22 LTC3122 Package Description MSE Package 12-Lead Plastic MSOP , Exposed Die Pad MSE Package (Reference LTC DWG # 12-Lead Plastic MSOP,05-08-1666 Exposed Rev DieF)Pad (Reference LTC DWG # 05-08-1666 Rev F) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ±0.102 (.112 ±.004) 5.23 (.206) MIN 2.845 ±0.102 (.112 ±.004) 0.889 ±0.127 (.035 ±.005) 6 1 1.651 ±0.102 (.065 ±.004) 1.651 ±0.102 3.20 – 3.45 (.065 ±.004) (.126 – .136) 12 0.65 0.42 ±0.038 (.0256) (.0165 ±.0015) BSC TYP RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 0.35 REF 4.039 ±0.102 (.159 ±.004) (NOTE 3) 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 7 NO MEASUREMENT PURPOSE 0.406 ±0.076 (.016 ±.003) REF 12 11 10 9 8 7 DETAIL “A” 0° – 6° TYP 3.00 ±0.102 (.118 ±.004) (NOTE 4) 4.90 ±0.152 (.193 ±.006) GAUGE PLANE 0.53 ±0.152 (.021 ±.006) DETAIL “A” 1.10 (.043) MAX 0.18 (.007) SEATING PLANE 0.22 – 0.38 (.009 – .015) TYP 1 2 3 4 5 6 0.650 (.0256) BSC NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 0.86 (.034) REF 0.1016 ±0.0508 (.004 ±.002) MSOP (MSE12) 0911 REV F 3122f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23 LTC3122 Typical Application Dual Supercapacitor Backup Power Supply, 0.5V to 5V L1 3.3µH VIN 0.5V TO 5V CIN 4.7µF VIN SW SC1 OFF ON 50F SD LTC3122 SC2 50F PWM/SYNC RT 57.6k C1 100nF FB VCC VC PGND CVCC 4.7µF VOUT 20mV/DIV AC-COUPLED SW 5V/DIV R1 383k CAP RT SGND VOUT 5V VOUT COUT 100µF RC 43.2k CC 1nF R2 121k INDUCTOR CURRENT 500mA/DIV CF 68pF CIN, CVCC: 4.7µF, 16V, X7R, 1206 C1: 100nF, 16V, X7R, 1206 COUT: 100µF, 16V, X7R, 1812 L1: TDK SPM6530T-3R3M SC1, SC2: MAXWELL BCAP0050-P270 500ns/DIV VIN = 0.5V OUTPUT CURRENT = 50mA 3122 TA08b 3122 TA08a Related Parts PART NUMBER DESCRIPTION COMMENTS LTC3421 3A ISW, 3MHz, Synchronous Step-Up DC/DC Converter with Output Disconnect 95% Efficiency, VIN = 0.5V to 4.5V, VOUT(MAX) = 5.25V, IQ = 12μA, ISD < 1μA, QFN24 Package LTC3422 1.5A ISW, 3MHz Synchronous Step-Up DC/DC Converter with Output Disconnect 95% Efficiency, VIN = 0.5V to 4.5V, VOUT(MAX) = 5.25V, IQ = 25μA, ISD < 1μA, 3mm × 3mm DFN Package LTC3112 2.5A ISW, 750kHz, Synchronous Buck-Boost DC/DC Converter with Output Disconnect, Burst Mode Operation 95% Efficiency, VIN = 2.7V to 15V, VOUT(MAX) = 14V, IQ = 50μA, ISD < 1μA, 4mm × 5mm DFN and TSSOP Packages LTC3458 1.4A ISW, 1.5MHz, Synchronous Step-Up DC/DC Converter/ Output Disconnect/Burst Mode Operation 93% Efficiency, VIN = 1.5V to 6V, VOUT(MAX) = 7.5V, IQ = 15μA, ISD < 1μA, DFN12 Package LTC3528 1A ISW, 1MHz, Synchronous Step-Up DC/DC Converter with Output Disconnect/Burst Mode Operation 94% Efficiency, VIN = 700mV to 5.25V, VOUT(MAX) = 5.25V, IQ = 12µA, ISD < 1µA, 3mm × 2mm DFN Package LTC3539 2A ISW, 1MHz/2MHz, Synchronous Step-Up DC/DC Converters with Output Disconnect/Burst Mode Operation 94% Efficiency, VIN = 700mV to 5.25V, VOUT(MAX) = 5.25V, IQ = 10µA, ISD < 1µA, 3mm × 2mm DFN Package LTC3459 70mA ISW, 10V Micropower Synchronous Boost Converter/ Output Disconnect/Burst Mode Operation VIN = 1.5V to 5.5V, VOUT(MAX) = 10V, IQ = 10μA, ISD < 1μA, ThinSOT™ Package LTC3499 750mA Synchronous Step-Up DC/DC Converters with Reverse-Battery Protection 94% Efficiency, VIN = 1.8V to 5.5V, VOUT(MAX) = 6V, IQ = 20µA, ISD < 1µA, 3mm × 3mm DFN and MSOP Packages LTC3115-1 40V, 2A Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN = 2.7V to 40V, VOUT(MAX) = 40V, IQ = 50µA, ISD < 3µA, 4mm × 5mm DFN and TSSOP Packages 3122f 24 Linear Technology Corporation LT 0712 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2012