LINER LT3575

LT3798
Isolated No Opto-Coupler
Flyback Controller
with Active PFC
DESCRIPTION
FEATURES
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Isolated PFC Flyback with Minimum Number of
External Components
VIN and VOUT Limited Only by External Components
Active Power Factor Correction
Low Harmonic Distortion
No Opto-Coupler Required
Constant-Current and Constant-Voltage Regulation
Accurate Regulated Voltage and Current (±5% Typical)
Energy Star Compliant (<0.5W No Load Operation)
Thermally Enhanced 16-lead MSOP Package
The LT®3798 is a constant-voltage/constant-current isolated flyback controller that combines active power factor
correction (PFC) with no opto-coupler required for output
voltage feedback into a single-stage converter. A LT3798
based design can achieve a power factor of greater than 0.97
by actively modulating the input current, allowing compliance with most Harmonic Current Emission requirements.
The LT3798 is well suited for a wide variety of off-line
applications. The input range can be scaled up or down,
depending mainly on the choice of external components.
Efficiencies higher than 86% can be achieved with output
power levels up to 100W. In addition, the LT3798 can easily
be designed into high DC input applications.
APPLICATIONS
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Offline 5W to 100W+ Applications
High DC VIN Isolated Applications
Offline Bus Converter (12V, 24V or 48V Outputs)
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear
Technology Corporation. All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 5438499 and 7471522.
TYPICAL APPLICATION
Universal Input 24W PFC Bus Converter
VOUT vs IOUT
24.50
90V
TO 265V
AC
0.1μF
499k
100k
499k
100k
D2 20Ω
4:1:1
4.7pF
10μF
2k
D3
VIN
VIN_SENSE
1M
DCM
D4
90.9k
Z1
EN/UVLO
4.99k
221k
16.5k
CTRL3
SENSE
CTRL1
OVP
INTVCC
GND
2.2μF
0
0.2
0.4
0.6
IOUT (A)
0.8
1
0.05Ω
4.7μF
COMP
23.50
Z2
GATE
CTRL2
VC
VAC = 90V
VAC = 120V
VAC = 220V
VAC = 265V
3798 TA01b
20Ω
+
24V
1A
560μF
×2
22pF
D1
LT3798
VREF
40.2k
24.00
23.75
FB
95.3k
100k
VOUT (V)
24.25
2.2nF
COMP–
0.1μF
3798 TA01a
3798f
1
LT3798
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
EN/UVLO...................................................................30V
VIN ............................................................................42V
INTVCC ......................................................................12V
CTRL1, CTRL2, CTRL3 ................................................4V
FB, VREF, COMP + ........................................................3V
VC, OVP, COMP– .........................................................4V
SENSE ......................................................................0.4V
VIN_SENSE .................................................................1mA
DCM .......................................................................±3mA
Operating Temperature Range (Note 2)
LT3798E/LT3798I................................... –40°C to 125°C
Storage Temperature Range .................. –65°C to 150°C
TOP VIEW
CTRL1
CTRL2
CTRL3
VREF
OVP
VC
COMP+
COMP–
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
17
GND
VIN_SENSE
SENSE
GATE
INTVCC
EN/UVLO
VIN
DCM
FB
MSE PACKAGE
16-LEAD PLASTIC MSOP
θJA = 50°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3798EMSE#PBF
LT3798EMSE#TRPBF
3798
16-Lead Plastic MSOP
–40°C to 125°C
LT3798IMSE#PBF
LT3798IMSE#TRPBF
3798
16-Lead Plastic MSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, INTVCC = 11V, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
Input Voltage Range
TYP
10
MAX
UNITS
38
V
70
μA
μA
VIN Quiescent Current
VEN/UVLO = 0.2V
VEN/UVLO = 1.5V, Not Switching
VIN Quiescent Current, INTVCC Overdriven
VINTVCC = 11V
60
μA
VIN Shunt Regulator Voltage
I = 1mA
40
V
45
VIN Shunt Regulator Current Limit
60
70
8
INTVCC Quiescent Current
VEN/UVLO = 0.2V
VEN/UVLO = 1.5V, Not Switching
EN/UVLO Pin Threshold
EN/UVLO Pin Voltage Rising
EN/UVLO Pin Hysteresis Current
EN/UVLO=1V
VIN_SENSE Threshold
Turn Off
VREF Voltage
0 μA Load
200μA Load
CTRL1/CTRL2/CTRL3 Pin Bias Current
CTLR1/CTRL2/CTRL3 = 1V
l
mA
12.5
1.8
15.5
2.2
17.5
2.7
μA
mA
1.21
1.25
1.29
V
8
10
12
μA
1.97
1.95
2.0
1.98
27
l
l
μA
2.03
2.03
V
V
±30
nA
3798f
2
LT3798
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, INTVCC = 11V, unless otherwise noted.
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
96
102
107
mV
SENSE Current Limit Threshold
VIN_SENSE = 150μA
Minimum SENSE Current Limit Threshold
VIN_SENSE = 34μA
14
mV
Minimum SENSE Current Limit Threshold
VIN_SENSE = 21μA
4
mV
SENSE Input Bias Current
Current Out of Pin, SENSE = 0V
15
μA
l
FB Voltage
1.22
1.25
1.28
V
0.01
0.03
%/V
4.25
4.4
FB voltage Line Regulation
10V < VIN < 35V
FB Pin Bias Current
(Note 3), FB = 1V
FB Error Amplifier Voltage Gain
ΔVVC/ΔVFB, CTRL1=1V, CTRL2=2V, CTRL3=2V
180
V/V
FB Error Amplifier Transconductance
ΔI = 5μA
170
UMHOS
Current Error Amplifier Voltage Gain
ΔVCOMP+/ΔVCOMP–, CTRL1 = 1V, CTRL2 = 2V, CTRL3 = 2V
100
V/V
Current Error Amplifier Transconductance
ΔI = 5μA
50
UMHOS
Current Loop Voltage Gain
ΔVCTRL/ΔVSENSE,1000pF Cap from COMP+ to COMP–
21
V/V
DCM Current Turn-On Threshold
Current Out of Pin
80
μA
Maximum Oscillator Frequency
COMP+ = 0.95V, VIN_SENSE = 150μA
150
kHz
Minimum Oscillator Frequency
COMP+ = 0V, VFB <VOVP
4
kHz
Minimum Oscillator Frequency
COMP+ = 0V, VFB >VOVP
0.5
kHz
20
kHz
4.05
Backup Oscillator Frequency
μA
Linear Regulator
INTVCC Regulation Voltage
No Load
9.8
10
10.4
V
500
900
mV
Dropout (VIN-INTVCC)
IINTVCC = –10mA, VIN = 10V
Current Limit
Below Undervoltage Threshold
12
25
mA
Current Limit
Above Undervoltage Threshold
80
120
mA
Gate Driver
tr GATE Driver Output Rise Time
CL = 3300pF, 10% to 90%
18
ns
tf GATE Driver Output Fall Time
CL = 3300pF, 90% to 10%
18
ns
GATE Output Low (VOL)
GATE Output High (VOH)
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3798E is guaranteed to meet specified performance from
0°C to 125°C junction temperature. Specification over the –40°C and
0.01
V
INTVCC50mV
V
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
3798I is guaranteed to meet specified performance from –40°C to 125°C
operating junction temperature range.
Note 3: Current flows out of the FB pin.
3798f
3
LT3798
TYPICAL PERFORMANCE CHARACTERISTICS
EN/UVLO Threshold
vs Temperature
EN/UVLO Hysteresis Current
vs Temperature
RISING
1.26
1.24
FALLING
1.22
1.2
–50 –25
0
90
11.5
11
20
10
0
2.05
2.075
2.04
120
MAX ILIM
100
CURRENT LIMIT (mA)
VREF (V)
2.02
2.000
VIN = 24V WITH 200μA LOAD
2.01
NO LOAD
2
1.99
200μA LOAD
1.98
1.950
1.97
1.925
1.96
1.900
–50 –25
1.95
25 50 75 100 125 150
TEMPERATURE (°C)
60
40
20
15
25
VIN (V)
30
35
0
–50 –25
40
Backup Oscillator Frequency
vs Temperature
25
FREQUENCY (kHz)
20
VFB < VOVP
FREQUENCY (kHz)
4
25 50 75 100 125 150
TEMPERATURE (°C)
3798 G06
5
195
0
3798 G05
Minimum Oscillator Frequency
vs Temperature
220
MIN ILIM VIN_SENSE = 34μA
MIN ILIM VIN_SENSE = 21μA
10
Maximum Oscillator Frequency
vs Temperature
80
20
3798 G05
170
25 50 75 100 125 150
TEMPERATURE (°C)
SENSE Current Limit Threshold
vs Temperature
2.03
VIN = 24V WITH NO LOAD
0
3798 G03
VREF vs VIN
2.050
VREF (V)
0
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
3798 G02
2.100
0
50
30
10.5
VREF vs Temperature
1.975
60
40
3798 G01
2.025
VIN = 12V
70
10
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
VIN = 24V
80
IQ (μA)
EN/UVLO HYSTERESIS CURRENT (μA)
1.28
FREQUENCY (kHz)
VIN IQ vs Temperature
100
12
1.3
EN/UVLO (V)
TA = 25°C, unless otherwise noted.
3
2
15
10
145
5
1
VOVP > VFB
120
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3798 G06a
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3799 G07
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3798 G07a
3798f
4
LT3798
TYPICAL PERFORMANCE CHARACTERISTICS
INTVCC vs Temperature
TA = 25°C, unless otherwise noted.
INTVCC vs VIN
10.5
NO LOAD
10mA LOAD
25mA LOAD
VIN Shunt Voltage vs Temperature
10.2
42
10
41.5
VIN SHUNT VOLTAGE (V)
10.25
INTVCC (V)
INTVCC (V)
9.8
10
9.6
9.4
9.2
0
9
25 50 75 100 125 150
TEMPERATURE (°C)
5
10
25
20
VIN (V)
30
35
9
8
7
6
25 50 75 100 125 150
TEMPERATURE (°C)
25 50 75 100 125 150
TEMPERATURE (°C)
3798 G10
VOUT vs Temperature
2
25
PAGE 17 SCHEMATIC:
UNIVERSAL
1.8
1.6
24.5
1.4
1.2
1
0.8
VAC = 120V
24
VAC = 220V
0.6
23.5
0.4
0.2
0
0
120
60
40
80
100
20
SENSE CURRENT LIMIT THRESHOLD (mV)
23
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
3798 G012a
3798 G12
Output Voltage
vs Input Voltage
Output Current
vs Input Voltage
24.6
1.10
PAGE 17 SCHEMATIC:
UNIVERSAL
PAGE 17 SCHEMATIC:
UNIVERSAL
OUTPUT CURRENT (A)
24.4
VOUT (V)
0
3798 G09
3798 G11
24.2
24
23.8
23.6
39
–50 –25
40
VOUT (V)
LEAKAGE INDUCTANCE BLANKING TIME (μs)
10
SHUNT CURRENT (mA)
15
Leakage Inductance Blanking Time
vs SENSE Current Limit Threshold
Maximum Shunt Current
vs Temperature
0
40
39.5
3798 G08
5
–50 –25
41
40.5
9.75
9.5
–50 –25
ISHUNT = 1mA
90 110 130 150 170 190 210 230 250 270
VIN (VAC)
3798 G13
1.05
VOUT = 22V
1.00
0.95
0.90
90 110 130 150 170 190 210 230 250 270
VIN (VAC)
3798 G14
3798f
5
LT3798
TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25°C, unless otherwise noted.
Power Factor vs Input Voltage
Efficiency vs Input Voltage
100
1.00
PAGE 17 SCHEMATIC:
UNIVERSAL
0.99
90
0.97
EFFICIENCY (%)
POWER FACTOR
0.98
0.96
0.95
0.94
0.93
80
70
0.92
0.91 PAGE 17 SCHEMATIC:
UNIVERSAL
0.90
90 110 130 150 170 190 210 230 250 270
VIN (VAC)
3798 G15
60
90 110 130 150 170 190 210 230 250 270
VIN (VAC)
3798 G16
PIN FUNCTIONS
CTRL1, CTRL2, CTRL3 (Pin 1, Pin 2, Pin 3): Current
Output Adjustment Pins. These pins control the output
current. The lowest value out of the three CTRL inputs is
compared to negative input of the operational amplifier.
VREF (Pin 4): Voltage Reference Output Pin. Typically 2V.
This pin drives a resistor divider for the CTRL pin, either
for analog dimming or for temperature limit/compensation
of output load. Can supply up to 200μA.
OVP (Pin 5): Overvoltage Protection. This pin accepts a
DC voltage to compare to the sample and hold’s voltage
output information. When output voltage information is
above the OVP, the part divides the minimum switching
frequency by 8, around 500Hz. This protects devices connected to the output. This also allows the part to operate
with very little power consumption with no load to meet
energy star requirements.
VC (Pin 6): Compensation Pin for Internal Error Amplifier.
Connect a series RC from this pin to ground to compensate
the switching regulator. A 100pF capacitor in parallel helps
eliminate noise.
COMP+, COMP– (Pin 7, Pin 8): Compensation Pins for
Internal Error Amplifier. Connect a capacitor between these
two pins to compensate the internal feedback loop.
FB (Pin 9): Voltage Loop Feedback Pin. FB is used to
regulate the output voltage by sampling the third winding. If the converter is used in current mode, the FB pin
will normally be at a voltage level lower than 1.25V, and
will reach the steady state of 1.25V if it detects an open
output condition.
DCM (Pin 10): Discontinuous Conduction Mode Detection
Pin. Connect a capacitor and resistor in series with this
pin to the third winding.
VIN (Pin 11): Input Voltage. This pin supplies current to
the internal start-up circuitry and to the INTVCC LDO. This
pin must be locally bypassed with a capacitor. A 42V shunt
regulator is internally connected to this pin.
EN/UVLO (Pin 12): Enable/Undervoltage Lockout. A resistor divider connected to VIN is tied to this pin to program
the minimum input voltage at which the LT3798 will turn
on. When below 1.25V, the part will draw 60μA with most
of the internal circuitry disabled and a 10μA hysteresis
current will be pulled out of the EN/UVLO pin. When above
1.25V, the part will be enabled and begin to switch and the
10μA hysteresis current is turned off.
INTVCC (Pin 13): Regulated Supply for Internal Loads
and GATE Driver. Supplied from VIN and regulates to 10V
(typical). INTVCC must be bypassed with a 4.7μF capacitor
placed close to the pin.
3798f
6
LT3798
PIN FUNCTIONS
GATE (Pin 14): N-Channel FET Gate Driver Output. Switches
between INTVCC and GND. Driven to GND during shutdown
state and stays high during low voltage states.
VIN_SENSE (Pin 16): Line Voltage Sense Pin. The pin is
used for sensing the AC line voltage to perform power
factor correction. Connect a resistor in series with the
line voltage to this pin. If no PFC is needed, connect this
pin to INTVCC with a 25k resistor.
SENSE (Pin 15): The Current Sense Input for the Control
Loop. Kelvin connect this pin to the positive terminal of the
switch current sense resistor, RSENSE, in the source of the
NFET. The negative terminal of the current sense resistor
should be connected to the GND plane close to the IC.
GND (Exposed Pad Pin 17): Ground. The exposed pad
of the package provides both electrical contact to ground
and good thermal contact to the printed circuit board.
The exposed pad must be soldered to the circuit board
for proper operation.
BLOCK DIAGRAM
VRECTIFIED
D2
R3
t
R4
R1
R13
C2
L1C
C1
T1
R15
EN/UVLO
VIN_SENSE
+
A2
+ –
600mV
S&H
R9
COMP+
S
C5
GATE
R
Q
A3
FB
VOUT –
R11
+
A1
–
A9
C7
INTVCC
R7
CURRENT
COMPARATOR
–
OVP
–
A7
+
1.22V
ONE
SHOT
N:1
VIN
STARTUP
INTERNAL REG
VREF
L1B
R14
DCM
R8
VOUT +
C3
L1A
R5
D1
S
MASTER
LATCH
R12
1M
C6
COMP–
SW1
CTRL1
–
A5
+
CTRL2
MINIMUM
DRIVER
SENSE
A4
A6
MULTIPLIER
GND
LOW OUTPUT
CURRENT
OSCILLATOR
R6
CTRL3
S&H
FB
1.22V
VC
R10
+
A8
–
3798 BD
C4
3798f
7
LT3798
OPERATION
The LT3798 is a current mode switching controller IC
designed specifically for generating a constant current/
constant voltage supply in an isolated flyback topology.
The special problem normally encountered in such circuits
is that information relating to the output voltage and current on the isolated secondary side of the transformer
must be communicated to the primary side in order to
maintain regulation. Historically, this has been done with
an opto-isolator. The LT3798 uses a novel method of using
the external MOSFETs peak current information from the
sense resistor to calculate the output current of a flyback
converter without the need of an opto-coupler.
Active power factor correction is becoming a requirement
for offline power supplies and the power levels are decreasing. A power factor of one is achieved if the current
drawn is proportional to the input voltage. The LT3798
modulates the peak current limit with a scaled version
of the input voltage. This technique can provide power
factors of 0.97 or greater.
The Block Diagram shows an overall view of the system. The
external components are in a flyback topology configuration. The third winding senses the output voltage and also
supplies power to the part in steady-state operation. The
VIN pin supplies power to an internal LDO that generates
10V at the INTVCC pin. The novel control circuitry consists
of two error amplifiers, a minimum circuit, a multiplier,
a transmission gate, a current comparator, a low output
current oscillator and a master latch, which will be explained in the following sections. The part also features a
sample-and-hold to sample the output voltage from the
third winding. A comparator is used to detect discontinuous conduction mode (DCM) with a cap connected to the
third winding. The part features a 1.9A gate driver.
The LT3798 is designed for both off-line and DC applications. The EN/UVLO and a resistor divider can be configured
for a micropower hysteretic start-up. In the Block Diagram,
R3 is used to stand off the high voltage supply voltage.
The internal LDO starts to supply current to the INTVCC
when VIN is above 2.5V. The VIN and INTVCC capacitors are
charged by the current from R3. When VIN exceeds the
turn-on threshold and INTVCC is in regulation at 10V, the
part begins to switch. The VIN hysteresis is set by the EN/
UVLO resistor divider. The third winding provides power
to VIN when its voltage is higher than the VIN voltage. A
voltage shunt is provided for fault protection and can sink
8mA of current when VIN is over 40V.
During a typical cycle, the gate driver turns the external
MOSFET on and a current flows through the primary winding. This current increases at a rate proportional to the
input voltage and inversely proportional to the magnetizing
inductance of the transformer. The control loop determines
the maximum current and the current comparator turns
the switch off when the current level is reached. When the
switch turns off, the energy in the core of the transformer
flows out the secondary winding through the output diode,
D1. This current decreases at a rate proportional to the
output voltage. When the current decreases to zero, the
output diode turns off and voltage across the secondary
winding starts to oscillate from the parasitic capacitance
and the magnetizing inductance of the transformer. Since
all windings have the same voltage across them, the third
winding rings too. The capacitor connected to the DCM
pin, C1, trips the comparator A2, which serves as a dv/dt
detector, when the ringing occurs. This timing information
is used to calculate the output current and will be described
below. The dv/dt detector waits for the ringing waveform
to reach its minimum value and then the switch turns back
on. This switching behavior is similar to zero volt switching
and minimizes the amount of energy lost when the switch
is turned back on and improves efficiency as much as
5%. Since this part operates on the edge of continuous
conduction mode and discontinuous conduction mode,
the operating mode is called critical conduction mode (or
boundary conduction mode).
Primary Side Control Loops
The LT3798 achieves constant current/constant voltage
operation by using two separate error amplifiers. These
two amplifiers are then fed to a circuit that outputs the
lower voltage of the two, shown as the "minimum" block in
the Block Diagram. This voltage is converted to a current
before being fed into the multiplier.
3798f
8
LT3798
OPERATION
Primary Side Current Control Loop
The CTRL1/CTRL2/CTRL3 pins control the output current
of the flyback controller. To simplify the loop, let’s assume
the VIN_SENSE pin is held at a constant voltage above 1V
eliminating the multiplier from the control loop. The error
amplifier, A5, is configured as integrator with the external
capacitor C6. The COMP+ node voltage is converted to a
current into the multiplier with the V/I converter, A6. Since
A7’s output is constant, the output of the multiplier is
proportional to A6 and can be ignored. The output of the
multiplier controls the peak current with its connection to
the current comparator, A1. The output of the multiplier is
also connected to the transmission gate, SW1, and to a
1M resistor. The transmission gate, SW1, turns on when
the secondary current flows to the output capacitor. This
is called the flyback period when the output diode D1 is
on. The current through the 1M resistor gets integrated by
A5. The lowest CTRL input is equal to the negative input
of A5 in steady state.
A current output regulator normally uses a sense resistor
in series with the output current and uses a feedback loop
to control the peak current of the switching converter. In
this isolated case, the output current information is not
available so instead the LT3798 calculates it using the information available on the primary side of the transformer.
The output current may be calculated by taking the average
of the output diode current. As shown in Figure 1, the diode
current is a triangle waveform with a base of the flyback
time and a height of the peak secondary winding current.
In a flyback topology, the secondary winding current is N
times the primary winding current, where NPS is the primary
to secondary winding ratio. Instead of taking the area of
the triangle, let’s think of it as a pulse width modulation
(PWM) waveform. During the flyback time, the average
current is half the peak secondary winding current and
zero during the rest of the cycle. The equation to express
the output current is:
IOUT = 0.5 • IPK • NPS • D
where D is equal to the percentage of the cycle that the
flyback time represents. The LT3798 has access to the
primary winding current, the input to the current comparator, and when the flyback time starts and ends. Now
the output current can be calculated by averaging a PWM
IPK(sec)
SECONDARY
DIODE CURRENT
SWITCH
WAVEFORM
TFLYBACK
3798 F01
TPERIOD
Figure 1. Secondary Diode Current and Switch Waveforms
waveform with a height of the current limit and a duty cycle
of the flyback time over the entire cycle. In the feedback
loop described above, the input to the integrator is such
a waveform. The integrator adjusts the peak current until
calculated output current equals the control voltage. If the
calculated output current is low compared to the control
pin, the error amplifier increases the voltage on the COMP+
node thus increasing the current comparator input.
Primary Side Voltage Control
The output voltage is available through the third winding on
the primary side. A resistor divider attenuates the output
voltage for the voltage error amplifier. A sample-and-hold
circuit samples the attenuated output voltage and feeds it
to the error amplifier. The output of the error amplifier is
the VC pin. This node needs a capacitor to compensate
the output voltage control loop.
Power Factor Correction
When the VIN_SENSE voltage is connected to a resistor divider
of the supply voltage, the current limit is proportional to
the supply voltage. The minimum of the two error amplifier outputs is multiplied with the VIN_SENSE pin voltage. If
the LT3798 is configured with a fast control loop, slower
changes from the VIN_SENSE pin would not interfere with
the current limit or the output current. The COMP+ pin
would adjust to the changes of the VIN_SENSE. The only
way for the multiplier to function is to set the control loop
to be an order of magnitude slower than the fundamental
frequency of the VIN_SENSE signal. In an offline case, the
3798f
9
LT3798
OPERATION
fundamental frequency of the supply voltage is 120Hz so
the control loop unity gain frequency needs to be set less
than approximately 12Hz. Without a large amount of energy
storage on the secondary side, the output current will be
affected by the supply voltage changes, but the DC component of the output current will be accurate. For DC input
or non-PFC AC input applications, connect a 25k resistor
from VIN_SENSE to INTVCC instead of the AC line voltage.
VIN
R1
EN/UVLO
R2
LT3798
GND
3798 F02
Figure 2. Undervoltage Lockout (UVLO)
Startup
The LT3798 uses a hysteretic start-up to operate from
high offline voltages. A resistor connected to the supply
voltage protects the part from high voltages. This resistor
is connected to the VIN pin on the part and bypassed with
a capacitor. When the resistor charges the VIN pin to a
turn-on voltage set with the EN/UVLO resistor divider and
the INTVCC pin is at its regulation point, the part begins
to switch. The resistor cannot provide power for the part
in steady state, but relies on the capacitor to start-up the
part, then the third winding begins to provide power to the
VIN pin along with the resistor. An internal voltage clamp
is attached to the VIN pin to prevent the resistor current
from allowing VIN to go above the absolute maximum
voltage of the pin. The internal clamp is set at 40V and is
capable of 8mA(typical) of current at room temperature.
Programming Output Voltage
Setting the VIN Turn-On and Turn-Off Voltages
The temperature coefficient of the diode's forward drop
needs to be the opposite of the term, (R4 • ITC)/NST. By
taking the partial derivative with respect to temperature,
the value of R4 is found to be the following:
A large voltage difference between the VIN turn-on voltage
and the VIN turn-off voltage is preferred to allow time for the
third winding to power the part. The EN/UVLO sets these
two voltages. The pin has a 10μA current sink when the
pins voltage is below 1.25V and 0μA when above 1.25V.
The VIN pin connects to a resistor divider as shown in
Figure 2. The UVLO threshold for VIN rising is:
1.25V • (R1+ R2)
+ 10μA • R1
R2
The UVLO Threshold for VIN Falling is :
VIN(UVLO,RISING) =
VIN(UVLO,FALLING)
1.25V • (R1+ R2)
=
R2
The output voltage is set using a resistor divider from
the third winding to the FB pin. From the Block Diagram,
the resistors R4 and R5 form a resistor divider from the
third winding. The FB also has an internal current source
that compensates for the diode drop. This current source
causes an offset in the output voltage that needs to be accounted for when setting the output voltage. The output
voltage equation is:
VOUT = VBG (R4+R5)/(NST • R5)–(VF + (R4 • ITC)/NST)
where VBG is the internal reference voltage, NST is the
winding ratio between the secondary winding and the third
winding, VF is the forward drop of the output rectifying
diode, and ITC is the internal current source for the FB pin.
R4 = NST(1/(δITC /δT)(δVF /δT))
δITC /δT = 12.4nA/°C
ITC = 4.25μA
where δITC /δT is the partial derivative of the ITC current
source, and δVF /δT is the partial derivative of the forward
drop of the output rectifying diode.
With R4 set with the above equation, the resistor value
for R5 is found using the following:
R5 = (VBG • R4)/(NST(VOUT+VF)+R4 • ITC-VBG)
3798f
10
LT3798
OPERATION
Programming Output Current
The maximum output current depends on the supply voltage and the output voltage in a flyback topology. With the
VIN_SENSE pin connected to 100μA current source and a DC
supply voltage, the maximum output current is determined
at the minimum supply voltage, and the maximum output
voltage using the following equation:
NPS
IOUT(MAX) = 2•(1– D)•
42 • RSENSE
where
VOUT • NPS
D=
VOUT • NPS + VIN
The maximum control voltage to achieve this maximum
output current is 2V • (1-D).
It is suggested to operate at 95% of these values to give
margin for the part’s tolerances.
When designing for power factor correction, the output
current waveform is going to have a half sine wave squared
shape and will no longer be able to provide the above
currents. By taking the integral of a sine wave squared
over half a cycle, the average output current is found to
be half the value of the peak output current. In this case,
the recommended maximum average output current is
as follows:
NPS
IOUT(MAX) = 2•(1−D) •
• 47.5%
42 • RSENSE
where
D=
VOUT • NPS
VOUT • NPS + VIN
The maximum control voltage to achieve this maximum
output current is (1-D) • 47.5%.
For control voltages below the maximum, the output current is equal to the following equation:
NPS
IOUT = CTRL •
42 • RSENSE
The VREF pin supplies a 2V reference voltage to be used
with the control pins. To set an output current, a resistor
divider is used from the 2V reference to one of the control
pins. The following equation sets the output current with
a resistor divider:
⎛
⎞
2NPS
R1= R2 ⎜
– 1⎟
⎝ 42 •IOUT • RSENSE ⎠
where R1 is the resistor connected to the VREF pin and the
CTRL pin and R2 is the resistor connected to the CTRL
pin and ground.
Setting VIN_SENSE Resistor
The VIN_SENSE resistor sets the current feeding the internal
multiplier that modulates the current limit for power factor
correction. At the maximum line voltage, VMAX, the current
is set to 360μA. Under this condition, the resistor value is
equal to (VMAX/360μA).
For DC input or non-PFC AC input applications, connect
a 25k resistor from VIN_SENSE to INTVCC instead of the
AC line voltage.
Critical Conduction Mode Operation
Critical conduction mode is a variable frequency switching
scheme that always returns the secondary current to zero
with every cycle. The LT3798 relies on boundary mode and
discontinuous mode to calculate the critical current because
the sensing scheme assumes the secondary current returns
to zero with every cycle. The DCM pin uses a fast current
input comparator in combination with a small capacitor to
detect dv/dt on the third winding. To eliminate false tripping due to leakage inductance ringing, a blanking time of
between 600ns and 2μs is applied after the switch turns off,
depending on the current limit shown in the Leakage Inductance Blanking Time vs SENSE Current Limit Threshold
curve in the Typical Performance Characteristics section.
The detector looks for 80μA of current through the DCM
pin due to falling voltage on the third winding when the
secondary diode turns off. This detection is important since
the output current is calculated using this comparator’s
output. This is not the optimal time to turn the switch on
because the switch voltage is still close to VIN + VOUT • NPS
and would waste all the energy stored in the parasitic capacitance on the switch node. Discontinuous ringing begins
when the secondary current reaches zero and the energy
in the parasitic capacitance on the switch node transfers
3798f
11
LT3798
OPERATION
to the input capacitor. This is a second-order network
composed of the parasitic capacitance on the switch node
and the magnetizing inductance of the primary winding
of the transformer. The minimum voltage of the switch
node during this discontinuous ring is VIN – VOUT • NPS.
The LT3798 turns the switch back on at this time, during
the discontinuous switch waveform, by sensing when
the slope of the switch waveform goes from negative to
positive using the dv/dt detector. This switching technique
may increase efficiency by 5%.
Sense Resistor Selection
The resistor, RSENSE, between the source of the external
N-channel MOSFET and GND should be selected to provide
an adequate switch current to drive the application without
exceeding the current limit threshold.
For applications without power factor correction, select a
resistor according to:
RSENSE =
2(1– D)NPS
• 95%
IOUT • 42
VOUT • NPS
VOUT • NPS + VIN
For applications with power factor correction, select a
resistor according to:
RSENSE =
2(1– D)NPS
• 47.5%
IOUT • 42
where
D=
To help improve crossover distortion of the line input
current, a second minimum current limit of 6% becomes
active when the VIN_SENSE current is lower than 27μA. Since
the off-time becomes very short with this lower minimum
current limit, the sample-and-hold is deactivated.
Universal Input
The LT3798 operates over the universal input voltage
range of 90VAC to 265VAC. In the Typical Performance
Characteristics section, the Output Voltage vs VIN and the
Output Current vs VIN graphs, show the output voltage
and output current line regulation for the first application
picture in the Typical Applications section.
Selecting Winding Turns Ratio
where
D=
very high frequency. The output voltage sensing circuitry
needs a minimum amount of flyback waveform time to
sense the output voltage on the third winding. The time
needed is 350ns. The minimum current limit allows the
use of smaller transformers since the magnetizing primary
inductance does not need to be as high to allow proper
time to sample the output voltage information.
VOUT • NPS
VOUT • NPS + VIN
Minimum Current Limit
The LT3798 features a minimum current limit of approximately 18% of the peak current limit. This is necessary
when operating in critical conduction mode since low
current limits would increase the operating frequency to a
Boundary mode operation gives a lot of freedom in selecting
the turns ratio of the transformer. We suggest to keep the
duty cycle low, lower NPS, at the maximum input voltage
since the duty cycle will increase when the AC waveform
decreases to zero volts. A higher NPS increases the output
current while keeping the primary current limit constant.
Although this seems to be a good idea, it comes at the
expense of a higher RMS current for the secondary-side
diode which might not be desirable because of the primary
side MOSFET’s superior performance as a switch. A higher
NPS does reduce the voltage stress on the secondary-side
diode while increasing the voltage stress on the primaryside MOSFET. If switching frequency at full output load is
kept constant, the amount of energy delivered per cycle by
the transformer also stays constant regardless of the NPS.
Therefore, the size of the transformer remains the same at
practical NPS’s. Adjusting the turns ratio is a good way to
find an optimal MOSFET and diode for a given application.
3798f
12
LT3798
OPERATION
Switch Voltage Clamp Requirement
Leakage inductance of an offline transformer is high due
to the extra isolation requirement. The leakage inductance
energy is not coupled to the secondary but goes into
the drain node of the MOSFET. This is problematic since
400V and higher rated MOSFETs cannot always handle
this energy by avalanching. Therefore the MOSFET needs
protection. A transient voltage suppressor (TVS) and diode
are recommended for all offline application and connected,
as shown in Figure 3. The TVS device needs a reverse
breakdown voltage greater than (VOUT + VF) • NPS where
VOUT is the output voltage of the flyback converter, VF is
the secondary diode forward voltage, and NPS is the turns
ratio. An RCD clamp can be used in place of the TVS clamp.
VSUPPLY
VSUPPLY
period, as well. Similarly, initial values can be estimated
using stated switch capacitance and transformer leakage
inductance. Once the value of the drain node capacitance
and inductance is known, a series resistor can be added
to the snubber capacitance to dissipate power and critically dampen the ringing. The equation for deriving the
optimal series resistance using the observed periods
(tPERIOD, and tPERIOD(SNUBBED)) and snubber capacitance
(CSNUBBER) is below, and the resultant waveforms are
shown in Figure 4.
CPAR =
LPAR =
CSNUBBER
⎛ tPERIOD(SNUBBED) ⎞ 2
⎜
⎟ –1
tPERIOD
⎝
⎠
tPERIOD2
CPAR • 4π2
RSNUBBER =
GATE
LPAR
CPAR
GATE
90
80
70
3798 F03
In addition to clamping the spike, in some designs where
short circuit protection is desired, it will be necessary to
decrease the amount of ringing by using an RC snubber.
Leakage inductance ringing is at its worst during a short
circuit condition, and can keep the converter from cycling
on and off by peak charging the bias capacitor. On/off
cycling is desired to keep power dissipation down in the
output diode. Alternatively, a heat sink can be used to
manage diode temperature.
The recommended approach for designing an RC snubber
is to measure the period of the ringing at the MOSFET
drain when the MOSFET turns off without the snubber
and then add capacitance—starting with something in
the range of 100pF—until the period of the ringing is 1.5
to 2 times longer. The change in period will determine
the value of the parasitic capacitance, from which the
parasitic inductance can be determined from the initial
VDRAIN (V)
Figure 3. TVS & RCD Switch Voltage Clamps
60
50
40
30
NO SNUBBER
WITH SNUBBER
CAPACITOR
WITH RESISTOR
AND CAPACITOR
20
10
0
0
0.05
0.10
0.15 0.20
TIME (μs)
0.25
0.30
3798 F04
Figure 4. Observed Waveforms at MOSFET Drain when
Iteratively Implementing an RC Snubber
Note that energy absorbed by a snubber will be converted
to heat and will not be delivered to the load. In high voltage or high current applications, the snubber may need to
be sized for thermal dissipation. To determine the power
dissipated in the snubber resistor from capacitive losses,
measure the drain voltage immediately before the MOSFET turns on and use the following equation relating that
3798f
13
LT3798
OPERATION
voltage and the MOSFET switching frequency to determine
the expected power dissipation:
PSNUBBER = fSW • CSNUBBER • VDRAIN2/2
Decreasing the value of the capacitor will reduce the dissipated power in the snubber at the expense of increased
peak voltage on the MOSFET drain, while increasing the
value of the capacitance will decrease the overshoot.
with several leading magnetic component manufacturers
to produce predesigned flyback transformers for use with
the LT3798. Table 1 shows the details of several of these
transformers.
Loop Compensation
Transformer Design Considerations
The voltage feedback loop is a traditional GM error amplifier. The loop cross-over frequency is set much lower than
twice the line frequency for PFC to work properly.
Transformer specification and design is a critical part of
successfully applying the LT3798. In addition to the usual
list of caveats dealing with high frequency isolated power
supply transformer design, the following information
should be carefully considered. Since the current on the
secondary side of the transformer is inferred by the current
sampled on the primary, the transformer turns ratio must
be tightly controlled to ensure a consistent output current.
The current output feedback loop is an integrator configuration with the compensation capacitor between the
negative input and output of the operational amplifier.
This is a one-pole system therefore a zero is not needed
in the compensation. For offline applications with PFC,
the crossover should be set an order of magnitude lower
than the line frequency of 120Hz or 100Hz. In a typical
application, the compensation capacitor is 0.1μF.
A tolerance of ±5% in turns ratio from transformer to
transformer could result in a variation of more than ±5% in
output regulation. Fortunately, most magnetic component
manufacturers are capable of guaranteeing a turns ratio
tolerance of 1% or better. Linear Technology has worked
In non-PFC applications, the crossover frequency may be
increased to improve transient performance. The desired
crossover frequency needs to be set an order of magnitude
below the switching frequency for optimal performance.
Table 1. Predesigned Transformers—Typical Specifications, Unless Otherwise Noted
TRANSFORMER SIZE
PART NUMBER (L × W × H)
LPRI
(μH)
NPSA
(NP:NS:NA)
RPRI
(mΩ)
RSEC
(mΩ)
MANUFACTURER
TARGET
APPLICATION
(VOUT /IOUT)
JA4429
400
1:0.24:0.24
252
126
Coilcraft
22V/1A
21.1mm × 21.1mm × 17.3mm
7508110210
15.75mm × 15mm × 18.5mm
2000
6.67:1:1.67
5100
165
Würth Elektronik
10V/0.4A
750813002
15.75mm × 15mm × 18.5mm
2000
20:1.0:5.0
6100
25
Würth Elektronik
3.8V/1.1A
750811330
43.2mm × 39.6mm × 30.5mm
300
6:1.0:1.0
150
25
Würth Elektronik
18V/5A
750813144
16.5mm × 18mm × 18mm
600
4:1:0.71
2400
420
Würth Elektronik
28V/0.5A
750813134
16.5mm × 18mm × 18mm
600
8:1:1.28
1850
105
Würth Elektronik
14V/1A
750811291
31mm × 31mm × 25mm
400
1:1:0.24
550
1230
Würth Elektronik
85V/0.4A
750813390
43.18mm × 39.6mm × 30.48mm
100
1:1:0.22
150
688
Würth Elektronik
90V/1A
750811290
31mm × 31mm × 25mm
460
1:1:0.17
600
560
Würth Elektronik
125V/0.32A
X-11181-002
23.5mm × 21.4mm × 9.5mm
500
72:16:10
1000
80
Premo
30V/0.5A
750811248
31mm × 31mm × 25mm
300
4:1.0:1.0
280
25
Würth Elektronik
24V/2A
S001621
25mm × 22.2mm × 16mm
820
16:1.0:4.0
1150
10
Renco
5V/4A
750312872
43.2mm × 39.6mm × 30.5mm
14
1:1:0.8
11
11
Würth Elektronik
28V/4A
3798f
14
LT3798
OPERATION
MOSFET and Diode Selection
Power Factor Correction/Harmonic Content
With a strong 1.9A gate driver, the LT3798 can effectively
drive most high voltage MOSFETs. A low Qg MOSFET is
recommended to maximize efficiency. In most applications,
the RDS(ON) should be chosen to limit the temperature rise
of the MOSFET. The drain of the MOSFET is stressed to
VOUT • NPS + VIN during the time the MOSFET is off and
the secondary diode is conducting current. But in most
applications, the leakage inductance voltage spike exceeds
this voltage. The voltage of this stress is determined by the
switch voltage clamp. Always check the switch waveform
with an oscilloscope to make sure the leakage inductance
voltage spike is below the breakdown voltage of the MOSFET. A transient voltage suppressor and diode are slower
than the leakage inductance voltage spike, therefore causing
a higher voltage than calculated.
The LT3798 attains high power factor and low harmonic
content by making the peak current of the main power
switch proportional to the line voltage by using an internal
multiplier. A power factor of >0.97 is easily attainable for
most applications by following the design equations in
this data sheet. With proper design, LT3798 applications
can easily meet most harmonic standards.
The secondary diode stress may be as much as VOUT + 2
• VIN/NPS due to the anode of the diode ringing with the
secondary leakage inductance. An RC snubber in parallel
with the diode eliminates this ringing, so that the reverse
voltage stress is limited to VOUT + VIN/NPS. With a high
NPS and output current greater than 3A, the IRMS through
the diode can become very high and a low forward drop
Schottky is recommended.
Discontinuous Mode Detection
The discontinuous mode detector uses AC-coupling to
detect the ringing on the third winding. A 22pF capacitor
with a 30k resistor in series is recommended in most
designs. Depending on the amount of leakage inductance
ringing, an additional current may be needed to prevent
false tripping from the leakage inductance ringing. A resistor from INTVCC to the DCM pin adds this current. Up to
an additional 100μA of current may be needed in some
cases. The DCM pin is roughly 0.7V, therefore the resistor
value is selected using the following equation:
R=
Operation Under Light Output Loads
The LT3798 detects output overvoltage conditions by
looking at the voltage on the third winding. The third
winding voltage is proportional to the output voltage when
the main power switch is off and the secondary diode is
conducting current. Sensing the output voltage requires
delivering power to the output. When the output current is
very low, this periodic delivery of output current can exceed
the load current. The OVP pin sets the output overvoltage threshold. When the output of the sample-and-hold
is above this voltage, the minimum switching frequency
is divided by 8 as shown in Figure 5. This OVP threshold
needs to be set above 1.35V and should be set out of the
way of output voltage transients. The output clamp point
is set with the following formula:
VOUT = VOVP(R4 + R5)/(NST • R5)–(VF + (R4•ITC)/NST)
The VOVP pin voltage may be provided by a resistor divider
from the VREF pin. This frequency division greatly reduces
the output current delivered to the output but a Zener or
resistor is required to dissipate the remaining output current. The Zener diode’s voltage needs to be 5% higher than
the output voltage set by the resistor divider connected to
the FB pin. Multiple Zener diodes in series may be needed
for higher output power applications to keep the Zener’s
temperature within the specification.
10V – 0.7V
I
where I is equal to the additional current into the DCM pin.
3798f
15
LT3798
OPERATION
Protection from Shorted Output Conditions
Usage with DC Input Voltage
During a shorted output condition as shown in Figure 6, the
LT3798 operates at the minimum operating frequency. In
normal operation, the third winding provides power to the
IC, but the third winding voltage is zero during a shorted
condition. This causes the part’s VIN UVLO to shutdown
switching. The part starts switching again when VIN has
reached its turn-on voltage.
The LT3798 is flexible enough to operate well from low
voltage to very high voltage DC input voltage applications.
When the supply voltage is less than 40V, the startup resistor is not needed and the part's VIN can be connected
directly to the supply voltage. The startup sequence for
voltages higher than 40V is the same as what is described
for high voltage offline supply voltages.
The loop compensation component values can be chosen
to provide faster loop response since the LT3798 does
not have to provide PFC for the slow 50Hz/60Hz AC input
voltage. For DC input applications, connect a 25k resistor
from VIN_SENSE to INTVCC.
V3RD WINDING
20V/DIV
VOUT
10V/DIV
IOUT
1A/DIV
1ms/DIV
3798 F05
Figure 5. Switching Waveforms When Output
Open-Circuits or at Very Light Load Conditions
VIN
20V/DIV
V3RD WINDING
50V/DIV
IPRI
1A/DIV
100ms/DIV
3798 F06
Figure 6. Switching Waveforms When Output Short-Circuits
3798f
16
LT3798
TYPICAL APPLICATIONS
Universal Input 24W PFC Bus Converter
L2
800μH
L1
33mH
90V
TO 265V
AC
BR1
C1
0.068μF
C2
0.1μF
R3
499k
R4
499k
R5
1M
R7
100k
R17
D2 20Ω
R8
100k
C4
C5 4.7pF
10μF
4:1:1
R13
2k
D3
VIN
DCM
VIN_SENSE
FB
560μF
×2
R16
20Ω
LT3798
GATE
VREF
R9
40.2k
CTRL3
SENSE
CTRL2
INTVCC
CTRL1
R10
16.5k
OVP
VC
GND
COMP+
C3
2.2μF
24V
1A
+ C10
C6
22pF
D1
R6
95.3k
R12
221k
D4
Z1
R15
4.99k
EN/UVLO
R11
100k
R14
90.9k
COMP–
M1
C9
4.7μF
Z2
RS
0.05Ω C8
2.2nF
“Y1 CAP”
C7, 0.1μF
3798 TA02
BR1: DIODES, INC. HD06
C8:
VISHAY 440LD22-R
D1:
CENTRAL SEMICONDUCTOR CMR1U-06M
D2, D3: DIODES INC. BAV20W
D4:
CENTRAL SEMICONDUCTOR CMR1U-02M
M1: FAIRCHILD FDPF15N65
T1:
COILCRAFT JA4429-AL
Z1:
FAIRCHILD SMBJ170A
Z2:
CENTRAL SEMICONDUCTOR CMZ5937B
3798f
17
LT3798
TYPICAL APPLICATIONS
Universal Input 48W PFC Application
L1
1mH
L2
27mH
BR1
C1
0.1μF
90V
TO 265V
AC
C2
0.22μF
R3
499k
R7
100k
R4
499k
R8
100k
R17
D2 47Ω
C4
22pF
C5
10μF
R13
33k
D3
R5
2.4M
VIN
R14
100k
DCM
FB
VIN_SENSE
C6
22pF
EN/UVLO
R6
301k
LT3798
R9
40.2k
R16
20Ω
CTRL3
GATE
CTRL2
SENSE
CTRL1
INTVCC
OVP
GND
R12
221k
R15
5.49k
D4
Z1
1000μF
×2
R10
31.6k
COMP+
VC
C3
1μF
COMP–
M1
C9
4.7μF
24V
2A
+ C10
D1
VREF
R11
100k
4:1:1
C11
10μF
×2
Z2
RS
0.03Ω C8
2.2nF
“Y1 CAP”
C7, 0.1μF
3798 TA03
BR1: DIODES, INC. HD06
C8:
VISHAY 440LD22-R
C11: MURATA GRM32ER7YA106KA12L
D1:
CENTRAL SEMICONDUCTOR CMR1U-06M
D2, D3: DIODES INC. BAV20W
D4:
DIODES INC. SBR20A200CTB
M1: INFINEON IPB60R165CP
T1:
WÜRTH ELEKTRONIK 750811248
Z1:
FAIRCHILD SMBJ170A
Z2:
CENTRAL SEMICONDUCTOR CMZ5937B
3798f
18
LT3798
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
MSE Package
16-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1667 Rev E)
BOTTOM VIEW OF
EXPOSED PAD OPTION
2.845 ± 0.102
(.112 ± .004)
5.23
(.206)
MIN
2.845 ± 0.102
(.112 ± .004)
0.889 ± 0.127
(.035 ± .005)
8
1
1.651 ± 0.102
(.065 ± .004)
1.651 ± 0.102 3.20 – 3.45
(.065 ± .004) (.126 – .136)
0.305 ± 0.038
(.0120 ± .0015)
TYP
16
0.50
(.0197)
BSC
4.039 ± 0.102
(.159 ± .004)
(NOTE 3)
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
0.35
REF
0.12 REF
DETAIL “B”
CORNER TAIL IS PART OF
DETAIL “B” THE LEADFRAME FEATURE.
FOR REFERENCE ONLY
9
NO MEASUREMENT PURPOSE
0.280 ± 0.076
(.011 ± .003)
REF
16151413121110 9
DETAIL “A”
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
0° – 6° TYP
GAUGE PLANE
0.53 ± 0.152
(.021 ± .006)
1234567 8
DETAIL “A”
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
0.50
NOTE:
(.0197)
1. DIMENSIONS IN MILLIMETER/(INCH)
BSC
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL
NOT EXCEED 0.254mm (.010") PER SIDE.
0.86
(.034)
REF
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MSE16) 0911 REV E
3798f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LT3798
TYPICAL APPLICATION
112W Wide DC Input Industrial Power Supply
VIN
20V TO 60V
R17
D2 20Ω
R7
6.8k
C2
10μF
TO
INTVCC
R8
6.8k
R4
24k
R5
402k
C4
15pF
C5
10μF
1:1:0.8
R13
15k
D3
VIN
R14
100k
DCM
FB
VIN_SENSE
R15
5.9k
EN/UVLO
R6
51.1k
LT3798
D4
C6
22pF
Z1
Z3
D1
D5
C10
10μF
×4
28V
4A
VREF
R11
100k
R9
40.2k
CTRL3
GATE
CTRL2
SENSE
CTRL1
INTVCC
M1
C9
4.7μF
OVP
Z2
RS
0.004Ω
C8
2.2nF
GND
R12
221k
R10
34.8k
VC
COMP+
C3
0.1μF
R3
16.2k
COMP–
“Y1 CAP”
C7, 22nF
C2:
TDK C5750X7S2A106M
C8:
VISHAY 440LD22-R
C10: MURATA GRM32ER7YA106KA12L
D1:
DIODES INC. DFLS1150
D2, D3: DIODES INC. BAV20W
D4:
ON SEMICONDUCTOR MBR20200CT
D5:
DIODES INC. DFLS2100
3798 TA04
M1:
T1:
Z1:
Z2:
Z3:
FAIRCHILD FDP2532
WÜRTH ELEKTRONIK 750312872
DIODES INC. SMCJ60A
CENTRAL SEMICONDUCTOR CMZ59398
FAIRCHILD SMBJ170A
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DESCRIPTION
COMMENTS
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3798f
20 Linear Technology Corporation
LT 0212 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2012