® OPA646 OPA 646 Low Power, Wide Bandwidth OPERATIONAL AMPLIFIER FEATURES DESCRIPTION ● LOW POWER: 55mW ● UNITY-GAIN BANDWIDTH: 650MHz ● UNITY-GAIN STABLE The OPA646 is a low power, wideband voltage feedback operational amplifier. It features a high bandwidth of 650MHz as well as a 12-bit settling time of only 15ns. Its low input bias current and wide bandwidth allows it to be used for high speed integrator and active filter designs. Its low distortion gives exceptional performance for telecommunications, medical imaging and video applications. ● FAST 12-BIT SETTLING: 15ns (0.01%) ● LOW INPUT BIAS CURRENT: 2µA ● LOW HARMONICS: –82dBc at 5MHz ● LOW DIFFERENTIAL GAIN/PHASE ERRORS: 0.025%/0.08° APPLICATIONS ● TELECOMMUNICATIONS ● MEDICAL IMAGING ● CCD IMAGING ● PORTABLE EQUIPMENT ● ACTIVE FILTERS The OPA646 is internally compensated for unity-gain stability. This amplifier has a fully symmetrical differential input due to its “classical” operational amplifier circuit architecture. Its unusual combination of speed, accuracy and low power make it an ideal choice for many portable, multichannel and other high speed applications where power is at a premium. ● VIDEO AMPLIFICATION ● ADC/DAC GAIN AMPLIFIER ● HIGH SPEED INTEGRATORS +V S 7, 8 Non-Inverting Input 3 Inverting Input 2 Output Stage Current Mirror 6 VOUT CC 4, 5 –V S International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 • Twx: 910-952-1111 Internet: http://www.burr-brown.com/ • FAXLine: (800) 548-6133 (US/Canada Only) • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132 ® © 1993 Burr-Brown Corporation 1 PDS-1192C OPA646 Printed in U.S.A. March, 1998 SPECIFICATIONS ELECTRICAL At TA = +25°C, VS = ±5V, RL = 100Ω, CL = 2pF, RFB = 402Ω and all four power supply pins are used, unless otherwise noted. RFB = 25Ω for a gain of +1. OPA646U PARAMETER CONDITIONS OFFSET VOLTAGE Input Offset Voltage Average Drift Power Supply Rejection (+VS) (–VS) INPUT BIAS CURRENT Input Bias Current Over Specified Temperature Input Offset Current Over Specified Temperature TYP MAX ±8 50 45 ±3 ±20 70 55 2 3 0.4 0.9 5 7 1.5 3.0 VS = ±4.5 to ±5.5V VCM = 0V VCM = 0V NOISE Input Voltage Noise Noise Density: f = 100Hz f = 10kHz f = 1MHz f = 1MHz to 100MHz Voltage Noise, BW = 100Hz to 100MHz Input Bias Current Noise Current Noise Density, f = 0.1Hz to 20kHz Noise Figure (NF) RS = 10kΩ RS = 50Ω INPUT VOLTAGE RANGE Common-Mode Input Range Over Specified Temperature Common-Mode Rejection VCM = ±0.5V ±2.5 ±2.5 60 INPUT IMPEDANCE Differential Common-Mode OPEN-LOOP GAIN Open-Loop Voltage Gain Over Specified Temperature FREQUENCY RESPONSE Closed-Loop Bandwidth Slew Rate(1) At Minimum Specified Temperature Rise Time Fall Time Settling Time: 0.01% 0.1% 1% Over-Voltage Recovery(2) Spurious Free Dynamic Range Differential Gain Error at 3.58MHz Differential Phase Error at 3.58MHz Gain Flatness to 0.1dB OUTPUT Voltage Output Over Specified Temperature Voltage Output Over Specified Temperature Voltage Output Over Specified Temperature Current Output, +25°C to max Temp Over Specified Temperature Short Circuit Current Output Resistance POWER SUPPLY Specified Operating Voltage Operating Voltage Range Quiescent Current Over Specified Temperature TEMPERATURE RANGE Specification: U, UB Thermal Resistance U, UB 8-Pin SO-8 OPA646UB MIN MIN TYP MAX UNITS ±2.5 60 48 ±1 ±12 ✻ ✻ mV µV/°C dB dB ✻ ✻ ✻ ✻ 3.5 ✻ ✻ ✻ µA µA µA µA 23.2 7.5 7.1 7.2 141 ✻ ✻ ✻ ✻ ✻ nV/√Hz nV/√Hz nV/√Hz nV/√Hz µVrms 1.1 ✻ pA/√Hz 3.0 19.1 ✻ ✻ dB dB ✻ ✻ 90 V V dB ✻ ✻ kΩ || pF MΩ || pF 55 53 dB dB 650 160 45 22 180 155 5.3 5.9 15 11.5 6 65 82 ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ MHz MHz MHz MHz V/µs V/µs ns ns ns ns ns ns dBc 0.025 0.08 100 ✻ ✻ ✻ % degrees MHz ±3.0 ±3.0 80 ✻ ✻ 75 15 || 1 1.6 || 1 VO = ±2V, RL = 100Ω 45 43 G = +1V/V G = +2V/V G = +5V/V G = +10V/V G = +1, 2V Step 1V Step 1V Step G = +1, 2V Step G = +1, 2V Step G = +1, 2V Step G = +1, f = 5.0MHz VO = 2Vp-p, RL = 402Ω G = +2V/V, VO = 0 to 1.4V, RL = 150Ω G = +2V/V, VO = 0 to 1.4V, RL = 150Ω No Load RL = 250Ω RL = 100Ω Ambient θJA, Junction to Ambient 47 45 ±2.5 ±2.75 ✻ ✻ ±2.5 ±2.7 ✻ ✻ V V V ±2.0 ±40 ±30 ±2.5 ±52 ±48 60 0.2 ✻ ✻ ✻ ✻ ✻ ✻ ✻ ✻ V mA mA mA Ω 1MHz, G = +1V/V TMIN to TMAX TMIN to TMAX 51 49 ±4.5 ±5 ±5.25 ±6.5 –40 125 ✻ ±5.5 ±6.5 ±7.5 ✻ +85 ✻ ✻ ✻ ✻ ✻ ✻ ✻ V V mA mA ✻ °C °C/W ✻ Specification same as OPA646U. NOTE: (1) Slew rate is rate of change from 10% to 90% of output voltage step. (2) Recovery time to linear operation from the input overdrive midpoint. ® OPA646 2 PIN CONFIGURATION ABSOLUTE MAXIMUM RATINGS Top View Power Supply .............................................................................. ±5.5VDC Internal Power Dissipation .......................... See Thermal Considerations Differential Input Voltage .................................................................. ±1.2V Input Voltage Range ............................................................................ ±VS Storage Temperature Range: U, UB ............................ –40°C to +125°C Lead Temperature (soldering, 10s) .............................................. +300°C (soldering, SO-8 3s) ....................................... +260°C Junction Temperature (TJ ) ............................................................ +175°C SO-8 NC 1 8 +VS2(1) Inverting Input 2 7 +VS1 Non-Inverting Input 3 6 Output –VS1 4 5 –VS2(1) ELECTROSTATIC DISCHARGE SENSITIVITY NOTE: (1) Making use of all four power supply pins is highly recommended, although not required. Using these four pins, instead of just pins 4 and 7, will lower the effective pin impedance and substantially lower distortion. This integrated circuit can be damaged by ESD. Burr-Brown recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. PACKAGE /ORDERING INFORMATION PRODUCT OPA646U, UB PACKAGE PACKAGE DRAWING NUMBER(1) SO-8 Surface Mount 182 ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. NOTE: (1) For detailed drawing and dimension table, please see end of data sheet, or Appendix C of Burr-Brown IC Data Book. (2) The “B” grade of the SO-8 will be marked with a “B” by pin 8. The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems. ® 3 OPA646 TYPICAL PERFORMANCE CURVES At TA = +25°C, VS = ±5V, RL = 100Ω, CL = 2pF, RFB = 402Ω and all four power supply pins are used, unless otherwise noted. RFB = 25Ω for a gain of +1. COMMON-MODE REJECTION vs INPUT COMMON-MODE VOLTAGE AOL , PSR, AND CMR vs TEMPERATURE 110 Common-Mode Rejection (dB) 90 CMR AOL , PSR, CMR (dB) 80 PSR+ 70 60 AOL 50 –75 PSR– –50 –25 0 +25 +50 +75 +100 100 90 80 70 –5 +125 –4 –3 INPUT BIAS CURRENT vs TEMPERATURE Supply Current (±mA) Input Bias Current (µA) +1 +2 +3 +4 +5 –50 –25 0 +25 +50 +75 +100 7 6 5 4 –75 +125 –50 Ambient Temperature (°C) –25 0 +25 +50 +75 +100 +125 Ambient Temperature (°C) VOLTAGE NOISE vs FREQUENCY OUTPUT CURRENT vs TEMPERATURE 100 Voltage Noise (nV/√Hz) 70 Output Current (±mA) 0 8 2 I O+ 60 I O– 50 40 –60 –1 SUPPLY CURRENT vs TEMPERATURE 4 0 –75 –2 Common-Mode Voltage (V) Temperature (°C) 80 60 40 20 0 –40 –20 0 20 40 60 80 10 100 120 140 ® OPA646 100 1k 10k 100k Frequency (Hz) Ambient Temperature (°C) 4 1M 10M TYPICAL PERFORMANCE CURVES (CONT) At TA = +25°C, VS = ±5V, RL = 100Ω, CL = 2pF, RFB = 402Ω and all four power supply pins are used, unless otherwise noted. RFB = 25Ω for a gain of +1. SMALL SIGNAL TRANSIENT RESPONSE (G = +1, RL = 100Ω) RECOMMENDED ISOLATION RESISTANCE vs CAPACITIVE LOAD 200 25 120 Voltage Output (mV) Isolation Resistance (Ω) 160 20 15 10 80 40 0 –40 –80 –120 5 –160 –200 0 50 0 100 Time (5ns/div) 150 Capacitive Load (pF) LARGE SIGNAL TRANSIENT RESPONSE (G = +1, RL = 100Ω) G = +1 CLOSED-LOOP SMALL SIGNAL BANDWIDTH 2.0 9 1.6 6 Gain 3 0 0.4 0 –0.4 –3 –0.8 –9 –1.2 –12 –1.6 –15 –2.0 0 –6 –45 Closed-Loop Phase –90 Bandwidth = 625MHz –18 Time (5ns/div) 10M 1M 100M Phase Shift (°) 0.8 Gain (dB) Voltage Output (V) 1.2 –135 –180 –225 1GHz Frequency (Hz) G = –1V/V CLOSED-LOOP SMALL SIGNAL BANDWIDTH G = +2 CLOSED-LOOP SMALL SIGNAL BANDWIDTH 15 12 –180 –6 –225 Closed-Loop Phase –9 –270 Gain (dB) Gain (dB) –3 9 Phase Shift (°) Bandwidth = 165MHz 0 Gain Bandwidth = 185MHz 6 0 3 0 –12 –315 –15 –360 –45 Closed-Loop Phase –90 –3 –135 –6 1M 10M 100M 1GHz 1M Frequency (Hz) Phase Shift (°) Gain 10M 100M 1GHz Frequency (Hz) ® 5 OPA646 TYPICAL PERFORMANCE CURVES (CONT) At TA = +25°C, VS = ±5V, RL = 100Ω, CL = 2pF, RFB = 402Ω and all four power supply pins are used, unless otherwise noted. RFB = 25Ω for a gain of +1. G = +5 CLOSED-LOOP SMALL SIGNAL BANDWIDTH 23 Gain (dB) 17 Gain Bandwidth = 625MHz 14 0 11 8 –45 Closed-Loop Phase –90 100M 5 10M 1M 3fO –80 2fO HARMONIC DISTORTION vs FREQUENCY (G = +2, VO = 2Vp-p, RL = 100Ω) Harmonic Distortion (dBc) –80 2fO 1M 10M –60 3fO –80 2fO –100 100k 100M 10M 100M Frequency (Hz) HARMONIC DISTORTION vs FREQUENCY (G = +5, VO = 2Vp-p, RL = 100Ω) HARMONIC DISTORTION vs TEMPERATURE (G = +1, VO = 2Vp-p, RL = 100Ω, fO = 5MHz) 3fO 2fO –100 100k 1M Frequency (Hz) –60 –80 100M HARMONIC DISTORTION vs FREQUENCY (G = –1, VO = 2Vp-p, RL = 100Ω) 3fO –60 10M Frequency (Hz) –60 –40 1M Frequency (Hz) –40 –100 100k Harmonic Distortion (dBc) –60 –100 100k Harmonic Distortion (dBc) Harmonic Distortion (dBc) –40 Phase Shift (°) 20 Harmonic Distortion (dBc) –40 HARMONIC DISTORTION vs FREQUENCY (G = +1, VO = 2Vp-p, RL = 100Ω) 1M 10M 2fO –80 –90 –75 100M Frequency (Hz) –50 –25 0 25 50 Temperature (°C) ® OPA646 3fO –70 6 75 100 125 TYPICAL PERFORMANCE CURVES (CONT) At TA = +25°C, VS = ±5V, RL = 100Ω, CL = 2pF, RFB = 402Ω and all four power supply pins are used, unless otherwise noted. RFB = 25Ω for a gain of +1. 5MHz HARMONIC DISTORTION vs OUTPUT SWING (G = +1, RL = 100Ω) –60 10MHz HARMONIC DISTORTION vs OUTPUT SWING (G = +1, RL = 100Ω) –50 Harmonic Distortion (dBc) Harmonic Distortion (dBc) 3fO 2fO –70 –80 3fO –60 2fO –70 –80 –90 0 1.0 2.0 3.0 0 4.0 1.0 2.0 3.0 4.0 Output Swing (Vp-p) Output Swing (Vp-p) APPLICATIONS INFORMATION DISCUSSION OF PERFORMANCE The OPA646 provides a level of speed and precision not previously attainable in monolithic form. Unlike current feedback amplifiers, the OPA646’s design uses a “classical” operational amplifier architecture and can therefore be used in all traditional operational amplifier applications. While it is true that current feedback amplifiers can provide wider bandwidth at higher gains, they offer some disadvantages. The asymmetrical input characteristics of current feedback amplifiers (i.e., one input is a low impedance) prevents them from being used in a variety of applications. In addition, unbalanced inputs make input bias current errors difficult to correct. Cancelling offset errors (due to input bias currents) through matching of inverting and non-inverting input resistors is impossible because the input bias currents are uncorrelated. Current noise is also asymmetrical and is usually significantly higher on the inverting input. Perhaps most important, settling time to 0.01% is often extremely poor due to internal design tradeoffs. Many current feedback designs exhibit settling times to 0.01% in excess of 10 microseconds even though 0.1% settling times are reasonable. Such amplifiers are completely inadequate for fast settling 12-bit applications. The OPA646’s “classical” operational amplifier architecture employs true differential and fully symmetrical inputs to eliminate these troublesome problems. All traditional circuit configurations and op amp theory apply to the OPA646. amplifiers when they are improperly used. In general, all printed circuit board conductors should be wide to provide low resistance, low impedance signal paths. They should also be as short as possible. The entire physical circuit should be as small as practical. Stray capacitances should be minimized, especially at high impedance nodes, such as the amplifier’s input terminals. Stray signal coupling from the output or power supplies to the inputs should be minimized. All circuit element leads should be no longer than 1/4 inch (6mm) to minimize lead inductance, and low values of resistance should be used. This will minimize time constants formed with the circuit capacitances and will eliminate stray, parasitic circuits. Grounding is the most important application consideration for the OPA646, as it is with all high-frequency circuits. Oscillations at high frequencies can easily occur if good grounding techniques are not used. A heavy ground plane (2 oz. copper recommended) should connect all unused areas on the component side. Good ground planes can reduce stray signal pickup, provide a low resistance, low inductance common return path for signal and power, and can conduct heat from active circuit package pins into ambient air by convection. Supply bypassing is extremely critical and must always be used, especially when driving high current loads. Both power supply leads should be bypassed to ground as close as possible to the amplifier pins. Tantalum capacitors (2.2µF) with very short leads are recommended. A parallel 0.01µF ceramic must also be added. Surface-mount bypass capacitors will produce excellent results due to their low lead inductance. Additionally, suppression filters can be used to WIRING PRECAUTIONS Maximizing the OPA646’s capability requires some wiring precautions and high-frequency layout techniques. Oscillation, ringing, poor bandwidth and settling, gain peaking, and instability are typical problems plaguing all high-speed ® 7 OPA646 7) Don’t forget that these amplifiers use ±5V supplies. Although they will operate perfectly well with +5V and –5.2V, use of ±15V supplies will destroy the part. 8) Standard commercial test equipment has not been designed to test devices in the OPA646’s speed range. Benchtop op amp testers and ATE systems will require a special test head to successfully test these amplifiers. 9) Terminate transmission line loads. Unterminated lines, such as coaxial cable, can appear to the amplifier to be a capacitive or inductive load. By terminating a transmission line with its characteristic impedance, the amplifier’s load then appears purely resistive. 10) Plug-in prototype boards and wire-wrap boards will not be satisfactory. A clean layout using RF techniques is essential; there are no shortcuts. isolate noisy supply lines. Properly bypassed and modulation-free power supply lines allow full amplifier output and optimum settling time performance. Points to Remember 1) Making use of all four power supply pins will lower the effective power supply impedance seen by the input and output stages. This will improve the AC performance including lower distortion. The lowest distortion is achieved when running separated traces to VS1 and VS2. Power supply bypassing with 0.01µF and 2.2µF surface-mount capacitors on the topside of the PC Board is recommended. It is essential to keep the 0.01µF capacitor very close to the power supply pins. Refer to the DEM-OPA64X Data Sheet for the recommended layout and component placements. 2) Whenever possible, use surface mount. Don’t use pointto-point wiring as the increase in wiring inductance will be detrimental to AC performance. However, if it must be used, very short, direct signal paths are required. The input signal ground return, the load ground return, and the power supply common should all be connected to the same physical point to eliminate ground loops, which can cause unwanted feedback. 3) Surface mount on backside of PC Board. Good component selection is essential. Capacitors used in critical locations should be a low inductance type with a high quality dielectric material. Likewise, diodes used in critical locations should be Schottky barrier types, such as HP50822835 for fast recovery and minimum charge storage. Ordinary diodes will not be suitable in RF circuits. 4) Use a small feedback resistor (usually 25Ω) in unity-gain voltage follower applications for the best performance. For gain configurations, resistors used in feedback networks should have values of a few hundred ohms for best performance. Shunt capacitance problems limit the acceptable resistance range to about 1kΩ on the high end and to a value that is within the amplifier’s output drive limits on the low end. Metal film and carbon resistors will be satisfactory, but wirewound resistors (even “non-inductive” types) are absolutely unacceptable in high-frequency circuits. Feedback resistors should be placed directly between the output and the inverting input on the backside of the PC board. This placement allows for the shortest feedback path and the highest bandwidth. Refer to the demonstration board layout at the end of the data sheet. A longer feedback path than this will decrease the realized bandwidth substantially. 5) Surface-mount components (chip resistors, capacitors, etc.) have low lead inductance and are therefore strongly recommended. Circuits using all surface-mount components with the OPA646U (SO-8 package) will offer the best AC performance. OFFSET VOLTAGE ADJUSTMENT If additional offset adjustment is needed, the circuit in Figure 1 can be used without degrading offset drift with temperature. Avoid external adjustment whenever possible since extraneous noise, such as power supply noise, can be inadvertently coupled into the amplifier’s inverting input terminal. Remember that additional offset errors can be created by the amplifier’s input bias currents. Whenever possible, match the impedance seen by both inputs as is shown with R3. This will reduce input bias current errors to the amplifier’s offset current. +VCC RTrim 20kΩ 47kΩ –VCC OPA646 10µF (1) R1 R3 = R1 || R2 VIN or Ground Output Trim Range ≅ +VCC R2 RTrim to –VCC R2 RTrim NOTE: (1) R3 is optional and can be used to cancel offset errors due to input bias currents. FIGURE 1. Offset Voltage Trim. 6) Avoid overloading the output. Remember that output current must be provided by the amplifier to drive its own feedback network as well as to drive its load. Lowest distortion is achieved with high impedance loads. INPUT PROTECTION Static damage has been well recognized for MOSFET devices, but any semiconductor device deserves protection ® OPA646 R2 8 from this potentially damaging source. The OPA646 incorporates on-chip ESD protection diodes as shown in Figure 2. This eliminates the need for the user to add external protection diodes, which can add capacitance and degrade AC performance. All pins on the OPA646 are internally protected from ESD by means of a pair of back-to-back reverse-biased diodes to either power supply as shown. These diodes will begin to conduct when the input voltage exceeds either power supply by about 0.7V. This situation can occur with loss of the amplifier’s power supplies while a signal source is still present. The diodes can typically withstand a continuous current of 30mA without destruction. To insure long term reliability, however, diode current should be externally limited to 10mA or so whenever possible. The internal power dissipation is given by the equation PD = PDQ + PDL, where PDQ is the quiescent power dissipation and PDL is the power dissipation in the output stage due to the load. (For ±VCC = ±5V, PDQ = 10V x 7.5mA = 75mW, max). For the case where the amplifier is driving a grounded load (RL) with a DC voltage (±VOUT) the maximum value of PDL occurs at ±VOUT = ±VCC/2, and is equal to PDL, max = (±VCC)2/4RL. Note that it is the voltage across the output transistor, and not the load, that determines the power dissipated in the output stage. 100 AV = +1V/V Output Impedance (Ω) +V CC ESD Protection diodes internally connected to all pins. External Pin 10 Internal Circuitry 1 0.1 0.01 –V CC 0.001 100k 10k FIGURE 2. Internal ESD Protection. 1M 10M 100M Frequency (Hz) The OPA646 utilizes a fine geometry high speed process that withstands 500V using the Human Body Model and 100V using the Machine Model. However, static damage can cause subtle changes in amplifier input characteristics without necessarily destroying the device. In precision operational amplifiers, this may cause a noticeable degradation of offset voltage and drift. Therefore, static protection is strongly recommended when handling the OPA646. FIGURE 3. Small-Signal Output Impedance vs Frequency. A short-circuit condition represents the maximum amount of internal power dissipation that can be generated. The variation of output current with temperature is shown in Figure 4. OUTPUT DRIVE CAPABILITY The OPA646 has been optimized to drive 75Ω and 100Ω resistive loads. The device can drive 2Vp-p into a 75Ω load. This high-output drive capability makes the OPA646 an ideal choice for a wide range of RF, IF, and video applications. In many cases, additional buffer amplifiers are unneeded. Many demanding high-speed applications such as ADC/DAC buffers require op amps with low wideband output impedance. For example, low output impedance is essential when driving the signal-dependent capacitances at the inputs of flash A/D converters. As shown in Figure 3, the OPA646 maintains very low closed-loop output impedance over frequency. Closed-loop output impedance increases with frequency since loop gain is decreasing with frequency. Output Current (±mA) 70 I O+ 60 I O– 50 40 –60 –40 –20 0 20 40 60 80 100 120 140 Ambient Temperature (°C) FIGURE 4. Output Current vs Temperature. THERMAL CONSIDERATIONS The OPA646 does not require a heat sink for operation in most environments. At extreme temperatures and under full load conditions a heat sink may be necessary. ® 9 OPA646 CAPACITIVE LOADS The OPA646’s output stage has been optimized to drive low resistive loads. Capacitive loads, however, will decrease the amplifier’s phase margin which may cause high frequency peaking or oscillations. Capacitive loads greater than 10pF should be buffered by connecting a small resistance, usually 5Ω to 25Ω, in series with the output as shown in Figure 5. This is particularly important when driving high capacitance loads such as flash A/D converters. Increasing the gain from +1 will improve the capacitive load drive due to increased phase margin. the phase margin and avoid peaking by keeping the break frequency of this zero sufficiently high. When high closedloop gains are required, a three-resistor attenuator (tee network) is recommended to avoid using large value resistors with large time constants. SETTLING TIME Settling time is defined as the total time required, from the input signal step, for the output to settle to within the specified error band around the final value. This error band is expressed as a percentage of the value of the output transition, a 2V step. Thus, settling time to 0.01% requires an error band of ±200µV centered around the final value of 2V. Settling time, specified in an inverting gain of one, occurs in only 15ns to 0.01% for a 2V step, making the OPA646 one of the fastest settling monolithic amplifiers commercially available. Settling time increases with closed-loop gain and output voltage change as described in the Typical Performance Curves. Preserving settling time requires critical attention to the details as mentioned under “Wiring Precautions.” The amplifier also recovers quickly from input overloads. Overload recovery time to linear operation from a 50% overload is typically only 65ns. In practice, settling time measurements on the OPA646 prove to be very difficult to perform. Accurate measurement is next to impossible in all but the very best equipped labs. Among other things, a fast flat-top generator and high speed oscilloscope are needed. Unfortunately, fast flat-top generators, which settle to 0.01% in sufficient time, are scarce and expensive. Fast oscilloscopes, however, are more commonly available. For best results a sampling oscilloscope is recommended. Sampling scopes typically have bandwidths that are greater than 1GHz and very low capacitance inputs. They also exhibit faster settling times in response to signals that would tend to overload a real-time oscilloscope. (RS typically 5Ω to 25Ω) RS OPA646 RL CL FIGURE 5. Driving Capacitive Loads. In general, capacitive loads should be minimized for optimum high frequency performance. Coax lines can be driven if the cable is properly terminated. The capacitance of coax cable (29pF/foot for RG-58) will not load the amplifier when the coaxial cable or transmission line is terminated in its characteristic impedance. COMPENSATION The OPA646 is internally compensated and is stable in unity gain with a phase margin of approximately 60°. However, the unity gain buffer is the most demanding circuit configuration for loop stability and oscillations are most likely to occur in this gain. If possible, use the device in a noise gain of two or greater to improve phase margin and reduce the susceptibility to oscillation. (Note that, from a stability standpoint, an inverting gain of –1V/V is equivalent to a noise gain of 2.) Gain and phase response for other gains are shown in the Typical Performance Curves. The high-frequency response of the OPA646 in a good layout is very flat with frequency. However, some circuit configurations such as those where large feedback resistances are used, can produce high-frequency gain peaking. This peaking can be minimized by connecting a small capacitor in parallel with the feedback resistor. This capacitor compensates for the closed-loop, high frequency, transfer function zero that results from the time constant formed by the input capacitance of the amplifier (typically 2pF after PC board mounting), and the input and feedback resistors. The selected compensation capacitor may be a trimmer, a fixed capacitor, or a planned PC board capacitance. The capacitance value is strongly dependent on circuit layout and closed-loop gain. Using small resistor values will preserve DIFFERENTIAL GAIN AND PHASE Differential Gain (DG) and Differential Phase (DP) are among the more important specifications for video applications. DG is defined as the percent change in closed-loop gain over a specified change in output voltage level. DP is defined as the change in degrees of the closed-loop phase over the same output voltage change. Both DG and DP are specified at the NTSC sub-carrier frequency of 3.58MHz. DG and DP increase with closed-loop gain and output voltage transition. All measurements were performed using a Tektronix model VM700 Video Measurement Set. ® OPA646 10 SPICE MODELS Computer simulation using SPICE is often useful when analyzing the performance of analog circuits and systems. This is particularly true for Video and RF amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. SPICE models are available for the OPA646. Contact Burr-Brown Applications Department to receive a spice diskette. DISTORTION The OPA646’s harmonic distortion characteristics into a 100Ω load are shown vs frequency and power output in the Typical Performance Curves. Distortion can be significantly improved by increasing the load resistance as illustrated in Figure 6. Remember to include the contribution of the feedback resistance when calculating the effective load resistance seen by the amplifier. DEMONSTRATION BOARDS Demonstration boards to speed prototyping are available. Refer to the DEM-OPA64X Data Sheet for details. –40 Harmonic Distortion (dBc) (G = +1, fO = 5MHz) –50 –60 3fO –70 2fO –80 –90 –100 10 20 50 100 200 500 1K Load Resistance (Ω) FIGURE 6. 5MHz Harmonic Distortion vs Load Resistance with RF = 402Ω. NOISE FIGURE The OPA646 voltage and current noise spectral densities are specified in the Typical Performance Curves. For RF applications, however, Noise Figure (NF) is often the preferred noise specification since it allows system noise performance to be more easily calculated. The OPA646’s Noise Figure vs Source Resistance is shown in Figure 7. 25 NF = 10 LOG 1 + Noise Figure (dB) 20 en2 + (InRS)2 4KTRS 15 10 5 0 10 100 1k 10k 100k Source Resistance (Ω) FIGURE 7. Noise Figure vs Source Resistance. ® 11 OPA646 APPLICATIONS 390Ω 390Ω 75Ω Transmission Line 75Ω V OUT OPA646 Video Input 75Ω 75Ω FIGURE 8. Low Power Video Amplifier. +5V R3 2kΩ (–) D OPA646 (+) R4 2kΩ (1) (1) J1 D J2 S S 2N5911 2 C2 1000pF R5 158Ω 3 7 OPA646 6 V OUT 4 R1 15.8kΩ VIN R2 158Ω (1)R VOUT OPA646 1 2kΩ (1)R 2 2kΩ C1 1000pF –5V fC = 1MHz BW = 20kHz at –3dB Q = 50 NOTE: (1) Select J1, J2 and R1, R2 to set input stage current for optimum performance. FIGURE 9. High-Q 1MHz Bandpass Filter. Input Bias Current: 1pA FIGURE 10. Low Power, Wideband FET Input Op Amp. 50Ω or 75Ω Transmission Line 50Ω or 75Ω OPA646 50Ω or 75Ω Differential Input 50Ω or 75Ω RF 402Ω RG 806Ω Differential Output RF 402Ω 50Ω or 75Ω Transmission Line OPA646 50Ω or 75Ω 50Ω or 75Ω Differential Voltage Gain = 2V/V = 1 + 2RF/RG FIGURE 11. Differential Line Driver for 50Ω or 75Ω Systems. ® OPA646 12 50Ω or 75Ω OPA646 RF 402Ω 402Ω 402Ω RG 806Ω RF 402Ω OPA646 402Ω 402Ω OPA646 Differential Voltage Gain = 2V/V = 1 + 2RF/RG FIGURE 12. Wideband, Fast-Settling Instrumentation Amplifier. C 402Ω 402Ω Differential Input R VIN SingleEnded Output OPA646 402Ω VO OPA646 402Ω R VO = – 1 VIN RC ∫ FIGURE 15. A High Speed Integrator. FIGURE 13. Unity Gain Difference Amplifier. +VS High Speed ADC R +VS VS VOUT = 2 VS 2 ROUT VAC OPA646 RS R Input OPA646 499Ω + AV VAC RL RF 402Ω 499Ω RG 402Ω 499Ω FIGURE 14. ADC Input Buffer Amplifier (G = +2V/V). AV = 1 + RF RG FIGURE 16. Single Supply Operation. ® 13 OPA646