CA3140, CA3140A Data Sheet September 1998 4.5MHz, BiMOS Operational Amplifier with MOSFET Input/Bipolar Output The CA3140A and CA3140 are integrated circuit operational amplifiers that combine the advantages of high voltage PMOS transistors with high voltage bipolar transistors on a single monolithic chip. The CA3140A and CA3140 BiMOS operational amplifiers feature gate protected MOSFET (PMOS) transistors in the input circuit to provide very high input impedance, very low input current, and high speed performance. The CA3140A and CA3140 operate at supply voltage from 4V to 36V (either single or dual supply). These operational amplifiers are internally phase compensated to achieve stable operation in unity gain follower operation, and additionally, have access terminal for a supplementary external capacitor if additional frequency roll-off is desired. Terminals are also provided for use in applications requiring input offset voltage nulling. The use of PMOS field effect transistors in the input stage results in common mode input voltage capability down to 0.5V below the negative supply terminal, an important attribute for single supply applications. The output stage uses bipolar transistors and includes built-in protection against damage from load terminal short circuiting to either supply rail or to ground. The CA3140 Series has the same 8-lead pinout used for the “741” and other industry standard op amps. The CA3140A and CA3140 are intended for operation at supply voltages up to 36V (±18V). Ordering Information PART NUMBER (BRAND) TEMP. RANGE (oC) 957.4 Features • MOSFET Input Stage - Very High Input Impedance (ZIN) -1.5TΩ (Typ) - Very Low Input Current (Il) -10pA (Typ) at ±15V - Wide Common Mode Input Voltage Range (VlCR) - Can be Swung 0.5V Below Negative Supply Voltage Rail - Output Swing Complements Input Common Mode Range • Directly Replaces Industry Type 741 in Most Applications Applications • Ground-Referenced Single Supply Amplifiers in Automobile and Portable Instrumentation • Sample and Hold Amplifiers • Long Duration Timers/Multivibrators (µseconds-Minutes-Hours) • Photocurrent Instrumentation • Peak Detectors • Active Filters • Comparators • Interface in 5V TTL Systems and Other Low Supply Voltage Systems • All Standard Operational Amplifier Applications • Function Generators • Tone Controls • Power Supplies • Portable Instruments • Intrusion Alarm Systems Pinouts PACKAGE PKG. NO. CA3140AE -55 to 125 8 Ld PDIP E8.3 CA3140AM (3140A) -55 to 125 8 Ld SOIC M8.15 CA3140AS -55 to 125 8 Pin Metal Can T8.C CA3140AT -55 to 125 8 Pin Metal Can T8.C CA3140E -55 to 125 8 Ld PDIP E8.3 CA3140M (3140) -55 to 125 8 Ld SOIC M8.15 CA3140M96 (3140) -55 to 125 8 Ld SOIC Tape and Reel CA3140T -55 to 125 8 Pin Metal Can 1 File Number CA3140 (METAL CAN) TOP VIEW TAB STROBE 8 OFFSET NULL 1 + INV. 2 INPUT NON-INV. INPUT 7 3 V+ 6 OUTPUT 5 OFFSET NULL 4 V- AND CASE CA3140 (PDIP, SOIC) TOP VIEW T8.C OFFSET NULL 1 INV. INPUT 2 NON-INV. INPUT 3 V- 4 + 8 STROBE 7 V+ 6 OUTPUT 5 OFFSET NULL CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Copyright © Intersil Corporation 1999 CA3140, CA3140A Absolute Maximum Ratings Thermal Information DC Supply Voltage (Between V+ and V- Terminals) . . . . . . . . . 36V Differential Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . 8V DC Input Voltage . . . . . . . . . . . . . . . . . . . . . . (V+ +8V) To (V- -0.5V) Input Terminal Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1mA Output Short Circuit Duration (Note 2). . . . . . . . . . . . . . . . Indefinite Thermal Resistance (Typical, Note 1) θJA (oC/W) θJC (oC/W) PDIP Package . . . . . . . . . . . . . . . . . . . 100 N/A SOIC Package . . . . . . . . . . . . . . . . . . . 160 N/A Metal Can Package . . . . . . . . . . . . . . . 170 85 Maximum Junction Temperature (Metal Can Package). . . . . . . 175oC Maximum Junction Temperature (Plastic Package) . . . . . . . 150oC Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC (SOIC - Lead Tips Only) Operating Conditions Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . -55oC to 125oC CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTES: 1. θJA is measured with the component mounted on an evaluation PC board in free air. 2. Short circuit may be applied to ground or to either supply. VSUPPLY = ±15V, TA = 25oC Electrical Specifications TYPICAL VALUES PARAMETER SYMBOL Input Offset Voltage Adjustment Resistor TEST CONDITIONS Typical Value of Resistor Between Terminals 4 and 5 or 4 and 1 to Adjust Max VIO CA3140 CA3140A UNITS 4.7 18 kΩ Input Resistance RI 1.5 1.5 TΩ Input Capacitance CI 4 4 pF Output Resistance RO 60 60 Ω Equivalent Wideband Input Noise Voltage (See Figure 27) eN BW = 140kHz, RS = 1MΩ 48 48 µV Equivalent Input Noise Voltage (See Figure 35) eN RS = 100Ω f = 1kHz 40 40 nV/√Hz f = 10kHz 12 12 nV/√Hz IOM+ Source 40 40 mA IOM- Sink 18 18 mA Short Circuit Current to Opposite Supply Gain-Bandwidth Product, (See Figures 6, 30) fT 4.5 4.5 MHz Slew Rate, (See Figure 31) SR 9 9 V/µs 220 220 µA Rise Time 0.08 0.08 µs Overshoot 10 10 % To 1mV 4.5 4.5 µs To 10mV 1.4 1.4 µs Sink Current From Terminal 8 To Terminal 4 to Swing Output Low Transient Response (See Figure 28) tr OS Settling Time at 10VP-P, (See Figure 5) tS RL = 2kΩ CL = 100pF RL = 2kΩ CL = 100pF Voltage Follower For Equipment Design, at VSUPPLY = ±15V, TA = 25oC, Unless Otherwise Specified Electrical Specifications CA3140 PARAMETER CA3140A SYMBOL MIN TYP MAX MIN TYP MAX UNITS Input Offset Voltage |VIO| - 5 15 - 2 5 mV Input Offset Current |IIO| - 0.5 30 - 0.5 20 pA II - 10 50 - 10 40 pA AOL 20 100 - 20 100 - kV/V 86 100 - 86 100 - dB Input Current Large Signal Voltage Gain (Note 3) (See Figures 6, 29) 2 CA3140, CA3140A For Equipment Design, at VSUPPLY = ±15V, TA = 25oC, Unless Otherwise Specified (Continued) Electrical Specifications CA3140 PARAMETER Common Mode Rejection Ratio (See Figure 34) CA3140A SYMBOL MIN TYP MAX MIN TYP MAX UNITS CMRR - 32 320 - 32 320 µV/V 70 90 - 70 90 - dB Common Mode Input Voltage Range (See Figure 8) VICR -15 -15.5 to +12.5 11 -15 -15.5 to +12.5 12 V Power-Supply Rejection Ratio, ∆VIO/∆VS (See Figure 36) PSRR - 100 150 - 100 150 µV/V 76 80 - 76 80 - dB Max Output Voltage (Note 4) (See Figures 2, 8) VOM+ +12 13 - +12 13 - V VOM- -14 -14.4 - -14 -14.4 - V Supply Current (See Figure 32) I+ - 4 6 - 4 6 mA Device Dissipation PD - 120 180 - 120 180 mW ∆VIO/∆T - 8 - - 6 - µV/oC Input Offset Voltage Temperature Drift NOTES: 3. At VO = 26VP-P , +12V, -14V and RL = 2kΩ. 4. At RL = 2kΩ. For Design Guidance At V+ = 5V, V- = 0V, TA = 25oC Electrical Specifications TYPICAL VALUES PARAMETER SYMBOL CA3140 CA3140A UNITS Input Offset Voltage |VIO| 5 2 mV Input Offset Current |IIO| 0.1 0.1 pA Input Current II 2 2 pA Input Resistance RI 1 1 TΩ AOL 100 100 kV/V 100 100 dB 32 32 µV/V 90 90 dB -0.5 -0.5 V 2.6 2.6 V PSRR ∆VIO/∆VS 100 100 µV/V 80 80 dB VOM+ 3 3 V VOM- 0.13 0.13 V Source IOM+ 10 10 mA Sink I OM- 1 1 mA Slew Rate (See Figure 31) SR 7 7 V/µs Gain-Bandwidth Product (See Figure 30) fT 3.7 3.7 MHz Supply Current (See Figure 32) I+ 1.6 1.6 mA Device Dissipation PD 8 8 mW 200 200 µA Large Signal Voltage Gain (See Figures 6, 29) Common Mode Rejection Ratio CMRR Common Mode Input Voltage Range (See Figure 8) VICR Power Supply Rejection Ratio Maximum Output Voltage (See Figures 2, 8) Maximum Output Current: Sink Current from Terminal 8 to Terminal 4 to Swing Output Low 3 CA3140, CA3140A Block Diagram 2mA 4mA 7 V+ BIAS CIRCUIT CURRENT SOURCES AND REGULATOR 200µA + 1.6mA 3 - 2µA A≈ 10,000 A ≈ 10 INPUT 200µA 2mA A≈1 6 OUTPUT 2 C1 12pF 5 1 OFFSET NULL 4 VSTROBE 8 Schematic Diagram BIAS CIRCUIT INPUT STAGE SECOND STAGE OUTPUT STAGE DYNAMIC CURRENT SINK 7 V+ D1 D7 Q1 Q3 Q2 D8 R10 1K Q4 Q5 Q6 Q20 R9 50Ω Q19 R11 20Ω Q7 R12 12K R14 20K Q21 Q17 R1 8K R13 5K R8 Q8 1K Q 18 6 OUTPUT D2 D3 D4 D5 INVERTING INPUT 2 NON-INVERTING INPUT 3 - Q9 + Q10 C1 R3 500Ω R2 500Ω 12pF Q14 Q11 R4 500Ω Q12 1 OFFSET NULL NOTE: All resistance values are in ohms. 4 Q16 D6 R6 50Ω R5 500Ω 5 Q15 Q13 R7 30Ω 8 4 STROBE V- CA3140, CA3140A Application Information Circuit Description As shown in the block diagram, the input terminals may be operated down to 0.5V below the negative supply rail. Two class A amplifier stages provide the voltage gain, and a unique class AB amplifier stage provides the current gain necessary to drive low-impedance loads. A biasing circuit provides control of cascoded constant current flow circuits in the first and second stages. The CA3140 includes an on chip phase compensating capacitor that is sufficient for the unity gain voltage follower configuration. Input Stage The schematic diagram consists of a differential input stage using PMOS field-effect transistors (Q9, Q10) working into a mirror pair of bipolar transistors (Q11, Q12) functioning as load resistors together with resistors R2 through R5. The mirror pair transistors also function as a differential-to-single-ended converter to provide base current drive to the second stage bipolar transistor (Q13). Offset nulling, when desired, can be effected with a 10kΩ potentiometer connected across Terminals 1 and 5 and with its slider arm connected to Terminal 4. Cascode-connected bipolar transistors Q2, Q5 are the constant current source for the input stage. The base biasing circuit for the constant current source is described subsequently. The small diodes D3, D4, D5 provide gate oxide protection against high voltage transients, e.g., static electricity. Second Stage Most of the voltage gain in the CA3140 is provided by the second amplifier stage, consisting of bipolar transistor Q13 and its cascode connected load resistance provided by bipolar transistors Q3, Q4. On-chip phase compensation, sufficient for a majority of the applications is provided by C1. Additional Miller-Effect compensation (roll off) can be accomplished, when desired, by simply connecting a small capacitor between Terminals 1 and 8. Terminal 8 is also used to strobe the output stage into quiescence. When terminal 8 is tied to the negative supply rail (Terminal 4) by mechanical or electrical means, the output Terminal 6 swings low, i.e., approximately to Terminal 4 potential. Output Stage The CA3140 Series circuits employ a broad band output stage that can sink loads to the negative supply to complement the capability of the PMOS input stage when operating near the negative rail. Quiescent current in the emitter-follower cascade circuit (Q17, Q18) is established by transistors (Q14, Q15) whose base currents are “mirrored” to current flowing through diode D2 in the bias circuit section. When the CA3140 is operating such that output Terminal 6 is sourcing current, transistor Q18 functions as an emitter-follower to source current from the V+ bus (Terminal 7), via D7, R9, and R11. Under these conditions, the collector potential of Q13 is sufficiently high to permit the necessary flow of base current to emitter follower Q17 which, in turn, drives Q18. 5 When the CA3140 is operating such that output Terminal 6 is sinking current to the V- bus, transistor Q16 is the current sinking element. Transistor Q16 is mirror connected to D6, R7, with current fed by way of Q21, R12, and Q20. Transistor Q20, in turn, is biased by current flow through R13, zener D8, and R14. The dynamic current sink is controlled by voltage level sensing. For purposes of explanation, it is assumed that output Terminal 6 is quiescently established at the potential midpoint between the V+ and V- supply rails. When output current sinking mode operation is required, the collector potential of transistor Q13 is driven below its quiescent level, thereby causing Q17, Q18 to decrease the output voltage at Terminal 6. Thus, the gate terminal of PMOS transistor Q21 is displaced toward the V- bus, thereby reducing the channel resistance of Q21. As a consequence, there is an incremental increase in current flow through Q20, R12, Q21, D6, R7, and the base of Q16. As a result, Q16 sinks current from Terminal 6 in direct response to the incremental change in output voltage caused by Q18. This sink current flows regardless of load; any excess current is internally supplied by the emitter-follower Q18. Short circuit protection of the output circuit is provided by Q19, which is driven into conduction by the high voltage drop developed across R11 under output short circuit conditions. Under these conditions, the collector of Q19 diverts current from Q4 so as to reduce the base current drive from Q17, thereby limiting current flow in Q18 to the short circuited load terminal. Bias Circuit Quiescent current in all stages (except the dynamic current sink) of the CA3140 is dependent upon bias current flow in R1. The function of the bias circuit is to establish and maintain constant current flow through D1, Q6, Q8 and D2. D1 is a diode connected transistor mirror connected in parallel with the base emitter junctions of Q1, Q2, and Q3. D1 may be considered as a current sampling diode that senses the emitter current of Q6 and automatically adjusts the base current of Q6 (via Q1) to maintain a constant current through Q6, Q8, D2. The base currents in Q2, Q3 are also determined by constant current flow D1. Furthermore, current in diode connected transistor Q2 establishes the currents in transistors Q14 and Q15. Typical Applications Wide dynamic range of input and output characteristics with the most desirable high input impedance characteristics is achieved in the CA3140 by the use of an unique design based upon the PMOS Bipolar process. Input common mode voltage range and output swing capabilities are complementary, allowing operation with the single supply down to 4V. The wide dynamic range of these parameters also means that this device is suitable for many single supply applications, such as, for example, where one input is driven below the potential of Terminal 4 and the phase sense of the output signal must be maintained – a most important consideration in comparator applications. CA3140, CA3140A Output Circuit Considerations level shifting circuitry usually associated with the 741 series of operational amplifiers. Excellent interfacing with TTL circuitry is easily achieved with a single 6.2V zener diode connected to Terminal 8 as shown in Figure 1. This connection assures that the maximum output signal swing will not go more positive than the zener voltage minus two base-to-emitter voltage drops within the CA3140. These voltages are independent of the operating supply voltage. Figure 4 shows some typical configurations. Note that a series resistor, RL, is used in both cases to limit the drive available to the driven device. Moreover, it is recommended that a series diode and shunt diode be used at the thyristor input to prevent large negative transient surges that can appear at the gate of thyristors, from damaging the integrated circuit. V+ 5V TO 36V 8 2 Offset Voltage Nulling LOGIC SUPPLY 5V 7 6.2V 6 CA3140 TYPICAL TTL GATE ≈ 5V 3 The input offset voltage can be nulled by connecting a 10kΩ potentiometer between Terminals 1 and 5 and returning its wiper arm to terminal 4, see Figure 3A. This technique, however, gives more adjustment range than required and therefore, a considerable portion of the potentiometer rotation is not fully utilized. Typical values of series resistors (R) that may be placed at either end of the potentiometer, see Figure 3B, to optimize its utilization range are given in the Electrical Specifications table. 4 OUTPUT STAGE TRANSISTOR (Q15, Q16) SATURATION VOLTAGE (mV) FIGURE 1. ZENER CLAMPING DIODE CONNECTED TO TERMINALS 8 AND 4 TO LIMIT CA3140 OUTPUT SWING TO TTL LEVELS 1000 100 An alternate system is shown in Figure 3C. This circuit uses only one additional resistor of approximately the value shown in the table. For potentiometers, in which the resistance does not drop to 0Ω at either end of rotation, a value of resistance 10% lower than the values shown in the table should be used. SUPPLY VOLTAGE (V-) = 0V TA = 25oC SUPPLY VOLTAGE (V+) = +5V +15V +30V Low Voltage Operation Operation at total supply voltages as low as 4V is possible with the CA3140. A current regulator based upon the PMOS threshold voltage maintains reasonable constant operating current and hence consistent performance down to these lower voltages. 10 1 0.01 0.1 1.0 LOAD (SINKING) CURRENT (mA) 10 The low voltage limitation occurs when the upper extreme of the input common mode voltage range extends down to the voltage at Terminal 4. This limit is reached at a total supply voltage just below 4V. The output voltage range also begins to extend down to the negative supply rail, but is slightly higher than that of the input. Figure 8 shows these characteristics and shows that with 2V dual supplies, the lower extreme of the input common mode voltage range is below ground potential. FIGURE 2. VOLTAGE ACROSS OUTPUT TRANSISTORS (Q15 AND Q16) vs LOAD CURRENT Figure 2 shows output current sinking capabilities of the CA3140 at various supply voltages. Output voltage swing to the negative supply rail permits this device to operate both power transistors and thyristors directly without the need for V+ V+ 2 2 V+ 7 2 7 6 CA3140 CA3140 6 CA3140 6 3 3 7 4 4 3 5 5 R 1 R 1 10kΩ 10kΩ 4 5 1 10kΩ R V- V- FIGURE 3B. IMPROVED RESOLUTION FIGURE 3A. BASIC V- FIGURE 3C. SIMPLER IMPROVED RESOLUTION FIGURE 3. THREE OFFSET VOLTAGE NULLING METHODS 6 CA3140, CA3140A RS V+ LOAD 30V NO LOAD 2 MT2 7 120VAC +HV 7 LOAD 6 CA3140 RL 2 3 CA3140 4 6 MT1 RL 3 4 FIGURE 4. METHODS OF UTILIZING THE VCE(SAT) SINKING CURRENT CAPABILITY OF THE CA3140 SERIES FOLLOWER +15V 7 0.1µF 3 SIMULATED LOAD 10kΩ 6 CA3140 2kΩ 100pF 2 4 0.1µF -15V 2kΩ LOAD RESISTANCE (RL) = 2kΩ LOAD CAPACITANCE (CL) = 100pF 0.05µF SUPPLY VOLTAGE: VS = ±15V TA = 25oC 10 1mV INVERTING 5kΩ 1mV 8 INPUT VOLTAGE (V) 6 10mV +15V 10mV 7 4 2 FOLLOWER INVERTING CA3140 200Ω -2 6 100pF 3 2kΩ 4 -4 0.1µF 4.99kΩ -6 -10 0.1 SIMULATED LOAD 5kΩ 0 -8 0.1µF 2 1mV 10mV 10mV 1.0 SETTLING TIME (µs) 5.11kΩ -15V 1mV SETTLING POINT 10 D2 D1 1N914 FIGURE 5A. WAVEFORM 1N914 FIGURE 5B. TEST CIRCUITS FIGURE 5. SETTLING TIME vs INPUT VOLTAGE Bandwidth and Slew Rate For those cases where bandwidth reduction is desired, for example, broadband noise reduction, an external capacitor connected between Terminals 1 and 8 can reduce the open loop -3dB bandwidth. The slew rate will, however, also be proportionally reduced by using this additional capacitor. Thus, a 20% reduction in bandwidth by this technique will also reduce the slew rate by about 20%. Figure 5 shows the typical settling time required to reach 1mV or 10mV of the final value for various levels of large signal inputs for the voltage follower and inverting unity gain amplifiers. The exceptionally fast settling time characteristics 7 are largely due to the high combination of high gain and wide bandwidth of the CA3140; as shown in Figure 6. Input Circuit Considerations As mentioned previously, the amplifier inputs can be driven below the Terminal 4 potential, but a series current limiting resistor is recommended to limit the maximum input terminal current to less than 1mA to prevent damage to the input protection circuitry. Moreover, some current limiting resistance should be provided between the inverting input and the output when the CA3140 is used as a unity gain voltage follower. This resistance prevents the possibility of extremely large input CA3140, CA3140A The typical input current is on the order of 10pA when the inputs are centered at nominal device dissipation. As the output supplies load current, device dissipation will increase, raising the chip temperature and resulting in increased input current. Figure 7 shows typical input terminal current versus ambient temperature for the CA3140. -75 SUPPLY VOLTAGE: VS = ±15V TA = 25oC 100 φOL RL = 2kΩ, CL = 0pF -90 -105 -120 -135 80 -150 OPEN LOOP PHASE (DEGREES) OPEN LOOP VOLTAGE GAIN (dB) It is well known that MOSFET devices can exhibit slight changes in characteristics (for example, small changes in input offset voltage) due to the application of large 60 RL = 2kΩ, CL = 100pF 40 differential input voltages that are sustained over long periods at elevated temperatures. Both applied voltage and temperature accelerate these changes. The process is reversible and offset voltage shifts of the opposite polarity reverse the offset. Figure 9 shows the typical offset voltage change as a function of various stress voltages at the maximum rating of 125oC (for metal can); at lower temperatures (metal can and plastic), for example, at 85oC, this change in voltage is considerably less. In typical linear applications, where the differential voltage is small and symmetrical, these incremental changes are of about the same magnitude as those encountered in an operational amplifier employing a bipolar transistor input stage. 10K INPUT CURRENT (pA) signal transients from forcing a signal through the input protection network and directly driving the internal constant current source which could result in positive feedback via the output terminal. A 3.9kΩ resistor is sufficient. SUPPLY VOLTAGE: VS = ±15V 1K 100 10 20 0 101 102 103 104 105 106 FREQUENCY (Hz) 107 1 -60 108 RL = ∞ 0 +VICR AT TA = 125oC +VICR AT TA = 25oC +VICR AT TA = -55oC -0.5 -1.0 +VOUT AT TA = 125oC +VOUT AT TA = 25oC +VOUT AT TA = -55oC -1.5 -2.0 -2.5 -3.0 0 5 10 15 SUPPLY VOLTAGE (V+, V-) 20 -20 0 20 40 60 80 TEMPERATURE (oC) 100 120 140 FIGURE 7. INPUT CURRENT vs TEMPERATURE 25 INPUT AND OUTPUT VOLTAGE EXCURSIONS FROM TERMINAL 4 (V-) INPUT AND OUTPUT VOLTAGE EXCURSIONS FROM TERMINAL 7 (V+) FIGURE 6. OPEN LOOP VOLTAGE GAIN AND PHASE vs FREQUENCY -40 1.5 -VICR AT TA = 125oC 1.0 -VICR AT TA = 25oC 0.5 -VOUT FOR TA = -55oC to 125oC 0 -VICR AT TA = -55oC -0.5 -1.0 -1.5 0 5 10 15 SUPPLY VOLTAGE (V+, V-) 20 25 FIGURE 8. OUTPUT VOLTAGE SWING CAPABILITY AND COMMON MODE INPUT VOLTAGE RANGE vs SUPPLY VOLTAGE All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification. Intersil semiconductor products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see web site http://www.intersil.com 8 CA3140, CA3140A OFFSET VOLTAGE SHIFT (mV) 7 TA = 125oC FOR METAL CAN PACKAGES DIFFERENTIAL DC VOLTAGE (ACROSS TERMINALS 2 AND 3) = 2V OUTPUT STAGE TOGGLED 6 5 4 3 2 DIFFERENTIAL DC VOLTAGE (ACROSS TERMINALS 2 AND 3) = 0V OUTPUT VOLTAGE = V+ / 2 1 0 0 500 1000 1500 2000 2500 3000 3500 4000 4500 TIME (HOURS) FIGURE 9. TYPICAL INCREMENTAL OFFSET VOLTAGE SHIFT vs OPERATING LIFE Super Sweep Function Generator A function generator having a wide tuning range is shown in Figure 10. The 1,000,000/1 adjustment range is accomplished by a single variable potentiometer or by an auxiliary sweeping signal. The CA3140 functions as a noninverting readout amplifier of the triangular signal developed across the integrating capacitor network connected to the output of the CA3080A current source. Buffered triangular output signals are then applied to a second CA3080 functioning as a high speed hysteresis switch. Output from the switch is returned directly back to the input of the CA3080A current source, thereby, completing the positive feedback loop The triangular output level is determined by the four 1N914 level limiting diodes of the second CA3080 and the resistor divider network connected to Terminal No. 2 (input) of the CA3080. These diodes establish the input trip level to this switching stage and, therefore, indirectly determine the amplitude of the output triangle. Compensation for propagation delays around the entire loop is provided by one adjustment on the input of the CA3080. This adjustment, which provides for a constant generator amplitude output, is most easily made while the generator is sweeping. High frequency ramp linearity is adjusted by the single 7pF to 60pF capacitor in the output of the CA3080A. It must be emphasized that only the CA3080A is characterized for maximum output linearity in the current generator function. Meter Driver and Buffer Amplifier Figure 11 shows the CA3140 connected as a meter driver and buffer amplifier. Low driving impedance is required of the CA3080A current source to assure smooth operation of the Frequency Adjustment Control. This low-driving impedance requirement is easily met by using a CA3140 connected as a voltage follower. Moreover, a meter may be 9 placed across the input to the CA3080A to give a logarithmic analog indication of the function generator’s frequency. Analog frequency readout is readily accomplished by the means described above because the output current of the CA3080A varies approximately one decade for each 60mV change in the applied voltage, VABC (voltage between Terminals 5 and 4 of the CA3080A of the function generator). Therefore, six decades represent 360mV change in VABC. Now, only the reference voltage must be established to set the lower limit on the meter. The three remaining transistors from the CA3086 Array used in the sweep generator are used for this reference voltage. In addition, this reference generator arrangement tends to track ambient temperature variations, and thus compensates for the effects of the normal negative temperature coefficient of the CA3080A VABC terminal voltage. Another output voltage from the reference generator is used to insure temperature tracking of the lower end of the Frequency Adjustment Potentiometer. A large series resistance simulates a current source, assuring similar temperature coefficients at both ends of the Frequency Adjustment Control. To calibrate this circuit, set the Frequency Adjustment Potentiometer at its low end. Then adjust the Minimum Frequency Calibration Control for the lowest frequency. To establish the upper frequency limit, set the Frequency Adjustment Potentiometer to its upper end and then adjust the Maximum Frequency Calibration Control for the maximum frequency. Because there is interaction among these controls, repetition of the adjustment procedure may be necessary. Two adjustments are used for the meter. The meter sensitivity control sets the meter scale width of each decade, while the meter position control adjusts the pointer on the scale with negligible effect on the sensitivity adjustment. Thus, the meter sensitivity adjustment control calibrates the meter so that it deflects 1/6 of full scale for each decade change in frequency. Sine Wave Shaper The circuit shown in Figure 12 uses a CA3140 as a voltage follower in combination with diodes from the CA3019 Array to convert the triangular signal from the function generator to a sine-wave output signal having typically less than 2% THD. The basic zero crossing slope is established by the 10kΩ potentiometer connected between Terminals 2 and 6 of the CA3140 and the 9.1kΩ resistor and 10kΩ potentiometer from Terminal 2 to ground. Two break points are established by diodes D1 through D4. Positive feedback via D5 and D6 establishes the zero slope at the maximum and minimum levels of the sine wave. This technique is necessary because the voltage follower configuration approaches unity gain rather than the zero gain required to shape the sine wave at the two extremes. CA3140, CA3140A CENTERING -15V 10kΩ 7.5kΩ +15V 360Ω 3 7 + 15kΩ - 2 4 5 2MΩ 51 pF 7-60 pF -15V +15V 39kΩ 120Ω -15V + CA3140 3 HIGH FREQ. SHAPE -15V 2kΩ FREQUENCY ADJUSTMENT 10kΩ 0.1 µF 5.1kΩ 62kΩ 10kΩ EXTERNAL OUTPUT EXTERNAL OUTPUT 7 - 2 11kΩ 11kΩ 10kΩ 4 +15V 5 6 - 2 100kΩ FROM BUFFER METER DRIVER (OPTIONAL) 0.1 µF 7 6 CA3080A 360Ω SYMMETRY -15V +15V HIGH FREQUENCY LEVEL 910kΩ 7-60pF 6 CA3080 + 4 3 2.7kΩ -15V 13kΩ TO OUTPUT AMPLIFIER TO SINE WAVE SHAPER 1N914 OUTPUT AMPLIFIER +15V THIS NETWORK IS USED WHEN THE OPTIONAL BUFFER CIRCUIT IS NOT USED FIGURE 10A. CIRCUIT FREQUENCY ADJUSTMENT Top Trace: Output at junction of 2.7Ω and 51Ω resistors; 5V/Div., 500ms/Div. +15V METER DRIVER AND BUFFER AMPLIFIER Center Trace: External output of triangular function generator; 2V/Div., 500ms/Div. Bottom Trace: Output of “Log” generator; 10V/Div., 500ms/Div. POWER SUPPLY ±15V M -15V FUNCTION GENERATOR FIGURE 10B. FIGURE FUNCTION GENERATOR SWEEPING WIDEBAND LINE DRIVER SINE WAVE SHAPER 51Ω SWEEP GENERATOR FINE RATE GATE DC LEVEL SWEEP ADJUST OFF INT. COARSE RATE EXT. EXTERNAL INPUT SWEEP LENGTH 1V/Div., 1s/Div. Three tone test signals, highest frequency ≥0.5MHz. Note the slight asymmetry at the three second/cycle signal. This asymmetry is due to slightly different positive and negative integration from the CA3080A and from the PC board and component leakages at the 100pA level. FIGURE 10C. FUNCTION GENERATOR WITH FIXED FREQUENCIES V- FIGURE 10D. INTERCONNECTIONS FIGURE 10. FUNCTION GENERATOR 10 V- CA3140, CA3140A 500kΩ FREQUENCY ADJUSTMENT 10kΩ FREQUENCY CALIBRATION MAXIMUM 620kΩ 7 51kΩ 3 + 6 CA3140 SWEEP IN 3MΩ - 2 4.7kΩ 4 +15V 2kΩ 12kΩ FREQUENCY 2.4kΩ CALIBRATION MINIMUM 2.5 kΩ 0.1µF 5.1kΩ 510Ω 2kΩ 7 10kΩ D1 D4 9.1kΩ 6 5 9 3.6kΩ 13 8 D6 D3 12 METER POSITION ADJUSTMENT 7 -15V R3 10kΩ EXTERNAL OUTPUT R1 10kΩ 14 10 2 D2 1 3 4 D CA3019 5 DIODE ARRAY 3/ OF CA3086 5 -15V FIGURE 11. METER DRIVER AND BUFFER AMPLIFIER FIGURE 12. SINE WAVE SHAPER 750kΩ “LOG” SAWTOOTH 18MΩ 1N914 100kΩ 100kΩ FINE RATE 1MΩ 22MΩ SAWTOOTH SYMMETRY 1N914 0.47µF 0.047µF 8.2kΩ +15V SAWTOOTH AND RAMP LOW LEVEL SET (-14.5V) COARSE RATE 4700pF 50kΩ 75kΩ 470pF 7 2 - 3 CA3140 + 4 51kΩ SAWTOOTH +15V 0.1 µF “LOG”+15V 3 100kΩ TO OUTPUT 2 AMPLIFIER 30kΩ 0.1 µF 50kΩ LOG RATE ADJUST -15V 43kΩ 10kΩ 10kΩ +15V 36kΩ TRIANGLE 6 7 - CA3140 + 4 6 10kΩ -15V EXTERNAL OUTPUT TO FUNCTION GENERATOR “SWEEP IN” SWEEP WIDTH -15V 3 6 CA3140 2 LOGVIO +15V 7 + 5 1 51kΩ 4 6.8kΩ 91kΩ 10kΩ TRIANGLE 25kΩ 3.9Ω -15V 5 1 4 2 100Ω 390Ω TRANSISTORS FROM CA3086 ARRAY 3 FIGURE 13. SWEEPING GENERATOR 11 TO WIDEBAND OUTPUT AMPLIFIER SUBSTRATE OF CA3019 1MΩ 100 kΩ -15V 6 4 +15V 9 8 - 0.1µF 1kΩ 200µA M METER 510Ω 6 CA3140 620Ω 11 5.6kΩ 7.5kΩ 7 + 3 2 METER SENSITIVITY ADJUSTMENT 0.1µF -15V +15V TO CA3080A OF FUNCTION CA3080A GENERATOR (FIGURE 10) 4 5 SAWTOOTH “LOG” GATE PULSE OUTPUT 430Ω R2 1kΩ CA3140, CA3140A This circuit can be adjusted most easily with a distortion analyzer, but a good first approximation can be made by comparing the output signal with that of a sine wave generator. The initial slope is adjusted with the potentiometer R1, followed by an adjustment of R2. The final slope is established by adjusting R3, thereby adding additional segments that are contributed by these diodes. Because there is some interaction among these controls, repetition of the adjustment procedure may be necessary. REFERENCE VOLTAGE VOLTAGE ADJUSTMENT 3 + 7 CA3140 INPUT 2 - 6 REGULATED OUTPUT 4 Sweeping Generator Figure 13 shows a sweeping generator. Three CA3140s are used in this circuit. One CA3140 is used as an integrator, a second device is used as a hysteresis switch that determines the starting and stopping points of the sweep. A third CA3140 is used as a logarithmic shaping network for the log function. Rates and slopes, as well as sawtooth, triangle, and logarithmic sweeps are generated by this circuit. Wideband Output Amplifier Figure 14 shows a high slew rate, wideband amplifier suitable for use as a 50Ω transmission line driver. This circuit, when used in conjunction with the function generator and sine wave shaper circuits shown in Figures 10 and 12 provides 18VP-P output open circuited, or 9VP-P output when terminated in 50Ω. The slew rate required of this amplifier is 28V/µs (18VP-P x π x 0.5MHz). +15V + SIGNAL LEVEL ADJUSTMENT 2.5kΩ 3 7 + 2 - 4 -15V 200Ω + 2.4pF 2pF 1.8kΩ 2N3053 1N914 2.7Ω 1N914 2.7Ω 51Ω 50µF 25V 2.2 kΩ 2W 2N4037 -15V NOMINAL BANDWIDTH = 10MHz tr = 35ns FIGURE 14. WIDEBAND OUTPUT AMPLIFIER Power Supplies High input impedance, common mode capability down to the negative supply and high output drive current capability are key factors in the design of wide range output voltage supplies that use a single input voltage to provide a regulated output voltage that can be adjusted from essentially 0V to 24V. Unlike many regulator systems using comparators having a bipolar transistor input stage, a high impedance reference voltage divider from a single supply can be used in connection with the CA3140 (see Figure 15). 12 Essentially, the regulators, shown in Figures 16 and 17, are connected as non inverting power operational amplifiers with a gain of 3.2. An 8V reference input yields a maximum output voltage slightly greater than 25V. As a voltage follower, when the reference input goes to 0V the output will be 0V. Because the offset voltage is also multiplied by the 3.2 gain factor, a potentiometer is needed to null the offset voltage. Series pass transistors with high ICBO levels will also prevent the output voltage from reaching zero because there is a finite voltage drop (VCESAT) across the output of the CA3140 (see Figure 2). This saturation voltage level may indeed set the lowest voltage obtainable. The high impedance presented by Terminal 8 is advantageous in effecting current limiting. Thus, only a small signal transistor is required for the current-limit sensing amplifier. Resistive decoupling is provided for this transistor to minimize damage to it or the CA3140 in the event of unusual input or output transients on the supply rail. OUT 8 1 OUTPUT DC LEVEL +15V ADJUSTMENT 3kΩ 2.2 kΩ 6 CA3140 200Ω 50µF 25V FIGURE 15. BASIC SINGLE SUPPLY VOLTAGE REGULATOR SHOWING VOLTAGE FOLLOWER CONFIGURATION Figures 16 and 17, show circuits in which a D2201 high speed diode is used for the current sensor. This diode was chosen for its slightly higher forward voltage drop characteristic, thus giving greater sensitivity. It must be emphasized that heat sinking of this diode is essential to minimize variation of the current trip point due to internal heating of the diode. That is, 1A at 1V forward drop represents one watt which can result in significant regenerative changes in the current trip point as the diode temperature rises. Placing the small signal reference amplifier in the proximity of the current sensing diode also helps minimize the variability in the trip level due to the negative temperature coefficient of the diode. In spite of those limitations, the current limiting point can easily be adjusted over the range from 10mA to 1A with a single adjustment potentiometer. If the temperature stability of the current limiting system is a serious consideration, the more usual current sampling resistor type of circuitry should be employed. A power Darlington transistor (in a metal can with heatsink), is used as the series pass element for the conventional current limiting system, Figure 16, because high power Darlington dissipation will be encountered at low output voltage and high currents. CA3140, CA3140A A small heat sink VERSAWATT transistor is used as the series pass element in the fold back current system, Figure 17, since dissipation levels will only approach 10W. In this system, the D2201 diode is used for current sampling. Foldback is provided by the 3kΩ and 100kΩ divider network connected to the base of the current sensing transistor. Both regulators provide better than 0.02% load regulation. Because there is constant loop gain at all voltage settings, the +30V 3 2N6385 CURRENT POWER DARLINGTON LIMITING ADJUST D2201 2 75Ω 1kΩ OUTPUT 0.1 ⇒ 24V AT 1A 8 +30V 1 2 8 1kΩ 56pF 2.7kΩ 10µF + 5 - 2.2kΩ - 10 11 1 2 9 3 8 7 5 6 4 14 + 2.7kΩ 10µF 4 5µF 50kΩ 5 - 1 100kΩ 82kΩ 3 4 INPUT VOLTAGE ADJUST 100kΩ + - 0.01µF + 2.2kΩ 250µF 12 - 10 11 1 2 9 3 8 7 5 6 4 5µF 50kΩ 14 VOLTAGE ADJUST 100kΩ + - 250µF 12 0.01µF 13 13 CA3086 CA3086 1kΩ 1kΩ 62kΩ 62kΩ HUM AND NOISE OUTPUT <200µVRMS (MEASUREMENT BANDWIDTH ~10MHz) LINE REGULATION 0.1%/V 180kΩ 1kΩ CA3140 6 82kΩ INPUT + 56pF 2 3 1 100kΩ 1kΩ 7 180kΩ 1kΩ CA3140 3kΩ 2N2102 3 2 6 100kΩ 100kΩ 2N2102 7 OUTPUT ⇒ 0V TO 25V 25V AT 1A “FOLDS BACK” TO 40mA “FOLDBACK” CURRENT LIMITER 2N5294 D2201 2 3 1 1kΩ 100Ω Figure 18A shows the turn ON and turn OFF characteristics of both regulators. The slow turn on rise is due to the slow rate of rise of the reference voltage. Figure 18B shows the transient response of the regulator with the switching of a 20Ω load at 20V output. 1kΩ 200Ω 1kΩ 3kΩ 1 regulation also remains constant. Line regulation is 0.1% per volt. Hum and noise voltage is less than 200µV as read with a meter having a 10MHz bandwidth. LOAD REGULATION (NO LOAD TO FULL LOAD) <0.02% FIGURE 16. REGULATED POWER SUPPLY 5V/Div., 1s/Div. FIGURE 18A. SUPPLY TURN-ON AND TURNOFF CHARACTERISTICS HUM AND NOISE OUTPUT <200µVRMS (MEASUREMENT BANDWIDTH ~10MHz) LINE REGULATION 0.1%/V LOAD REGULATION (NO LOAD TO FULL LOAD) <0.02% FIGURE 17. REGULATED POWER SUPPLY WITH “FOLDBACK” CURRENT LIMITING Top Trace: Output Voltage; 200mV/Div., 5µs/Div. Bottom Trace: Collector of load switching transistor, load = 1A; 5V/Div., 5µs/Div. FIGURE 18B. TRANSIENT RESPONSE FIGURE 18. WAVEFORMS OF DYNAMIC CHARACTERISTICS OF POWER SUPPLY CURRENTS SHOWN IN FIGURES 16 AND 17 13 CA3140, CA3140A Bass treble boost and cut are ±15dB at 100Hz and 10kHz, respectively. Full peak-to-peak output is available up to at least 20kHz due to the high slew rate of the CA3140. The amplifier gain is 3dB down from its “flat” position at 70kHz. Tone Control Circuits High slew rate, wide bandwidth, high output voltage capability and high input impedance are all characteristics required of tone control amplifiers. Two tone control circuits that exploit these characteristics of the CA3140 are shown in Figures 19 and 20. Figure 19 shows another tone control circuit with similar boost and cut specifications. The wideband gain of this circuit is equal to the ultimate boost or cut plus one, which in this case is a gain of eleven. For 20dB boost and cut, the input loading of this circuit is essentially equal to the value of the resistance from Terminal No. 3 to ground. A detailed analysis of this circuit is given in “An IC Operational Transconductance Amplifier (OTA) With Power Capability” by L. Kaplan and H. Wittlinger, IEEE Transactions on Broadcast and Television Receivers, Vol. BTR-18, No. 3, August, 1972. The first circuit, shown in Figure 20, is the Baxandall tone control circuit which provides unity gain at midband and uses standard linear potentiometers. The high input impedance of the CA3140 makes possible the use of lowcost, low-value, small size capacitors, as well as reduced load of the driving stage. FOR SINGLE SUPPLY NOTES: +30V 2.2MΩ 5. 20dB Flat Position Gain. 3 7 + CA3140 2 - 0.005µF 5.1 MΩ 6. ±15dB Bass and Treble Boost and Cut at 100Hz and 10kHz, respectively. 0.1µF 6 7. 25VP-P output at 20kHz. 8. -3dB at 24kHz from 1kHz reference. 4 BOOST 2.2MΩ 0.1 µF 0.012µF TREBLE CUT 200kΩ (LINEAR) 0.001µF 18kΩ 100 pF FOR DUAL SUPPLIES +15V 3 7 + CA3140 2 - 0.005µF 100pF 5.1MΩ 4 0.022µF 2µF - + 0.0022µF 0.1µF 6 0.1µF -15V 10kΩ 1MΩ 100kΩ CCW (LOG) BOOST BASS CUT TONE CONTROL NETWORK TONE CONTROL NETWORK FIGURE 19. TONE CONTROL CIRCUIT USING CA3130 SERIES (20dB MIDBAND GAIN) FOR SINGLE SUPPLY BOOST 0.047µF BASS CUT (LINEAR) 240kΩ 5MΩ 240kΩ FOR DUAL SUPPLIES 2.2MΩ 750 pF +32V 750 pF +15V 7 3 2.2MΩ 0.1 µF 2.2 MΩ 2 + CA3140 51kΩ 5MΩ 51kΩ (LINEAR) BOOST TREBLE CUT TONE CONTROL NETWORK 6 0.047µF 4 20pF 0.1 µF TONE CONTROL NETWORK 7 + CA3140 2 - NOTES: 9. ±15dB Bass and Treble Boost and Cut at 100Hz and 10kHz, Respectively. 10. 25VP-P Output at 20kHz. 11. -3dB at 70kHz from 1kHz Reference. 12. 0dB Flat Position Gain. FIGURE 20. BAXANDALL TONE CONTROL CIRCUIT USING CA3140 SERIES 14 3 4 -15V 0.1µF 6 0.1µF CA3140, CA3140A Wien Bridge Oscillator Another application of the CA3140 that makes excellent use of its high input impedance, high slew rate, and high voltage qualities is the Wien Bridge sine wave oscillator. A basic Wien Bridge oscillator is shown in Figure 21. When R1 = R2 = R and C1 = C2 = C, the frequency equation reduces to the familiar f = 1/(2πRC) and the gain required for oscillation, AOSC is equal to 3. Note that if C2 is increased by a factor of four and R2 is reduced by a factor of four, the gain required for oscillation becomes 1.5, thus permitting a potentially higher operating frequency closer to the gain bandwidth product of the CA3140. C2 R2 NOTES: + OUTPUT RF C1 R1 1000pF 3 C1 2 1000 pF R1 C1 R2 A OSC = 1 + ------- + ------C2 R1 FIGURE 21. BASIC WIEN BRIDGE OSCILLATOR CIRCUIT USING AN OPERATIONAL AMPLIFIER Oscillator stabilization takes on many forms. It must be precisely set, otherwise the amplitude will either diminish or reach some form of limiting with high levels of distortion. The element, RS, is commonly replaced with some variable resistance element. Thus, through some control means, the value of RS is adjusted to maintain constant oscillator output. A FET channel resistance, a thermistor, a lamp bulb, or other device whose resistance increases as the output amplitude is increased are a few of the elements often utilized. 7 + CA3140 4 As the frequency is increased, the output amplitude must be reduced to prevent the output signal from becoming slewrate limited. An output frequency of 180kHz will reach a slew rate of approximately 9V/µs when its amplitude is 16VP-P. 15 CA3109 DIODE ARRAY 6 SUBSTRATE OF CA3019 8 9 1 2 6 0.1µF -15V 3 7.5kΩ 5 4 R1 = R2 = R 50Hz, R = 100Hz, R = 1kHz, R = 10kHz, R = 30kHz, R = 3.3MΩ 1.6MΩ 160MΩ 16MΩ 5.1MΩ 3.6kΩ 500Ω FIGURE 22. WIEN BRIDGE OSCILLATOR CIRCUIT USING CA3140 Simple Sample-and-Hold System Figure 23 shows a very simple sample-and-hold system using the CA3140 as the readout amplifier for the storage capacitor. The CA3080A serves as both input buffer amplifier and low feed-through transmission switch (see Note 13). System offset nulling is accomplished with the CA3140 via its offset nulling terminals. A typical simulated load of 2kΩ and 30pF is shown in the schematic. 0 30kΩ SAMPLE STROBE -15 1N914 HOLD +15V 1N914 INPUT +15V 5 2kΩ + 3 - 0.1µF 7 CA3080A 2 Figure 22 shows another means of stabilizing the oscillator with a zener diode shunting the feedback resistor (RF of Figure 21). As the output signal amplitude increases, the zener diode impedance decreases resulting in more feedback with consequent reduction in gain; thus stabilizing the amplitude of the output signal. Furthermore, this combination of a monolithic zener diode and bridge rectifier circuit tends to provide a zero temperature coefficient for this regulating system. Because this bridge rectifier system has no time constant, i.e., thermal time constant for the lamp bulb, and RC time constant for filters often used in detector networks, there is no lower frequency limit. For example, with 1µF polycarbonate capacitors and 22MΩ for the frequency determining network, the operating frequency is 0.007Hz. 0.1µF 7 0.1µF 1 f = ------------------------------------------2π R 1 C 1 R 2 C 2 RF A CL = 1 + -------RS RS R2 C2 OUTPUT 19VP-P TO 22VP-P THD <0.3% +15V 6 4 0.1µF 2kΩ 3.5kΩ 7 3 + CA3140 2 - 6 0.1 µF 4 5 1 100kΩ -15V -15V 200pF 2kΩ C1 200pF 2kΩ 400Ω 0.1µF 30pF SIMULATED LOAD NOT REQUIRED FIGURE 23. SAMPLE AND HOLD CIRCUIT In this circuit, the storage compensation capacitance (C1) is only 200pF. Larger value capacitors provide longer “hold” periods but with slower slew rates. The slew rate is: dv I ------ = ---- = 0.5mA ⁄ 200pF = 2.5V ⁄ µs dt C NOTE: 13. AN6668 “Applications of the CA3080 and CA 3080A High Performance Operational Transconductance Amplifiers”. CA3140, CA3140A Pulse “droop” during the hold interval is 170pA/200pF which is 0.85µV/µs; (i.e., 170pA/200pF). In this case, 170pA represents the typical leakage current of the CA3080A when strobed off. If C1 were increased to 2000pF, the “hold-droop” rate will decrease to 0.085µV/µs, but the slew rate would decrease to 0.25V/µs. The parallel diode network connected between Terminal 3 of the CA3080A and Terminal 6 of the CA3140 prevents large input signal feedthrough across the input terminals of the CA3080A to the 200pF storage capacitor when the CA3080A is strobed off. Figure 24 shows dynamic characteristic waveforms of this sample-and-hold system. Current Amplifier The low input terminal current needed to drive the CA3140 makes it ideal for use in current amplifier applications such as the one shown in Figure 25 (see Note 14). In this circuit, low current is supplied at the input potential as the power supply to load resistor RL. This load current is increased by the multiplication factor R2/R1, when the load current is monitored by the power supply meter M. Thus, if the load current is 100nA, with values shown, the load current presented to the supply will be 100µA; a much easier current to measure in many systems. R1 10kΩ +15V IL x R2 R1 0.1µF 7 + CA3140 3 M - 2 POWER SUPPLY 5 4 1 6 0.1µF R2 10MΩ IL RL 100kΩ Top Trace: Output; 50mV/Div., 200ns/Div. Bottom Trace: Input; 50mV/Div., 200ns/Div. 4.3kΩ -15V FIGURE 25. BASIC CURRENT AMPLIFIER FOR LOW CURRENT MEASUREMENT SYSTEMS Note that the input and output voltages are transferred at the same potential and only the output current is multiplied by the scale factor. Top Trace: Output Signal; 5V/Div, 2µs/Div. Center Trace: Difference of Input and Output Signals through Tektronix Amplifier 7A13; 5mV/Div., 2µs/Div. Bottom Trace: Input Signal; 5V/Div., 2µs/Div. LARGE SIGNAL RESPONSE AND SETTLING TIME SAMPLING RESPONSE Top Trace: Output; 100mV/Div., 500ns/Div. Bottom Trace: Input; 20V/Div., 500ns/Div. FIGURE 24. SAMPLE AND HOLD SYSTEM DYNAMIC CHARACTERISTICS WAVEFORMS 16 The dotted components show a method of decoupling the circuit from the effects of high output load capacitance and the potential oscillation in this situation. Essentially, the necessary high frequency feedback is provided by the capacitor with the dotted series resistor providing load decoupling. Full Wave Rectifier Figure 26 shows a single supply, absolute value, ideal fullwave rectifier with associated waveforms. During positive excursions, the input signal is fed through the feedback network directly to the output. Simultaneously, the positive excursion of the input signal also drives the output terminal (No. 6) of the inverting amplifier in a negative going excursion such that the 1N914 diode effectively disconnects the amplifier from the signal path. During a negative going excursion of the input signal, the CA3140 functions as a normal inverting amplifier with a gain equal to -R2/R1. When the equality of the two equations shown in Figure 26 is satisfied, the full wave output is symmetrical. NOTE: 14. “Operational Amplifiers Design and Applications”, J. G. Graeme, McGraw-Hill Book Company, page 308, “Negative Immittance Converter Circuits”. CA3140, CA3140A +15V R2 5kΩ +15V 0.1µF 0.1µF R1 10kΩ + 8 3 5 1 6 4 2 1N914 10kΩ R3 100kΩ OFFSET ADJUST PEAK ADJUST 10kΩ 6 - 100pF 4 2kΩ 0.1µF -15V 2kΩ R2 R3 GAIN = ------- = X = ----------------------------R1 R1 R2 + R3 SIMULATED LOAD + CA3140 CA3140 3 INPUT 7 - 2 7 10kΩ BW (-3dB) = 4.5MHz SR = 9V/µs 0.05µF FIGURE 28A. TEST CIRCUIT 2 X+X R 3 = ----------------- R 1 1–X R2 5kΩ FOR X = 0.5 --------------- = ------10kΩ R1 0.75 R 3 = 10kΩ ----------- = 15kΩ 0.5 20VP-P Input BW (-3dB) = 290kHz, DC Output (Avg) = 3.2V OUTPUT 0 INPUT 0 Top Trace: Output; 50mV/Div., 200ns/Div. Bottom Trace: Input; 50mV/Div., 200ns/Div. FIGURE 28B. SMALL SIGNAL RESPONSE FIGURE 26. SINGLE SUPPLY, ABSOLUTE VALUE, IDEAL FULL WAVE RECTIFIER WITH ASSOCIATED WAVEFORMS +15V 7 RS 3 1MΩ 0.01µF + 6 CA3140 2 - NOISE VOLTAGE OUTPUT 4 0.01µF 30.1kΩ -15V BW (-3dB) = 140kHz TOTAL NOISE VOLTAGE (REFERRED TO INPUT ) = 48µV (TYP) 1kΩ FIGURE 27. TEST CIRCUIT AMPLIFIER (30dB GAIN) USED FOR WIDEBAND NOISE MEASUREMENT 17 (Measurement made with Tektronix 7A13 differential amplifier.) Top Trace: Output Signal; 5V/Div., 5µs/Div. Center Trace: Difference Signal; 5mV/Div., 5µs/Div. Bottom Trace: Input Signal; 5V/Div., 5µs/Div. FIGURE 28C. INPUT-OUTPUT DIFFERENCE SIGNAL SHOWING SETTLING TIME FIGURE 28. SPLIT SUPPLY VOLTAGE FOLLOWER TEST CIRCUIT AND ASSOCIATED WAVEFORMS CA3140, CA3140A Typical Performance Curves 20 GAIN BANDWIDTH PRODUCT (MHz) OPEN-LOOP VOLTAGE GAIN (dB) RL = 2kΩ TA = -55oC 25oC 125 125oC 100 75 50 25 RL = 2kΩ CL = 100pF 10 TA = -55oC 1 0 0 5 10 15 20 25 0 5 SUPPLY VOLTAGE (V) QUIESCENT SUPPLY CURRENT (mA) 125oC TA = -55oC SLEW RATE (V/µs) 15 10 5 0 10 15 SUPPLY VOLTAGE (V) 20 25 COMMON-MODE REJECTION RATIO (dB) 25 20 15 10 5 100K 1M FREQUENCY (Hz) FIGURE 33. MAXIMUM OUTPUT VOLTAGE SWING vs FREQUENCY 18 7 6 TA = -55oC 25oC 5 125oC 4 3 2 1 0 0 5 10 15 20 25 FIGURE 32. QUIESCENT SUPPLY CURRENT vs SUPPLY VOLTAGE AND TEMPERATURE SUPPLY VOLTAGE: VS = ±15V TA = 25oC OUTPUT SWING (VP-P) 25 SUPPLY VOLTAGE (V) FIGURE 31. SLEW RATE vs SUPPLY VOLTAGE AND TEMPERATURE 0 10K 20 RL = ∞ 25oC 5 15 FIGURE 30. GAIN BANDWIDTH PRODUCT vs SUPPLY VOLTAGE AND TEMPERATURE RL = 2kΩ CL = 100pF 0 10 SUPPLY VOLTAGE (V) FIGURE 29. OPEN-LOOP VOLTAGE GAIN vs SUPPLY VOLTAGE AND TEMPERATURE 20 25oC 125oC 4M 120 SUPPLY VOLTAGE: VS = ±15V TA = 25oC 100 80 60 40 20 0 101 102 103 104 105 FREQUENCY (Hz) 106 107 FIGURE 34. COMMON MODE REJECTION RATIO vs FREQUENCY CA3140, CA3140A Typical Performance Curves SUPPLY VOLTAGE: VS = ±15V TA = 25oC SUPPLY VOLTAGE: VS = ±15V TA = 25oC POWER SUPPLY REJECTION RATIO (dB) EQUIVALENT INPUT NOISE VOLTAGE (nV/√Hz) 1000 (Continued) 100 10 1 1 101 102 103 FREQUENCY (Hz) 104 100 60 40 -PSRR 20 POWER SUPPLY REJECTION RATIO (PSRR) = ∆VIO/∆VS 0 101 105 FIGURE 35. EQUIVALENT INPUT NOISE VOLTAGE vs FREQUENCY +PSRR 80 102 103 104 105 FREQUENCY (Hz) 61 10 20 30 40 50 60 65 60 50 40 58-66 (1.473-1.676) 30 20 10 0 4-10 (0.102-0.254) 62-70 (1.575-1.778) Dimensions in parenthesis are in millimeters and are derived from the basic inch dimensions as indicated. Grid graduations are in mils (10-3 inch). The photographs and dimensions represent a chip when it is part of the wafer. When the wafer is cut into chips, the cleavage angles are 57o instead of 90ο with respect to the face of the chip. Therefore, the isolated chip is actually 7 mils (0.17mm) larger in both dimensions. 19 107 FIGURE 36. POWER SUPPLY REJECTION RATIO vs FREQUENCY Metallization Mask Layout 0 106