19-0480; Rev 3; 4/97 KIT ATION EVALU LE B A IL A AV Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers ____________________________Features The MAX1630–MAX1635 are buck-topology, step-down, switch-mode, power-supply controllers that generate logic-supply voltages in battery-powered systems. These high-performance, dual/triple-output devices include onboard power-up sequencing, power-good signaling with delay, digital soft-start, secondary winding control, lowdropout circuitry, internal frequency-compensation networks, and automatic bootstrapping. Up to 96% efficiency is achieved through synchronous rectification and Maxim’s proprietary Idle Mode™ control scheme. Efficiency is greater than 80% over a 1000:1 load-current range, which extends battery life in systemsuspend or standby mode. Excellent dynamic response corrects output load transients caused by the latest dynamic-clock CPUs within five 300kHz clock cycles. Strong 1A on-board gate drivers ensure fast external N-channel MOSFET switching. These devices feature a logic-controlled and synchronizable, fixed-frequency, pulse-width-modulation (PWM) operating mode. This reduces noise and RF interference in sensitive mobile communications and pen-entry applications. Asserting the SKIP pin enables fixed-frequency mode, for lowest noise under all load conditions. The MAX1630–MAX1635 include two PWM regulators, adjustable from 2.5V to 5.5V with fixed 5.0V and 3.3V modes. All these devices include secondary feedback regulation, and the MAX1630/MAX1632/MAX1633/ MAX1635 each contain 12V/120mA linear regulators. The MAX1631/MAX1634 include a secondary feedback input (SECFB), plus a control pin (STEER) that selects which PWM (3.3V or 5V) receives the secondary feedback signal. SECFB provides a method for adjusting the secondary winding voltage regulation point with an external resistor divider, and is intended to aid in creating auxiliary voltages other than fixed 12V. The MAX1630/MAX1631/MAX1632 contain internal output overvoltage and undervoltage protection features. ♦ ♦ ♦ ♦ ________________________Applications ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ ♦ 96% Efficiency +4.2V to +30V Input Range 2.5V to 5.5V Dual Adjustable Outputs Selectable 3.3V and 5V Fixed or Adjustable Outputs (Dual Mode™) 12V Linear Regulator Adjustable Secondary Feedback (MAX1631/MAX1634) 5V/50mA Linear Regulator Output Precision 2.5V Reference Output Programmable Power-Up Sequencing Power-Good (RESET) Output Output Overvoltage Protection (MAX1630/MAX1631/MAX1632) Output Undervoltage Shutdown (MAX1630/MAX1631/MAX1632) 200kHz/300kHz Low-Noise, Fixed-Frequency Operation Low-Dropout, 99% Duty-Factor Operation 2.5mW Typical Quiescent Power (+12V input, both SMPSs on) 4µA Typical Shutdown Current 28-Pin SSOP Package _______________Ordering Information PART TEMP. RANGE PIN-PACKAGE MAX1630CAI 0°C to +70°C 28 SSOP MAX1630EAI -40°C to +85°C 28 SSOP Ordering Information continued on last page. ________________Functional Diagram INPUT +12V +5V (RTC) Notebook and Subnotebook Computers 5V LINEAR 12V LINEAR 3.3V SMPS 5V SMPS POWER-UP SEQUENCE POWERGOOD PDAs and Mobile Communicators Desktop CPU Local DC-DC Converters +3.3V +5V Pin Configurations and Selector Guide appear at end of data sheet. ON/OFF Idle Mode and Dual Mode are trademarks of Maxim Integrated Products. RESET ________________________________________________________________ Maxim Integrated Products 1 For free samples & the latest literature: http://www.maxim-ic.com, or phone 1-800-998-8800. For small orders, phone 408-737-7600 ext. 3468. MAX1630–MAX1635 ________________General Description MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers ABSOLUTE MAXIMUM RATINGS V+ to GND ..............................................................-0.3V to +36V PGND to GND.....................................................................±0.3V VL to GND ................................................................-0.3V to +6V BST3, BST5 to GND ...............................................-0.3V to +36V LX3 to BST3..............................................................-6V to +0.3V LX5 to BST5..............................................................-6V to +0.3V REF, SYNC, SEQ, STEER, SKIP, TIME/ON5, SECFB, RESET to GND ............................................-0.3V to +6V VDD to GND ............................................................-0.3V to +20V RUN/ON3, SHDN to GND.............................-0.3V to (V+ + 0.3V) 12OUT to GND ...........................................-0.3V to (VDD + 0.3V) DL3, DL5 to PGND........................................-0.3V to (VL + 0.3V) DH3 to LX3 ...............................................-0.3V to (BST3 + 0.3V) DH5 to LX5 ...............................................-0.3V to (BST5 + 0.3V) VL, REF Short to GND ................................................Momentary 12OUT Short to GND..................................................Continuous REF Current...........................................................+5mA to -1mA VL Current.........................................................................+50mA 12OUT Current ...............................................................+200mA VDD Shunt Current ............................................................+15mA Operating Temperature Ranges MAX163_CAI.......................................................0°C to +70°C MAX163_EAI ....................................................-40°C to +85°C Storage Temperature Range .............................-65°C to +160°C Continuous Power Dissipation (TA = +70°C) SSOP (derate 9.52mW/°C above +70°C) ....................762mW Lead Temperature (soldering, 10sec) .............................+300°C Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS (V+ = 15V, both PWMs on, SYNC = VL, VL load = 0mA, REF load = 0mA, SKIP = 0V, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER CONDITIONS MIN TYP MAX UNITS 30.0 V MAIN SMPS CONTROLLERS Input Voltage Range 4.2 3V Output Voltage in Adjustable Mode V+ = 4.2V to 30V, CSH3–CSL3 = 0V, CSL3 tied to FB3 2.42 2.5 2.58 V 3V Output Voltage in Fixed Mode V+ = 4.2V to 30V, 0mV < CSH3–CSL3 < 80mV, FB3 = 0V 3.20 3.39 3.47 V 5V Output Voltage in Adjustable Mode V+ = 4.2V to 30V, CSH5–CSL5 = 0V, CSL5 tied to FB5 2.42 2.5 2.58 V 5V Output Voltage in Fixed Mode V+ = 5.2V to 30V, 0mV < CSH–CSL5 < 80mV, FB5 = 0V 4.85 5.13 5.25 V Output Voltage Adjust Range Either SMPS REF 5.5 V Adjustable-Mode Threshold Voltage Dual Mode comparator 0.5 1.1 Load Regulation Either SMPS, 0V < CSH_–CSL_ < 80mV Line Regulation Either SMPS, 5.2V < V+ < 30V -2 0.03 %/V CSH3–CSL3 or CSH5–CSL5 80 100 120 SKIP = VL or VDD < 13V or SECFB < 2.44V -50 -100 -150 Idle Mode Threshold SKIP = 0V, not tested 10 25 40 Soft-Start Ramp Time From enable to 95% full current limit with respect to fOSC (Note 1) Current-Limit Threshold Oscillator Frequency Maximum Duty Factor 2 512 mV mV clks SYNC = VL 270 300 330 SYNC = 0V 170 200 230 SYNC = VL 97 98 SYNC = 0V (Note 2) 98 99 _______________________________________________________________________________________ V % kHz % Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers (V+ = 15V, both PWMs on, SYNC = VL, VL load = 0mA, REF load = 0mA, SKIP = 0V, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER CONDITIONS MIN SYNC Input High Pulse Width Not tested 200 SYNC Input Low Pulse Width Not tested 200 SYNC Rise/Fall Time Not tested SYNC Input Frequency Range Current-Sense Input Leakage Current TYP UNITS ns ns 240 V+ = VL = 0V, CSL3 = CSH3 = CSL5 = CSH5 = 5.5V MAX 0.01 200 ns 350 kHz 10 µA 14 V FLYBACK CONTROLLER VDD Regulation Threshold Falling edge (Note 3) 13 SECFB Regulation Threshold Falling edge (MAX1631/MAX1634) DL Pulse Width VDD < 13V or SECFB < 2.44V VDD Shunt Threshold Rising edge, hysteresis = 1% (Note 3) 18 VDD Shunt Sink Current VDD = 20V (Note 3) 10 VDD Leakage Current VDD = 5V, off mode (Notes 3, 4) 2.44 2.60 1 V µs 20 V mA 30 µA 12V LINEAR REGULATOR (Note 3) 12OUT Output Voltage 13V < VDD < 18V, 0mA < ILOAD < 120mA 12OUT Current Limit 12OUT forced to 11V, VDD = 13V 11.65 12.1 150 Quiescent VDD Current VDD = 18V, run mode, no 12OUT load 50 12.50 V mA 100 µA 5.1 V 3.7 V INTERNAL REGULATOR AND REFERENCE VL Output Voltage SHDN = V+, RUN/ON3 = TIME/ON5 = 0V, 5.3V < V+ < 30V, 0mA < ILOAD < 50mA 4.7 VL Undervoltage Lockout Fault Threshold Falling edge, hysteresis = 1% 3.5 VL Switchover Threshold Rising edge of CSL5, hysteresis = 1% 4.2 4.5 4.7 V REF Output Voltage No external load (Note 5) 2.45 2.5 2.55 V REF Load Regulation 3.6 0µA < ILOAD < 50µA 12.5 0mA < ILOAD < 5mA 100.0 REF Sink Current 10 mV µA REF Fault Lockout Voltage Falling edge 2.4 V V+ Operating Supply Current VL switched over to CSL5, 5V SMPS on 1.8 5 50 µA V+ Standby Supply Current V+ = 5.5V to 30V, both SMPSs off, includes current into SHDN 30 60 µA V+ Standby Supply Current in Dropout V+ = 4.2V to 5.5V, both SMPSs off, includes current into SHDN 50 200 µA V+ Shutdown Supply Current V+ = 4V to 24V, SHDN = 0V 4 10 µA Both SMPSs enabled, FB3 = FB5 = 0V, (Note 3) CSL3 = CSH3 = 3.5V, MAX1631/ CSL5 = CSH5 = 5.3V MAX1634 2.5 4 Quiescent Power Consumption 1.5 4 mW _______________________________________________________________________________________ 3 MAX1630–MAX1635 ELECTRICAL CHARACTERISTICS (continued) MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers ELECTRICAL CHARACTERISTICS (continued) (V+ = 15V, both PWMs on, SYNC = VL, VL load = 0mA, REF load = 0mA, SKIP = 0V, TA = TMIN to TMAX, unless otherwise noted. Typical values are at TA = +25°C.) PARAMETER CONDITIONS MIN TYP MAX 7 10 UNITS FAULT DETECTION (MAX1630/MAX1631/MAX1632) Overvoltage Trip Threshold With respect to unloaded output voltage Overvoltage-Fault Propagation Delay CSL_ driven 2% above overvoltage trip threshold Output Undervoltage Threshold With respect to unloaded output voltage Output Undervoltage Lockout Time From each SMPS enabled, with respect to fOSC Thermal Shutdown Threshold Typical hysteresis = +10°C 4 1.5 % µs 60 70 80 % 5000 6144 7000 clks 150 °C RESET RESET Trip Threshold With respect to unloaded output voltage, falling edge; typical hysteresis = 1% RESET Propagation Delay Falling edge, CSL_ driven 2% below RESET trip threshold RESET Delay Time With respect to fOSC -7 -5.5 -4 1.5 27,000 % µs 32,000 37,000 clks 1 50 nA 0.6 V INPUTS AND OUTPUTS Feedback Input Leakage Current FB3, FB5; SECFB = 2.6V Logic Input Low Voltage RUN/ON3, SKIP, TIME/ON5 (SEQ = REF), SHDN, STEER, SYNC Logic Input High Voltage RUN/ON3, SKIP, TIME/ON5 (SEQ = REF), SHDN, STEER, SYNC Input Leakage Current RUN/ON3, SKIP, TIME/ON5 (SEQ = REF), SHDN, STEER, SYNC, SEQ; VPIN = 0V or 3.3V ±1 µA Logic Output Low Voltage RESET, ISINK = 4mA 0.4 V Logic Output High Current RESET = 3.5V TIME/ON5 Input Trip Level SEQ = 0V or VL 2.4 TIME/ON5 Source Current TIME/ON5 = 0V, SEQ = 0V or VL 2.5 TIME/ON5 On-Resistance Gate Driver Sink/Source Current Gate Driver On-Resistance High or low 2.4 V 1 mA 2.6 V 3 3.5 µA TIME/ON5; RUN/ON3 = 0V, SEQ = 0V or VL 15 80 Ω DL3, DH3, DL5, DH5; forced to 2V 1 7 Ω 1.5 A Note 1: Each of the four digital soft-start levels is tested for functionality; the steps are typically in 20mV increments. Note 2: High duty-factor operation supports low input-to-output differential voltages, and is achieved at a lowered operating frequency (see Overload and Dropout Operation section). Note 3: MAX1630/MAX1632/MAX1633/MAX1635 only. Note 4: Off mode for the 12V linear regulator occurs when the SMPS that has flyback feedback (VDD) steered to it is disabled. In situations where the main outputs are being held up by external keep-alive supplies, turning off the 12OUT regulator prevents a leakage path from the output-referred flyback winding, through the rectifier, and into VDD. Note 5: Since the reference uses VL as its supply, the reference’s V+ line-regulation error is insignificant. 4 _______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers EFFICIENCY vs. 3.3V OUTPUT CURRENT 90 V+ = 15V EFFICIENCY (%) 80 70 ON5 = 5V ON3 = 0V f = 300kHz MAX1631/MAX1634 60 ON3 = ON5 = 5V f = 300kHz MAX1631/MAX1634 0.01 0.1 1 0.001 10 600 5V LOAD = 0A 400 5V LOAD = 3A 200 0.01 0.1 1 0 10 5 15 10 20 5V OUTPUT CURRENT (A) 3.3V OUTPUT CURRENT (A) SUPPLY VOLTAGE (V) MAX1630/MAX1633 MAXIMUM 15V VDD OUTPUT CURRENT vs. SUPPLY VOLTAGE PWM MODE INPUT CURRENT vs. INPUT VOLTAGE IDLE MODE INPUT CURRENT vs. INPUT VOLTAGE 25 INPUT CURRENT (mA) 400 3.3V LOAD = 0A 300 200 3.3V LOAD = 3A 100 ON3 = ON5 = 5V SKIP = VL NO LOAD 10 ON3 = ON5 = 5V SKIP = 0V NO LOAD INPUT CURRENT (mA) VDD > 13V 3.3V REGULATING MAX1630/35-05 30 MAX 1630/35-04 500 20 15 10 0 0 5 15 10 1 0.1 5 0 0.01 0 20 5 10 15 20 25 30 0 5 10 15 20 25 30 SUPPLY VOLTAGE (V) INPUT VOLTAGE (V) INPUT VOLTAGE (V) STANDBY INPUT CURRENT vs. INPUT VOLTAGE SHUTDOWN INPUT CURRENT vs. INPUT VOLTAGE MINIMUM VIN TO VOUT DIFFERENTIAL vs. 5V OUTPUT CURRENT 8 INPUT CURRENT (µA) 1000 SHDN = 0V 100 10 6 4 2 0 1 5 10 15 20 INPUT VOLTAGE (V) 25 30 1000 MAX1630/35-09 10 MIN VIN TO VOUT DIFFERENTIAL (mV) ON3 = ON5 = 0V NO LOAD MAX1630/35-07 10,000 0 VDD > 13V 5V REGULATING 0 50 0.001 MAXIMUM OUTPUT CURRENT (mA) V+ = 15V 70 60 50 INPUT CURRENT (µA) 80 MAX1630/35-08 EFFICIENCY (%) 90 800 MAX 1630/35-03 V+ = 6V MAX1630/35-06 V+ = 6V MAX1630/35-02 MAX1630/35-01 100 MAXIMUM OUTPUT CURRENT (mA) EFFICIENCY vs. 5V OUTPUT CURRENT 100 MAX1632/MAX1635 MAXIMUM 15V VDD OUTPUT CURRENT vs. SUPPLY VOLTAGE 100 10 5V, 3A CIRCUIT VOUT > 4.8V f = 300kHz 1 0 5 10 15 20 INPUT VOLTAGE (V) 25 30 0.001 0.01 0.1 1 10 5V OUTPUT CURRENT (A) _______________________________________________________________________________________ 5 MAX1630–MAX1635 __________________________________________Typical Operating Characteristics (Circuit of Figure 1, 3A Table 1 components, TA = +25°C, unless otherwise noted.) ____________________________________Typical Operating Characteristics (continued) (Circuit of Figure 1, 3A Table 1 components, TA = +25°C, unless otherwise noted.) VL REGULATOR OUTPUT VOLTAGE vs. OUTPUT CURRENT SWITCHING FREQUENCY vs. LOAD CURRENT VL OUTPUT VOLTAGE (V) 100 +5V, VIN = 15V 10 +3.3V, VIN = 15V +3.3V, VIN = 6V 1 MAX 1630/35-11 5.00 MAX1630/35-10 SWITCHING FREQUENCY (kHz) 1000 4.98 4.96 4.94 4.92 VIN = 15V ON3 = ON5 = 0V +5V, VIN = 6V 4.90 0.1 10 1 100 0 1000 10 20 30 40 50 LOAD CURRENT (mA) OUTPUT CURRENT (mA) REF OUTPUT VOLTAGE vs. OUTPUT CURRENT START-UP WAVEFORMS MAX 1630/35-12 2.510 2.505 2.500 60 MAX1630/35-13 0.1 REF OUTPUT VOLTAGE (V) MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers RUN 5V/div 3.3V OUTPUT 2V/div 2.495 TIME 5V/div 2.490 2.485 5V OUTPUT 5V/div VIN = 15V ON3 = ON5 = 0V 2.480 0 1 2 3 4 5 2ms/div 6 SEQ = VL, 0.015µF CAPACITOR ON-TIME OUTPUT CURRENT (mA) __________________________________________________________________________Pin Description PIN NAME 1 CSH3 Current-Sense Input for the 3.3V SMPS. Current-limit level is 100mV referred to CSL3. 2 CSL3 Current-Sense Input. Also serves as the feedback input in fixed-output mode. 3 FB3 Feedback Input for the 3.3V SMPS; regulates at FB3 = REF (approx. 2.5V) in adjustable mode. FB3 is a Dual Mode input that also selects the 3.3V fixed output voltage setting when tied to GND. Connect FB3 to a resistor divider for adjustable-output mode. 12OUT (MAX1630/ 32/33/35) 12V/120mA Linear Regulator Output. Input supply comes from VDD. Bypass 12OUT to GND with 1µF minimum. STEER (MAX1631/ MAX1634) Logic-Control Input for secondary feedback. Selects the PWM that uses a transformer and secondary feedback signal (SECFB): STEER = GND: 3.3V SMPS uses transformer STEER = VL: 5V SMPS uses transformer 4 6 FUNCTION _______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers PIN NAME FUNCTION VDD (MAX1630/ 32/33/35) Supply Voltage Input for the 12OUT Linear Regulator. Also connects to an internal resistor divider for secondary winding feedback, and to an 18V overvoltage shunt regulator clamp. SECFB (MAX1631/ MAX1634) Secondary Winding Feedback Input. Normally connected to a resistor divider from an auxiliary output. SECFB regulates at VSECFB = 2.5V (see Secondary Feedback Regulation Loop section). Tie to VL if not used. 6 SYNC Oscillator Synchronization and Frequency Select. Tie to VL for 300kHz operation; tie to GND for 200kHz operation. Can be driven at 240kHz to 350kHz for external synchronization. 7 TIME/ON5 8 9 10 GND REF 11 RESET Active-Low Timed Reset Output. RESET swings GND to VL. Goes high after a fixed 32,000 clock-cycle delay following power-up. 12 FB5 Feedback Input for the 5V SMPS; regulates at FB5 = REF (approx. 2.5V) in adjustable mode. FB5 is a Dual Mode input that also selects the 5V fixed output voltage setting when tied to GND. Connect FB5 to a resistor divider for adjustable-output mode. 13 CSL5 14 CSH5 15 SEQ 16 DH5 17 18 19 20 LX5 BST5 DL5 PGND 21 VL 5V Internal Linear-Regulator Output. VL is also the supply voltage rail for the chip. After the 5V SMPS output has reached +4.5V (typical), VL automatically switches to the output voltage via CSL5 for bootstrapping. Bypass to GND with 4.7µF. VL supplies up to 25mA for external loads. 22 V+ Battery Voltage Input, +4.2V to +30V. Bypass V+ to PGND close to the IC with a 0.22µF capacitor. Connects to a linear regulator that powers VL. 23 SHDN Shutdown Control Input, active low. Logic threshold is set at approximately 1V. For automatic start-up, connect SHDN to V+ through a 220kΩ resistor and bypass SHDN to GND with a 0.01µF capacitor. 24 25 26 DL3 BST3 LX3 27 DH3 Gate-Drive Output for the low-side synchronous-rectifier MOSFET. Swings 0V to VL. Boost Capacitor Connection for high-side gate drive (0.1µF) Switching Node (inductor) Connection. Can swing 2V below ground without hazard. Gate-Drive Output for the 3.3V, high-side N-channel switch. DH3 is a floating driver output that swings from LX3 to BST3, riding on the LX3 switching node voltage. 28 RUN/ON3 5 SKIP Dual-Purpose Timing Capacitor Pin and ON/OFF Control Input. See Power-Up Sequencing and ON/OFF Controls section. Low-Noise Analog Ground and Feedback Reference Point 2.5V Reference Voltage Output. Bypass to GND with 1µF minimum. Logic-Control Input that disables Idle Mode when high. Connect to GND for normal use. Current-Sense Input for the 5V SMPS. Also serves as the feedback input in fixed-output mode, and as the bootstrap supply input when the voltage on CSL5/VL is > 4.5V. Current-Sense Input for the 5V SMPS. Current-limit level is 100mV referred to CSL5. Pin-Strap Input that selects the SMPS power-up sequence: SEQ = GND: 5V before 3.3V, RESET output determined by both outputs SEQ = REF: Separate ON3/ON5 controls, RESET output determined by 3.3V output SEQ = VL: 3.3V before 5V, RESET output determined by both outputs Gate-Drive Output for the 5V, high-side N-channel switch. DH5 is a floating driver output that swings from LX5 to BST5, riding on the LX5 switching node voltage. Switching Node (inductor) Connection. Can swing 2V below ground without hazard. Boost capacitor connection for high-side gate drive (0.1µF) Gate-Drive Output for the low-side synchronous-rectifier MOSFET. Swings 0V to VL. Power Ground ON/OFF Control Input. See Power-Up Sequencing and ON/OFF Controls section. _______________________________________________________________________________________ 7 MAX1630–MAX1635 _________________________________________________Pin Description (continued) MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers ON/OFF INPUT +5V ALWAYS ON C3 10Ω 4.7µF 0.1µF 0.1µF V+ SHDN SECFB VL 4.7µF Q1 0.1µF R1 +5V OUTPUT C1 BST3 DH5 Q3 DH3 0.1µF 0.1µF L1 LX5 Q2 * SYNC BST5 0.1µF L2 LX3 DL5 MAX1631 MAX1634 DL3 R2 +3.3V OUTPUT * C2 Q4 PGND CSH5 CSH3 CSL5 CSL3 FB3 FB5 RESET 5V ON/OFF TIME/ON5 3.3V ON/OFF RUN/ON3 RESET OUTPUT SKIP STEER GND SEQ REF +2.5V ALWAYS ON 1µF *1A SCHOTTKY DIODE REQUIRED FOR THE MAX1631 (SEE OUTPUT OVERVOLTAGE PROTECTION SECTION). Figure 1. Standard 3.3V/5V Application Circuit (MAX1631/MAX1634) _______Standard Application Circuit The basic MAX1631/MAX1634 dual-output 3.3V/5V buck converter (Figure 1) is easily adapted to meet a wide range of applications with inputs up to 28V by substituting components from Table 1. These circuits represent a good set of tradeoffs between cost, size, and efficiency, while staying within the worst-case specification limits for stress-related parameters, such as capacitor ripple current. Don’t change the frequency 8 of these circuits without first recalculating component values (particularly inductance value at maximum battery voltage). Adding a Schottky rectifier across each synchronous rectifier improves the efficiency of these circuits by approximately 1%, but this rectifier is otherwise not needed because the MOSFETs required for these circuits typically incorporate a high-speed silicon diode from drain to source. Use a Schottky rectifier rated at a DC current equal to at least one-third of the load current. _______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers LOAD CURRENT COMPONENT 2A 3A 4A Input Range 4.75V to 18V 4.75V to 28V 4.75V to 24V Application PDA Notebook Workstation Frequency 300kHz 300kHz 200kHz Q1, Q3 High-Side MOSFETs 1/2 IR IRF7301; 1/2 Siliconix Si9925DQ; or 1/2 Motorola MMDF3N03HD or MMDF4N01HD (10V max) IR IRF7403 or IRF7401 (18V max); Siliconix Si4412DY; or Motorola MMSF5N03HD or MMSF5N02HD (18V max) IR IRF7413 or Siliconix Si4410DY Q2, Q4 Low-Side MOSFETs 1/2 IR IRF7301; 1/2 Siliconix Si9925DQ; or 1/2 Motorola MMDF3N03HD or MMDF4N01HD (10V max) IR IRF7403 or IRF7401 (18V max); Siliconix Si4412DY; or Motorola MMSF5N03HD or MMSF5N02HD (18V max) IR IRF7413 or Siliconix Si4410DY C3 Input Capacitor 10µF, 30V Sanyo OS-CON; 22µF, 35V AVX TPS; or Sprague 594D 2 x 10µF, 30V Sanyo OS-CON; 2 x 22µF, 35V AVX TPS; or Sprague 594D 3 x 10µF, 30V Sanyo OS-CON; 4 x 22µF, 35V AVX TPS; or Sprague 595D C1, C2 Output Capacitors 220µF, 10V AVX TPS or Sprague 595D 2 x 220µF, 10V AVX TPS or Sprague 595D 4 x 220µF, 10V AVX TPS or Sprague 595D R1, R2 Resistors 0.033Ω IRC LR2010-01-R033 or Dale WSL2010-R033-F 0.02Ω IRC LR2010-01-R020 or Dale WSL2010-R020-F 0.012Ω Dale WSL2512-R012-F 15µH, 2.4A Ferrite Coilcraft DO3316P-153 or Sumida CDRH125-150 10µH, 4A Ferrite Coilcraft DO3316P-103 or Sumida CDRH125-100 4.7µH, 5.5A Ferrite Coilcraft DO3316-472 or 5.2µH, 6.5A Ferrite Sumida CDRH127-5R2MC L1, L2 Inductors Table 2. Component Suppliers COMPANY AVX FACTORY FAX (COUNTRY CODE) (1) 803-626-3123 USA PHONE (803) 946-0690 COMPANY FACTORY FAX (COUNTRY CODE) USA PHONE Motorola (1) 602-994-6430 (602) 303-5454 Murata-Erie (1) 814-238-0490 (814) 237-1431 NIEC (81) 3-3494-7414 (805) 867-2555* Central Semiconductor (1) 516-435-1824 (516) 435-1110 Coilcraft (1) 847-639-1469 (847) 639-6400 Sanyo (81) 7-2070-1174 (619) 661-6835 Coiltronics (1) 561-241-9339 (561) 241-7876 Siliconix (1) 408-970-3950 (408) 988-8000 Dale (1) 605-665-1627 (605) 668-4131 Sprague (1) 603-224-1430 (603) 224-1961 Sumida (81) 3-3607-5144 (847) 956-0666 TDK (1) 847-390-4428 (847) 390-4373 Transpower Technologies (1) 702-831-3521 (702) 831-0140 International Rectifier (IR) (1) 310-322-3332 (310) 322-3331 IRC (1) 512-992-3377 (512) 992-7900 Matsuo (1) 714-960-6492 (714) 969-2491 *Distributor _______________________________________________________________________________________ 9 MAX1630–MAX1635 Table 1. Component Selection for Standard 3.3V/5V Application MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers INPUT V+ SHDN SYNC CSL5 + MAX1632 4.5V ON/OFF +5V ALWAYS ON 12V LINEAR REG 5V LINEAR REG VL REF VDD + - 13V DH3 +3.3V +12V IN SECFB 2.5V REF BST3 12OUT BST5 RAW +15V DH5 LX3 VL DL3 3.3V PWM LOGIC 5V PWM LOGIC 200kHz TO 300kHz OSC LX5 +5V VL DL5 PGND OV/UV FAULT + REF - REF LPF 60kHz + LPF 60kHz 1.75V 2.68V CSH5 CSL5 - CSH3 CSL3 2.388V FB3 - R3 FB5 OUTPUTS UP - - R2 - + + R1 + R4 + 0.6V 0.6V VL REF - POWER-ON SEQUENCE LOGIC + SEQ + + 1V RUN/ON3 RESET 2.6V - TIME/ON5 TIMER GND Figure 2. MAX1632 Block Diagram 10 ______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers The MAX1630 is a dual, BiCMOS, switch-mode powersupply controller designed primarily for buck-topology regulators in battery-powered applications where high efficiency and low quiescent supply current are critical. Lightload efficiency is enhanced by automatic Idle Mode™ operation, a variable-frequency pulse-skipping mode that reduces transition and gate-charge losses. Each stepdown, power-switching circuit consists of two N-channel MOSFETs, a rectifier, and an LC output filter. The output voltage is the average AC voltage at the switching node, which is regulated by changing the duty cycle of the MOSFET switches. The gate-drive signal to the N-channel high-side MOSFET must exceed the battery voltage, and is provided by a flying-capacitor boost circuit that uses a 100nF capacitor connected to BST_. Devices in the MAX1630 family contain ten major circuit blocks (Figure 2). The two pulse-width modulation (PWM) controllers each consist of a Dual Mode™ feedback network and multiplexer, a multi-input PWM comparator, high-side and low-side gate drivers, and logic. MAX1630/MAX1631/ MAX1632 contain fault-protection circuits that monitor the main PWM outputs for undervoltage and overvoltage. A power-on sequence block controls the powerup timing of the main PWMs and determines whether one or both of the outputs are monitored for undervoltage faults. The MAX1630/MAX1632/MAX1633/ MAX1635 include a secondary feedback network and 12V linear regulator to generate a 12V output from a coupled-inductor flyback winding. The MAX1631/ MAX1634 have a secondary feedback input (SECFB) instead, which allows a quasi-regulated, adjustableoutput, coupled-inductor flyback winding to be attached to either the 3.3V or the 5V main inductor. Bias generator blocks include the 5V IC internal rail (VL) linear regulator, 2.5V precision reference, and automatic bootstrap switchover circuit. The PWMs share a common 200kHz/300kHz synchronizable oscillator. These internal IC blocks aren’t powered directly from the battery. Instead, the 5V VL linear regulator steps down the battery voltage to supply both VL and the gate drivers. The synchronous-switch gate drivers are directly powered from VL, while the high-side switch gate drivers are indirectly powered from VL via an external diode-capacitor boost circuit. An automatic bootstrap circuit turns off the +5V linear regulator and powers the IC from the 5V PWM output voltage if the output is above 4.5V. PWM Controller Block The two PWM controllers are nearly identical. The only differences are fixed output settings (3.3V vs. 5V), the VL/CSL5 bootstrap switch connected to the +5V PWM, and SECFB. The heart of each current-mode PWM controller is a multi-input, open-loop comparator that sums three signals: the output voltage error signal with respect to the reference voltage, the current-sense signal, and the slope compensation ramp (Figure 3). The PWM controller is a direct-summing type, lacking a traditional error amplifier and the phase shift associated with it. This direct-summing configuration approaches ideal cycle-by-cycle control over the output voltage. When SKIP = low, Idle Mode circuitry automatically optimizes efficiency throughout the load current range. Idle Mode dramatically improves light-load efficiency by reducing the effective frequency, which reduces switching losses. It keeps the peak inductor current above 25% of the full current limit in an active cycle, allowing subsequent cycles to be skipped. Idle Mode transitions seamlessly to fixed-frequency PWM operation as load current increases. With SKIP = high, the controller always operates in fixed-frequency PWM mode for lowest noise. Each pulse from the oscillator sets the main PWM latch that turns on the high-side switch for a period determined by the duty factor (approximately VOUT/VIN). As the high-side switch turns off, the synchronous rectifier latch sets; 60ns later, the low-side switch turns on. The low-side switch stays on until the beginning of the next clock cycle. Table 3. SKIP PWM Table SKIP LOAD CURRENT MODE DESCRIPTION Low Light Idle Pulse-skipping, supply current = 250µA at VIN = 12V, discontinuous inductor current Low Heavy PWM Constant-frequency PWM, continuous inductor current High Light PWM Constant-frequency PWM, continuous inductor current High Heavy PWM Constant-frequency PWM, continuous inductor current ______________________________________________________________________________________ 11 MAX1630–MAX1635 _______________Detailed Description MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers CSH_ 1X CSL_ REF FROM FEEDBACK DIVIDER MAIN PWM COMPARATOR BST_ R LEVEL SHIFT Q S DH_ LX_ SLOPE COMP OSC 30mV SKIP CURRENT LIMIT DAC SHOOTTHROUGH CONTROL CK COUNTER SHDN SOFT-START SYNCHRONOUS RECTIFIER CONTROL R -100mV S VL Q LEVEL SHIFT DL_ PGND REF 1µs SINGLE-SHOT SECFB Figure 3. PWM Controller Detailed Block Diagram 12 ______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers The output filter capacitors (Figure 1, C1 and C2) set a dominant pole in the feedback loop that must roll off the loop gain to unity before encountering the zero introduced by the output capacitor’s parasitic resistance (ESR) (see Design Procedure section). A 60kHz polezero cancellation filter provides additional rolloff above the unity-gain crossover. This internal 60kHz lowpass compensation filter cancels the zero due to filter capacitor ESR. The 60kHz filter is included in the loop in both fixed-output and adjustable-output modes. Synchronous Rectifier Driver (DL) Synchronous rectification reduces conduction losses in the rectifier by shunting the normal Schottky catch diode with a low-resistance MOSFET switch. Also, the synchronous rectifier ensures proper start-up of the boost gatedriver circuit. If the synchronous power MOSFETs are omitted for cost or other reasons, replace them with a small-signal MOSFET, such as a 2N7002. If the circuit is operating in continuous-conduction mode, the DL drive waveform is simply the complement of the DH high-side drive waveform (with controlled dead time to prevent cross-conduction or “shootthrough”). In discontinuous (light-load) mode, the synchronous switch is turned off as the inductor current falls through zero. The synchronous rectifier works The MAX1630 family uses a relatively low loop gain, allowing the use of lower-cost output capacitors. The relative gains of the voltage-sense and current-sense inputs are weighted by the values of current sources that bias three differential input stages in the main PWM comparator (Figure 4). The relative gain of the voltage comparator to the current comparator is internally fixed at K = 2:1. The low loop gain results in the 2% typical load-regulation error. The low value of loop gain helps reduce output filter capacitor size and cost by shifting the unity-gain crossover frequency to a lower level. VL R1 R2 TO PWM LOGIC UNCOMPENSATED HIGH-SPEED LEVEL TRANSLATOR AND BUFFER OUTPUT DRIVER FB_ I1 I2 I3 VBIAS REF CSH_ CSL_ SLOPE COMPENSATION Figure 4. Main PWM Comparator Block Diagram ______________________________________________________________________________________ 13 MAX1630–MAX1635 In PWM mode, the controller operates as a fixedfrequency current-mode controller where the duty ratio is set by the input/output voltage ratio. The currentmode feedback system regulates the peak inductor current value as a function of the output-voltage error signal. In continuous-conduction mode, the average inductor current is nearly the same as the peak current, so the circuit acts as a switch-mode transconductance amplifier. This pushes the second output LC filter pole, normally found in a duty-factor-controlled (voltagemode) PWM, to a higher frequency. To preserve innerloop stability and eliminate regenerative inductor current “staircasing,” a slope compensation ramp is summed into the main PWM comparator to make the apparent duty factor less than 50%. MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers under all operating conditions, including Idle Mode. The SECFB signal further controls the synchronous switch timing in order to improve multiple-output crossregulation (see Secondary Feedback Regulation Loop section). turns on the high-side MOSFET by closing an internal switch between BST_ and DH_. This provides the necessary enhancement voltage to turn on the high-side switch, an action that “boosts” the 5V gate-drive signal above the battery voltage. Internal VL and REF Supplies Ringing at the high-side MOSFET gate (DH3 and DH5) in discontinuous-conduction mode (light loads) is a natural operating condition. It is caused by residual energy in the tank circuit, formed by the inductor and stray capacitance at the switching node, LX. The gate-drive negative rail is referred to LX, so any ringing there is directly coupled to the gate-drive output. An internal regulator produces the +5V supply (VL) that powers the PWM controller, logic, reference, and other blocks within the IC. This 5V low-dropout linear regulator supplies up to 25mA for external loads, with a reserve of 25mA for supplying gate-drive power. Bypass VL to GND with 4.7µF. Important: Ensure that VL does not exceed 6V. Measure VL with the main output fully loaded. If it is pumped above 5.5V, either excessive boost diode capacitance or excessive ripple at V+ is the probable cause. Use only small-signal diodes for the boost circuit (10mA to 100mA Schottky or 1N4148 are preferred), and bypass V+ to PGND with 4.7µF directly at the package pins. The 2.5V reference (REF) is accurate to ±2% over temperature, making REF useful as a precision system reference. Bypass REF to GND with 1µF minimum. REF can supply up to 5mA for external loads. (Bypass REF with a minimum 1µF/mA reference load current.) However, if extremely accurate specifications for both the main output voltages and REF are essential, avoid loading REF more than 100µA. Loading REF reduces the main output voltage slightly, because of the reference load-regulation error. When the 5V main output voltage is above 4.5V, an internal P-channel MOSFET switch connects CSL5 to VL, while simultaneously shutting down the VL linear regulator. This action bootstraps the IC, powering the internal circuitry from the output voltage, rather than through a linear regulator from the battery. Bootstrapping reduces power dissipation due to gate charge and quiescent losses by providing that power from a 90%-efficient switch-mode source, rather than from a much less efficient linear regulator. Boost High-Side Gate-Drive Supply (BST3 and BST5) Gate-drive voltage for the high-side N-channel switches is generated by a flying-capacitor boost circuit (Figure 2). The capacitor between BST_ and LX_ is alternately charged from the VL supply and placed parallel to the high-side MOSFET’s gate-source terminals. On start-up, the synchronous rectifier (low-side MOSFET) forces LX_ to 0V and charges the boost capacitors to 5V. On the second half-cycle, the SMPS 14 Current-Limiting and Current-Sense Inputs (CSH and CSL) The current-limit circuit resets the main PWM latch and turns off the high-side MOSFET switch whenever the voltage difference between CSH and CSL exceeds 100mV. This limiting is effective for both current flow directions, putting the threshold limit at ±100mV. The tolerance on the positive current limit is ±20%, so the external low-value sense resistor (R1) must be sized for 80mV/IPEAK, where IPEAK is the required peak inductor current to support the full load current, while components must be designed to withstand continuous current stresses of 120mV/R1. For breadboarding or for very-high-current applications, it may be useful to wire the current-sense inputs with a twisted pair, rather than PC traces. (This twisted pair needn’t be anything special; two pieces of wire-wrap wire twisted together are sufficient.) This reduces the possible noise picked up at CSH_ and CSL_, which can cause unstable switching and reduced output current. The CSL5 input also serves as the IC’s bootstrap supply input. Whenever VCSL5 > 4.5V, an internal switch connects CSL5 to VL. Oscillator Frequency and Synchronization (SYNC) The SYNC input controls the oscillator frequency. Low selects 200kHz; high selects 300kHz. SYNC can also be used to synchronize with an external 5V CMOS or TTL clock generator. SYNC has a guaranteed 240kHz to 350kHz capture range. A high-to-low transition on SYNC initiates a new cycle. 300kHz operation optimizes the application circuit for component size and cost. 200kHz operation provides increased efficiency, lower dropout, and improved load-transient response at low input-output voltage differences (see Low-Voltage Operation section). ______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers Power-Up Sequencing and ON/OFF Controls Start-up is controlled by RUN/ON3 and TIME/ON5 in conjunction with SEQ. With SEQ tied to REF, the two control inputs act as separate ON/OFF controls for each supply. With SEQ tied to VL or GND, RUN/ON3 becomes the master ON/OFF control input and TIME/ON5 becomes a timing pin, with the delay between the two supplies determined by an external capacitor. The delay is approximately 800µs/nF. The +3.3V supply powers-up first if SEQ is tied to VL, and the +5V supply is first if SEQ is tied to GND. When driving TIME/ON5 as a control input with external logic, always place a resistor (>1kΩ) in series with the input. This prevents possible crowbar current due to the internal discharge pull-down transistor, which turns on in standby mode and momentarily at the first power-up or in shutdown mode. RESET Power-Good Voltage Monitor The power-good monitor generates a system RESET signal. At first power-up, RESET is held low until both the 3.3V and 5V SMPS outputs are in regulation. At this point, an internal timer begins counting oscillator pulses, and RESET continues to be held low until 32,000 cycles have elapsed. After this timeout period (107ms at 300kHz or 160ms at 200kHz), RESET is actively pulled up to VL. If SEQ is tied to REF (for separate ON3/ON5 controls), only the 3.3V SMPS is monitored—the 5V SMPS is ignored. Output Undervoltage Shutdown Protection (MAX1630/MAX1631/MAX1632) The output undervoltage lockout circuit is similar to foldback current limiting, but employs a timer rather than a variable current limit. Each SMPS has an undervoltage protection circuit that is activated 6144 clock cycles after the SMPS is enabled. If either SMPS output is under 70% of the nominal value, both SMPSs are latched off and their outputs are clamped to ground by the synchronous rectifier MOSFETs (see Output Overvoltage Protection section). They won’t restart until SHDN or RUN/ON3 is toggled, or until V+ power is cycled below 1V. Note that undervoltage protection can make prototype troubleshooting difficult, since you have only 20ms or 30ms to figure out what might be wrong with the circuit before both SMPSs are latched off. In extreme cases, it may be useful to substitute the MAX1633/MAX1634/MAX1635 into the prototype breadboard until the prototype is working properly. Output Overvoltage Protection (MAX1630/MAX1631/MAX1632) Both SMPS outputs are monitored for overvoltage. If either output is more than 7% above the nominal regulation point, both low-side gate drivers (DL_) are latched high until SHDN or RUN/ON3 is toggled, or until V+ power is cycled below 1V. This action turns on the synchronous rectifiers with 100% duty, in turn rapidly discharging the output capacitors and forcing both SMPS outputs to ground. The DL outputs are also kept high whenever the corresponding SMPS is disabled, and in shutdown if VL is sustained. Table 4. Operating Modes SHDN SEQ RUN/ON3 TIME/ON5 MODE DESCRIPTION All circuit blocks turned off. Supply current = 4µA. Low X X X Shutdown High Ref Low Low Standby High Ref High Low Run 3.3V SMPS enabled/5V off High Ref Low High Run 5V SMPS enabled/3.3V off High Ref High High Run Both SMPSs enabled High GND Low Timing capacitor Standby High GND High Timing capacitor Run High VL Low Timing capacitor Standby High VL High Timing capacitor Run Both SMPSs off. Supply current = 30µA. Both SMPSs off. Supply current = 30µA. Both SMPSs enabled. 5V enabled before 3.3V. Both SMPSs off. Supply current = 30µA. Both SMPSs enabled. 3.3V enabled before 5V. X = Don’t Care ______________________________________________________________________________________ 15 MAX1630–MAX1635 Shutdown Mode Holding SHDN low puts the IC into its 4µA shutdown mode. SHDN is logic input with a threshold of about 1V (the VTH of an internal N-channel MOSFET). For automatic start-up, bypass SHDN to GND with a 0.01µF capacitor and connect it to V+ through a 220kΩ resistor. MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers Discharging the output capacitor through the main inductor causes the output to momentarily go below GND. Clamp this negative pulse with a back-biased 1A Schottky diode across the output capacitor (Figure 1). To ensure overvoltage protection on initial power-up, connect signal diodes from both output voltages to VL (cathodes to VL) to eliminate the VL power-up delay. This circuitry protects the load from accidental overvoltage caused by a short-circuit across the high-side power MOSFETs. This scheme relies on the presence of a fuse, in series with the battery, which is blown by the resulting crowbar current. Note that the overvoltage circuitry will interfere with external keep-alive supplies that hold up the outputs (such as lithium backup or hotswap power supplies); in such cases, the MAX1633, MAX1634, or MAX1635 should be used. Low-Noise Operation (PWM Mode) PWM mode (SKIP = high) minimizes RF and audio interference in noise-sensitive applications (such as hifi multimedia-equipped systems), cellular phones, RF communicating computers, and electromagnetic penentry systems. See the summary of operating modes in Table 2. SKIP can be driven from an external logic signal. Interference due to switching noise is reduced in PWM mode by ensuring a constant switching frequency, thus concentrating the emissions at a known frequency outside the system audio or IF bands. Choose an oscillator frequency for which switching frequency harmonics don’t overlap a sensitive frequency band. If necessary, synchronize the oscillator to a tight-tolerance external clock generator. To extend the output-voltage-regulation range, constant operating frequency is not maintained under overload or dropout conditions (see Overload and Dropout Operation section.) PWM mode (SKIP = high) forces two changes upon the PWM controllers. First, it disables the minimum-current comparator, ensuring fixed-frequency operation. Second, it changes the detection threshold for reversecurrent limit from 0mV to -100mV, allowing the inductor current to reverse at light loads. This results in fixedfrequency operation and continuous inductor-current flow. This eliminates discontinuous-mode inductor ringing and improves cross regulation of transformercoupled multiple-output supplies, particularly in circuits that don’t use additional secondary regulation via SECFB or VDD. In most applications, tie SKIP to GND to minimize quiescent supply current. VL supply current with SKIP high is typically 20mA, depending on external MOSFET gate capacitance and switching losses. 16 Internal Digital Soft-Start Circuit Soft-start allows a gradual increase of the internal current-limit level at start-up to reduce input surge currents. Both SMPSs contain internal digital soft-start circuits, each controlled by a counter, a digital-to-analog converter (DAC), and a current-limit comparator. In shutdown or standby mode, the soft-start counter is reset to zero. When an SMPS is enabled, its counter starts counting oscillator pulses, and the DAC begins incrementing the comparison voltage applied to the currentlimit comparator. The DAC output increases from 0mV to 100mV in five equal steps as the count increases to 512 clocks. As a result, the main output capacitor charges up relatively slowly. The exact time of the output rise depends on output capacitance and load current, and is typically 1ms with a 300kHz oscillator. Dropout Operation Dropout (low input-output differential operation) is enhanced by stretching the clock pulse width to increase the maximum duty factor. The algorithm follows: If the output voltage (VOUT) drops out of regulation without the current limit having been reached, the SMPS skips an off-time period (extending the on-time). At the end of the cycle, if the output is still out of regulation, the SMPS skips another off-time period. This action can continue until three off-time periods are skipped, effectively dividing the clock frequency by as much as four. The typical PWM minimum off-time is 300ns, regardless of the operating frequency. Lowering the operating frequency raises the maximum duty factor above 98%. Adjustable-Output Feedback (Dual Mode FB) Fixed, preset output voltages are selected when FB_ is connected to ground. Adjusting the main output voltage with external resistors is simple for any of the MAX1630 family ICs, through resistor dividers connected to FB3 and FB5 (Figure 2). Calculate the output voltage with the following formula: VOUT = VREF (1 + R1 / R2) where VREF = 2.5V nominal. The nominal output should be set approximately 1% or 2% high to make up for the MAX1630’s -2% typical load-regulation error. For example, if designing for a 3.0V output, use a resistor ratio that results in a nominal output voltage of 3.05V. This slight offsetting gives the best possible accuracy. Recommended normal values for R2 range from 5kΩ to 100kΩ. To achieve a 2.5V nominal output, simply connect FB_ directly to CSL_. ______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers Secondary Feedback Regulation Loop (SECFB or VDD) A flyback-winding control loop regulates a secondary winding output, improving cross-regulation when the primary output is lightly loaded or when there is a low input-output differential voltage. If V DD or SECFB falls below its regulation threshold, the low-side switch is turned on for an extra 1µs. This reverses the inductor (primary) current, pulling current from the output filter capacitor and causing the flyback transformer to operate in forward mode. The low impedance presented by the transformer secondary in forward mode dumps current into the secondary output, charging up the secondary capacitor and bringing VDD or SECFB back into regulation. The secondary feedback loop does not improve secondary output accuracy in normal flyback mode, where the main (primary) output is heavily loaded. In this condition, secondary output accuracy is determined by the secondary rectifier drop, transformer turns ratio, and accuracy of the main output voltage. A linear post-regulator may still be needed to meet strict output-accuracy specifications. Devices with a 12OUT linear regulator have a VDD pin that regulates at a fixed 13.5V, set by an internal resistor divider. The MAX1631/MAX1634 have an adjustable secondary output voltage set by an external resistor divider on SECFB (Figure 5). Ordinarily, the secondary regulation point is set 5% to 10% below the voltage normally produced by the flyback effect. For example, if the output voltage as determined by turns ratio is 15V, set the feedback resistor ratio to produce 13.5V. Otherwise, the SECFB one-shot might be triggered unintentionally, unnecessarily increasing supply current and output noise. R2 SECFB 1-SHOT TRIG R1 2.5V REF POSITIVE SECONDARY OUTPUT V+ DH_ MAIN OUTPUT MAX1631 MAX1634 DL_ R1 +VTRIP = VREF 1 + ––– R2 ( ) WHERE VREF (NOMINAL) = 2.5V Figure 5. Adjusting the Secondary Output Voltage with SECFB +12V OUTPUT 200mA 12OUT 0.1µF 10µF VDD 2N3906 0.1µF MAX1630 MAX1632 MAX1633 MAX1635 DH_ V+ 10Ω 0.1µF VDD OUTPUT 2.2µF MAIN OUTPUT DL_ Figure 6. Increased 12V Linear Regulator Output Current 12V Linear Regulator Output (MAX1630/MAX1632/MAX1633/MAX1635) The MAX1630/MAX1632/MAX1633/MAX1635 include a 12V linear regulator output capable of delivering 120mA of output current. Typically, greater current is available at the expense of output accuracy. If an accurate output of more than 120mA is needed, an external pass tran- ______________________________________________________________________________________ 17 MAX1630–MAX1635 Remote output-voltage sensing, while not possible in fixed-output mode due to the combined nature of the voltage-sense and current-sense inputs (CSL3 and CSL5), is easy to do in adjustable mode by using the top of the external resistor divider as the remote sense point. When using adjustable mode, it is a good idea to always set the “3.3V output” to a lower voltage than the “5V output.” The 3.3V output must always be less than VL, so that the voltage on CSH3 and CSL3 is within the common-mode range of the current-sense inputs. While VL is nominally 5V, it can be as low as 4.7V when linearly regulating, and as low as 4.2V when automatically bootstrapped to CSH5. MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers sistor can be added. Figure 6’s circuit delivers more than 200mA. Total output current is constrained by the V+ input voltage and the transformer primary load (see Maximum 15V VDD Output Current vs. Supply Voltage graphs in the Typical Operating Characteristics). for high-efficiency, battery-powered applications. See Appendix A in Maxim’s Battery Management and DCDC Converter Circuit Collection for crossover-point and discontinuous-mode equations. Discontinuous conduction doesn’t affect normal Idle Mode operation. __________________Design Procedure Three key inductor parameters must be specified: inductance value (L), peak current (IPEAK), and DC resistance (RDC). The following equation includes a constant, LIR, which is the ratio of inductor peak-topeak AC current to DC load current. A higher LIR value allows smaller inductance, but results in higher losses and higher ripple. A good compromise between size and losses is found at a 30% ripple-current to loadcurrent ratio (LIR = 0.3), which corresponds to a peak inductor current 1.15 times higher than the DC load current. The three predesigned 3V/5V standard application circuits (Figure 1 and Table 1) contain ready-to-use solutions for common application needs. Also, two standard flyback transformer circuits support the 12OUT linear regulator in the Applications Information section. Use the following design procedure to optimize these basic schematics for different voltage or current requirements. But before beginning a design, firmly establish the following: Maximum input (battery) voltage, V IN(MAX) . This value should include the worst-case conditions, such as no-load operation when a battery charger or AC adapter is connected but no battery is installed. VIN(MAX) must not exceed 30V. Minimum input (battery) voltage, V IN(MIN) . This should be taken at full load under the lowest battery conditions. If VIN(MIN) is less than 4.2V, use an external circuit to externally hold VL above the VL undervoltage lockout threshold. If the minimum input-output difference is less than 1.5V, the filter capacitance required to maintain good AC load regulation increases (see LowVoltage Operation section). Inductor Value The exact inductor value isn’t critical and can be freely adjusted to make trade-offs between size, cost, and efficiency. Lower inductor values minimize size and cost, but reduce efficiency due to higher peak-current levels. The smallest inductor is achieved by lowering the inductance until the circuit operates at the border between continuous and discontinuous mode. Further reducing the inductor value below this crossover point results in discontinuous-conduction operation even at full load. This helps lower output filter capacitance requirements, but efficiency suffers due to high I2R losses. On the other hand, higher inductor values mean greater efficiency, but resistive losses due to extra wire turns will eventually exceed the benefit gained from lower peak-current levels. Also, high inductor values can affect load-transient response (see the VSAG equation in the Low-Voltage Operation section). The equations that follow are for continuous-conduction operation, since the MAX1630 family is intended mainly L = VOUT (VIN(MAX) - VOUT ) VIN(MAX) x f x IOUT x LIR where: f = switching frequency, normally 200kHz or 300kHz IOUT = maximum DC load current LIR = ratio of AC to DC inductor current, typically 0.3; should be selected for >0.15 The nominal peak inductor current at full load is 1.15 x IOUT if the above equation is used; otherwise, the peak current can be calculated by: IPEAK = ILOAD + VOUT (VIN(MAX) - VOUT ) 2 x f x L x VIN(MAX) The inductor’s DC resistance should be low enough that RDC x IPEAK < 100mV, as it is a key parameter for efficiency performance. If a standard off-the-shelf inductor is not available, choose a core with an LI2 rating greater than L x IPEAK2 and wind it with the largest-diameter wire that fits the winding area. For 300kHz applications, ferrite core material is strongly preferred; for 200kHz applications, Kool-Mu® (aluminum alloy) or even powdered iron is acceptable. If light-load efficiency is unimportant (in desktop PC applications, for example), then low-permeability iron-powder cores, such as the Micrometals type found in Pulse Engineering’s 2.1µH PE-53680, may be acceptable even at 300kHz. For high-current applications, shielded-core geometries, such as toroidal or pot core, help keep noise, EMI, and switching-waveform jitter low. Kool-Mu is a registered trademark of Magnetics Div., Spang & Co. 18 ______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers Bypassing VL Bypass the VL output with a 4.7µF tantalum capacitor paralleled with a 0.1µF ceramic capacitor, close to the device. 80mV RSENSE = IPEAK The output filter capacitor values are generally determined by the ESR and voltage rating requirements, rather than actual capacitance requirements for loop stability. In other words, the low-ESR electrolytic capacitor that meets the ESR requirement usually has more output capacitance than is required for AC stability. Use only specialized low-ESR capacitors intended for switching-regulator applications, such as AVX TPS, Sprague 595D, Sanyo OS-CON, or Nichicon PL series. To ensure stability, the capacitor must meet both minimum capacitance and maximum ESR values as given in the following equations: Use IPEAK from the second equation in the Inductor Value section Use the calculated value of RSENSE to size the MOSFET switches and specify inductor saturation-current ratings according to the worst-case high-current-limit threshold voltage: IPEAK(MAX) = 120mV RSENSE Low-inductance resistors, such as surface-mount metal-film, are recommended. Input Capacitor Value Connect low-ESR bulk capacitors and small ceramic capacitors (0.1µF) directly to the drains on the highside MOSFETs. The bulk input filter capacitor is usually selected according to input ripple current requirements and voltage rating, rather than capacitor value. Electrolytic capacitors with low enough effective series resistance (ESR) to meet the ripple current requirement invariably have sufficient capacitance values. Aluminum electrolytic capacitors, such as Sanyo OS-CON or Nichicon PL, are superior to tantalum types, which carry the risk of power-up surge-current failure, especially when connecting to robust AC adapters or low-impedance batteries. RMS input ripple current (IRMS) is determined by the input voltage and load current, with the worst case occurring at VIN = 2 x VOUT: IRMS = ILOAD x VOUT (VIN - VOUT ) VIN Therefore, when VIN is 2 x VOUT : I IRMS = LOAD 2 Bypassing V+ Bypass the V+ input with a 4.7µF tantalum capacitor paralleled with a 0.1µF ceramic capacitor, close to the IC. A 10Ω series resistor to VIN is also recommended. Output Filter Capacitor Value COUT > VREF (1 + VOUT / VIN(MIN) ) VOUT x RSENSE x f R x VOUT RESR < SENSE VREF (can be multiplied by 1.5; see text below) These equations are worst case, with 45 degrees of phase margin to ensure jitter-free, fixed-frequency operation and provide a nicely damped output response for zero to full-load step changes. Some costconscious designers may wish to bend these rules with less-expensive capacitors, particularly if the load lacks large step changes. This practice is tolerable if some bench testing over temperature is done to verify acceptable noise and transient response. No well-defined boundary exists between stable and unstable operation. As phase margin is reduced, the first symptom is a bit of timing jitter, which shows up as blurred edges in the switching waveforms where the scope won’t quite sync up. Technically speaking, this jitter (usually harmless) is unstable operation, since the duty factor varies slightly. As capacitors with higher ESRs are used, the jitter becomes more pronounced, and the load-transient output voltage waveform starts looking ragged at the edges. Eventually, the load-transient waveform has enough ringing on it that the peak noise levels exceed the allowable output voltage tolerance. Note that even with zero phase margin and gross instability present, the output voltage noise never gets much worse than IPEAK x RESR (under constant loads). Designers of RF communicators or other noise-sensitive analog equipment should be conservative and stay within the guidelines. Designers of notebook computers and similar commercial-temperature-range digital ______________________________________________________________________________________ 19 MAX1630–MAX1635 Current-Sense Resistor Value The current-sense resistor value is calculated according to the worst-case-low current-limit threshold voltage (from the Electrical Characteristics table) and the peak inductor current: MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers systems can multiply the RESR value by a factor of 1.5 without hurting stability or transient response. The output voltage ripple is usually dominated by the filter capacitor’s ESR, and can be approximated as IRIPPLE x RESR. There is also a capacitive term, so the full equation for ripple in continuous-conduction mode is V NOISE (p-p) = I RIPPLE x [R ESR + 1/(2 x π x f x COUT)]. In Idle Mode, the inductor current becomes discontinuous, with high peaks and widely spaced pulses, so the noise can actually be higher at light load (compared to full load). In Idle Mode, calculate the output ripple as follows: VNOISE(p-p) = 0.02 x RESR + RSENSE [ ] 0.0003 x Lx 1 / VOUT + 1 / (VIN - VOUT ) 2 (RSENSE ) x COUT where: V SEC = the minimum required rectified secondary output voltage VFWD = the forward drop across the secondary rectifier V OUT(MIN) = the minimum value of the main output voltage (from the Electrical Characteristics) VRECT = the on-state voltage drop across the synchronous rectifier MOSFET VSENSE = the voltage drop across the sense resistor In positive-output applications, the transformer secondary return is often referred to the main output voltage, rather than to ground, to reduce the needed turns ratio. In this case, the main output voltage must first be subtracted from the secondary voltage to obtain VSEC. Selecting Other Components Transformer Design (for Auxiliary Outputs Only) Buck-plus-flyback applications, sometimes called “coupled-inductor” topologies, need a transformer to generate multiple output voltages. Performing the basic electrical design is a simple task of calculating turns ratios and adding the power delivered to the secondary to calculate the current-sense resistor and primary inductance. However, extremes of low input-output differentials, widely different output loading levels, and high turns ratios can complicate the design due to parasitic transformer parameters such as interwinding capacitance, secondary resistance, and leakage inductance. For examples of what is possible with realworld transformers, see the Maximum Secondary Current vs. Input Voltage graph in the Typical Operating Characteristics section. Power from the main and secondary outputs is combined to get an equivalent current referred to the main output voltage (see the Inductor Value section for parameter definitions). Set the current-sense resistor resistor value at 80mV / ITOTAL. PTOTAL = The sum of the output power from all outputs ITOTAL = PTOTAL / VOUT = The equivalent output current referred to VOUT L(primary) = VOUT (VIN(MAX) - VOUT ) VIN(MAX) x f x ITOTAL x LIR Turns Ratio N = 20 VSEC + VFWD VOUT(MIN) + VRECT + VSENSE MOSFET Switches The high-current N-channel MOSFETs must be logic-level types with guaranteed on-resistance specifications at VGS = 4.5V. Lower gate threshold specifications are better (i.e., 2V max rather than 3V max). Drain-source breakdown voltage ratings must at least equal the maximum input voltage, preferably with a 20% derating factor. The best MOSFETs will have the lowest on-resistance per nanocoulomb of gate charge. Multiplying RDS(ON) x QG provides a good figure for comparing various MOSFETs. Newer MOSFET process technologies with dense cell structures generally perform best. The internal gate drivers tolerate >100nC total gate charge, but 70nC is a more practical upper limit to maintain best switching times. In high-current applications, MOSFET package power dissipation often becomes a dominant design factor. I2R power losses are the greatest heat contributor for both high-side and low-side MOSFETs. I2R losses are distributed between Q1 and Q2 according to duty factor (see the following equations). Generally, switching losses affect only the upper MOSFET, since the Schottky rectifier clamps the switching node in most cases before the synchronous rectifier turns on. Gatecharge losses are dissipated by the driver and don’t heat the MOSFET. Calculate the temperature rise according to package thermal-resistance specifications to ensure that both MOSFETs are within their maximum junction temperature at high ambient temperature. The worst-case dissipation for the high-side MOSFET occurs at both extremes of input voltage, and the worst-case dissipation for the low-side MOSFET occurs at maximum input voltage. ______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers V x CRSS + VIN x ILOAD x f x IN + 20ns IGATE PD(lower FET) = (ILOAD )2 x RDS(ON) x (1 - DUTY) DUTY = (VOUT + VQ2 ) / (VIN - VQ1) where: on-state voltage drop VQ_ = ILOAD x RDS(ON) CRSS = MOSFET reverse transfer capacitance IGATE =DH driver peak output current capability (1A typical) 20ns = DH driver inherent rise/fall time Under output short-circuit, the MAX1633/MAX1634/ MAX1635’s synchronous rectifier MOSFET suffers extra stress because its duty factor can increase to greater than 0.9. It may need to be oversized to tolerate a continuous DC short circuit. During short circuit, the MAX1630/MAX1631/MAX1632’s output undervoltage shutdown protects the synchronous rectifier under output short-circuit conditions. To reduce EMI, add a 0.1µF ceramic capacitor from the high-side switch drain to the low-side switch source. Rectifier Clamp Diode The rectifier is a clamp across the low-side MOSFET that catches the negative inductor swing during the 60ns dead time between turning one MOSFET off and each low-side MOSFET on. The latest generations of MOSFETs incorporate a high-speed silicon body diode, which serves as an adequate clamp diode if efficiency is not of primary importance. A Schottky diode can be placed in parallel with the body diode to reduce the forward voltage drop, typically improving efficiency 1% to 2%. Use a diode with a DC current rating equal to onethird of the load current; for example, use an MBR0530 (500mA-rated) type for loads up to 1.5A, a 1N5819 type for loads up to 3A, or a 1N5822 type for loads up to 10A. The rectifier’s rated reverse breakdown voltage must be at least equal to the maximum input voltage, preferably with a 20% derating factor. Boost-Supply Diode D2 A signal diode such as a 1N4148 works well in most applications. If the input voltage can go below +6V, use a small (20mA) Schottky diode for slightly improved efficiency and dropout characteristics. Don’t use large power diodes, such as 1N5817 or 1N4001, since high junction capacitance can pump up VL to excessive voltages. Rectifier Diode D3 (Transformer Secondary Diode) The secondary diode in coupled-inductor applications must withstand flyback voltages greater than 60V, which usually rules out most Schottky rectifiers. Common silicon rectifiers, such as the 1N4001, are also prohibited because they are too slow. This often makes fast silicon rectifiers such as the MURS120 the only choice. The flyback voltage across the rectifier is related to the VIN - VOUT difference, according to the transformer turns ratio: VFLYBACK = VSEC + (VIN - VOUT ) x N where: N = the transformer turns ratio SEC/PRI V SEC = the maximum secondary DC output voltage VOUT = the primary (main) output voltage Subtract the main output voltage (VOUT) from VFLYBACK in this equation if the secondary winding is returned to VOUT and not to ground. The diode reverse breakdown rating must also accommodate any ringing due to leakage inductance. D3’s current rating should be at least twice the DC load current on the secondary output. Low-Voltage Operation Low input voltages and low input-output differential voltages each require extra care in their design. Low absolute input voltages can cause the VL linear regulator to enter dropout and eventually shut itself off. Low input voltages relative to the output (low VIN-VOUT differential) can cause bad load regulation in multi-output flyback applications (see the design equations in the Transformer Design section). Also, low VIN-VOUT differentials can also cause the output voltage to sag when the load current changes abruptly. The amplitude of the sag is a function of inductor value and maximum duty factor (an Electrical Characteristics parameter, 98% guaranteed over temperature at f = 200kHz), as follows: VSAG = (ISTEP )2 x L 2 x COUT x (VIN(MAX) x DMAX - VOUT ) The cure for low-voltage sag is to increase the output capacitor’s value. For example, at VIN = +5.5V, VOUT = +5V, L = 10µH, f = 200kHz, ISTEP = 3A, a total capacitance of 660µF keeps the sag less than 200mV. Note that only the capacitance requirement increases, and the ESR requirements don’t change. Therefore, the added capacitance can be supplied by a low-cost bulk capacitor in parallel with the normal low-ESR capacitor. ______________________________________________________________________________________ 21 MAX1630–MAX1635 PD(upper FET) = (ILOAD )2 x RDS(ON) x DUTY MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers Table 5. Low-Voltage Troubleshooting Chart SYMPTOM CONDITION ROOT CAUSE SOLUTION Sag or droop in VOUT under step-load change Low VIN-VOUT differential, <1.5V Limited inductor-current slew rate per cycle. Increase bulk output capacitance per formula (see Low-Voltage Operation section). Reduce inductor value. Dropout voltage is too high (VOUT follows VIN as VIN decreases) Low VIN-VOUT differential, <1V Maximum duty-cycle limits exceeded. Reduce operation to 200kHz. Reduce MOSFET on-resistance and coil DCR. Unstable—jitters between different duty factors and frequencies Low VIN-VOUT differential, <0.5V Normal function of internal low-dropout circuitry. Increase the minimum input voltage or ignore. Secondary output won’t support a load Low VIN-VOUT differential, VIN < 1.3 x VOUT (main) Not enough duty cycle left to initiate forward-mode operation. Small AC current in primary can’t store energy for flyback operation. Reduce operation to 200kHz. Reduce secondary impedances; use a Schottky diode, if possible. Stack secondary winding on the main output. Poor efficiency Low input voltage, <5V VL linear regulator is going into dropout and isn’t providing good gate-drive levels. Use a small 20mA Schottky diode for boost diode D2. Supply VL from an external source. Won’t start under load or quits before battery is completely dead Low input voltage, <4.5V VL output is so low that it hits the VL UVLO threshold. Supply VL from an external source other than VIN, such as the system +5V supply. ________________Applications Information Heavy-Load Efficiency Considerations The major efficiency-loss mechanisms under loads are, in the usual order of importance: • P(I2R) = I2R losses • P(tran) = transition losses • P(gate) = gate-charge losses • P(diode) = diode-conduction losses • P(cap) = capacitor ESR losses • P(IC) = losses due to the IC’s operating supply supply current Inductor core losses are fairly low at heavy loads because the inductor’s AC current component is small. Therefore, they aren’t accounted for in this analysis. Ferrite cores are preferred, especially at 300kHz, but powdered cores, such as Kool-Mu, can work well. Efficiency = POUT / PIN x 100% = POUT / (POUT + PTOTAL ) x 100% PTOTAL = P(I2R) + P(tran) + P(gate) + P(diode) + P(cap) + P(IC) P = (I2R) = (ILOAD )2 x (RDC + RDS(ON) + RSENSE ) 22 where RDC is the DC resistance of the coil, RDS(ON) is the MOSFET on-resistance, and RSENSE is the currentsense resistor value. The RDS(ON) term assumes identical MOSFETs for the high-side and low-side switches, because they time-share the inductor current. If the MOSFETs aren’t identical, their losses can be estimated by averaging the losses according to duty factor. 3 PD(tran) = transition loss = VIN x ILOAD x f x x 2 [(VIN x CRSS / IGATE ) + 20ns] where CRSS is the reverse transfer capacitance of the high-side MOSFET (a data-sheet parameter), IGATE is the DH gate-driver peak output current (1.5A typical), and 20ns is the rise/fall time of the DH driver (20ns typical). P(gate) = qG x f x VL where VL is the internal-logic-supply voltage (+5V), and qG is the sum of the gate-charge values for low-side and highside switches. For matched MOSFETs, qG is twice the data-sheet value of an individual MOSFET. If VOUT is set to less than 4.5V, replace VL in this equation with VBATT. In this case, efficiency can be improved by connecting VL to an efficient 5V source, such as the system +5V supply. P(diode) = diode - conduction losses = ILOAD x VFWD x t D x f ______________________________________________________________________________________ Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers P(cap) = input capacitor ESR loss = (IRMS )2 x RESR where IRMS is the input ripple current as calculated in the Design Procedure and Input Capacitor Value sections. Light-Load Efficiency Considerations Under light loads, the PWM operates in discontinuous mode, where the inductor current discharges to zero at some point during the switching cycle. This makes the inductor current’s AC component high compared to the load current, which increases core losses and I2R losses in the output filter capacitors. For best light-load efficiency, use MOSFETs with moderate gate-charge levels, and use ferrite, MPP, or other low-loss core material. Avoid powdered-iron cores; even Kool-Mu (aluminum alloy) is not as good as ferrite. PC Board Layout Considerations Good PC board layout is required in order to achieve specified noise, efficiency, and stability performance. The PC board layout artist must be given explicit instructions, preferably a pencil sketch showing the placement of power-switching components and highcurrent routing. See the PC board layout in the MAX1630 Evaluation Kit manual for examples. A ground plane is essential for optimum performance. In most applications, the circuit will be located on a multilayer board, and full use of the four or more copper layers is recommended. Use the top layer for high-current connections, the bottom layer for quiet connections (REF, SS, GND), and the inner layers for an uninterrupted ground plane. Use the following step-by-step guide: 1) Place the high-power components (Figure1, C1, C3, Q1, Q2, D1, L1, and R1) first, with any grounded connections adjacent. Priority 1: Minimize current-sense resistor trace lengths and ensure accurate current sensing with Kelvin connections (Figure 7). Priority 2: Minimize ground trace lengths in the high-current paths (discussed below). Priority 3: Minimize other trace lengths in the highcurrent paths. Use >5mm-wide traces C IN to high-side MOSFET drain: 10mm max length Rectifier diode cathode to low-side MOSFET: 5mm max length LX node (MOSFETs, rectifier cathode, inductor): 15mm max length Ideally, surface-mount power components are butted up to one another with their ground terminals almost touching. These high-current grounds are then connected to each other with a wide filled zone of top-layer copper so they don’t go through vias. The resulting toplayer “sub-ground-plane” is connected to the normal inner-layer ground plane at the output ground terminals, which ensures that the IC’s analog ground is sensing at the supply’s output terminals without interference from IR drops and ground noise. Other highcurrent paths should also be minimized, but focusing primarily on short ground and current-sense connections eliminates about 90% of all PC board layout problems (see the PC board layouts in the MAX1630 Evaluation Kit manual for examples). 2) Place the IC and signal components. Keep the main switching nodes (LX nodes) away from sensitive analog components (current-sense traces and REF capacitor). Place the IC and analog components on the opposite side of the board from the powerswitching node. Important: the IC must be no farther than 10mm from the current-sense resistors. Keep the gate-drive traces (DH_, DL_, and BST_) shorter than 20mm and route them away from CSH_, CSL_, and REF. 3) Use a single-point star ground where the input ground trace, power ground (sub-ground-plane), and normal ground plane meet at the supply’s output ground terminal. Connect both IC ground pins and all IC bypass capacitors to the normal ground plane. HIGH CURRENT PATH SENSE RESISTOR MAX1630 Figure 7. Kelvin Connections for the Current-Sense Resistors ______________________________________________________________________________________ 23 MAX1630–MAX1635 where tD is the diode-conduction time (120ns typical) and VFWD is the forward voltage of the diode. This power is dissipated in the MOSFET body diode if no external Schottky diode is used. MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers _______________________________________________________________________________Application Circuits * TO +3.3V OUTPUT * TO +5V OUTPUT INPUT +5.2V TO +24V C3 C4 10Ω 4.7µF 0.1µF ON/OFF 23 22 6 0.1µF 21 +5V ALWAYS ON 4.7µF SHDN V+ SYNC VL 5 25 2.7µF Q1 27 VDD 12OUT BST3 BST5 4 +12V AT 120mA 2.2µF 18 16 Q3 DH3 DH5 LX3 LX5 17 DL5 19 Q4 0.1µF C1 +3.3V OUTPUT (3A) L2 * R1 T1 1:4 0.1µF 26 0.1µF Q2 24 MAX1630 MAX1633 DL3 PGND 1N5819 1 2 3 3V ON/OFF 28 5V ON/OFF 7 CSH3 CSH5 CSL3 CSL5 FB3 FB5 RUN/ON3 SEQ REF 10 1N5819 14 13 12 15 9 +2.5V REF 1µF RESET R1 = R2 = 20mΩ L2 = 10µH SUMIDA CDRH125-100 T1 = 10µH 1:4 TRANSFORMER TRANSPOWER TECHNOLOGIES TTI-5902 Q1–Q4 = Si4410DY or IRF7413 C1 = 3 x 220µF 10V SPRAGUE 594D227X0010D2T C2 = 2 x 220µF 10V SPRAGUE 594D227X0010D2T C3 = C4 = 2 x 10µF 30V SANYO OS-CON 30SC10M +5V OUTPUT (3A) * 20 TIME/ON5 SKIP R2 0.1µF 11 POWER-GOOD GND 8 *VL DIODES AND OUTPUT SCHOTTKY DIODES REQUIRED FOR THE MAX1630 ONLY (SEE OUTPUT OVERVOLTAGE PROTECTION AND OUTPUT UNDERVOLTAGE SHUTDOWN PROTECTION SECTIONS). Figure 8. Triple-Output Application for Low-Voltage Batteries (MAX1630/MAX1633) 24 ______________________________________________________________________________________ C2 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers * TO +3.3V OUTPUT * TO +5V OUTPUT INPUT +6.5V TO +28V C3 C4 10Ω 4.7µF 0.1µF +5V ALWAYS ON ON/OFF 23 22 6 0.1µF 21 4.7µF SHDN V+ SYNC VL 12OUT VDD D1 25 Q1 27 L1 +3.3V OUTPUT (3A) C1 * R1 BST5 BST3 DH5 DH3 +12V AT 120mA 4 2.2µF 5 D2 D5 18 2.2µF 16 Q3 R2 0.1µF 26 0.1µF 24 Q2 MAX1632 MAX1635 LX3 DL3 1N5819 1 2 3 3V ON/OFF 28 5V ON/OFF 7 LX5 17 DL5 19 PGND 20 CSH3 CSH5 CSL3 CSL5 FB3 FB5 RUN/ON3 SEQ REF SKIP 10 T2 0.1µF 1:2.2 +5V OUTPUT (3A) * C2 Q4 1N5819 14 13 12 15 9 +2.5V REF 1µF TIME/ON5 RESET R1 = R2 = 20mΩ L1 = 10µH SUMIDA CDRH125-100 T2 = 10µH 1:2.2 TRANSFORMER TRANSPOWER TECHNOLOGIES TTI-5870 Q1–Q4 = Si4410DY or IRF7413 C1 = 3 x 220µF 10V SPRAGUE 594D227X0010D2T C2 = 2 x 220µF 10V SPRAGUE 594D227X0010D2T C3 = C4 = 2 x 10µF 30V SANYO OS-CON 30SC10M 0.1µF 11 POWER-GOOD GND 8 *VL DIODES AND OUTPUT SCHOTTKY DIODES REQUIRED FOR THE MAX1632 ONLY (SEE OUTPUT OVERVOLTAGE PROTECTION AND OUTPUT UNDERVOLTAGE SHUTDOWN PROTECTION SECTIONS). Figure 9. Triple-Output Application for High-Voltage Batteries (MAX1632/MAX1635) ______________________________________________________________________________________ 25 MAX1630–MAX1635 _____________________________________________________________Application Circuits (continued) MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers _____________________________________________________________Application Circuits (continued) ON/OFF * * INPUT +6V TO +24V 5V ALWAYS ON C3 10Ω 22 0.1µF 23 5 21 4.7µF V+ SHDN SECFB VL 4.7µF 18 16 Q1 2.5V OUTPUT 0.1µF L1 R1 BST5 BST3 DH5 DH3 Q2 C1 1N5819 0.1µF 27 0.1µF Q3 0.1µF LX3 26 17 LX5 * 25 19 20 14 MAX1631 24 DL3 MAX1634 DL5 L2 R2 Q4 C2 1N5819 PGND CSH5 CSH3 1 13 CSL5 CSL3 2 0Ω OPEN 12 OPEN ON/OFF ON/OFF 7 28 FB3 FB5 RESET TIME/ON5 SKIP RUN/ON3 STEER 8 GND REF SYNC 9 6 3 11 RESET OUTPUT 0Ω 10 4 SEQ 15 1µF R1 = R2 = 15mΩ L1 = L2 = 6.8µH SUMIDA CDRH 127-6R8MC Q1 = Q4 = Si4410DY or 1RF7413 C1 = C2 = 2X SANYO OS-CON 10 SA220M C3 = 4X SANYO OS-CON 30SC10M *VL DIODES AND OUTPUT SCHOTTKY DIODES REQUIRED FOR THE MAX1631 ONLY (SEE OUTPUT OVERVOLTAGE PROTECTION AND OUTPUT UNDERVOLTAGE SHUTDOWN PROTECTION SECTIONS). Figure 10. Dual, 4A, Notebook Computer Power Supply 26 +3.3V OUTPUT ______________________________________________________________________________________ * Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers TOP VIEW CSH3 1 28 RUN/ON3 CSH3 1 28 RUN/ON3 CSL3 2 27 DH3 CSL3 2 27 DH3 FB3 3 26 LX3 FB3 3 26 LX3 12OUT 4 VDD 5 SYNC 6 TIME/ON5 7 MAX1630 MAX1632 MAX1633 MAX1635 25 BST3 STEER 4 24 DL3 SECFB 5 23 SHDN SYNC 6 25 BST3 MAX1631 MAX1634 24 DL3 23 SHDN 22 V+ TIME/ON5 7 22 V+ GND 8 21 VL GND 8 21 VL REF 9 20 PGND REF 9 20 PGND SKIP 10 19 DL5 SKIP 10 19 DL5 RESET 11 18 BST5 RESET 11 18 BST5 FB5 12 17 LX5 16 DH5 CSL5 13 16 DH5 15 SEQ CSH5 14 15 SEQ FB5 12 17 LX5 CSL5 13 CSH5 14 SSOP SSOP _______________________________________________________________Selector Guide DEVICE AUXILIARY OUTPUT SECONDARY FEEDBACK OVER/UNDERVOLTAGE PROTECTION MAX1630 12V Linear Regulator Feeds into the 3.3V SMPS Yes MAX1631 None (SECFB input) Selectable (STEER pin) Yes MAX1632 12V Linear Regulator Feeds into the 5V SMPS Yes MAX1633 12V Linear Regulator Feeds into the 3.3V SMPS No MAX1634 None (SECFB input) Selectable (STEER pin) No MAX1635 12V Linear Regulator Feeds into the 5V SMPS No ______________________________________________________________________________________ 27 MAX1630–MAX1635 ________________________________________________________________________________Pin Configurations __Ordering Information (continued) PART TEMP. RANGE PIN-PACKAGE MAX1631CAI 0°C to +70°C 28 SSOP MAX1631EAI -40°C to +85°C 28 SSOP MAX1632CAI 0°C to +70°C 28 SSOP MAX1632EAI -40°C to +85°C 28 SSOP MAX1633CAI 0°C to +70°C 28 SSOP MAX1633EAI -40°C to +85°C 28 SSOP MAX1634CAI 0°C to +70°C 28 SSOP MAX1634EAI -40°C to +85°C 28 SSOP MAX1635CAI 0°C to +70°C 28 SSOP MAX1635EAI -40°C to +85°C 28 SSOP ________________________________________________________Package Information SSOP.EPS MAX1630–MAX1635 Multi-Output, Low-Noise Power-Supply Controllers for Notebook Computers Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 28 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 1997 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.