TI TPA0222PWP

TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
D
D
D
D
D
D
D
D
D
D
D
Compatible With PC 99 Desktop Line-Out
Into 10-kΩ Load
Compatible With PC 99 Portable Into 8-Ω
Load
Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
2-W/Ch Output Power Into 3-Ω Load
Input MUX Select Terminal
PC-Beep Input
Depop Circuitry
Stereo Input MUX
Fully Differential Input
Low Supply Current and Shutdown Current
Surface-Mount Power Packaging
24-Pin TSSOP PowerPAD
PWP PACKAGE
(TOP VIEW)
GND
GAIN0
GAIN1
LOUT+
LLINEIN
LHPIN
PVDD
RIN
LOUT–
LIN
BYPASS
GND
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
GND
RLINEIN
SHUTDOWN
ROUT+
RHPIN
VDD
PVDD
HP/LINE
ROUT–
SE/BTL
PC-BEEP
GND
description
The TPA0222 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-Ω loads. This device minimizes the number of
external components needed, simplifying the design, and freeing up board space for other features. When
driving 1 W into 8-Ω speakers, the TPA0222 has less than 0.5% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in
the speakers.
Amplifier gain is internally configured and controlled by two terminals (GAIN0 and GAIN1). BTL gain settings
of 2, 6, 12, and 24 V/V are provided, while SE gain is always configured as 1 V/V for headphone drive. An internal
input MUX allows two sets of stereo inputs to the amplifier. The HP/LINE terminal allows the user to select which
MUX input is active regardless of whether the amplifier is in SE or BTL mode. In notebook applications, where
internal speakers are driven as BTL and the line outputs (often headphone drive) are required to be SE, the
TPA0222 automatically switches into SE mode when the SE/BTL input is activated, and reduces the gain to 1
V/V.
The TPA0222 consumes only 18 mA of supply current during normal operation. A miserly shutdown mode
reduces the supply current to less than 150 µA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are truly realized in multilayer PCB
applications. This allows the TPA0222 to operate at full power into 8-Ω loads at an ambient temperature of 85°C.
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments Incorporated.
Copyright  1999, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
1
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
functional block diagram
RHPIN
RLINEIN
R
MUX
–
ROUT+
+
RIN
–
ROUT–
+
PC-BEEP
GAIN0
GAIN1
SE/BTL
PCBeep
Gain/
MUX
Control
Depop
Circuitry
LLINEIN
PVDD
VDD
BYPASS
SHUTDOWN
GND
HP/LINE
LHPIN
Power
Management
L
MUX
–
LOUT+
+
LIN
–
LOUT–
+
2
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• DALLAS, TEXAS 75265
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
AVAILABLE OPTIONS
TA
PACKAGED DEVICE
TSSOP†
(PWP)
– 40°C to 85°C
TPA0222PWP
† The PWP package is available taped and
reeled. To order a taped and reeled part, add
the suffix R to the part number (e.g.,
TPA0222PWPR).
Terminal Functions
TERMINAL
NAME
NO.
I/O
DESCRIPTION
BYPASS
11
GAIN0
2
I
Bit 0 of gain control
GAIN1
3
I
Bit 1 of gain control
GND
Tap to voltage divider for internal mid-supply bias generator
1, 12,
13, 24
Ground connection for circuitry. Connected to the thermal pad
LHPIN
6
I
Left channel headphone input, selected when SE/BTL is held high
LIN
10
I
Common left input for fully differential input. AC ground for single-ended inputs
LLINEIN
5
I
Left channel line input, selected when SE/BTL is held low
LOUT+
4
O
Left channel positive output in BTL mode and positive output in SE mode
LOUT–
9
O
Left channel negative output in BTL mode and high-impedance in SE mode
PC-BEEP
14
I
The input for PC Beep mode. PC-BEEP is enabled when a > 1-V (peak-to-peak) square wave is input
to PC-BEEP or PCB ENABLE is high.
HP/LINE
17
I
HP/LINE is the input MUX control input. When the HP/LINE terminal is held high, the headphone inputs
(LHPIN or RHPIN [6, 20]) are active. When the HP/LINE terminal is held low, the line BTL inputs (LLINEIN
or RLINEIN [5, 23]) are active.
7, 18
I
Power supply for output stage
PVDD
RHPIN
20
I
Right channel headphone input, selected when SE/BTL is held high
RIN
8
I
Common right input for fully differential input. AC ground for single-ended inputs
RLINEIN
23
I
Right channel line input, selected when SE/BTL is held low
ROUT+
21
O
Right channel positive output in BTL mode and positive output in SE mode
ROUT–
16
O
Right channel negative output in BTL mode and high-impedance in SE mode
SHUTDOWN
22
I
Places entire IC in shutdown mode when held low, except PC-BEEP remains active
SE/BTL
15
I
Hold SE/BTL low for BTL mode and hold high for SE mode.
VDD
19
I
Analog VDD input supply. This terminal needs to be isolated from PVDD to achieve highest performance.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
3
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)†
Supply voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VDD +0.3 V
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 85°C
Operating junction temperature range, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 150°C
Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
† Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
TA ≤ 25°C
2.7 W‡
PACKAGE
PWP
DERATING FACTOR
21.8 mW/°C
TA = 70°C
1.7 W
TA = 85°C
1.4 W
‡ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.
recommended operating conditions
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Supply voltage, VDD
High level input voltage,
High-level
voltage VIH
MIN
MAX
4.5
5.5
SE/BTL, HP/LINE
4
SHUTDOWN
2
SE/BTL, HP/LINE
UNIT
V
V
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Low level input voltage,
Low-level
voltage VIL
SHUTDOWN
0.8
Operating free-air temperature, TA
– 40
85
V
°C
electrical characteristics at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise
noted)
PARAMETER
TEST CONDITIONS
|VOS|
Output offset voltage (measured differentially)
PSRR
Power supply rejection ratio
VI = 0,
AV = –2 V/V
VDD = 4.9 V to 5.1 V
|IIH|
High-level input current
VDD = 5.5 V,
VI = VDD
|IIL|
Low-level input current
VDD = 5.5 V,
VI = 0 V
IDD
Supply current
IDD(SD)
Supply current, shutdown mode
4
MIN
TYP
25
77
BTL mode
18
SE mode
9
150
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
MAX
UNIT
mV
dB
900
nA
900
nA
mA
300
µA
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
operating characteristics, VDD = 5 V, TA = 25°C, RL = 8 Ω, Gain = –2 V/V, BTL mode
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PARAMETER
PO
Output power
THD + N
Total harmonic distortion plus noise
BOM
Maximum output power bandwidth
Supply ripple rejection ratio
SNR
Signal-to-noise ratio
Vn
Noise output voltage
ZI
Input impedance
TEST CONDITIONS
THD = 1%,
RL = 4 Ω
f = 1 kHz,
PO = 1 W,
THD = 5%
f = 20 Hz to 15 kHz
f = 1 kHz,
CB = 0.47 µF
CB = 0.47 µ
µF,,
f = 20 Hz to 20 kHz
BTL mode
MIN
TYP
MAX
UNIT
1.9
W
0.5%
>15
kHz
68
dB
105
dB
BTL mode
16
SE mode
30
µV RMS
See Table 1
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
THD+N
Total harmonic distortion plus noise
vs Output power
1, 4–7, 10–13,
16–19, 21
vs Frequency
2, 3, 8, 9, 14,
15, 20, 22
vs Output voltage
Vn
SNR
Output noise voltage
vs Bandwidth
24
Supply ripple rejection ratio
vs Frequency
25, 26
Crosstalk
vs Frequency
27–29
Shutdown attenuation
vs Frequency
30
Signal-to-noise ratio
vs Bandwidth
31
Closed loop respone
PO
PD
23
32–35
Output power
Power dissipation
vs Load resistance
36, 37
vs Output power
38, 39
vs Ambient temperature
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40
5
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
AV = 2 V/V
f = 1 kHz
BTL
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
RL = 4 Ω
1%
RL = 8 Ω
RL = 3 Ω
0.1%
0.01%
0.5 0.75
1
1.25 1.5 1.75
2
2.25 2.5 2.75
PO = 1.75 W
RL = 3 Ω
BTL
1%
AV = –24 V/V
AV = –12 V/V
AV = –2 V/V
0.1%
AV = –6 V/V
0.01%
20
3
PO – Output Power – W
10%
RL = 3 Ω
AV = –2 V/V
BTL
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
1%
PO = 1.0 W
PO = 0.5 W
0.1%
PO = 1.75 W
100
1k
10k 20k
f = 15 kHz
1%
f = 1 kHz
0.1%
f = 20 Hz
RL = 3 Ω
AV = –2 V/V
BTL
0.01%
0.01
f – Frequency – Hz
Figure 3
6
10k 20k
Figure 2
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
0.01%
20
1k
f – Frequency – Hz
Figure 1
10%
100
0.1
1
PO – Output Power – W
Figure 4
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
10
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
f = 15 kHz
1%
f = 1 kHz
f = 20 Hz
0.1%
0.01%
0.01
RL = 3 Ω
AV = –6 V/V
BTL
0.1
1
PO – Output Power – W
f = 15 kHz
1%
f = 1 kHz
f = 20 Hz
0.1%
0.01%
0.01
10
RL = 3 Ω
AV = –12 V/V
BTL
0.1
1
PO – Output Power – W
Figure 5
Figure 6
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
f = 15 kHz
1%
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
f = 1 kHz
f = 20 Hz
0.1%
0.01%
0.01
10
RL = 3 Ω
AV = –24 V/V
BTL
0.1
1
PO – Output Power – W
10
PO = 1.5 W
RL = 4 Ω
BTL
AV = –24 V/V
1%
AV = –12 V/V
0.1%
AV = –2 V/V
AV = –6 V/V
0.01%
20
100
1k
10k 20k
f – Frequency – Hz
Figure 7
Figure 8
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• DALLAS, TEXAS 75265
7
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
RL = 4 Ω
AV = –2 V/V
BTL
1%
0.1%
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
PO = 1.5 W
PO = 0.25 W
PO = 1.0 W
0.01%
20
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
100
1k
10k 20k
RL = 4 Ω
AV = –2 V/V
BTL
f = 15 kHz
1%
f = 1 kHz
0.1%
f = 20 Hz
0.01%
0.01
0.1
1
PO – Output Power – W
f – Frequency – Hz
Figure 9
Figure 10
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
f = 15 kHz
1%
f = 1 kHz
0.1%
f = 20 Hz
RL = 4 Ω
AV = –6 V/V
BTL
0.01%
0.01
0.1
1
PO – Output Power – W
10
f = 15 kHz
1%
f = 1 kHz
f = 20 Hz
0.1%
RL = 4 Ω
AV = –12 V/V
BTL
0.01%
0.01
Figure 11
8
10
0.1
1
PO – Output Power – W
Figure 12
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
10
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
f = 15 kHz
1%
f = 1 kHz
f = 20 Hz
0.1%
RL = 4 Ω
AV = –24 V/V
BTL
0.01%
0.01
0.1
1
PO – Output Power – W
10
RL = 8 Ω
AV = –2 V/V
BTL
1%
PO = 0.25 W
0.1%
PO = 1.0 W
0.01%
PO = 0.5 W
0.001%
20
100
Figure 13
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
10%
PO = 1 W
RL = 8 Ω
BTL
AV = –24 V/V
1%
AV = –12 V/V
0.1%
AV = –2 V/V
0.01%
AV = –6 V/V
100
1k
10k 20k
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10k 20k
Figure 14
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
0.001%
20
1k
f – Frequency – Hz
RL = 8 Ω
AV = –2 V/V
BTL
1%
f = 15 kHz
f = 1 kHz
0.1%
f = 20 Hz
0.01%
0.001%
0.01
f – Frequency – Hz
Figure 15
0.1
1
PO – Output Power – W
10
Figure 16
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• DALLAS, TEXAS 75265
9
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL = 8 Ω
AV = –6 V/V
BTL
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
f = 15 kHz
1%
f = 1 kHz
0.1%
f = 20 Hz
0.01%
0.01
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
0.1
1
PO – Output Power – W
f = 15 kHz
1%
f = 1 kHz
0.1%
f = 20 Hz
RL = 8 Ω
AV = –12 V/V
BTL
0.01%
0.01
10
0.1
1
PO – Output Power – W
Figure 17
Figure 18
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
f = 15 kHz
1%
f = 1 kHz
f = 20 Hz
0.1%
RL = 8 Ω
AV = –24 V/V
BTL
0.01%
0.01
0.1
1
PO – Output Power – W
10
RL = 32 Ω
AV = –1 V/V
SE
1%
PO = 25 mW
0.1%
PO = 75 mW
PO = 50 mW
0.01%
0.001%
20
100
1k
f – Frequency – Hz
Figure 19
10
10
Figure 20
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10k 20k
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
RL = 32 Ω
AV = –1 V/V
SE
THD+N –Total Harmonic Distortion + Noise
THD+N –Total Harmonic Distortion + Noise
10%
1%
f = 15 kHz
0.1%
f = 1 kHz
0.01%
f = 20 Hz
0.001%
0.01
0.1
PO – Output Power – W
RL = 10 kΩ
AV = –1 V/V
SE
1%
0.1%
VO = 1 VRMS
0.01%
0.001%
20
1
100
Figure 21
OUTPUT NOISE VOLTAGE
vs
BANDWIDTH
100
RL = 10 kΩ
AV = –1 V/V
SE
1%
0.1%
f = 20 Hz
f = 15 kHz
0.01%
f = 1 kHz
0.6
0.8
1
1.2
1.4 1.6
80
70
AV = –24 V/V
60
50
AV = –12 V/V
40
30
AV = –6 V/V
20
10
0.001%
0.4
VDD = 5 V
RL = 4Ω
90
V n – Output Noise Voltage – µ V
THD+N –Total Harmonic Distortion + Noise
10%
0.2
10k 20k
Figure 22
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT VOLTAGE
0
1k
f – Frequency – Hz
1.8
2
AV = –2 V/V
0
10
VO – Output Voltage – VRMS
100
1k
10k
BW - Bandwidth - Hz
Figure 23
Figure 24
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11
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
0
RL = 8 Ω
CB = 0.47 µF
BTL
–20
Supply Ripple Rejection Ratio – dB
Supply Ripple Rejection Ratio – dB
0
–40
–60
AV = –24 V/V
–80
–100
AV = –2 V/V
–120
20
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
100
1k
–20
RL = 32 Ω
CB = 0.47 µF
SE
–40
AV = –1 V/V
–60
–80
–100
–120
20
10k 20k
100
1k
f – Frequency – Hz
f – Frequency – Hz
Figure 25
Figure 26
CROSSTALK
vs
FREQUENCY
CROSSTALK
vs
FREQUENCY
0
0
PO = 1 W
RL = 8 Ω
Av = –2 V/V
BTL
–20
–40
Crosstalk – dB
Crosstalk – dB
–20
10k 20k
–60
–80
PO = 1 W
RL = 8 Ω
Av = –24 V/V
BTL
–40
–60
LEFT TO RIGHT
–80
LEFT TO RIGHT
–100
–100
RIGHT TO LEFT
RIGHT TO LEFT
–120
20
100
1k
10k 20k
–120
20
f – Frequency – Hz
1k
f – Frequency – Hz
Figure 27
12
100
Figure 28
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10k 20k
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
CROSSTALK
vs
FREQUENCY
SHUTDOWN ATTENUATION
vs
FREQUENCY
0
VI = 1 VRMS
–20
RL = 10 kΩ, SE
–60
LEFT TO RIGHT
–80
–40
Attenuation – dB
–40
–60
RL = 32 Ω, SE
–80
–100
RL = 8 Ω, BTL
–100
RIGHT TO LEFT
–120
20
100
1k
–120
20
10k 20k
100
f – Frequency – Hz
1k
10k 20k
f – Frequency – Hz
Figure 29
Figure 30
SIGNAL-TO-NOISE RATIO
vs
BANDWIDTH
120
115
SNR – Signal-To-Noise Ratio – dB
Crosstalk – dB
–20
0
VO = 1 VRMS
RL = 10 kΩ
Av = –1 V/V
SE
PO = 1 W
RL = 8 Ω
BTL
AV = –24 V/V
AV = –12 V/V
110
105
100
AV = –2 V/V
95
AV = –6 V/V
90
85
80
20
100
1k
10k 20k
BW – Bandwidth – Hz
Figure 31
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13
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE
180°
10
7.5
Gain
Gain – dB
2.5
Phase
0°
0
–2.5
–5
RL = 8 Ω
AV = –2 V/V
BTL
φ m – Phase Margin
90°
5
–90°
–7.5
–10
10
100
1k
10k
100k
1M
–180°
2M
f – Frequency – Hz
Figure 32
CLOSED LOOP RESPONSE
180°
30
25
Gain
Gain – dB
15
Phase
0°
10
5
0
RL = 8 Ω
AV = –6 V/V
BTL
φ m – Phase Margin
90°
20
–90°
–5
–10
10
–180°
100
1k
10k
100k
1M
f – Frequency – Hz
Figure 33
14
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2M
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE
25
180°
RL = 8 Ω
AV = –12 V/V
BTL
90°
20
Gain – dB
Gain
15
Phase
0°
10
5
φ m – Phase Margin
30
–90°
0
–5
–10
10
–180°
100
1k
10k
100k
1M
2M
f – Frequency – Hz
Figure 34
CLOSED LOOP RESPONSE
30
25
180°
RL = 8 Ω
AV = –24 V/V
BTL
Gain
Gain – dB
15
Phase
0°
10
5
φ m – Phase Margin
90°
20
–90°
0
–5
–10
10
–180°
100
1k
10k
100k
1M
2M
f – Frequency – Hz
Figure 35
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15
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
TYPICAL CHARACTERISTICS
OUTPUT POWER
vs
LOAD RESISTANCE
OUTPUT POWER
vs
LOAD RESISTANCE
3.5
3
AV = –1 V/V
SE
1250
PO– Output Power – mW
PO – Output Power – W
1500
AV = –2 V/V
BTL
2.5
2
10% THD+N
1.5
1
1% THD+N
0.5
1000
750
10% THD+N
500
250
1% THD+N
0
0
8
16
24
32
40
48
RL – Load Resistance – Ω
56
0
64
0
48
16
24
32
40
RL – Load Resistance – Ω
8
Figure 36
POWER DISSIPATION
vs
OUTPUT POWER
1.8
0.4
3Ω
0.35
1.4
1.2
PD – Power Dissipation – W
PD – Power Dissipation – W
1.6
4Ω
1
0.8
0.6
8Ω
0.4
f = 1 kHz
BTL
Each Channel
0.2
0.5
1.5
1
PO – Output Power – W
2
0.3
4Ω
0.25
0.2
0.15
8Ω
0.1
2.5
0
0
f = 1 kHz
SE
Each Channel
32 Ω
0.05
0.1
Figure 38
16
64
Figure 37
POWER DISSIPATION
vs
OUTPUT POWER
0
0
56
0.4
0.5
0.6
0.2
0.3
PO – Output Power – W
Figure 39
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0.7
0.8
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
POWER DISSIPATION
vs
AMBIENT TEMPERATURE
7
ΘJA1 = 45.9°C/W
ΘJA2 = 45.2°C/W
ΘJA3 = 31.2°C/W
ΘJA4 = 18.6°C/W
ΘJA4
PD – Power Dissipation – W
6
5
4
ΘJA3
3
ΘJA1,2
2
1
0
–40 –20
20 40 60 80 100 120 140 160
0
TA – Ambient Temperature – °C
Figure 40
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17
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 41)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements (< 2 mm) of many of
today’s advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.
DIE
Side View (a)
Thermal
Pad
DIE
End View (b)
Bottom View (c)
Figure 41. Views of Thermally Enhanced PWP Package
Figure 42 and Figure 43 are schematic diagrams of typical notebook computer application circuits.
18
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TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
Right CIRHP
Head– 0.47 µF
phone
Input
Signal
20
CIRLINE
Right 0.47 µF
Line
Input
Signal
23
RHPIN
RLINEIN
R
MUX
–
+
8
ROUT+
21
RIN
CRIN
0.47 µF
PC BEEP
14
Input
Signal CPCB
0.47 µF
COUTR
330 µF
PC–BEEP
–
+
PC–
Beep
ROUT–
16
VDD
1 kΩ
100 kΩ
2
3
GAIN0
15
SE/BTL
Left CILHP
Head– 0.47 µF 17
phone
Input
Signal
6
CILLINE
Left 0.47 µF
Line
Input
Signal
5
10
GAIN1
Gain/
MUX
Control
Depop
Circuitry
Power
Management
HP/LINE
LHPIN
LLINEIN
PVDD
18
VDD
19
BYPASS
SHUT–
DOWN
11
GND
L
MUX
See Note A
VDD
CSR
0.1 µF
VDD
CSR
0.1 µF
22
To
System
Control
–
+
LOUT+
4
–
+
LOUT–
9
CBYP
0.47 µF
1 kΩ
1,12,
13,24
COUTL
330 µF
LIN
CLIN
0.47 µF
100 kΩ
NOTE A:
A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower–frequency noise signals, a larger
electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 42. Typical TPA0222 Application Circuit Using Single-Ended Inputs and Input MUX
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19
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
N/C
20
CIRIN–
Right 0.47 µF
Negative
23
Differential
Input
Signal
CIRIN+
Right 0.47 µF
Positive
8
Differential
Input
Signal
PC BEEP
14
Input
Signal C
PCB
0.47 µF
RHPIN
RLINEIN
–
+
21
COUTR
330 µF
PC–BEEP
–
+
PC–
Beep
ROUT–
16
VDD
1 kΩ
100 kΩ
GAIN0
15
SE/BTL
GAIN1
Gain/
MUX
Control
Depop
Circuitry
Power
Management
HP/LINE
N/C
6
LHPIN
5
LLINEIN
CILIN–
Left
0.47 µF
Negative
Differential
Input
Signal
CILIN+
Left 0.47 µF
Positive
10
Differential
Input
Signal
ROUT+
RIN
2
3
17
R
MUX
PVDD
18
VDD
19
BYPASS
SHUT–
DOWN
11
GND
L
MUX
See Note A
VDD
CSR
0.1 µF
VDD
CSR
0.1 µF
22
To
System
Control
–
+
LOUT+
4
–
+
LOUT–
9
CBYP
0.47 µF
1 kΩ
1,12,
13,24
COUTL
330 µF
LIN
100 kΩ
NOTE A:
A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower–frequency noise signals, a larger
electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 43. Typical TPA0222 Application Circuit Using Differential Inputs
20
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
gain setting via GAIN0 and GAIN1 inputs
The gain of the TPA0222 is set by two input terminals, GAIN0 and GAIN1.
Table 1. Gain Settings
GAIN0
GAIN1
SE/BTL
AV
0
0
0
–2 V/V
0
1
0
–6 V/V
1
0
0
–12 V/V
1
1
0
–24 V/V
X
X
1
–1 V/V
The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This
causes the input impedance, ZI, to be dependant on the gain setting. The actual gain settings are controlled
by ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input
impedance will shift by 30% due to shifts in the actual resistance of the input impedance.
For design purposes, the input network (discussed in the next section) should be designed assuming an input
impedance of 10 kΩ, which is the absolute minimum input impedance of the TPA0222. At the higher gain
settings, the input impedance could increase as high as 115 kΩ.
input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the –3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.
ZF
C
IN
Input
Signal
ZI
R
The typical input resistance at each gain setting is given in the table below:
Av
ZI
–24 V/V
14 kΩ
–12 V/V
26 kΩ
–6 V/V
45.5 kΩ
–2 V/V
91 kΩ
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21
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
The –3 dB frequency can be calculated using equation 1:
ƒ –3 dB
+
ǒø Ǔ
1
2p C R R
(1)
I
If the filter must be more accurate, the value of the capacitor should be increased while the value of the resistor
to ground should be decreased. In addition, the order of the filter could be increased.
input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, ZI, form a
high-pass filter with the corner frequency determined in equation 2.
–3 dB
+ 2 p Z1 C
f c(highpass)
(2)
I I
fc
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where ZI is 710 kΩ and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.
CI
+ 2 p Z1 fc
(3)
I
In this example, CI is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at VDD/2, which is likely higher
than the source dc level. Note that it is important to confirm the capacitor polarity in the application.
22
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TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
power supply decoupling, CS
The TPA0222 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 µF placed as close as possible to the device VDD lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near
the audio power amplifier is recommended.
midrail bypass capacitor, CBYP
The midrail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor, CBYP, values of 0.47 µF to 1 µF ceramic or tantalum low-ESR capacitors are recommended
for the best THD and noise performance.
output coupling capacitor, CC
In the typical single-supply SE configuration, an output coupling capacitor (CC) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.
–3 dB
f c(high)
+ 2 p R1 C
(4)
L C
fc
The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher, degrading the bass response. Large values of CC are required to pass low
frequencies into the load. Consider the example where a CC of 330 µF is chosen and loads vary from 3 Ω,
4 Ω, 8 Ω, 32 Ω, 10 kΩ, to 47 kΩ. Table 2 summarizes the frequency response characteristics of each
configuration.
POST OFFICE BOX 655303
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23
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode
RL
CC
330 µF
Lowest Frequency
3Ω
4Ω
330 µF
120 Hz
60 Hz
161 Hz
8Ω
330 µF
32 Ω
330 µF
15 Hz
10,000 Ω
330 µF
0.05 Hz
47,000 Ω
330 µF
0.01 Hz
As Table 2 indicates, most of the bass response is attenuated into a 4-Ω load, an 8-Ω load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.
using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.
bridged-tied load versus single-ended mode
Figure 44 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0222 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration, but initially consider power to the load. The differential drive to the speaker
means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the
voltage swing on the load as compared to a ground referenced load. Plugging 2 × VO(PP) into the power
equation, where voltage is squared, yields 4× the output power from the same supply rail and load impedance
(see equation 5).
V (rms)
+ O(PP)
2 Ǹ2
Power
+
V
24
V (rms)
(5)
2
RL
POST OFFICE BOX 655303
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TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
VDD
VO(PP)
2x VO(PP)
RL
VDD
–VO(PP)
Figure 44. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-Ω speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement —
which is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 45. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to 1000 µF)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
fc
+ 2 p R1 C
(6)
L C
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD
–3 dB
VO(PP)
CC
RL
VO(PP)
fc
Figure 45. Single-Ended Configuration and Frequency Response
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
25
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4× the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.
single-ended operation
In SE mode (see Figure 44 and Figure 45), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier’s gain to 1 V/V.
input MUX operation
The input MUX allows two separate inputs to be applied to the amplifier. This allows the designer to choose
which input is active independent of the state of the SE/BTL terminal. When the HP/LINE terminal is held high,
the headphone inputs are active. When the HP/LINE terminal is held low, the line BTL inputs are active.
BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
VDD. The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 46).
IDD
VO
IDD(avg)
V(LRMS)
Figure 46. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
26
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
Efficiency of a BTL amplifier
+ PP L
(7)
SUP
Where:
PL
+ VLRrms
L
and P SUP
2
, and V LRMS
+ VDD IDDavg
+ VǸ2P ,
2
therefore, P L
+ ŕ
1
I DDavg
and
p
p
0
+ 2VRP
VP
RL
L
sin(t) dt
+ 1p
VP
RL
[cos(t)]
p
0
+ p2VRP
L
Therefore,
P SUP
+ 2 VpDDR VP
L
substituting PL and PSUP into equation 7,
2
Efficiency of a BTL amplifier
Where:
VP
+2 V
h BTL
+p
PL = Power devilered to load
PSUP = Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL = Load resistance
VP = Peak voltage on BTL load
IDDavg = Average current drawn from
the power supply
VDD = Power supply voltage
ηBTL = Efficiency of a BTL amplifier
p VP
DD V P
+4V
p RL
+ Ǹ2 PL RL
Therefore,
VP
2 RL
DD
Ǹ
2 PL RL
4 V DD
(8)
Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-Ω loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 3. Efficiency Vs Output Power in 5-V 8-Ω BTL Systems
Output Power
(W)
Efficiency
(%)
Peak Voltage
(V)
Internal Dissipation
(W)
0.25
31.4
2.00
0.55
0.50
44.4
2.83
0.62
1.00
62.8
4.00
4.47†
0.59
1.25
70.2
† High peak voltages cause the THD to increase.
0.53
A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, VDD is in the denominator. This
indicates that as VDD goes down, efficiency goes up.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
27
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the internal
dissipated power at the average output power level must be used. From the TPA0222 data sheet, one can see
that when the TPA0222 is operating from a 5-V supply into a 3-Ω speaker 4-W peaks are available. Converting
watts to dB:
P dB
+ 10 Log PPW + 10 Log 41 WW + 6 dB
(9)
ref
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
6 dB – 15 dB = –9 dB (15 dB crest factor)
6 dB – 12 dB = –6 dB (12 dB crest factor)
6 dB – 9 dB = –3 dB (9 dB crest factor)
6 dB – 6 dB = 0 dB (6 dB crest factor)
6 dB – 3 dB = 3 dB (3 dB crest factor)
Converting dB back into watts:
PW
+ 10PdBń10 Pref
+ 63 mW (18 dB crest factor)
+ 125 mW (15 dB crest factor)
+ 250 mW (9 dB crest factor)
+ 500 mW (6 dB crest factor)
+ 1000 mW (3 dB crest factor)
+ 2000 mW (15 dB crest factor)
(10)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-Ω system, the internal dissipation in the TPA0222 and
maximum ambient temperatures is shown in Table 4.
Table 4. TPA0222 Power Rating, 5-V, 3-Ω, Stereo
28
PEAK OUTPUT POWER
(W)
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE
4
2 W (3 dB)
1.7
– 3°C
4
1000 mW (6 dB)
1.6
6°C
4
500 mW (9 dB)
1.4
24°C
4
250 mW (12 dB)
1.1
51°C
4
125 mW (15 dB)
0.8
78°C
4
63 mW (18 dB)
0.6
96°C
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
crest factor and thermal considerations (continued)
Table 5. TPA0222 Power Rating, 5-V, 8-Ω, Stereo
PEAK OUTPUT POWER
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE
2.5 W
1250 mW (3 dB crest factor)
0.55
100°C
2.5 W
1000 mW (4 dB crest factor)
0.62
94°C
2.5 W
500 mW (7 dB crest factor)
0.59
97°C
2.5 W
250 mW (10 dB crest factor)
0.53
102°C
The maximum dissipated power, PDmax, is reached at a much lower output power level for an 8-Ω load than for
a 3-Ω load. As a result, this simple formula for calculating PDmax may be used for an 8-Ω application:
2V 2
P Dmax
+ p2RDD
(11)
L
However, in the case of a 3-Ω load, the PDmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the PDmax formula
for a 3 Ω load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to ΘJA:
Θ JA
1
+ Derating1 Factor + 0.022
+ 45°CńW
(12)
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given ΘJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0222 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
T A Max
+ TJ Max * ΘJA PD
+ 150 * 45 (0.6 2) + 96°C (15 dB crest factor)
(13)
NOTE:
Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.
TableS 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0222 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Tables 4 and 5 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-Ω speakers dramatically increases the thermal performance by increasing amplifier
efficiency.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
29
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
SE/BTL operation
The ability of the TPA0222 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0222, two separate amplifiers drive OUT+ and OUT–. The SE/BTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT– and ROUT– (terminals 9 and 16). When
SE/BTL is held low, the amplifier is on and the TPA0222 is in the BTL mode. When SE/BTL is held high, the OUT–
amplifiers are in a high output impedance state, which configures the TPA0222 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). IDD is reduced by approximately one-half in SE mode. Control of the SE/BTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 47.
20
23
RHPIN
RLINEIN
R
MUX
–
+
8
ROUT+
21
RIN
VDD
–
+
ROUT–
16
100 kΩ
SE/BTL
15
COUTR
330 µF
1 kΩ
100 kΩ
Figure 47. TPA0222 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 100-kΩ/1-kΩ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kΩ
resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT– amplifier is
shut down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.
30
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
APPLICATION INFORMATION
PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is activated automatically. When the PC BEEP input is active,
both of the LINEIN and HPIN inputs are deselected and both the left and right channels are driven in BTL mode
with the signal from PC BEEP. The gain from the PC BEEP input to the speakers is fixed at 0.3 V/V and is
independent of the volume setting. When the PC BEEP input is deselected, the amplifier will return to the
previous operating mode and volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC
BEEP will take the device out of shutdown and output the PC BEEP signal, then return the amplifier to shutdown
mode.
The preferred input signal is a square wave or pulse train with an amplitude of 1 Vpp or greater. To be accurately
detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall times of less than 0.1 µs and a
minimum of 8 rising edges. When the signal is no longer detected, the amplifier will return to its previous
operating mode and volume setting.
If it is desired to ac-couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy
equation 14:
C
PCB
w 2p ƒ
1
PCB
W
(14)
(100 k )
The PC BEEP input can also be dc-coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no signal is present.
shutdown modes
The TPA0222 employs a shutdown mode of operation designed to reduce supply current, IDD, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, IDD = 150 µA. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.
Table 6. HP/LINE, SE/BTL, and Shutdown Functions
INPUTS†
HP/LINE
SE/BTL
AMPLIFIER STATE
SHUTDOWN
INPUT
OUTPUT
X
X
Low
X
Mute
Low
Low
High
Line
BTL
Low
High
High
Line
SE
High
Low
High
HP
BTL
High
High
High
† Inputs should never be left unconnected.
X = do not care
HP
SE
POST OFFICE BOX 655303
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31
TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 – NOVEMBER 1999
MECHANICAL DATA
PWP (R-PDSO-G**)
PowerPAD PLASTIC SMALL-OUTLINE PACKAGE
20-PIN SHOWN
0,30
0,19
0,65
20
0,10 M
11
Thermal Pad
(See Note D)
4,50
4,30
0,15 NOM
6,60
6,20
Gage Plane
1
10
0,25
A
0°– 8°
0,75
0,50
Seating Plane
0,15
0,05
1,20 MAX
PINS **
0,10
14
16
20
24
28
A MAX
5,10
5,10
6,60
7,90
9,80
A MIN
4,90
4,90
6,40
7,70
9,60
DIM
4073225/E 03/97
NOTES: A.
B.
C.
D.
All linear dimensions are in millimeters.
This drawing is subject to change without notice.
Body dimensions do not include mold flash or protrusions.
The package thermal performance may be enhanced by bonding the thermal pad to an external thermal plane. This pad is electrically
and thermally connected to the backside of the die and terminals 1, 12, 13, and 24. The dimensions of the thermal pad are
2.40 mm × 4.70 mm (maximum). The pad is centered on the bottom of the package.
E. Falls within JEDEC MO-153
PowerPAD is a trademark of Texas Instruments Incorporated.
32
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
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Copyright  1999, Texas Instruments Incorporated