TI TPA6120A2_07

TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
HIGH FIDELITY HEADPHONE AMPLIFIER
FEATURES
•
•
•
•
•
•
•
•
•
•
DESCRIPTION
80 mW into 600 Ω From a ±12-V Supply at
0.00014% THD + N
Current-Feedback Architecture
Greater than 120 dB of Dynamic Range
SNR of 120 dB
Output Voltage Noise of 5 µVrms at
Gain = 2 V/V
Power Supply Range: ±5 V to ±15 V
1300 V/µs Slew Rate
Differential Inputs
Independent Power Supplies for Low
Crosstalk
Short Circuit and Thermal Protection
The TPA6120A2 is a high fidelity audio amplifier built
on a current-feedback architecture. This high
bandwidth, extremely low noise device is ideal for
high performance equipment. The better than 120 dB
of dynamic range exceeds the capabilities of the
human ear, ensuring that nothing audible is lost due
to the amplifier. The solid design and performance of
the TPA6120A2 ensures that music, not the amplifier,
is heard.
Three key features make current-feedback amplifiers
outstanding for audio. The first feature is the high
slew rate that prevents odd order distortion
anomalies. The second feature is current-on-demand
at the output that enables the amplifier to respond
quickly and linearly when necessary without risk of
output distortion. When large amounts of output
power are suddenly needed, the amplifier can respond extremely quickly without raising the noise
floor of the system and degrading the signal-to-noise
ratio. The third feature is the gain-independent frequency response that allows the full bandwidth of the
amplifier to be used over a wide range of gain
settings. The excess loop gain does not deteriorate at
a rate of 20 dB/decade.
APPLICATIONS
•
•
•
•
•
Professional Audio Equipment
Mixing Boards
Headphone Distribution Amplifiers
Headphone Drivers
Microphone Preamplifiers
Filter and I/V Gain Stage
1/2 OPA4134
CF
2.7 nF
RF
1 kΩ
Stereo Hi−Fi
Headphone Driver
AUDIO DAC
LRCK
PCM
Audio
Data
Source
TPA6120A2
IOUT L−
−IN A
RF
OUT A
+IN A
1 kΩ
BCK
+IN B
DATA
IOUT L+
SCK
OUT B
−IN B
PCM1792
or
DSD1792
RI
RI
1 kΩ
RF
1 kΩ
CF
2.7 nF
CF
2.7 nF
RF
1 kΩ
IOUT R+
MDI
+IN C
OUT C
−IN C
+IN D
MC
MDO
RO
LIN+
LOUT
10 Ω
RF
1 kΩ
RF
1 kΩ
ZEROR
Controller
LIN−
1/2 OPA4134
ZEROL
MS
1 kΩ
IOUT R−
RST
OUT D
−IN D
RI
RIN+
1 kΩ
RIN−
RI
1 kΩ
RF
1 kΩ
CF
2.7 nF
RO
10 Ω
RF
ROUT
DYR > 120 dB
for Whole
System!
1 kΩ
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2004, Texas Instruments Incorporated
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted)
(1)
TPA6120A2
Supply voltage, VCC+ to VCC-
33 V
Input voltage, VI (2)
± VCC
Differential input voltage, VID
6V
Minimum load impedance
8Ω
Continuous total power dissipation
See Dissipation Rating Table
Operating free–air temperature range, TA
Operating junction temperature range, TJ
- 40°C to 85°C
(3)
- 40°C to 150°C
Storage temperature range, Tstg
- 40°C to 125°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1)
(2)
(3)
235°C
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating
conditions” is not implied. Exposure to absolute–maximum–rated conditions for extended periods may affect device reliability.
When the TPA6120A2 is powered down, the input source voltage must be kept below 600-mV peak.
The TPA6120A2 incorporates an exposed PowerPAD on the underside of the chip. This acts as a heatsink and must be connected to a
thermally dissipating plane for proper power dissipation. Failure to do so may result in exceeding the maximum junction temperature that
could permanently damage the device. See TI Technical Brief SLMA002 for more information about utilizing the PowerPAD thermally
enhanced package.
DISSIPATION RATING TABLE
(1)
PACKAGE
θJA (1)
(°C/W)
θJC
(°C/W)
TA = 25°C
POWER RATING
DWP
44.4
33.8
2.8 W
The PowerPAD must be soldered to a thermal land on the printed-circuit board. See the PowerPAD
Thermally Enhanced Package application note (SLMA002)
AVAILABLE OPTIONS
(1)
TA
PACKAGE
PART NUMBER
SYMBOL
-40°C to 85°C
DWP (1)
TPA6120A2DWP
6120A2
The DWP package is available taped and reeled. To order a taped and reeled part, add the suffix R
to the part number (e.g., TPA6120A2DWPR).
RECOMMENDED OPERATING CONDITIONS
Supply voltage, VCC+ and VCCLoad impedance
Operating free–air temperature, TA
2
MIN
MAX
Split Supply
±5
±15
Single Supply
10
30
VCC = ±5 V or ±15 V
16
-40
UNIT
V
Ω
85
°C
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
ELECTRICAL CHARACTERISTICS
over operating free-air temperature range (unless otherwise noted)
PARAMETER
TEST CONDITIONS
|VIO|
Input offset voltage (measured differentially)
VCC = ±5 V or ±15 V
PSRR
Power supply rejection ratio
VCC = 2.5 V to 5.5 V
MIN
TYP
MAX
2
5
75
VCC = ±5 V
±3.6
±3.7
VCC = ±15 V
±13.4
±13.5
UNIT
mV
dB
VIC
Common mode input voltage
ICC
Supply current (each channel)
IO
Output current (per channel)
VCC= ±5 V to ±15 V
700
mA
Input offset voltage drift
VCC = ±5 V or ±15 V
20
µV/°C
300
kΩ
13
Ω
12.5 to
-12.2
V
ri
Input resistance
ro
Output resistance
VO
Output voltage swing
VCC = ±5 V
11.5
VCC= ±15 V
13
15
Open Loop
VCC = ±15 V, RL = 25 Ω
V
11.8 to
-11.5
mA
3
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
OPERATING CHARACTERISTICS (1)
TA = 25°C, RL = 25 Ω, Gain = 2 V/V (unless otherwise noted)
PARAMETER
Intermodulation distortion
(SMPTE)
IMD
THD+N
Total harmonic distortion
plus noise
TEST CONDITIONS
MIN
TYP
VCC = ±12 V to ±15 V,
RL = 32 Ω,
VI = 1 VPP
0.00014%
VCC = ±12 V to ±15 V,
RL = 64 Ω,
VI = 1 VPP
0.000095%
PO = 100 mW, RL = 32 Ω
f = 1 kHz
VCC = ±12 V
0.00055%
VCC = ±15 V
0.00060%
PO = 100 mW, RL = 64 Ω
f = 1 kHz
VCC = ±12 V
0.00038%
VCC = ±15 V
0.00029%
VCC = ±12 V, Gain = 3 V/V
RL = 600 Ω, f = 1 kHz
PO = 80 mW
0.00014%
PO = 40 mW
0.000065%
VCC = ±15 V, Gain = 3 V/V
RL = 600 Ω, f = 1 kHz
PO = 125 mW
0.00012%
PO = 62.5 mW
0.000061%
VCC = ±12 V,
Gain = 3 V/V
VO = 15 VPP,
RL = 10 kΩ
f = 1 kHz
0.000024%
VCC = ±15 V,
Gain = 3 V/V
VO = 15 VPP,
RL = 10 kΩ
f = 1 kHz
0.000021%
RL = 32 Ω
f = 10 Hz to 22 kHz
V(RIPPLE) = 1 VPP
VCC= ±12 V
-80
VCC= ±15 V
-83
RL = 64 Ω
f = 10 Hz to 22 kHz
V(RIPPLE) = 1 VPP
VCC= ±12 V
-76
VCC= ±15 V
-79
SMTPE ratio = 4:1,
Gain = 2 V/V,
IM frequency = 60 Hz
High frequency = 7 kHz
kSVR
Supply voltage rejection
ratio
CMRR
Common mode rejection
ratio (differential)
SR
Slew rate
5
Output noise voltage
VCC = ±12 V to ±15 V
RL = 32 Ω to 64 Ω
f = 1 kHz
Gain = 2 V/V
Vn
Gain = 100 V/V
50
125
Signal-to-noise ratio
VCC = ±12 V to ±15 V
RL = 32 Ω to 64 Ω
f = 1 kHz
Gain = 2 V/V
SNR
Gain = 100 V/V
104
VCC = ±12 V
123
VCC = ±15 V
125
VCC = ±12 V
124
VCC = ±15 V
126
VI = 1 VRMS
RF = 1 kΩ
-90
VCC = ±5 V or ±15 V
100
VCC = ±15 V, Gain = 5 V/V, VO = 20 VPP
1300
VCC = ±5 V, Gain = 2 V/V, VO = 5 VPP
900
RL = 32 Ω, f = 1 kHz
Dynamic range
RL = 64 Ω, f = 1 kHz
Crosstalk
(1)
4
VCC = ±12 V to ±15 V
RL = 32 Ω to 64 Ω
f = 1 kHz
For IMD, THD+N, kSVR, and crosstalk, the bandwidth of the measurement instruments was set to 80 kHz.
MAX
UNIT
dB
dB
V/µs
µVrms
dB
dB
dB
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
DEVICE INFORMATION
Thermally Enhansed SOIC (DWP)
PowerPAD™ Package
Top View
1
2
3
4
5
6
7
8
9
10
LVCC−
LOUT
LVCC+
LIN+
LIN−
NC
NC
NC
NC
NC
20
19
18
17
16
15
14
13
12
11
RVCC−
ROUT
RVCC+
RIN+
RIN−
NC
NC
NC
NC
NC
NC − No internal connection
TERMINAL FUNCTIONS
PIN NAME
PIN NUMBER
I/O
DESCRIPTION
LVCC-
1
I
Left channel negative power supply – must be kept at the same potential as
RVCC-.
LOUT
2
O
Left channel output
LVCC+
3
I
Left channel positive power supply
LIN+
4
I
Left channel positive input
LIN-
5
I
Left channel negative input
NC
6,7,8,9,10,11,12,13,14,15
-
Not internally connected
RIN-
16
I
Right channel negative input
RIN+
17
I
Right channel positive input
RVCC+
18
I
Right channel positive power supply
ROUT
19
O
Right channel output
RVCC-
20
I
Right channel negative power supply - must be kept at the same potential as
LVCC-.
-
-
Connect to ground. The thermal pad must be soldered down in all
applications to properly secure device on the PCB.
Thermal Pad
5
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Frequency
Total harmonic distortion + noise
vs Output voltage
5
vs Output power
6, 7, 8
Power dissipation
vs Output power
Supply voltage rejection ratio
vs Frequency
Intermodulation distortion
1, 2, 3, 4
9
10, 11
vs High frequency
12
vs IM Amplitude
13
Crosstalk
vs Frequency
Signal-to-noise ratio
vs Gain
15, 16
14
Slew rate
vs Output step
17, 18
Small and large signal frequency response
19, 20
400-mV step response
21
10-V step response
22
20-V step response
23
TOTAL HARMONIC DISTORTION + NOISE
vs
FREQUENCY
0.01
RL = 10 k,
Gain = 3 V/V,
RF = 2 k,
RI = 1 k,
BW = 80 kHz
0.001
VCC = 15 VO = 15 VPP
VCC = 12 VO = 15 VPP
0.0001
0.00001
10
VCC = 12 VO = 12 VPP
VCC = 15 VO = 23 VPP
100
1k
f − Frequency − Hz
Figure 1.
6
THD+N −Total Harmonic Distortion + Noise − %
THD+N −Total Harmonic Distortion + Noise − %
0.01
TOTAL HARMONIC DISTORTION + NOISE
vs
FREQUENCY
RL = 600 ,
Gain = 3 V/V,
RF = 2 k,
RI = 1 k,
BW = 80 kHz
0.001
VCC = 15 V,
PO = 125 mW
VCC = 12 V,
PO = 80 mW
0.0001
10 k
50 k
10
100
1k
f − Frequency − Hz
Figure 2.
10 k
50 k
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
TYPICAL CHARACTERISTICS (continued)
TOTAL HARMONIC DISTORTION + NOISE
vs
FREQUENCY
1
RL = 64 ,
Gain = 2 V/V,
RF = 1 k,
RI = 1 k,
BW = 80 kHz
THD+N −Total Harmonic Distortion + Noise − %
THD+N −Total Harmonic Distortion + Noise − %
0.1
TOTAL HARMONIC DISTORTION + NOISE
vs
FREQUENCY
0.01
VCC = 15 V, PO = 700 mW
VCC = 15 V, PO = 1.35 W
VCC = 12 V, PO = 425 mW
0.001
VCC = 12 V, PO = 500 mW
0.0001
10
100
1k
10 k
RL = 32 ,
Gain = 2 V/V,
RF = 1 k,
RI = 1 k,
BW = 80 kHz
0.1
VCC = 15 V, PO = 1.5 W
0.01
VCC = 15 V, PO = 1.25 W
VCC = 12 V, PO = 950 mW
VCC = 12 V, PO = 800 mW
0.001
0.0001
10
50 k
100
f − Frequency − Hz
Figure 4.
TOTAL HARMONIC DISTORTION + NOISE
vs
OUTPUT VOLTAGE
TOTAL HARMONIC DISTORTION + NOISE
vs
OUTPUT POWER
50 k
10
THD+N −Total Harmonic Distortion + Noise − %
THD+N −Total Harmonic Distortion + Noise − %
10 k
Figure 3.
10
1
1k
f − Frequency − Hz
RL = 10 k,
Gain = 3 V/V,
f = 1 kHz,
RF = 2 k,
RI = 1 k,
BW = 80 kHz
0.1
0.01
VCC = 12 V
0.001
0.0001
VCC = 15 V
0.00001
3 5
10
15
20
25
VO − Output Voltage − VPP
Figure 5.
30
35
1
RL = 600 ,
Gain = 3 V/V,
f = 1 kHz,
RF = 2 k,
RI = 1 k,
BW = 80 kHz
0.1
VCC = 12 V
0.01
VCC = 15 V
0.001
0.0001
0.00001
0.01
0.1
0.2
PO − Output Power − W
Figure 6.
7
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
TYPICAL CHARACTERISTICS (continued)
TOTAL HARMONIC DISTORTION + NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION + NOISE
vs
OUTPUT POWER
1
10
RL = 64 ,
Gain = 2 V/V,
f = 1 kHz,
RF = 1 k,
RI = 1 k,
BW = 80 kHz
THD+N −Total Harmonic Distortion + Noise − %
THD+N −Total Harmonic Distortion + Noise − %
10
VCC = 12 V
0.1
VCC = 15 V
0.01
0.001
0.0001
0.01
0.1
1
1
RL = 32 ,
Gain = 2 V/V,
f = 1 kHz,
RF = 1 k,
RI = 1 k,
BW = 80 kHz
VCC = 12 V
0.1
VCC = 15 V
0.01
0.001
0.0001
0.01
2
0.1
PO − Output Power − W
Figure 7.
Figure 8.
POWER DISSIPATION
vs
OUTPUT POWER
SUPPLY VOLTAGE REJECTION RATIO
vs
FREQUENCY
Mono Operation
k SVR − Supply Voltage Rejection Ratio − dB
2
VCC = 15 V, RL = 32 PD − Power Dissipation − W
1.8
1.6
1.4
VCC = 12 V, RL = 32 1.2
1
VCC = 15 V,
RL = 64 0.8
0.6
VCC = 12 V,
RL = 64 0.4
0.2
0
8
1
2
3 4
PO − Output Power − W
0
VCC = 12 V,
V(ripple) = 1 VPP,
Gain = 2 V/V
BW = 80 kHz
−10
−20
−30
Representative of both positive and
negative supplies.
−40
−50
−60
64 32 −70
−80
−90
0
0.5
1
1.5
2
2.5
3
3.5
PO − Output Power − W
10
100
1k
f − Frequency − Hz
Figure 9.
Figure 10.
10 k
50 k
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
TYPICAL CHARACTERISTICS (continued)
SUPPLY VOLTAGE REJECTION RATIO
vs
FREQUENCY
INTERMODULATION DISTORTION
vs
HIGH FREQUENCY
0.1
VCC = 15 V,
V(ripple) = 1 VPP,
Gain = 2 V/V
BW = 80 kHz
−10
−20
−30
Intermodulation Distortion − %
k SVR − Supply Voltage Rejection Ratio − dB
−0
Representative of both positive and
negative supplies.
−40
−50
−60
−70
64 32 0.01
0.001
VCC = 12 V,
RL = 32 1k
f − Frequency − Hz
10 k
2k
50 k
Figure 12.
INTERMODULATION DISTORTION
vs
IM AMPLITUDE (AT INPUT)
CROSSTALK
vs
FREQUENCY
−60
4:1 SMPTE Ratio
Gain = 3 V/V,
High Frequency = 7 kHz
IM Frequency = 60 Hz
Crosstalk − dB
VCC = 12 V, RL = 32 VCC = 12 V, RL = 64 0.01
VCC = 15 V, RL = 32 VCC = 15 V,
RL = 64 VCC = 12 V,
RL = 64 −80
−90
50 k
RF = 1 k,
Gain = 2 V/V,
BW = 80 kHz
−70
0.1
VCC = 15 V,
RL = 64 10 k
f − High Frequency − Hz
Figure 11.
10
1
VCC = 12 V,
RL = 64 0.00001
100
VCC = 15 V,
RL = 32 0.0001
−80
−90
10
Intermodulation Distortion − %
4:1 SMPTE Ratio
VI = 1 VPP,
Gain = 2 V/V,
IM Frequency = 60 Hz
VCC = 15 V,
RL = 32 −100
VCC = 12 V,
RL = 32 0.001
−110
0.0001
0.00001
0
VCC = 15 V, RL = 64 1
2
3
4
5
6
7
IM Amplitude (At Input) − VPP
Figure 13.
8
9
10
−120
10
100
1k
10 k
50 k
f − Frequency − Hz
Figure 14.
9
TPA6120A2
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SLOS431 – MARCH 2004
TYPICAL CHARACTERISTICS (continued)
SIGNAL-TO-NOISE RATIO
vs
GAIN
SIGNAL-TO-NOISE RATIO
vs
GAIN
130
130
VCC = 12 V
125
120
115
110
125
120
115
110
RI = 32 105
100
10
20
30
40
50
60
70
80
100
90 100
1
10
80
90 100
1
2
3
4
Output Step (Peak−To−Peak) − V
5
20
30
40
50
60
Gain − V/V
Gain − V/V
Figure 15.
Figure 16.
SLEW RATE
vs
OUTPUT STEP
SLEW RATE
vs
OUTPUT STEP
1500
70
1000
VCC = ± 15 V
Gain = 5 V/V
RF = 1 kΩ
RL = 25 Ω
1300
VCC = ± 5 V
Gain = 2 V/V
RF = 1 kΩ
RL = 25 Ω
900
+SR
800
−SR
+SR
Slew Rate − V/µ s
1100
Slew Rate − V/µ s
THD+N, RI = 32 105
1
VCC = 15 V
THD+N, RI = 64 Signal−To−Noise Ratio − dB
Signal−To−Noise Ratio − dB
RI = 64 900
700
700
−SR
600
500
400
500
300
300
200
100
100
0
5
10
15
Output Step (Peak−To−Peak) − V
Figure 17.
10
20
0
Figure 18.
TPA6120A2
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SLOS431 – MARCH 2004
TYPICAL CHARACTERISTICS (continued)
SMALL AND LARGE SIGNAL
FREQUENCY RESPONSE
SMALL AND LARGE SIGNAL
FREQUENCY RESPONSE
−3
3
VI = 500 mV
−6
−3
Output Level − dBV
−9
Output Level − dBV
VI = 500 mV
0
VI = 250 mV
−12
−15
VI = 125 mV
−18
−21
VI = 250 mV
−6
−9
VI = 125 mV
−12
−15
VI = 62.5 mV
−24
−27
−30
10
VI = 62.5 mV
−18
Gain = 1 V/V
VCC = ± 15 V
RF = 820 Ω
RL = 25 Ω
100
1k
−21
10k
100k
1M
−24
10
10M 100M 500M
Gain = 2 V/V
VCC = ± 15 V
RF = 680 Ω
RL = 25 Ω
100
f − Frequency − Hz
300
6
200
4
VCC = ±15 V
Gain = 2 V/V
RL = 25 Ω
RF = 1 kΩ
tr/tf= 300 ps
See Figure 3
−200
−300
−400
1M
10M 100M 500M
10-V STEP RESPONSE
8
VO − Output Voltage − V
VO − Output Voltage − mV
400-mV STEP RESPONSE
−100
100k
Figure 20.
400
0
10k
f − Frequency − Hz
Figure 19.
100
1k
2
0
−2
VCC = ±15 V
Gain = 2 V/V
RL = 25 Ω
RF = 1 kΩ
tr/tf= 5 ns
See Figure 3
−4
−6
−8
0
50
100 150 200 250 300 350 400 450 500
0
50
100 150 200 250 300 350 400 450 500
t − Time − ns
t − Time − ns
Figure 21.
Figure 22.
11
TPA6120A2
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SLOS431 – MARCH 2004
TYPICAL CHARACTERISTICS (continued)
20-V STEP RESPONSE
16
VCC = ±15 V
Gain = 5 V/V
RL = 25 Ω
RF = 2 kΩ
tr/tf= 5 ns
See Figure 3
VO − Output Voltage − V
12
8
4
0
−4
−8
−12
−16
0
50
100 150 200 250 300 350 400 450 500
t − Time − ns
Figure 23.
12
TPA6120A2
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SLOS431 – MARCH 2004
APPLICATION INFORMATION
Current-Feedback Amplifiers
The TPA6120A2 is a current-feedback amplifier with differential inputs and single-ended outputs.
Current-feedback results in low voltage noise, high open-loop gain throughout a large frequency range, and low
distortion. It can be used in a similar fashion as voltage-feedback amplifiers. The low distortion of the
TPA6120A2 results in a signal-to-noise ratio of 120 dB as well as a dynamic range of 120 dB.
Independent Power Supplies
The TPA6120A2 consists of two independent high-fidelity amplifiers. Each amplifier has its own voltage supply.
This allows the user to leave one of the amplifiers off, saving power, and reducing the heat generated. It also
reduces crosstalk.
Although the power supplies are independent, there are some limitations. When both amplifiers are used, the
same voltage must be applied to each amplifier. For example, if the left channel amplifier is connected to a ±12-V
supply, the right channel amplifier must also be connected to a ±12-V supply. If it is connected to a different
supply voltage, the device may not operate properly and consistently.
When the use of only one amplifier is preferred, it must be the left amplifier. The voltage supply to the left
amplifier is also responsible for internal start-up and bias circuitry of the device. Regardless of whether one or
both amplifiers are used, the VCC- pins of both amplifiers must always be at the same potential.
To power down the right channel amplifier, disconnect the VCC+ pin from the power source.
The two independent power supplies can be tied together on the board to receive their power from the same
source.
Power Supply Decoupling
As with any design, proper power supply decoupling is essential. It prevents noise from entering the device via
the power traces and provides the extra power the device can sometimes require in a rapid fashion. This
prevents the device from being momentarily current starved. Both of these functions serve to reduce distortion,
leaving a clean, uninterrupted signal at the output.
Bulk decoupling capacitors should be used where the main power is brought to the board. Smaller capacitors
should be placed as close as possible to the actual power pins of the device. Because the TPA6120A2 has four
power pins, use four surface mount capacitors. Both types of capacitors should be low ESR.
Resistor Values
RF = 1 k
VCC−
RI = 1 k
−
VI
+
RS = 50 RO = 10 RL
VCC+
Figure 24. Single-Ended Input With a Noninverting Gain of 2 V/V
In the most basic configuration (see Figure 24), four resistors must be considered, not including the load
impedance. The feedback and input resistors, RF and RI, respectively, determine the closed-loop gain of the
amplifier. RO is a series output resistor designed to protect the amplifier from any capacitance on the output path,
including board and load capacitance. RS is a series input resistor. Because the TPA6120A2 is a
current-feedback amplifier, take care when choosing the feedback resistor.
13
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
APPLICATION INFORMATION (continued)
The value of the feedback resistor should be chosen by using Figure 27 through Figure 32 as guidelines. The
gain can then be set by adjusting the input resistor. The smaller the feedback resistor, the less noise is
introduced into the system. However, smaller values move the dominant pole to higher and higher frequencies,
making the device more susceptible to oscillations. Higher feedback resistor values add more noise to the
system, but pull the dominant pole down to lower frequencies, making the device more stable. Higher impedance
loads tend to make the device more unstable. One way to combat this problem is to increase the value of the
feedback resistor. It is not recommended that the feedback resistor exceed a value of 10 kΩ. The typical value
for the feedback resistor for the TPA6120A2 is 1 kΩ. In some cases, where a high-impedance load is used along
with a relatively large gain and a capacitive load, it may be necessary to increase the value of the feedback
resistor from 1 kΩ to 2 kΩ, thus adding more stability to the system. Another method to deal with oscillations is to
increase the size of RO.
CAUTION:
Do not place a capacitor in the feedback path. Doing so can cause oscillations.
Capacitance at the outputs can cause oscillations. Capacitance from some sources, such as layout, can be
minimized. Other sources, such as those from the load (e.g., the inherent capacitance in a pair of headphones),
cannot be easily minimized. In this case, adjustments to RO and/or RF may be necessary.
The series output resistor should be kept at a minimum of 10 Ω. It is small enough so that the effect on the load
is minimal, but large enough to provide the protection necessary such that the output of the amplifier sees little
capacitance. The value can be increased to provide further isolation, up to 100 Ω.
The series resistor, RS, should be used for two reasons:
1. It prevents the positive input pin from being exposed to capacitance from the line and source.
2. It prevents the source from seeing the input capacitance of the TPA6120A2.
The 50-Ω resistor was chosen because it provides ample protection without interfering in any noticeable way with
the signal. Not shown is another 50-Ω resistor that can be placed on the source side of RS to ground. In that
capacity, it serves as an impedance match to any 50-Ω source.
RF = 1 k
VCC−
RI = 1 k
VI
−
RO = 10 +
RL
VCC+
Figure 25. Single-Ended Input With a Noninverting Gain of -1 V/V
RF = 1 k
VCC−
RI= 1 k
VI−
−
VI+
+
RI = 1 k
RO = 10 RL
VCC+
RF = 1 k
Figure 26. Differential Input With a Noninverting Gain of 2 V/V
Figure 26 shows the TPA6120A2 connected with differential inputs. Differential inputs are useful because they
take the greatest advantage of the device's high CMRR. The two feedback resistor values must be kept the
same, as do the input resistor values.
14
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
APPLICATION INFORMATION (continued)
Special note regarding mono operation:
• If both amplifiers are powered on, but only one channel is to be used, the unused amplifier MUST have a
feedback resistor from the output to the negative input. Additionally, the positive input should be grounded as
close to the pin as possible. Terminate the output as close to the output pin as possible with a 25-Ω load to
ground.
• These measures should be followed to prevent the unused amplifier from oscillating. If it oscillates, and the
power pins of both amplifiers are tied together, the performance of the amplifier could be seriously degraded.
Checking for Oscillations and Instability
Checking the stability of the amplifier setup is recommended. High frequency oscillations in the megahertz region
can cause undesirable effects in the audio band.
Sometimes, the oscillations can be quite clear. An unexpectedly large draw from the power supply may be an
indication of oscillations. These oscillations can be seen with an oscilloscope. However, if the oscillations are not
obvious, or there is a chance that the system is stable but close to the edge, placing a scope probe with 10 pF of
capacitance can make the oscillations worse, or actually cause them to start.
A network analyzer can be used to determine the inherent stability of a system. An output vs frequency curve
generated by a network analyzer can be a good indicator of stability. At high frequencies, the curve shows
whether a system is oscillating, close to oscillation, or stable. Looking at Figure 27 through Figure 32, several
different phenomena occur. In one scenario, the system is stable because the high frequency rolloff is smooth
and has no peaking. Increasing RF decreases the frequency at which this rolloff occurs (see the Resistor Values
section). Another scenario shows some peaking at high frequency. If the peaking is 2 dB, the amplifier is stable
as there is still 45 degrees of phase margin. As the peaking increases, the phase margin shrinks, the amplifier
and the system, move closer to instability. The same system that has a 2-dB peak has an increased peak when
a capacitor is added to the output. This indicates the system is either on the verge of oscillation or is oscillating,
and corrective action is required.
3
3
RF = 620 Ω
RF = 820 Ω
1
0
−1
RF = 1 kΩ
−2
−3
−4
−5
−6
−7
10
VCC = ±15 V
RL = 100 Ω
Gain = 1 V/V
VI = 200 mV
100
1k
10k
100k
1M
10M 100M 500M
f − Frequency − Hz
Figure 27. Normalized Output Response vs Frequency
2
Normalized Output Response − dB
Normalized Output Response − dB
2
RF = 430 Ω
1
0
−1
RF = 620 Ω
−2
−3
−4
−5
−6
10
RF = 1 kΩ
VCC = ±15 V
RL = 100 Ω
Gain = 2 V/V
VI = 200 mV
100
1k
10k
100k
1M
10M 100M 500M
f − Frequency − Hz
Figure 28. Normalized Output Response vs Frequency
15
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
APPLICATION INFORMATION (continued)
1
RL = 200 Ω
0
0
−1
−1
Normalized Output Response − dB
Normalized Output Response − dB
1
−2
RL = 100 Ω
−3
−4
RL = 50 Ω
−5
RL = 25 Ω
−6
−7
−8
−9
10
VCC = ±15 V
RF = 1 kΩ
Gain = 1 V/V
VI = 200 mV
100
1k
−2
−3
−5
RL = 200 Ω
−6
RL = 100 Ω
−7
−8
10k
100k
1M
−9
10
10M 100M 500M
RL = 25 Ω
−4
VCC = ±15 V
RF = 1 kΩ
Gain = 2 V/V
VI = 200 mV
100
1k
f − Frequency − Hz
10k
100k
1M
10M 100M 500M
f − Frequency − Hz
Figure 29. Normalized Output Response vs Frequency
Figure 30. Normalized Output Response vs Frequency
3
9
2
8
RF = 510 Ω
1
0
−1
RF = 1 kΩ
−2
RF = 1.5 kΩ
−3
−4
−5
VCC = ± 5 V
Gain = 1 V/V
RL = 25 Ω
VI = 200 mV
−6
10
100
1k
Output Amplitude − dB
RF = 620 Ω
Output Amplitude − dB
RL = 50 Ω
7
6
4
1
Figure 31. Output Amplitude vs Frequency
RF = 1.2 kΩ
3
2
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
RF = 820 Ω
5
0
10
VCC = ± 5 V
Gain = 2 V/V
RL = 25 Ω
VI = 200 mV
100
1k
10k 100k 1M 10M 100M 500M
f − Frequency − Hz
Figure 32. Output Amplitude vs Frequency
PCB Layout
Proper board layout is crucial to getting the maximum performance out of the TPA6120A2.
A ground plane should be used on the board to provide a low inductive ground connection. Having a ground
plane underneath traces adds capacitance, so care must be taken when laying out the ground plane on the
underside of the board (assuming a 2-layer board). The ground plane is necessary on the bottom for thermal
reasons. However, certain areas of the ground plane should be left unfilled. The area underneath the device
where the PowerPAD is soldered down should remain, but there should be no ground plane underneath any of
the input and output pins. This places capacitance directly on those pins and leads to oscillation problems. The
underside ground plane should remain unfilled until it crosses the device side of the input resistors and the
output series resistor. Thermal reliefs should be avoided if possible because of the inductance they introduce.
16
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
APPLICATION INFORMATION (continued)
Despite the removal of the ground plane in critical areas, stray capacitance can still make its way onto the
sensitive outputs and inputs. Place components as close as possible to the pins and reduce trace lengths. See
Figure 33 and Figure 34. It is important for the feedback resistor to be extremely close to the pins, as well as the
series output resistor. The input resistor should also be placed close to the pin. If the amplifier is to be driven in a
noninverting configuration, ground the input close to the device so the current has a short, straight path to the
PowerPAD (gnd).
Too Long
Too Long
RF
RI
VI
−
+
TPA6120A2
Too Long
Too Long
RO
RL
Figure 33. Layout That Can Cause Oscillation
Minimized Length of
Feedback Path
Short Trace
Before Resistors
VI
RF
RO
−
RI
+
RL
TPA6120A2
Ground as Close to
the Pin as Possible
Minimized Length of
the Trace Between
Output Node and RO
Figure 34. Layout Designed To Reduce Capacitance On Critical Nodes
Thermal Considerations
Amplifiers can generate quite a bit of heat. Linear amplifiers, as opposed to Class-D amplifiers, are extremely
inefficient, and heat dissipation can be a problem. There is no one to one relationship between output power and
heat dissipation, so the following equations must be used:
PL
Efficiency of an amplifier P SUP
(1)
Where
2
PL P SUP
2
VLRMS
V
V
, and VLRMS P , therefore, P L P per channel
2
RL
2RL
VCC I CCavg VCC I CC(q)
I CCavg
(3)
2 VP
VP
V
1
sin(t) dt [cos(t)] 2 P
RL
R L
RL
0
0
(2)
(4)
Where
17
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
APPLICATION INFORMATION (continued)
VP 2 PL R L
(5)
Therefore,
V V
P SUP CC P V CC I CC(q)
RL
(6)
PL = Power delivered to load (per channel)
PSUP = Power drawn from power supply
VLRMS = RMS voltage on the load
RL = Load resistance
VP = Peak voltage on the load
ICCavg = Average current drawn from the power supply
ICC(q) = Quiescent current (per channel)
VCC = Power supply voltage (total supply voltage = 30 V if running on a ±15-V power supply
η = Efficiency of a SE amplifier
For stereo operation, the efficiency does not change because both PL and PSUP are doubled. This effects the
amount of power dissipated by the package in the form of heat.
A simple formula for calculating the power dissipated, PDISS, is shown in Equation 7:
P DISS (1 ) P SUP
(7)
In stereo operation, PSUP is twice the quantity that is present in mono operation.
The maximum ambient temperature, TA, depends on the heat-sinking ability of the system. θJA for a 20-pin DWP,
whose thermal pad is properly soldered down, is shown in the dissipation rating table.
T A Max T J Max ΘJA P Diss
(8)
2
Mono Operation
VCC = 15 V, RL = 32 PD − Power Dissipation − W
1.8
1.6
1.4
VCC = 12 V, RL = 32 1.2
1
VCC = 15 V,
RL = 64 0.8
0.6
VCC = 12 V,
RL = 64 0.4
0.2
0
0
0.5
1
1.5
2
2.5
3
3.5
PO − Output Power − W
Figure 35. Power Dissipation vs Output Power
18
TPA6120A2
www.ti.com
SLOS431 – MARCH 2004
Application Circuit
OPA4134
12 V
10 µF
−12 V
10 µF
10 µF
100 µF
100 µF
+
0.1 µF
+
−5 V
0.1 µF
VCC−
+
10 µF
+
+
5V
V−
+
V+
TPA6120A2
VCC+
CF 2.7 nF
RF 1 k
V−
−INA
2
3
−
RF
1 k
11
OUTA
1
VCC−
+
RI
1 k
4
5V
V+
0.1 µF
2
ZEROL
ZEROR
VCC2L
AGND3L
28
+
1
10 µF
LIN−
LIN+
CF 2.7 nF
RI
1 k
27
RF 1 k
4
PCM
Audio
Data
Source
IOUTL− 26
LRCK
5
DATA
6
BCK
7
0.1 µF
MSEL
IOUTL+
SCK
8
DGND
9
VDD
PCM1792
25
AGND2
24
VCC1
23
VCOML
22
VCOMR
21
IREF
20
AGND1
19
5V
6
5
Controller
11 MDI
IOUTR−
18
12 MC
IOUTR+
17
13 MDO
+
VCC2R
0.1 F
VCC+
7
+
OUTB
V+
CF 2.7 nF
47 µF
RF 1 k
V−
9
−INC
0.1 µF
10
11
−
VCC−
RI
1 k
4
5V
RF
1 k
OUTC
8
+
V+
16
0.1 F
20
16
−
ROUT
RIN−
CF 2.7 nF
15
RIN+
10 µF
RI
1 k
RF 1 k
3.3 V
13
−IND
12
17
RF
1 k
V−
+
10 µF
RO 10 3
11
−
47 µF 10 µF
+
+
14 RST
AGND3R
2
+
4
10 kΩ
10 MS
LOUT
4
RF
1 k
V−
−INB
−
+
3
0.1 F
4
5
+
19
18
RO 10 0.1 F
VCC+
11
−
14
+
OUTD
4
V+
Figure 36. Typical Application Circuit
In many applications, the audio source is digital. It must go through a digital-to-analog converter (DAC) so that
traditional analog amplifiers can drive the speakers or headphones.
Figure 36 shows a complete circuit schematic for such a system. The digital audio is fed into a high performance
DAC. The PCM1792, a Burr-Brown product from TI, is a 24-bit, stereo DAC.
The output of the PCM1792 is current, not voltage, so the OPA4134 is used to convert the current input to a
voltage output. The OPA4134, a Burr-Brown product from TI, is a low-noise, high-speed, high-performance
operational amplifier. CF and RF are used to set the cutoff frequency of the filter. The RC combination in
Figure 36 has a cutoff frequency of 59 kHz. All four amplifiers of the OPA4134 are used so the TPA6120A2 can
be driven differentially.
19
TPA6120A2
SLOS431 – MARCH 2004
www.ti.com
The output of the OPA4134 goes into the TPA6120A2. The TPA6120A2 is configured for use with differential
inputs, stereo use, and a gain of 2V/V. Note that the 0.1-uF capacitors are placed at every supply pin of the
TPA6120A2, as well as the 10-Ω series output resistor.
Each output goes to one channel of a pair of stereo headphones, where the listener enjoys crisp, clean, virtually
noise free music with a dynamic range greater than the human ear is capable of detecting.
20
PACKAGE OPTION ADDENDUM
www.ti.com
18-Apr-2006
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
TPA6120A2DWP
ACTIVE
SO
Power
PAD
DWP
20
25
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPA6120A2DWPG4
ACTIVE
SO
Power
PAD
DWP
20
25
Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
TPA6120A2DWPR
ACTIVE
SO
Power
PAD
DWP
20
2000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
Lead/Ball Finish
MSL Peak Temp (3)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
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incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
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In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
Addendum-Page 1
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