® OPA2607 OPA 260 7 OPA 260 7 For most current data sheet and other product information, visit www.burr-brown.com Dual, High Output, Current-Feedback OPERATIONAL AMPLIFIER TM FEATURES DESCRIPTION ● WIDEBAND ±12V OPERATION: 25MHz (G = +8) ● UNITY GAIN STABLE: 35MHz (G = +1) The OPA2607 provides a high output voltage swing and low distortion required for low turns ratio ADSL upstream driver applications. Operating on a ±12V supply, the OPA2607 consumes a low 8.0mA/channel quiescent current to deliver a very high 250mA peak output current. Guaranteed output current of 180mA supports even the most demanding ADSL CPE requirements with low harmonic distortion. Differential driver applications will deliver < –75dBc distortion at the peak upstream power levels of full rate ADSL. Using a differential driver design, as shown below, the OPA2607 can deliver a high 38Vp-p voltage swing into a 1:0.8 step-down transformer to meet the ADSL CPE upstream power requirements. This low turns ratio actually provides a step up to the much weaker downstream signal arriving on the line side of this transformer, extending the DSL modem’s reach. ● ● ● ● ● ● ● HIGH OUTPUT CURRENT: 250mA OUTPUT VOLTAGE SWING: ±10.5V (VS = ±12V) HIGH SLEW RATE: 600V/µs LOW SUPPLY CURRENT: 8mA/channel FLEXIBLE POWER CONTROL (SO-14) ±6V TO ±16V SUPPLY RANGE POWER PACKAGING APPLICATIONS ● ● ● ● xDSL LINE DRIVER LOW-NOISE ADSL RECEIVER LOW-COST VIDEO DA LOW-COST UPGRADE TO LT1207/AD812 +12V 1/2 OPA2607 1.21kΩ 78.7Ω 1:0.8 100nF 4.8Vp-p 38Vp-p 348Ω 1.21kΩ 100Ω 15Vp-p 78.7Ω Power control features are included in the SO-14 package version to allow system power to be minimized. Two logic control lines allow four quiescent power settings. These include full power, power cutback for short loops, idle state for no signal transmission but line match maintenance, and shutdown for power off with a high impedance output. An additional I ADJ pin allows the maximum supply current to be adjusted ±25% from the nominal value. Connecting this pin to +VCC will increase the full power quiescent to 20mA, increasing the peak output current available, while connecting this pin to –VCC will decrease the full power quiescent to 12mA where a lower peak output current is required. The digital control lines continue to scale the total quiescent current from these new maximum levels in the same proportional steps as before. The OPA2607 is available in three package styles. For power package driver applications, a thermally enhanced 1/2 OPA2607 –12V Low Turns Ratio ADSL Upstream Driver with a heat slug is available in both SO-8 and SO-14 pinouts. For lower power receiver applications, a standard SO-8 package is available. OPA2607 RELATED PRODUCTS SINGLES DUALS TRIPLES NOTES OPA681 — OPA2681 OPA2677 OPA3681 — Single +12V Capable Single +12V Capable International Airport Industrial Park • Mailing Address: PO Box 11400, Tucson, AZ 85734 • Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 • Tel: (520) 746-1111 Twx: 910-952-1111 • Internet: http://www.burr-brown.com/ • Cable: BBRCORP • Telex: 066-6491 • FAX: (520) 889-1510 • Immediate Product Info: (800) 548-6132 ® © 2000 Burr-Brown Corporation SBOS128 PDS-1615A 1 Printed in U.S.A. August, 2000 OPA2607 SPECIFICATIONS: VS = ±12V RF = 1.21kΩ, RL = 100Ω, and G = +8, unless otherwise noted. OPA2607H, U, N TYP PARAMETER AC PERFORMANCE (see Figure 1) Small-Signal Bandwidth (VO = 0.5Vp-p) Bandwidth for 0.1dB Gain Flatness Large-Signal Bandwidth Slew Rate Rise/Fall Time Spurious Free Dynamic Range(4) Input Voltage Noise Non-Inverting Input Current Noise Inverting Input Current Noise Differential Gain Differential Phase Channel-to-Channel Crosstalk DC PERFORMANCE(5) Open-Loop Transimpedance Gain Input Offset Voltage Average Offset Voltage Drift Non-Inverting Input Bias Current Average Non-Inverting Input Bias Current Drift Inverting Input Bias Current Average Inverting Input Bias Current Drift INPUT Common-Mode Input Range (CMIR) Common-Mode Rejection Ratio (CMRR) Non-Inverting Input Impedance Inverting Input Resistance OUTPUT Voltage Output Swing Current Output, Sourcing Current Output, Sinking Closed-Loop Output Impedance Power Control (SO-14 only) Maximum Logic 0 Minimum Logic 1 Logic Input Current Supply Current at Full Power Supply Current at Power Cutback Supply Current at Idle Power Supply Current at Shutdown Output Impedance in Idle Power Output Impedance in Shutdown Shutdown Isolation Maximum Adjusted Quiescent Current Minimum Adjusted Quiescent Current POWER SUPPLY Minimum Operating Voltage Specified Operating Voltage Maximum Operating Voltage Maximum Quiescent Current Minimum Quiescent Current Power Supply Rejection Ratio (PSRR) TEMPERATURE RANGE Specification: H, U, N Thermal Resistance, θJA H PSO-8 Power Package(6) U SO-8 N PSO-14 Power Package(6) CONDITIONS +25°C G = +1, RF = 1.50kΩ G = +2, RF = 1.43kΩ G = +4, RF = 1.37kΩ G = +8, RF = 1.21kΩ G = +8, VO = 0.5Vp-p G = +8, VO = 20Vp-p G = +8, VO = 20V Step G = +8, VO = 0.5V Step VO = 2Vp-p, 1MHz, RL = 100Ω VO = 20Vp-p, 150kHz, RL = 150Ω 35 28 25 25 6 13 600 14 77 75 1.7 11 15 0.01 0.01 –60 GUARANTEED +25°C(2) 0°C to 70°C(3) –40°C to +85°C(3) UNITS MIN/ TEST MAX LEVEL(1) MHz MHz MHz MHz MHz MHz V/µs ns dB dB nV/√Hz pA/√Hz pA/√Hz % degrees dB typ typ typ min typ min min min min min max max max typ typ typ C C C B C B B B B B B B B C C C 19 18 17 10.6 470 18 66 70 2.0 13 17 9.0 400 20 60 58 2.6 13 17 7.9 350 21 57 57 2.7 13 17 950 ±1.5 440 ±7 ±3 ±12 ±4 ±40 390 ±8 –20 ±15 –70 ±58 –380 310 ±8.5 –25 ±20 –100 ±70 –425 kΩ mV µV/°C µA nA/°C µA nA/°C min max max max max max max A A B A B A B ±10.0 53 ±9.9 52 ±9.8 51 Open-Loop ±10.3 64 250 || 4 33 V dB kΩ || pF Ω min min typ typ A A C C No Load, Hard Limit RL = 100Ω, Hard Limit RL = 150Ω, SFDR > 67dB, 150kHz VO = 0 VO = 0 G = +8, f ≤ 10kHz ±11.2 ±10.5 ±10.2 310 250 0.02 ±10.9 ±9.9 ±10.8 ±9.8 ±10.7 ±9.7 210 180 175 150 140 110 V V V mA mA Ω min min typ min min typ A A C A A C V V µA mA mA mA mA Ω kΩ || pF dB mA mA max min max typ typ typ typ typ typ typ typ typ C C C C C C C C C C C C V V V mA mA dB min typ max max min min B C A A A A –40 to +85 °C typ C 50 125 45 °C/W °C/W °C/W typ typ typ C C C NTSC, G = +2, RL = 150Ω NTSC, G = +2, RL = 150Ω f = 1MHz VO = 0V, RL = 100Ω VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V VCM = 0V DIG_REF = Gnd A0, A1 A0, A1 0V to 4.5V A0 = 1, A1 = 1, IADJ = open A0 = 0, A1 = 1, IADJ = open A0 = 1, A1 = 0, IADJ = open A0 = 0, A1 = 0, IADJ = open Closed-Loop, f < 1MHz 0.8 2 60 16 13 3.8 1.3 0.7 350 || 17 75 20 12 G = +8, 1MHz A0 = 1, A1 = 1, IADJ at +VS A0 = 1, A1 = 1, IADJ at –VS ±12 Total Both Channels, Full Power Total Both Channels, Full Power f ≤ 10kHz 16 16 68 Junction-to-Ambient ±6 ±6 ±6 ±16 16.8 15.2 61 ±16 17 13.8 59 ±16 17.5 13.3 57 NOTES: (1) Test Levels: (A) 100% tested at 25°C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation. (C) Typical value only for information. (2) Junction temperature = ambient for 25°C guaranteed specifications. (3) Junction temperature = ambient at low temperature limit: junction temperature = ambient +40°C at high temperature limit for over temperature guaranteed specifications. (4) Single amplifier SFDR limited by 2nd Harmonic. Differential SFDR will be limited by 3rd Harmonic and will be > 15dB higher. (5) Current is considered positive out of node. VCM is the input common-mode voltage. (6) Slug in power package connected to –VS plane at least 2" x 2" (50mm x 50mm) in size. See the Board Layout Guidelines Section. ® OPA2607 2 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATIONS Power Supply ............................................................................. ±16.5VDC Internal Power Dissipation(1) ............................ See Thermal Information Differential Input Voltage ..................................................................... ±5V Input Voltage Range ............................................................................ ±VS Storage Temperature Range: U, N ................................ –40°C to +125°C Lead Temperature (soldering, 10s) .............................................. +260°C Junction Temperature (TJ ) ........................................................... +175°C ESD Rating (Human Body Model) .................................................. 4000V (Machine Model) ........................................................... 300V Top View Out A 1 8 +VS –In A 2 7 Out B +In A 3 6 –In B –VS 4 5 +In B NOTE:: (1) Packages must be derated based on specified θJA. Maximum TJ must be observed. SO-8, PSO-8 ELECTROSTATIC DISCHARGE SENSITIVITY Electrostatic discharge can cause damage ranging from performance degradation to complete device failure. Burr-Brown Corporation recommends that all integrated circuits be handled and stored using appropriate ESD protection methods. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet published specifications. –In A 1 14 Out A +In A 2 13 NC A0 3 12 DIG_REF –VS 4 A1 5 10 IADJ +In B 6 9 NC –In B 7 8 Out B Power Control 11 +VS PSO-14 NC = No Connection PACKAGE/ORDERING INFORMATION PRODUCT PACKAGE PACKAGE DRAWING NUMBER OPA2607H " OPA2607U " OPA2607N " PSO-8 " SO-8 " PSO-14 " 182-1 " 182 " 235-1 " SPECIFIED TEMPERATURE RANGE PACKAGE MARKING ORDERING NUMBER(1) TRANSPORT MEDIA –40°C to +85°C " " " " " OPA2607H " OPA2607U " OPA2607N " OPA2607H OPA2607H/2K5 OPA2607U OPA2607U/2K5 Contact Factory Contact Factory Rails Tape and Reel Rails Tape and Reel Rails Tape and Reel NOTE: (1) Models with a slash (/) are available only as Tape and Reel in the quantity indicated after the slash (e.g. /2K5 indicates 2500 devices per reel). Ordering 2500 pieces of the OPA2607U/2K5 will get a single 2500-piece Tape and Reel. The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes no responsibility for the use of this information, and all use of such information shall be entirely at the user’s own risk. Prices and specifications are subject to change without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant any BURR-BROWN product for use in life support devices and/or systems. ® 3 OPA2607 TYPICAL PERFORMANCE CURVES: VS = ±12V At TA = +25°C, G = +8, RF = 1.21kΩ, and RL = 100Ω, unless otherwise noted. See Figure 1 for AC performance only. NON-INVERTING SMALL-SIGNAL FREQUENCY RESPONSE 6 0 G = +2 RF = 1.43kΩ –3 G = +4 RF = 1.37kΩ –6 –9 G = +8 RF = 1.21kΩ –12 –15 G = –8 RF = 1.18kΩ 0 G = –1 RF = 1.43kΩ –3 –6 G = –2 RF = 1.40kΩ –9 G = –4 RF = 1.30kΩ –12 –15 –18 –18 1M 10M 100M 1M 10M 100M Frequency (Hz) Frequency (Hz) NON-INVERTING LARGE-SIGNAL FREQUENCY RESPONSE INVERTING LARGE-SIGNAL FREQUENCY RESPONSE 6 6 G = +8 VO = 0.5Vp-p 3 G = –8 0 Normalized Gain (dB) 3 VO = 2Vp-p –3 VO = 8Vp-p –6 VO = 16Vp-p –9 –12 –15 VO = 0.5Vp-p VO = 2Vp-p 0 –3 VO = 16Vp-p –6 –9 –12 VO = 8Vp-p –15 –18 –18 10M 100M 1M 10M Frequency (Hz) Frequency (Hz) NON-INVERTING PULSE RESPONSE INVERTING PULSE RESPONSE G = –8 Output Voltage (2V/div) VO = 0.5Vp-p Small Signal Output Voltage (100mV/div) G = +8 VO = 20Vp-p Large Signal VO = 20Vp-p Large Signal VO = 0.5Vp-p Small Signal Time (100ns/div) Time (100ns/div) ® OPA2607 100M 4 Output Voltage (100mV/div) 1M Output Voltage (2V/div) Normalized Gain (dB) VO = 0.5Vp-p 3 Normalized Gain (dB) Normalized Gain (dB) 6 G = +1 RF = 1.50kΩ VO = 0.5Vp-p 3 INVERTING SMALL-SIGNAL FREQUENCY RESPONSE TYPICAL PERFORMANCE CURVES: VS = ±12V (Cont.) At TA = +25°C, G = +8, RF = 1.21kΩ, and RL = 100Ω, unless otherwise noted. See Figure 1 for AC performance only. HARMONIC DISTORTION vs OUTPUT VOLTAGE HARMONIC DISTORTION vs FREQUENCY –50 –50 f = 1MHz Harmonic Distortion (dBc) Harmonic Distortion (dBc) VO = 2Vp-p –60 –70 3rd-Harmonic 2nd-Harmonic –80 –60 2nd-Harmonic –70 –80 3rd-Harmonic –90 –90 100k 1M 0.1 10M 1 Frequency (Hz) HARMONIC DISTORTION vs NON-INVERTING GAIN –50 VO = 2Vp-p f = 1MHz Harmonic Distortion (dBc) VO = 2Vp-p f = 1MHz Harmonic Distortion (dBc) 20 HARMONIC DISTORTION vs INVERTING GAIN –50 –60 –70 3rd-Harmonic –80 2nd-Harmonic –90 –60 –70 3rd-Harmonic –80 2nd-Harmonic –90 0.1 10 0.1 10 Gain Magnitude (V/V) Gain Magnitude (–V/V) HARMONIC DISTORTION vs DUAL SUPPLY VOLTAGE HARMONIC DISTORTION vs LOAD RESISTANCE –50 –50 VO = 2Vp-p fO = 1MHz Harmonic Distortion (dBc) Harmonic Distortion (dBC) 10 Output Voltage (Vp-p) –60 3rd-Harmonic –70 2nd-Harmonic –80 VO = 2Vp-p fO = 1MHz –60 –70 3rd-Harmonic –80 2nd-Harmonic –90 –90 10 100 6 1000 8 10 12 14 16 Dual Supply Voltage (±V) Load Resistance (Ω) ® 5 OPA2607 TYPICAL PERFORMANCE CURVES: VS = ±12V (Cont.) At TA = +25°C, G = +8, RF = 1.21kΩ, and RL = 100Ω, unless otherwise noted. See Figure 1 for AC performance only. TWO-TONE,3RD-ORDER INTERMODULATION SPURIOUS –50 3rd-Order Spurious Level (dBc) Load Power at Matched 50Ω Load –60 –70 fO = 2MHz –80 fO = 1MHz fO ≤ 500kHz –90 –5 0 5 10 15 20 Single-Tone Load Power (dBm) APPLICATIONS INFORMATION WIDEBAND CURRENT FEEDBACK OPERATION The OPA2607 gives the exceptional AC performance of a wideband current feedback op amp with a highly linear, high power output stage. Requiring only 8.0mA/chan quiescent current, the OPA2607 will swing to within 2V of either supply rail and deliver in excess of 180mA guaranteed at room temperature. This low output headroom requirement, along with supply-voltage independent biasing, gives remarkable single (+15V) supply operation. Previous boosted output stage amplifiers have typically suffered from very poor crossover distortion as the output current goes through zero. The OPA2607 achieves a comparable power gain with much better linearity. The primary advantage of a currentfeedback op amp over a voltage-feedback op amp is that AC performance (bandwidth and distortion) is relatively independent of signal gain. 0.1µF 2.2µF + 50Ω Source VI 49.9Ω VO 49.9Ω 1/2 OPA2607 50Ω Load RF 1.21kΩ RG 169Ω Figure 1 shows the DC-coupled, gain of +8, dual powersupply circuit configuration used as the basis of the ±12V Specifications and Typical Performance Curves. For test purposes, the input impedance is set to 50Ω with a resistor to ground and the output impedance is set to 50Ω with a series output resistor. Voltage swings reported in the specifications are taken directly at the input and output pins while load powers (dBm) are defined at a matched 50Ω load. For the circuit of Figure 1, the total effective load will be 100Ω || 1379Ω = 93Ω. + 2.2µF 0.1µF –VS –12V FIGURE 1. DC-Coupled, G = +8, Bipolar Supply, Specification and Test Circuit. ® OPA2607 +12V +VS 6 DESIGN-IN TOOLS DEMONSTRATION BOARDS VI Several PC boards are available to assist in the initial evaluation of circuit performance using the OPA2607 in its 3 package styles. All are available free as an unpopulated PC board delivered with descriptive documentation. The summary information for these boards is shown in Table I. α VO RI IERR PRODUCT PACKAGE DEMO BOARD NUMBER OPA2607U OPA2607N OPA2607H SO-8 SO-14 SO-Cool SO-8 SO-Cool DEM-OPA268xU DEM-OPA2607N DEM-OPA2607H ORDERING NUMBER MKT-352 MKT-367 MKT-366 Z(S) IERR RF RG TABLE I. FIGURE 2. Current-Feedback Transfer Function Analysis Circuit. Contact the Burr-Brown applications support line to request any of these boards. The buffer gain is typically very close to 1.00 and is normally neglected from signal gain considerations. It will, however set the CMRR for a single op amp differentialamplifier configuration. For a buffer gain α < 1.0, the CMRR = –20 • log (1– α) dB. MACROMODELS AND APPLICATIONS SUPPORT Computer simulation of circuit performance using SPICE is often useful when analyzing the performance of analog circuits and systems. This is particularly true for video and RF amplifier circuits where parasitic capacitance and inductance can have a major effect on circuit performance. SPICE models for some op amps are available through the BurrBrown web site (http://www.burr-brown.com). These models do a good job of predicting small-signal AC and transient performance under a wide variety of operating conditions. They do not do as well in predicting the harmonic distortion, dG/dP, or temperature characteristics. These models do not attempt to distinguish between the package types in their small-signal AC performance, nor do they attempt to simulate channel-to-channel coupling. RI, the buffer output impedance, is a critical portion of the bandwidth control equation. The OPA2607 has an RI typically about 33Ω. A current-feedback op amp senses an error current in the inverting node (as opposed to a differential input error voltage for a voltage-feedback op amp) and passes this on to the output through an internal frequency dependent transimpedance gain. The typical performance curves show this open-loop transimpedance response. This is analogous to the open-loop voltage gain curve for a voltage-feedback op amp. Developing the transfer function for the circuit of Figure 2 gives Equation 1: OPERATING SUGGESTIONS SETTING RESISTOR VALUES TO OPTIMIZE BANDWIDTH A current-feedback op amp like the OPA2607 can hold an almost constant bandwidth over signal gain settings with the proper adjustment of the external resistor values. This is shown in the Typical Performance Curves; the small-signal bandwidth decreases only slightly with increasing gain. Those curves also show that the feedback resistor has been changed for each gain setting. The resistor “values” on the inverting side of the circuit for a current-feedback op amp can be treated as frequency- response compensation elements while their “ratios” set the signal gain. Figure 2 shows the smallsignal frequency-response analysis circuit for the OPA2607. VO = VI 1+ R α 1 + F R α NG G = R F 1 + R F + R I NG R F + R I 1 + Z (S) RG Z (S) (1) RF NG ≡ 1 + R G This is written in a loop-gain analysis format where the errors arising from a finite open-loop gain are shown in the denominator. If Z(s) were infinite over all frequencies, the denominator of Equation 1 would reduce to 1 and the ideal desired signal gain shown in the numerator would be achieved. The fraction in the denominator of Equation 1 determines the frequency response. Equation 2 shows this as the loopgain equation: The key elements of this current feedback op amp model are: α → Buffer Gain from the Non-inverting Input to the Inverting Input RI → Buffer Output Impedance Z (S) iERR → Feedback Error Current Signal R F + R I NG Z(s) → Frequency Dependent Open Loop Transimpedance Gain from iERR to VO = Loop Gain (2) ® 7 OPA2607 If 20 • log (RF + NG • RI) were drawn on top of the openloop transimpedance plot, the difference between the two would be the loop gain at a given frequency. Eventually, Z(s) rolls off to equal the denominator of Equation 2 at which point the loop gain has reduced to 1 (and the curves have intersected). This point of equality is where the amplifier’s closed-loop frequency response given by Equation 1 will start to roll off, and is exactly analogous to the frequency at which the noise gain equals the open-loop voltage gain for a voltage-feedback op amp. The difference here is that the total impedance in the denominator of Equation 2 may be controlled somewhat separately from the desired signal gain (or NG). The total impedance going into the inverting input may be used to adjust the closed-loop signal bandwidth. Inserting a series resistor between the inverting input and the summing junction will increase the feedback impedance (denominator of Equation 2), decreasing the bandwidth. The internal buffer output impedance for the OPA2607 is slightly influenced by the source impedance looking out of the noninverting input terminal. High-source resistors will have the effect of increasing RI, decreasing the bandwidth. For those single-supply applications which develop a midpoint bias at the non-inverting input through high-valued resistors, the decoupling capacitor is essential for power-supply ripple rejection, non-inverting input-noise current shunting, and to minimize the high frequency value for RI in Figure 2. The OPA2607 is internally compensated to give a maximally flat frequency response for RF = 1.21kΩ at NG = 8 on ±12V supplies. Evaluating the denominator of Equation 2 (which is the feedback transimpedance) gives an optimal target of 1.44kΩ. As the signal gain changes, the contribution of the NG x RI term in the feedback transimpedance will change, but the total can be held constant by adjusting RF. Equation 3 gives an approximate equation for optimum RF over signal gain: R F = 1441Ω – NG R I INVERTING AMPLIFIER OPERATION Since the OPA2607 is a wideband, current-feedback op amp, most of the familiar op amp application circuits are available to the designer. Those dual op amp applications that require considerable flexibility in the feedback element (e.g. integrators, transimpedance, some filters) should consider the unity gain stable voltage-feedback OPA2680, since the feedback resistor is the compensation element for a current-feedback op amp. Wideband inverting operations (and especially summing) are particularly suited to the OPA2607. Figure 4 shows a typical inverting configuration where the I/O impedances are 50Ω. (3) As the desired signal gain increases, this equation will eventually predict a negative RF. A somewhat subjective limit to this adjustment can also be set by holding RG to a minimum value of 20Ω. Lower values will load both the buffer stage at the input and the output stage if RF gets too low—actually decreasing the bandwidth. Figure 3 shows the recommended RF versus NG. The values for RF versus Gain shown here are approximately equal to the values used to generate the typical performance curves. They differ in that the optimized values used in the typical performance curves are also correcting for board parasitics not considered in the simplified analysis leading to Equation 3. The values shown in Figure 3 give a good starting point for design where bandwidth optimization is desired. +12V Power supply de-coupling not shown 50Ω Load 1/2 OPA2607 50Ω Source VO 49.9Ω RF 1.21kΩ RG 150Ω VI RM 75.0Ω FEEDBACK RESISTOR vs NOISE GAIN –12V 1600 FIGURE 4. Inverting Gain of –8 with Impedance Matching. Feedback Resistor, RF (Ω) 1400 1200 In the inverting configuration, two key design considerations must be noted. The first is that the gain resistor (RG) becomes part of the signal-channel input impedance. If input impedance matching is desired (which is beneficial whenever the signal is coupled through a cable, twisted pair, long PC board trace, or other transmission line conductor), it is normally necessary to add an additional matching resistor (RM) to ground. RG by itself is normally not set to the required input impedance since its value, along with the desired gain, will determine an RF which may be nonoptimal from a frequency response standpoint. The total input impedance for the source becomes the parallel combination of RG and RM. 1000 800 600 400 200 0 0 5 10 15 20 Noise Gain, NG (V/V) FIGURE 3. Recommended Feedback Resistor vs Noise Gain. ® OPA2607 8 NOISE PERFORMANCE The second major consideration, touched on in the previous paragraph, is that the signal-source impedance becomes part of the noise-gain equation and will have slight effect on the bandwidth through Equation 1. The values shown in Figure 4 have accounted for this by slightly decreasing RF to reoptimize the bandwidth for the noise gain. In the example of Figure 4, the RM value combines in parallel with the external 50Ω source impedance, yielding an effective driving impedance of 50Ω || 75Ω = 30.0Ω. This impedance is added in series with RG for calculating the noise gain, which gives NG = 7.72 (instead of NG = 9.00 with a 0Ω source). This value, along with the RF of Figure 3 and the inverting input impedance of 33Ω, are inserted into Equation 3 to get RF = 1186Ω. Wideband current-feedback op amps generally have a higher output noise than comparable voltage-feedback op amps. The OPA2607 offers an excellent balance between voltage and current noise terms to achieve low output noise. The inverting current noise (15pA/√Hz) is significantly lower than competitive solutions, while the input voltage noise (1.7nV/√Hz) is lower than most unity gain stable, wideband, voltage-feedback op amps. This low input voltage noise was achieved at the price of higher non-inverting input current noise (11pA/√Hz). As long as the AC source impedance looking out of the noninverting node is less than 100Ω, this current noise will not contribute significantly to the total output noise. The op amp input voltage noise and the two input current noise terms combine to give low output noise under a wide variety of operating conditions. Figure 5 shows the op amp noise analysis model with all the noise terms included. In this model, all noise terms are taken to be noise voltage or current density terms in either nV/√Hz or pA/√Hz. Note that the non-inverting input in this bipolar-supply inverting application is connected directly to ground. It is often suggested that an additional resistor be connected to ground on the non-inverting input to achieve bias-current error cancellation at the output. The input bias currents for a current-feedback op amp are not generally matched in either magnitude or polarity. Connecting a resistor to ground on the non-inverting input of the OPA2607 in the circuit of Figure 4 will actually provide additional gain for that input’s bias and noise currents, but will not decrease the output DC error since the input bias currents are not matched. ENI 1/2 OPA2607 RS EO IBN ERS OUTPUT CURRENT AND VOLTAGE The OPA2607 provides outstanding output voltage and current capabilities. Under no-load conditions at 25°C, the output voltage typically swings within 0.8V of either supply rail; the guaranteed swing limit is within 1.1V of either rail. Into a 5Ω load (the minimum tested load), it is guaranteed to deliver more than ±180mA. RF ÷ 4kTRS IBI RG 4kT RG ÷ 4kTRF 4kT = 1.6 x 10–20J at 290°K FIGURE 5. Op Amp Noise Analysis Model. DISTORTION PERFORMANCE The OPA2607 provides good distortion performance into a 100Ω load on ±12V supplies. Relative to alternative solutions, it provides exceptional performance into lighter loads. Increasing the load impedance improves distortion directly. Remember that the total load includes the feedback network— in the non-inverting configuration (Figure 1) this is the sum of RF + RG, while in the inverting configuration it is just RF. Also, providing an additional supply decoupling capacitor (0.1µF) between the supply pins (for bipolar operation) improves the 2nd-order distortion slightly (3 to 6dB). The total output spot noise voltage can be computed as the square root of the sum of all squared output noise voltage contributors. Equation 4 shows the general form for the output noise voltage using the terms shown in Figure 5. (4) EO = (E 2 NI ) + (I BN R S ) + 4 kTR S NG 2 + (I BI R F ) + 4 kTR F NG 2 2 Dividing this expression by the noise gain (NG = (1+RF /RG )) will give the equivalent input referred spot noise voltage at the non-inverting input as shown in Equation 5. In most op amps, increasing the output voltage swing increases harmonic distortion directly. As the typical performance curves show, the spurious intermodulation powers do not increase as predicted by a traditional intercept model. As the fundamental power level increases, the dynamic range does not decrease significantly. For 2 tones centered at 1MHz, with 10dBm/tone into a matched 50Ω load (i.e. 2Vp-p for each tone at the load, which requires 8Vp-p for the overall 2-tone envelope at the output pin), the typical performance curves show 85dBc difference between the test-tone power and the 3rd-order intermodulation spurious levels. (5) 2 I R 4 kTR F 2 E N = E NI 2 + (I BN R S ) + 4 kTR S + BI F + NG NG Evaluating these two equations for the OPA2607 circuit and component values shown in Figure 1 will give a total output spot noise voltage of 27nV/√Hz and a total equivalent input spot noise voltage of 3.4nV/√Hz. This total input referred spot noise voltage is higher than the 1.7nV/√Hz specifica® 9 OPA2607 tion for the op amp voltage noise alone. This reflects the noise added to the output by the inverting current noise times the feedback resistor. If the feedback resistor is reduced in high-gain configurations (as suggested previously), the total input referred voltage noise given by Equation 5 will approach just the 1.7nV/√Hz of the op amp itself. For example, going to a gain of +20 using RF = 750Ω will give a total input referred noise of 2.0nV/√Hz . The ADSL Upstream Driver shown on the front page will be used as an example of these calculations. The ADSL (G.DMT) standard uses a spectrally-efficient coding technique, which produces a near-Gaussian output voltage distribution. Under these conditions, IO(AVE) = 0.8 IO(RMS). The maximum allowed crest factor in this standard is CF = 5.33Vpk/Vrms. We now calculate for each amplifier individually: (9) 348Ω 100Ω / 2 1.21kΩ + R L = 78.7Ω + 2 2 0.8 DC ACCURACY AND OFFSET CONTROL A current-feedback op amp like the OPA2607 provides exceptional bandwidth in high gains, giving fast pulse settling but only moderate DC accuracy. The typical specifications show an input offset voltage comparable to high-speed voltage-feedback amplifiers. However, the two input bias currents are somewhat higher and are unmatched. Whereas bias current cancellation techniques are very effective with most voltage-feedback op amps, they do not generally reduce the output DC offset for wideband current-feedback op amps. Since the two input bias currents are unrelated in both magnitude and polarity, matching the source impedance looking out of each input to reduce their error contribution to the output is ineffective. Evaluating the configuration of Figure 1, using worst case +25°C input offset voltage and the two input bias currents, gives a worst case output offset range equal to: = 141Ω I O( PK ) = I O( AVE ) (10) 2 • (12 V • 20.2 mA) – (25.3mA) • 141Ω 2 = 0.69 W TJ = 85°C + 0.69 W • 50°C / W = 120°C The junction temperature of this example is well below 175°C absolute maximum because the PSO-8 power package has such a low thermal impedance when properly connected to the –VS power plane (see the Board Layout Guidelines section). To help illustrate this point, the regular SO-8 package (OPA2607U) gives TJ = 171°C under the same conditions. = ±56.0mV ± 2.4mV ± 48.4mV = ±107mV THERMAL ANALYSIS Maximum desired junction temperature will set the maximum allowed internal power dissipating. In no case should the maximum junction temperature exceed 175°C. The operating junction temperature is given by: BOARD LAYOUT GUIDELINES Achieving optimum performance with a high-frequency amplifier like the OPA2607 requires careful attention to board layout parasitics and external component types. Recommendations that will optimize performance include: (6) a) Minimize parasitic capacitance to any AC ground for all of the signal I/O pins. Parasitic capacitance on the output and inverting input pins can cause instability. On the noninverting input it can react with the source impedance to cause unintentional bandlimiting. To reduce unwanted capacitance, a window around the signal I/O pins should be opened in all of the ground and power planes around those pins. Otherwise, ground and power planes should be unbroken elsewhere on the board. where TA is the ambient temperature, PD is the average power dissipation as calculated below, and θJA is the package thermal resistance shown in the specifications. The total internal power dissipation of a single amplifier, assuming bipolar supplies (±VS), is: (7) where IQ is the quiescent supply current, IO(AVE) is the average output current, IO(RMS) is the root-mean-square output current, and RL is the load seen by the output. Under absolute worst case conditions, with VO = VS/2, Equation 7 becomes: b) Minimize the distance (< 0.25") from the power supply pins to high frequency 0.1µF decoupling capacitors. At the device pins, the ground and power plane layout should not be in close proximity to the signal I/O pins. Avoid narrow power and ground traces to minimize inductance between the pins and the decoupling capacitors. The power supply (8) ® OPA2607 = PD = 2 • 12 V • 16 mA + = ± (8 • 7mV) ± (12µA • 25Ω • 8) ± (40µA • 1.21kΩ) PD = 2 VS I Q + VS2 / ( 4 R L ) I O( PK ) 38Vp − p / 2 = 135mA 141Ω Now calculate the typical junction temperature of both channels of the OPA2607H (PSO-8 Package) based on Equations 6 and 7: where NG = non-inverting signal gain PD = 2 VS I Q + VS I O( AVE ) – I O2 ( RMS) R L RL = 135mA = 25.3mA CF 5.33 = 0.8 I O( RMS) = 20.2 mA I O( RMS) = ± (NG • VOS(MAX)) ± (IBN • RS /2 • NG) ± (IBI • RF) TJ = TA + PD θ JA VO( PK ) 10 the destination device. Remember also that the terminating impedance will be the parallel combination of the shunt resistor and the input impedance of the destination device: this total effective impedance should be set to match the trace impedance. The high output voltage and current capability of the OPA2607 allows multiple destination devices to be handled as separate transmission lines, each with their own series and shunt terminations. If the 6dB attenuation of a doubly-terminated transmission line is unacceptable, a long trace can be series-terminated at the source end only. Treat the trace as a capacitive load in this case and set the series resistor value as shown in the plot of “RS vs Capacitive Load”. This will not preserve signal integrity as well as a doubly-terminated line. If the input impedance of the destination device is low, there will be some signal attenuation due to the voltage divider formed by the series output into the terminating impedance. connections (on pins 4 and 7) should always be decoupled with these capacitors. An optional supply decoupling capacitor across the two power supplies (for bipolar operation) will improve 2nd harmonic distortion performance. Larger (2.2µF to 6.8µF) decoupling capacitors, effective at lower frequency, should also be used on the main supply pins. These may be placed somewhat farther from the device and may be shared among several devices in the same area of the PC board. c) Careful selection and placement of external components will preserve the high-frequency performance of the OPA2607. Resistors should be a very low reactance type. Surface-mount resistors work best and allow a tighter overall layout. Metal film and carbon composition axially leaded resistors can also provide good high-frequency performance. Again, keep their leads and PC board trace length as short as possible. Never use wirewound-type resistors in a high-frequency application. Since the output pin and inverting input pin are the most sensitive to parasitic capacitance, always position the feedback and series output resistor, if any, as close as possible to the output pin. Other network components, such as non-inverting input termination resistors, should also be placed close to the package. Where double-side component mounting is allowed, place the feedback resistor directly under the package on the other side of the board between the output and inverting input pins. The frequency response is primarily determined by the feedback resistor value as described previously. Increasing its value will reduce the bandwidth, while decreasing it will give a more peaked frequency response. The 1.21kΩ feedback resistor used in the typical performance specifications at a gain of +8 on ±12V supplies is a good starting point for design. Note that a 1.50kΩ feedback resistor, rather than a direct short, is recommended for the unity-gain follower application. A current-feedback op amp requires a feedback resistor even in the unity-gain follower configuration to control stability. e) Do not socket a high speed part like the OPA2607. The additional lead length and pin-to-pin capacitance introduced by the socket can create an extremely troublesome parasitic network which can make it almost impossible to achieve a smooth, stable frequency response. Best results are obtained by soldering the OPA2607 onto the board. f) Use the –VS plane to conduct heat out of the PSO-8 and PSO-14 Power Packages (OPA2607H and OPA2607N). These packages attach the die directly to a metal slug in its bottom, which you should solder to the board. This slug needs to be connected electrically to the negative supply plane, which must have a minimum area of 2" x 2" (50mm x 50mm) to produce the θJA values in the specifications table. More details will be found in the data sheets that accompany the demo boards described in the Demonstration Boards section of this data sheet. INPUT AND ESD PROTECTION The OPA2607 is built using a very high-speed complementary bipolar process. All device pins have ESD protection using internal diodes to the power supplies as shown in Figure 6. The OPA2607 has an ESD rating of 4000V human body model, and 300V machine model. d) Connections to other wideband devices on the board may be made with short direct traces or through on-board transmission lines. For short connections, consider the trace and the input to the next device as a lumped capacitive load. Relatively wide traces (50 to 100mils) should be used, preferably with ground and power planes opened up around them. Estimate the total capacitive load and set RS from the plot of recommended “RS vs Capacitive Load”. Low parasitic capacitive loads (< 5pF) may not need an RS since the OPA2607 is nominally compensated to operate with a 2pF parasitic load. If a long trace is required, and the 6dB signal loss intrinsic to a doubly-terminated transmission line is acceptable, implement a matched-impedance transmission line using microstrip or stripline techniques (consult an ECL design handbook for microstrip and stripline layout techniques). A 50Ω environment is normally not necessary on board, and in fact a higher-impedance environment will improve distortion as shown in the “Distortion vs Load” plots. With a characteristic board-trace impedance defined based on board material and trace dimensions, a matching series resistor into the trace from the output of the OPA2607 is used as well as a terminating shunt resistor at the input of These diodes provide moderate protection to input overdrive voltages above the supplies as well. The protection diodes can typically support 30mA continuous current. Where higher currents are possible (e.g. in systems with ±30V supply parts driving into the OPA2607), current-limiting series resistors should be added into the two inputs. Keep these resistor values as low as possible since high values degrade both noise performance and frequency response. +V CC External Pin Internal Circuitry –V CC FIGURE 6. Internal ESD Protection. ® 11 OPA2607 PACKAGE OPTION ADDENDUM www.ti.com 9-Dec-2004 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty Lead/Ball Finish MSL Peak Temp (3) OPA2607H OBSOLETE HSOP DTJ 8 None Call TI Call TI OPA2607H/2K5 OBSOLETE HSOP DTJ 8 None Call TI Call TI OPA2607U OBSOLETE SOIC D 8 None Call TI Call TI OPA2607U/2K5 OBSOLETE SOIC D 8 None Call TI Call TI (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - May not be currently available - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. None: Not yet available Lead (Pb-Free). Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Green (RoHS & no Sb/Br): TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens, including bromine (Br) or antimony (Sb) above 0.1% of total product weight. (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. 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