LX1681/1682 V O LTA G E - M O D E P W M C O N T R O L L E R S T H E I N F I N I T E P O W E R I O F P N N O VA T I O N D R O D U C T I O N DESCRIPTION S H E E T K E Y F E AT U R E S ■ Fixed 200kHz Switching Frequency ■ Constant Frequency Voltage-Mode Control Requires NO External Compensation ■ Hiccup-Mode Over-Current Protection ■ High Efficiency ■ Output Voltage Set By Resistor Divider ■ Under-Voltage Lockout ■ Soft-Start And Enable ■ Synchronous Rectification (LX1682) ■ Non-Synchronous Rectification (LX1681) ■ Small, 8-pin Surface Mount Package Hiccup-mode fault protection reduces average power to the power elements during short-circuit conditions. Switching frequency is fixed at 200kHz for optimal cost and space. Under-voltage lockout and soft-start for optimal start-up performance. The LX1681/82 can be disabled by pulling the softstart pin to ground. Small 8-pin SOIC packaging reduces board space. Optimized for 5V-to-3.3V or 5V-to-2.5V conversion, the LX1681/82 can also be used for converting 12V to 5V, 3.3V or other voltages with high efficiency, eliminating the need for bulky heat sinks. The LX1681/1682 are monolithic, pulsewidth modulator controller ICs. They are designed to implement a flexible, low cost buck (step-down) regulator supply with minimal external components. The LX1681 is a non-synchronous controller; the LX1682 has a synchronous driver for higher efficiency. The output voltage is adjustable by means of a resistor divider to set the voltage between 1.25V and 4.5V. Short-circuit current limiting can be implemented without expensive current sense resistors. Current is sensed using the voltage drop across the RDS(ON) of the MOSFET — sensing is delayed for 1µs to eliminate MOSFET ringing errors. A T A A P P L I C AT I O N S ■ 5V to 3.3V Or Less Buck Regulators ■ FPGA Supplies ■ Microprocessor Chipset Supplies (e.g. Camino, Whitney, etc.) ■ Rambus® RIMMTM Supplies ■ Hard Disk Drives ■ Computer Add-on Cards NOTE: For current data & package dimensions, visit our web site: http://www.linfinity.com. PRODUCT HIGHLIGHT VBOOST 12V 1 2 VFB VCC SS CS 0.1µF 3 4 N.C. VC1 GND TDRV 1 8 CSS 2 7 0.1µF LX1681 6 Q1 IRL3103S VOUT L1 5µH 5 RSET VIN 5V C1 1500µFx2 C3 1µF C1 1500µFx3 C3 1µF CSS VBOOST 12V VIN 5V C2 1500µF x3 D2 MBR2545 R1 VCC VFB SS CS 8 7 LX1682 3 4 GND VC1 BDRV TDRV 6 Q1 IRL3103S L1 5µH 5 RSET R2 LX1681 NON-SYNCHRONOUS CONTROLLER VOUT Q2 IRL3103S C2 1500µF x3 R1 R2 LX1682 SYNCHRONOUS CONTROLLER PA C K A G E O R D E R I N F O R M A T I O N TA (°C) 0 to 70 Output SOIC DM Plastic 8-pin Non-Synchronous LX1681CDM Synchronous LX1682CDM Note: All surface-mount packages are available in Tape & Reel, append the letter "T" to part number. (i.e. LX1681CDMT) Copyright © 1999 Rev. 1.0 5/99 LINFINITY MICROELECTRONICS INC. 11861 WESTERN AVENUE, GARDEN GROVE, CA. 92841, 714-898-8121, FAX: 714-893-2570 1 PRODUCT DATABOOK 1996/1997 LX1681/1682 V O LTA G E - M O D E P W M C O N T R O L L E R S P R O D U C T I O N ABSOLUTE MAXIMUM RATINGS D A T A S H E E T (Note 1) Supply Voltage (VC1) ............................................................................................ 18V Supply Voltage (VCC) .............................................................................................. 7V Output Drive Peak Current Source (500ns) ....................................................... 1.0A Output Drive Peak Current Sink (500ns) ........................................................... 1.0A Input Voltage (SS/ENABLE Pin) ............................................................. -0.3V to 6V Operating Junction Temperature .................................................................... 150°C Storage Temperature Range ........................................................... -65°C to +150°C Lead Temperature (Soldering, 10 Seconds) .................................................... 300°C Note 1. Exceeding these ratings could cause damage to the device. All voltages are with respect to Ground. Currents are positive into, negative out of the specified terminal. PACKAGE PIN OUTS VFB SS N.C. / GND* GND / BDRV* 1 8 2 7 3 6 4 5 VCC CS VC1 TDRV DM PACKAGE (Top View) * Pin 3 = N.C. for LX1681, GND for LX1682 Pin 4 = GND for LX1681, BDRV for LX1682 THERMAL DATA DM PACKAGE: THERMAL RESISTANCE-JUNCTION TO AMBIENT, θJA 165°C/W Junction Temperature Calculation: TJ = TA + (PD x θJA). The θJA numbers are guidelines for the thermal performance of the device/pc-board system. All of the above assume no ambient airflow. ELECTRICAL CHARACTERISTICS (Unless otherwise specified, 4.75V < VCC < 5.25V and 10.8V < VC1 < 13.2V, 0°C ≤ TA ≤ 70°C. Test conditions: VCC = 5V, VC1 = 12V, T = 25°C.) Parameter Symbol Test Conditions LX1681/1682 Min. Typ. Max. Units Reference Reference Voltage VFB VOUT = VFB , TA = 25°C VOUT = VFB , 0°C ≤ TA ≤ 70°C 1.237 1.231 1.25 1.262 1.269 V V 170 190 1.25 230 kHz VPP Oscillator Frequency Ramp Amplitude FOSC VRAMP Error Amplifier Input Resistance RIN VOUT = VFB ISET VCS = VCC - 0.4V Reference to VCC 20 kΩ Current Sense Current Set VTRIP Current Sense Delayed 40 -460 TCSD 45 -400 1.1 -340 µA mV µsec Output Drivers Drive Rise Time, Fall Time Drive High Drive Low TRF VDH VDL CL = 3000pF ISOURCE = 20mA, VC1 = 12V ISINK = 20mA, VC1 = 12V 10 VST VC1 > 4.0V 4.0 50 11 0.1 ns V V 0.2 UVLO and Soft-Start (SS) VCC5 Start-Up Threshold Hysteresis SS Resistor SS Output Enable Hiccup Duty Cycle RSS VEN DCHIC 0.25 CSS = 0.1µF, FREQ = 100Hz 4.25 0.10 20 0.3 10 4.5 0.35 V V kΩ V % Supply Current VCC12 Dynamic Supply Current Static Supply Current 12V 5V 2 ICD IVC1 IVCC Out Freq = 200kHz, CL = 3000pF, Synch., VSS > 0.3V VSS < 0.3V VSS < 0.3V 24 5 10 28 7 12 mA mA mA Copyright © 1999 Rev. 1.0 5/99 PRODUCT DATABOOK 1996/1997 LX1681/1682 V O LTA G E - M O D E P W M C O N T R O L L E R S P R O D U C T I O N D A T A S H E E T BLOCK DIAGRAM ISET RSET 7 CS +12V CS Comp IRESET VTRIP 6 PWM VCC R Q S Q ISET 5 320k VFB 1 20k Error Comp Set 4 VIN (5V) VC1 TDRV BDRV CIN R2 R1 L VCORE ESR COUT VRESET Amplifier/ Compensation 3 GND VREF RSS +5V Hiccup 2 Hiccup Ramp Oscillator UVLO UVLO 8 VCC SS/ENABLE CSS FIGURE 1 — Block Diagram Copyright © 1999 Rev. 1.0 5/99 3 PRODUCT DATABOOK 1996/1997 LX1681/1682 V O LTA G E - M O D E P W M C O N T R O L L E R S P R O D U C T I O N D A T A S H E E T FUNCTIONAL PIN DESCRIPTION Pin Name 4 Description Pin Number LX1681 LX1682 VFB Voltage feedback — 1.25V reference is connected to a resistor divider to set desired output voltage. 1 1 SS Soft-start and hiccup capacitor pin. During start up the voltage of this pin controls the output voltage. An internal 20kΩ resistor and the external capacitor set the time constant for soft-startup. Soft-start does not begin until the supply voltage exceeds the UVLO threshold. When over-current occurs, this capacitor is used for timing hiccup. The PWM can be disabled by pulling the SS pin below 0.3V. 2 2 GND Ground for IC 4 3 TDRV Gate drive for upper MOSFET 5 4 BDRV Gate drive for lower MOSFET 5 VC1 Separate supply for MOSFET gate drive. Connect to 12V 6 6 CS Over-current set. Connect resistor between CS pin and the source of the upper MOSFET to set current-limit point. 7 7 VCC IC supply voltage (nominal 5V) and high side drain sense voltage. 8 8 Copyright © 1999 Rev. 1.0 5/99 PRODUCT DATABOOK 1996/1997 LX1681/1682 V O LTA G E - M O D E P W M C O N T R O L L E R S P R O D U C T I O N D A T A S H E E T T H E O R Y O F O P E R AT I O N GENERAL DESCRIPTION OVER-CURRENT PROTECTION (OCP) and HICCUP The LX1681/82 are voltage-mode pulse-width modulation controller integrated circuits. The internal oscillator and ramp generator frequency is fixed at 200kHz. The devices have internal compensation, so that no external compensation is required. The LX1681/1682 family uses the RDS(ON) of the upper MOSFET, together with a resistor (RSET) to set the actual current limit point. The comparator senses the current 1µs after the top MOSFET is switched on. Experiments have shown that the MOSFET drain voltage will ring for 200-500ns after the gate is turned on. In order to reduce inaccuracies due to ringing, a 1µs delay after gate turn-on is built into the current sense comparator. The comparator draws a current (ISET), whose magnitude is 45µA. The set resistor is selected to set the current limit for the application. When the sensed voltage across the RDS(ON) plus the set resistor exceeds the 400mV VTRIP threshold, the OCP comparator outputs a signal to reset the PWM latch and to start hiccup mode. The soft-start capacitor (CSS) is discharged slowly (10 times slower than when being charged up by RSS). When the voltage on the SS/ENABLE pin reaches a 0.3V threshold, hiccup finishes and the circuit soft-starts again. During hiccup, the top MOSFET is OFF and the bottom MOSFET remains ON. Hiccup is disabled during the soft-start interval, allowing the circuit to start up with the maximum current. If the rise speed of the output voltage is too fast, the required charging current to the output capacitor may be higher than the limit-current. In this case, the peak MOSFET current is regulated to the limit-current by the current-sense comparator. If the MOSFET current still reaches its limit after the soft-start finishes, the hiccup is triggered again. The hiccup ensures the average heat generation on both MOSFET’s and the average current to be much less than that in normal operation, if the output has a short circuit. Over-current protection can also be implemented using a sense resistor, instead of using the RDS(ON) of the upper MOSFET, for greater set-point accuracy. See Application Information section. POWER UP and INITIALIZATION At power up, the LX1681/82 monitors the supply voltage to both the +5V and the +12V pins (there is no special requirement for the sequence of the two supplies). Before both supplies reach their under-voltage lock-out (UVLO) thresholds, the soft-start (SS) pin is held low to prevent soft-start from beginning; the oscillator control is disabled and the top MOSFET is kept OFF. SOFT-START Once the supplies are above the UVLO threshold, the soft-start capacitor begins to be charged up by the reference through a 20kΩ internal resistor. The capacitor voltage at the SS pin rises as a simple RC circuit. The SS pin is connected to the amplifier's non-inverting input that controls the output voltage. The output voltage will follow the SS pin voltage if sufficient charging current is provided to the output capacitor. The simple RC soft-start allows the output to rise faster at the beginning and slower at the end of the soft-start interval. Thus, the required charging current into the output capacitor is less at the end of the soft-start interval so decreasing the possibility of an over-current. A comparator monitors the SS pin voltage and indicates the end of soft-start when SS pin voltage reaches 95% of VREF. OSCILLATOR FREQUENCY An internal oscillator sets the switching frequency at 200 kHz. Copyright © 1999 Rev. 1.0 5/99 5 PRODUCT DATABOOK 1996/1997 LX1681/1682 V O LTA G E - M O D E P W M C O N T R O L L E R S P R O D U C T I O N D A T A S H E E T A P P L I C AT I O N I N F O R M AT I O N OUTPUT INDUCTOR OUTPUT CAPACITOR The output inductor should be selected to meet the requirements of the output voltage ripple in steady-state operation and the inductor current slew-rate during transient. The peak-to-peak output voltage ripple is: Electrolytic capacitors can be used for the output capacitor, but are less stable with age than tantalum capacitors. As they age, their ESR degrades, reducing the system performance and increasing the risk of failure. It is recommended that multiple parallel capacitors be used, so that, as ESR increases with age, overall performance will still meet the processor’s requirements. There is frequently strong pressure to use the least expensive components possible, however, this could lead to degraded long-term reliability, especially in the case of filter capacitors. Linfinity’s demonstration boards use Sanyo MV-GX filter capacitors, which are aluminum electrolytic, and have demonstrated reliability. The Oscon series from Sanyo generally provides the very best performance in terms of long term ESR stability and general reliability, but at a substantial cost penalty. The MV-GX series provides excellent ESR performance at a reasonable cost. Beware of off-brand, very low-cost filter capacitors, which have been shown to degrade in both ESR and general electrolytic characteristics over time. VRIPPLE = ESR * IRIPPLE where IRIPPLE = (VIN - VOUT ) fSW * L * VOUT VIN IRIPPLE is the inductor ripple current, L is the output inductor value and ESR is the Effective Series Resistance of the output capacitor. IRIPPLE should typically be in the range of 20% to 40% of the maximum output current. Higher inductance results in lower output voltage ripple, allowing slightly higher ESR to satisfy the transient specification. Higher inductance also slows the inductor current slew rate in response to the load-current step change, ∆I, resulting in more output-capacitor voltage droop. The inductor-current rise and fall times are: TRISE = L * ∆I/(VIN – VOUT ) and TFALL = L * ∆I/VOUT When using electrolytic capacitors, the capacitor voltage droop is usually negligible, due to the large capacitance. OUTPUT CAPACITOR The output capacitor is sized to meet ripple and transient performance specifications. Effective Series Resistance (ESR) is a critical parameter. When a step load current occurs, the output voltage will have a step that equals the product of the ESR and the current step, ∆I. In an advanced microprocessor power supply, the output capacitor is usually selected for ESR instead of capacitance or RMS current capability. A capacitor that satisfies the ESR requirement usually has a larger capacitance and current capability than strictly needed. The allowed ESR can be found by: ESR * (IRIPPLE + ∆I ) < VEX where IRIPPLE is the inductor ripple current, ∆I is the maximum load current step change, and VEX is the allowed output voltage excursion in the transient. 6 (continued) INPUT CAPACITOR The input capacitor and the input inductor are to filter the pulsating current generated by the buck converter to reduce interference to other circuits connected to the same 5V rail. In addition, the input capacitor provides local de-coupling the buck converter. The capacitor should be rated to handle the RMS current requirement. The RMS current is: IRMS = IL √ d(1-d) where IL is the inductor current and the d is the duty cycle. The maximum value, when d = 50%, IRMS = 0.5IL. For 5V input and output in the range of 2 to 3V, the required RMS current is very close to 0.5IL. SOFT-START CAPACITOR The value of the soft-start capacitor determines how fast the output voltage rises and how large the inductor current is required to charge the output capacitor. The output voltage will follow the voltage at SS pin if the required inductor current does not exceed the maximum current in the inductor. The SS pin voltage can be expressed as: VSS = VSET (1-e-t/RssCss) where VSET is the reference voltage. RSS and CSS are soft start resistor and capacitor. The required inductor current for the output capacitor to follow the SS-pin voltage equals the required Copyright © 1999 Rev. 1.0 5/99 PRODUCT DATABOOK 1996/1997 LX1681/1682 V O LTA G E - M O D E P W M C O N T R O L L E R S P R O D U C T I O N D S A T A H E E T A P P L I C AT I O N I N F O R M AT I O N SOFT-START CAPACITOR OUTPUT ENABLE (continued) capacitor current plus the load current. The soft-start capacitor should be selected so that the overall inductor current does not exceed it maximum. The capacitor current to follow the SS-pin voltage is: ICout = COUT dV dt = COUT *e CSS -(t/RssCss ) where COUT is the output capacitance. The typical value of CSS should be in the range of 0.1 to 0.2µF. During the soft-start interval the load current from a microprocessor is negligible; therefore, the capacitor current is approximately the required inductor current. OVER-CURRENT PROTECTION ICL * RDS(ON) + ISET * RSET = VTRIP RSET = VTRIP - ICL * RDS(ON) ISET = 400mV - ICL * RDS(ON) 45µA Example: For 10A current limit, using IRL3303 MOSFET (26mΩ RDS(ON)): RSET = 0.4 - 10 * 0.026 45 * 10-6 Example: For 10A current limit, using a 5mΩ sense resistor: Copyright © 1999 Rev. 1.0 5/99 The output voltage is sensed by the feedback pin (VFB) which has a 1.25V reference. The output voltage can be set to any voltage above 1.25V (and lower than the input voltage) by means of a resistor divider (see Product Highlight). VOUT = VREF (1 + R1 /R2 ) Note: Keep R1 and R2 close to 100Ω (order of magnitude). To insure reliable operation, the operating junction temperature of the FET switches must be kept below certain limits. The Intel specification states that 115°C maximum junction temperature should be maintained with an ambient of 50°C. This is achieved by properly derating the part, and by adequate heat sinking. One of the most critical parameters for FET selection is the RDS(ON) resistance. This parameter directly contributes to the power dissipation of the FET devices, and thus impacts heat sink design, mechanical layout, and reliability. In general, the larger the current handling capability of the FET, the lower the RDS(ON) will be, since more die area is available. = 3.1kΩ Current Sensing Using Sense Resistor The method of current sensing using the RDS(ON) of the upper MOSFET is economical, but can have a large tolerance, since the RDS(ON) can vary with temperature, etc. A more accurate alternative is to use an external sense resistor (RSENSE). Since one input to the current sense comparator is the supply voltage to the IC (VCC - pin 8), the sense resistor could be a PCB trace (for construction details, see Application Note AN-10 or LX1668 data sheet). The over-current trip point is calculated as in the equations above, replacing RDS(ON) with RSENSE. RSET PROGRAMMING THE OUTPUT VOLTAGE FET SELECTION Current limiting occurs at current level ICL, when the voltage detected by the current sense comparator is greater than the current sense comparator threshold, VTRIP (400mV). So, The LX1681/82 FET driver outputs are driven to ground by pulling the soft-start pin below 0.3V. V - (ICL * RSENSE ) 0.4 - 10 * 0.005 = TRIP = = 7.8kΩ ISET 45 x 10-6 TABLE 1 - FET Selection Guide This table gives selection of suitable FETs from International Rectifier. Device RDS(ON) @ Ω) 10V (mΩ ID @ TC = 100°C Max. Breakdown Voltage IRL3803 IRL22203N IRL3103 IRL3102 IRL3303 IRL2703 6 7 14 13 26 40 83 71 40 56 24 17 30 30 30 20 30 30 All devices in TO-220 package. For surface mount devices (TO-263 / D2-Pak), add 'S' to part number, e.g. IRL3103S. Heat Dissipated In Upper MOSFET The heat dissipated in the top MOSFET will be: PD = (I2 * RDS(ON) * Duty Cycle) + (0.5 * I * VIN * tSW * fS ) Where tSW is switching transition line for body diode (~100ns) and fS is the switching frequency. For the IRL3102 (13mΩ RDS(ON)), converting 5V to 2.0V at 15A will result in typical heat dissipation of 1.92W. 7 PRODUCT DATABOOK 1996/1997 LX1681/1682 V O LTA G E - M O D E P W M C O N T R O L L E R S P R O D U C T I O N D A T A S H E E T A P P L I C AT I O N I N F O R M AT I O N FET SELECTION (continued) Synchronous Rectification – Lower MOSFET The lower pass element can be either a MOSFET or a Schottky diode. The use of a MOSFET (synchronous rectification) will result in higher efficiency, but at higher cost than using a Schottky diode (non-synchronous). Power dissipated in the bottom MOSFET will be: PD = I2 * RDS(ON) * [1 - Duty Cycle] = 3.51W 5V Input LX168x Output [IRL3303 or 1.76W for the IRL3102] Non-Synchronous Operation - Schottky Diode A typical Schottky diode, with a forward drop of 0.6V will dissipate 0.6 * 15 * [1 – 2/5] = 5.4W (compared to the 1.8 to 3.5W dissipated by a MOSFET under the same conditions). This power loss becomes much more significant at lower duty cycles. The use of a dual Schottky diode in a single TO-220 package (e.g. the MBR2535) helps improve thermal dissipation. Operation From A Single Power Supply The LX1681/1682 needs a secondary supply voltage (VC1) to provide sufficient drive to the upper MOSFET. In many applications with a 5V (VCC) and a 12V (VC1) supply are present. In situations where only 5V is present, VC1 can be generated using a bootstrap (charge pump) circuit, as shown in Figure 4 (Typical Applications section). The capacitor (C4) is alternatively charged up from VCC via the Schottky diode (D2), and then boosted up when the FET is turned on. This scheme provedes a VC1 voltage equal to 2 * VCC - VDS(D2), or approximately 9.5V with VCC = 5V. This voltage will provide sufficient gate drive to the external MOSFET in order to get a low RDS(ON). Note that using the bootstrap circuit in synchronous rectification mode is likely to result in faster turn-on than in nonsynchronous mode. LAYOUT GUIDELINES - THERMAL DESIGN A great deal of time and effort were spent optimizing the thermal design of the demonstration boards. Any user who intends to implement an embedded motherboard would be well advised to carefully read and follow these guidelines. If the FET switches have been carefully selected, external heatsinking is generally not required. However, this means that copper trace on the PC board must now be used. This is a potential trouble spot; as much copper area as possible must be dedicated to heatsinking the FET switches, and the diode as well if a non-synchronous solution is used. In our VRM module, heatsink area was taken from internal ground and VCC planes which were actually split and connected with VIAS to the power device tabs. The TO-220 and TO-263 cases are well suited for this application, and are the preferred packages. Remember to remove any conformal coating from all exposed PC traces which are involved in heatsinking. 8 GND FIGURE 2 — Enabling Linear Regulator General Notes As always, be sure to provide local capacitive decoupling close to the chip. Be sure use ground plane construction for all highfrequency work. Use low ESR capacitors where justified, but be alert for damping and ringing problems. High-frequency designs demand careful routing and layout, and may require several iterations to achieve desired performance levels. Power Traces To reduce power losses due to ohmic resistance, careful consideration should be given to the layout of traces that carry high currents. The main paths to consider are: ■ Input power from 5V supply to drain of top MOSFET. ■ Trace between top MOSFET and lower MOSFET or Schottky diode. ■ Trace between lower MOSFET or Schottky diode and ground. ■ Trace between source of top MOSFET and inductor and load. All of these traces should be made as wide and thick as possible, in order to minimize resistance and hence power losses. It is also recommended that, whenever possible, the ground, input and output power signals should be on separate planes (PCB layers). See Figure 2 – bold traces are power traces. Layout Assistance Please contact Linfinity’s Applications Engineers for assistance with any layout or component selection issues. A Gerber file with layout for the most popular devices is available upon request. Evaluation boards are also available upon request. Please check Linfinity's web site for further application notes. Copyright © 1999 Rev. 1.0 5/99 PRODUCT DATABOOK 1996/1997 LX1681/1682 V O LTA G E - M O D E P W M C O N T R O L L E R S P D R O D U C T I O N S A T A H E E T A P P L I C AT I O N I N F O R M AT I O N VBOOST 12V VIN 5V C3 1 CSS 2 VFB VCC SS CS C1 8 7 LX1681 0.1µF 3 4 N.C. VC1 GND TDRV RSENSE RSET 6 VOUT Q1 5 L1 C2 R1 D2 R2 FIGURE 3 — Current Sensing Using Sense Resistor VIN 5V C1 1 CSS 2 0.1µF VFB VCC SS CS 8 D2 7 LX1682 3 4 GND VC1 BDRV TDRV C4 6 VOUT 5 L1 Q1 C2 RSET Q2 R1 R2 FIGURE 4 — Bootstrap Circuit For 5V Only Operation PRODUCTION DATA - Information contained in this document is proprietary to LinFinity, and is current as of publication date. This document may not be modified in any way without the express written consent of LinFinity. Product processing does not necessarily include testing of all parameters. Linfinity reserves the right to change the configuration and performance of the product and to discontinue product at any time. Copyright © 1999 Rev. 1.0 5/99 9