TI TPS5401

TPS5401
www.ti.com
SLVSAB0 – DECEMBER 2010
0.5-A, 42-V Input, Step-Down SWIFT™ Converter
Check for Samples: TPS5401
FEATURES
•
•
•
1
•
•
•
2
•
•
•
•
•
•
•
3.5-V to 42-V Input Voltage Range
200-mΩ High-Side MOSFET
High Efficiency at Light Loads With a
Pulse-Skipping Eco-mode™ Control Scheme
116-mA Operating Quiescent Current
1.3-mA Shutdown Current
100-kHz to 2.5-MHz Switching Frequency
Synchronizes to External Clock
Adjustable Slow Start/Sequencing
UV and OV Power-Good Output
Adjustable UVLO Voltage and Hysteresis
•
0.8-V ±3.5% Internal Voltage Reference
MSOP10 With PowerPAD™ Package
Supported by SwitcherPro™ Software Tool
(http://focus.ti.com/docs/toolsw/folders/print/s
witcherpro.html)
For SWIFT™ Power Products Documentation,
See the TI Web Site at http://www.ti.com/swift
APPLICATIONS
•
•
12-V and 24-V Industrial and Commercial
Low-Power Systems
E-Meters
DESCRIPTION
The TPS5401 device is a 42-V, 0.5-A, step-down regulator with an integrated high-side MOSFET. Current-mode
control provides simple external compensation and flexible component selection. A low-ripple pulse-skip mode
reduces the supply current to 116 mA when outputting regulated voltage with no load. Using the enable pin,
shutdown supply current is reduced to 1.3 mA when the enable pin is low.
Undervoltage lockout is internally set at 2.5 V, but can be increased using the enable pin. The output voltage
start-up ramp is controlled by the slow-start pin that can also be configured for sequencing/tracking. An
open-drain power-good signal indicates the output is within 94% to 107% of its nominal voltage.
A wide switching-frequency range allows efficiency and external component size optimization. Frequency
foldback and thermal shutdown protect the part during an overload condition.
The TPS5401 is available in a 10-pin thermally enhanced MSOP PowerPAD package.
SPACER
SIMPLIFIED SCHEMATIC
VIN
VIN
CIN
EFFICIENCY
vs
LOAD CURRENT
PWRGD
TPS5401
100
90
CBOOT
BOOT
80
LOUT
PH
VOUT
SS/TR
COUT
RT/CLK
COMP
RT
R3
R1
C1
60
50
40
30
VSENSE
GND
VIN = 12 V
VOUT = 5 V
fSW = 700 kHz
20
R2
C2
CSS
70
Efficiency – %
EN
10
0
0
0.1
0.2
0.3
Load Current – A
0.4
0.5
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Eco-mode, PowerPAD, SwitcherPro, SWIFT are trademarks of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2010, Texas Instruments Incorporated
TPS5401
SLVSAB0 – DECEMBER 2010
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
Table 1. ORDERING INFORMATION (1)
(1)
(2)
TJ
PACKAGE
PART NUMBER (2)
–40°C to 150°C
10-pin MSOP
TPS5401DGQ
For the most-current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
Web site at www.ti.com.
The DGQ package is also available taped and reeled. Add an R suffix to the device type (i.e., TPS5401DGQR).
ABSOLUTE MAXIMUM RATINGS (1)
Over operating temperature range (unless otherwise noted).
VALUE
UNIT
MIN
MAX
VIN
–0.3
47
EN
–0.3
5
BOOT
Input voltage
55
VSENSE
–0.3
3
COMP
–0.3
3
PWRGD
–0.3
6
SS/TR
–0.3
3
RT/CLK
–0.3
3.6
–0.6
47
–2
47
BOOT – PH
Output voltage
8
PH
PH, 10-ns transient
Voltage difference
Source current
V
Thermal pad to GND
±200
EN
100
mA
BOOT
100
mA
10
mA
VSENSE
PH
Current limit
RT/CLK
mV
A
100
VIN
Sink current
V
Current limit
mA
A
COMP
100
mA
PWRGD
10
mA
SS/TR
200
mA
1
kV
Electrostatic discharge (HBM) QSS 009-105 (JESD22-A114A)
Electrostatic discharge (CDM) QSS 009-147 (JESD22-C101B.01)
500
V
Operating junction temperature
–40
150
°C
Storage temperature
–65
150
°C
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
THERMAL INFORMATION
THERMAL METRIC (1)
qJA
Junction-to-ambient thermal resistance
qJC(top)
Junction-to-case(top) thermal resistance
(1)
2
TPS5401
MSOP (10 PINS)
UNIT
68
°C/W
63.3
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
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THERMAL INFORMATION (continued)
TPS5401
THERMAL METRIC (1)
MSOP (10 PINS)
UNIT
qJB
Junction-to-board thermal resistance
44.6
°C/W
yJT
Junction-to-top characterization parameter
1.4
°C/W
yJB
Junction-to-board characterization parameter
22.8
°C/W
qJC(bottom)
Junction-to-case(bottom) thermal resistance
25.6
°C/W
ELECTRICAL CHARACTERISTICS
TJ = –40°C to 150°C, VIN = 3.5 V to 42 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE (VIN PIN)
VIN
Operating input voltage
3.5
42
Internal undervoltage lockout
threshold
No voltage hysteresis, rising and falling
2.5
Shutdown supply current
VEN = 0 V
1.3
4
Operating: nonswitching supply
current
VVSENSE = 0.83 V, VIN = 12 V, 25°C
116
136
1.25
1.55
V
V
mA
ENABLE AND UVLO (EN PIN)
Enable threshold voltage
Input current
No voltage hysteresis, rising and falling, 25°C
0.9
Enable threshold 50 mV
–3.8
Enable threshold –50 mV
–0.9
Hysteresis current
V
mA
–2.9
mA
VOLTAGE REFERENCE
Vref
Voltage reference
0.772
0.8
0.828
V
HIGH-SIDE MOSFET
On-resistance
VIN = 3.5 V, VBOOT – VPH = 3 V
300
VIN = 12 V, VBOOT – VPH = 6 V
200
410
mΩ
ERROR AMPLIFIER
Input current
gmEA
50
nA
Error amplifier transconductance
–2 mA < ICOMP < 2 mA, VCOMP = 1 V
97
mMhos
Error amplifier transconductance
during slow-start
–2 mA < ICOMP < 2 mA, VCOMP = 1 V,
VVSENSE = 0.4 V
26
mMhos
Error amplifier dc gain
VVSENSE = 0.8 V
Error amplifier bandwidth
Error amplifier source/sink
gmPS
VCOMP = 1 V, 100 mV overdrive
COMP to switch current
transconductance
10,000
V/V
2700
kHz
±7
mA
1.9
A/V
0.94
A
182
°C
CURRENT LIMIT
Current limit threshold
VIN = 12 V, TJ = 25°C
0.6
THERMAL SHUTDOWN
Thermal shutdown
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ELECTRICAL CHARACTERISTICS (continued)
TJ = –40°C to 150°C, VIN = 3.5 V to 42 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
2500
kHz
720
kHz
2200
kHz
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK PIN)
Switching-frequency range using RT
mode
fSW
Switching frequency
100
RT = 200 kΩ
450
Switching-frequency range using
CLK mode
581
300
Minimum CLK input pulse width
40
RT/CLK high threshold
1.9
RT/CLK low threshold
0.5
RT/CLK falling edge to PH rising
edge delay
Measured at 500 kHz with RT resistor in series
PLL lock-in time
Measured at 500 kHz
ns
2.2
V
0.7
V
60
ns
100
ms
SLOW-START AND TRACKING (SS/TR PIN)
Charge current
VSS/TR = 0.4 V
2
mA
SS/TR-to-VSENSE matching
VSS/TR = 0.4 V
45
mV
SS/TR-to-reference crossover
98% nominal
SS/TR discharge current (overload)
VVSENSE = 0 V, VSS/TR = 0.4 V
SS/TR discharge voltage
1
V
112
mA
VVSENSE = 0 V
54
mV
VVSENSE falling
92%
VVSENSE rising
94%
VVSENSE rising
109%
VVSENSE falling
107%
Hysteresis
VVSENSE falling
2%
Output-high leakage
VVSENSE = Vref, VPWRGD = 5.5 V, 25°C
On-resistance
IPWRGD = 3 mA, VVSENSE < 0.77 V
50
Minimum VIN for defined output
VPWRGD < 0.5 V, IPWRGD = 100 mA
0.95
POWER GOOD (PWRGD PIN)
VSENSE low threshold
VSENSE high threshold
4
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Vref
Vref
10
nA
Ω
1.5
V
Copyright © 2010, Texas Instruments Incorporated
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SLVSAB0 – DECEMBER 2010
DEVICE INFORMATION
PIN CONFIGURATION
MSOP10
(TOP VIEW)
BOOT
1
VIN
2
10
Thermal
Pad
(11)
PH
9
GND
8
COMP
EN
3
SS/TR
4
7
VSENSE
RT/CLK
5
6
PWRGD
PIN FUNCTIONS
PIN
I/O
DESCRIPTION
NAME
NO.
BOOT
1
O
A bootstrap capacitor is required between BOOT and PH. If the voltage on this capacitor is below the
minimum required by the output device, the output is forced to switch off until the capacitor is refreshed.
COMP
8
O
Error amplifier output, and input to the output switch current comparator. Connect frequency compensation
components to this pin.
EN
3
I
Enable pin, internal pullup current source. Pull below 1.2 V to disable. Float to enable. Adjust the input
undervoltage lockout with two resistors.
GND
9
–
Ground
PH
10
I
The source of the internal high-side power MOSFET
PWRGD
6
O
An open-drain output; asserts low if output voltage is low due to thermal shutdown, dropout, overvoltage or
EN shutdown.
RT/CLK
5
I
Resistor timing and external clock. An internal amplifier holds this pin at a fixed voltage when using an
external resistor to ground to set the switching frequency. If the pin is pulled above the PLL upper threshold,
a mode change occurs and the pin becomes a synchronization input. The internal amplifier is disabled and
the pin is a high-impedance clock input to the internal PLL. If clocking edges stop, the internal amplifier is
re-enabled and the mode returns to a resistor-set function.
SS/TR
4
I
Slow-start and tracking. An external capacitor connected to this pin sets the output rise time. Because the
voltage on this pin overrides the internal reference, it can be used for tracking and sequencing.
Thermal pad
(11)
–
GND pin must be electrically connected to the thermal pad on the printed circuit board for proper operation.
VIN
2
I
Input supply voltage, 3.5 V to 42 V.
VSENSE
7
I
Inverting node of the transconductance (gm) error amplifier.
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FUNCTIONAL BLOCK DIAGRAM
PWRGD
6
EN
3
VIN
2
Shutdown
Thermal
Shutdown
UV
UVLO
Enable
Comparator
Logic
Shutdown
Shutdown
Logic
OV
Enable
Threshold
Boot
Charge
Voltage
Reference
Boot
UVLO
Minimum
Clamp
Pulse
Skip
ERROR
AMPLIFIER
PWM
Comparator
VSENSE 7
Current
Sense
1 BOOT
Logic
And
PWM Latch
SS/TR 4
Shutdown
Slope
Compensation
10 PH
COMP 8
Frequency
Shift
Overload
Recovery
Maximum
Clamp
Oscillator
with PLL
11
TPS5401 Block Diagram
Thermal
Pad
9 GND
5
RT/CLK
6
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SLVSAB0 – DECEMBER 2010
TYPICAL CHARACTERISTICS
VOLTAGE REFERENCE vs JUNCTION TEMPERATURE
0.816
500
VIN = 12 V
VIN = 12 V
375
Vref – Voltage Reference – V
rDS(on) – Static Drain-Source On-State Resistance – mW
ON-RESISTANCE vs JUNCTION TEMPERATURE
VBOOT – VPH = 3 V
250
VBOOT – VPH = 6 V
125
0
–50
–25
0
25
50
75
100
TJ – Junction Temperature – °C
125
0.808
0.800
0.792
0.784
–50
150
0
25
50
75
100
TJ – Junction Temperature – °C
125
150
Figure 1.
Figure 2.
SWITCH-CURRENT LIMIT vs JUNCTION TEMPERATURE
SWITCHING FREQUENCY vs JUNCTION TEMPERATURE
1.1
610
VIN = 12 V
RT = 200 kW
fSW – Switching Frequency – kHz
VIN = 12 V
1
Switch Current – A
–25
0.9
0.8
600
590
580
570
560
0.7
–50
–25
0
25
50
75
100
125
550
–50
150
–25
TJ – Junction Temperature – °C
0
25
50
75
100
TJ – Junction Temperature – °C
150
Figure 3.
Figure 4.
SWITCHING FREQUENCY vs RT/CLK RESISTANCE
HIGH-FREQUENCY RANGE
SWITCHING FREQUENCY vs RT/CLK RESISTANCE
LOW-FREQUENCY RANGE
2500
500
fSW – Switching Frequency – kHz
VIN = 12 V
TJ = 25°C
fSW – Switching Frequency – kHz
125
2000
1500
1000
500
0
0
25
50
75
100
125
150
RT/CLK – Clock Setting Resistance – kW
175
200
VIN = 12 V
TJ = 25°C
400
300
200
100
0
200
300
Figure 5.
400
500
600 700
800
900 1000 1100
RT/CLK – Clock Setting Resistance – kW
1200
Figure 6.
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TYPICAL CHARACTERISTICS (continued)
EA TRANSCONDUCTANCE DURING SLOW-START vs
JUNCTION TEMPERATURE
EA TRANSCONDUCTANCE vs JUNCTION TEMPERATURE
40
150
VIN = 12 V
gm – EA Transconductance – mA/V
gm – EA Transconductance – mA/V
VIN = 12 V
30
20
10
–50
–25
0
25
50
75
100
TJ – Junction Temperature – °C
125
130
110
90
70
50
–50
150
–25
0
25
50
Figure 7.
150
–3.25
VIN = 12 V
VEN = Threshold + 50 mV
IEN – EN Pin Current – mA
VIN = 12 V
Enable Threshold Voltage – V
125
EN PIN CURRENT vs JUNCTION TEMPERATURE
1.4
1.3
1.2
1.1
–50
–25
0
25
50
75
100
125
150
–3.5
–3.75
–4
–4.25
–50
–25
0
TJ – Junction Temperature – °C
25
50
75
100
125
150
TJ – Junction Temperature – °C
Figure 9.
Figure 10.
EN PIN CURRENT vs JUNCTION TEMPERATURE
SS/TR CHARGE CURRENT vs JUNCTION TEMPERATURE
–1
–0.8
VIN = 12 V
VEN = Threshold – 50 mV
VIN = 12 V
ISS/TR – SS/TR Pin Charge Current – mA
IEN – EN Pin Current – mA
100
Figure 8.
ENABLE THRESHOLD VOLTAGE vs JUNCTION
TEMPERATURE
–0.85
–0.9
–0.95
–1
–50
–25
0
75
25
50
100
TJ – Junction Temperature –°C
125
150
–1.5
–2
–2.5
–3
–50
Figure 11.
8
75
TJ – Junction Temperature – °C
–25
0
25
50
75
100
TJ – Junction Temperature – °C
125
150
Figure 12.
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TYPICAL CHARACTERISTICS (continued)
SS/TR DISCHARGE CURRENT vs JUNCTION
TEMPERATURE
SWITCHING FREQUENCY vs VSENSE
120
100
VIN = 12 V
TJ = 25°C
80
115
% of Nominal fsw
ISS/TR – SS/TR Pin Discharge Current – mA
VIN = 12 V
110
60
40
105
20
100
–50
0
–25
0
25
50
75
100
TJ – Junction Temperature – °C
125
150
0
Figure 14.
SHUTDOWN SUPPLY CURRENT vs JUNCTION
TEMPERATURE
SHUTDOWN SUPPLY CURRENT vs INPUT VOLTAGE
2
IVIN – Shutdown Supply Current – mA
IVIN – Shutdown Supply Current – mA
VIN = 12 V
1.5
1
0.5
TJ = 25°C
1.5
1
0.5
0
–25
0
25
50
75
100
TJ – Junction Temperature – °C
125
150
0
10
20
30
VIN – Input Voltage – V
Figure 15.
VIN SUPPLY CURRENT vs JUNCTION TEMPERATURE
VIN SUPPLY CURRENT vs INPUT VOLTAGE
140
VIN = 12 V
VVSENSE = 0.83 V
TJ = 25oC,
VVSENSE = 0.83 V
130
IVIN – VIN Supply Current – mA
IVIN – VIN Supply Current – mA
40
Figure 16.
140
120
110
100
90
–50
0.8
Figure 13.
2
0
–50
0.2
0.4
0.6
VVSENSE – Voltage at VSENSE Pin – V
130
120
110
100
90
–25
0
25
50
75
100
TJ – Junction Temperature – °C
125
150
0
Figure 17.
20
VIN – Input Voltage – V
40
Figure 18.
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TYPICAL CHARACTERISTICS (continued)
PWRGD ON-RESISTANCE vs JUNCTION TEMPERATURE
PWRGD THRESHOLD vs JUNCTION TEMPERATURE
115
100
VIN = 12 V
PWRGD Threshold – % of Vref
VIN = 12 V
rDS(on) – W
80
60
40
20
VVSENSE Rising
110
VVSENSE Falling
105
100
VVSENSE Rising
95
VVSENSE Falling
90
0
–50
–25
0
25
50
75
100
125
85
–50
150
–25
0
25
50
75
100
TJ – Junction Temperature – °C
BOOT to PH UVLO THRESHOLD vs JUNCTION
TEMPERATURE
INPUT VOLTAGE UVLO vs JUNCTION TEMPERATURE
3
2.3
2.75
2
1.8
2.5
2.25
–25
0
25
50
75
100
TJ – Junction Temperature – °C
125
2
–50
150
–25
0
25
50
75
100
TJ – Junction Temperature – °C
Figure 21.
SS/TR TO VSENSE OFFSET vs VSENSE VOLTAGE
150
SS/TR TO VSENSE OFFSET vs TEMPERATURE
60
VSS/TR = 0.2 V
VIN = 12 V
VIN = 12 V
TJ = 25oC
55
400
50
Offset – mV
500
300
45
200
40
100
35
0
0
125
Figure 22.
600
Offset – mV
150
Figure 20.
2.5
1.5
–50
200
400
600
800
1000
30
–50
VVSENSE – mV
–25
0
25
50
75
100
125
150
TJ – Junction Temperature – °C
Figure 23.
10
125
Figure 19.
Input Voltage UVLO – V
BOOT to PH UVLO Threshold Voltage – V
TJ – Junction Temperature – °C
Figure 24.
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OVERVIEW
The TPS5401 device is a 42-V, 0.5-A, step-down (buck) regulator with an integrated high-side n-channel
MOSFET. To improve performance during line and load transients, the device implements a constant-frequency,
current-mode control which reduces output capacitance and simplifies external frequency compensation design.
The wide switching-frequency range of 100 kHz to 2500 kHz allows for efficiency and size optimization when
selecting the output filter components. The switching frequency is adjusted using a resistor to ground on the
RT/CLK pin. The device has an internal phase-lock loop (PLL) on the RT/CLK pin that is used to synchronize the
power-switch turnon to a falling edge of an external system clock.
The TPS5401 has a default start-up voltage of approximately 2.5 V. The EN pin has an internal pullup current
source that can be used to adjust the input voltage undervoltage lockout (UVLO) threshold with two external
resistors. In addition, the pullup current provides a default condition. When the EN pin is floating, the device
operates. The operating current is 116 mA when not switching and under no load. When the device is disabled,
the supply current is 1.3 mA.
The integrated 200-mΩ high-side MOSFET allows for high-efficiency power-supply designs capable of delivering
0.5 amperes of continuous current to a load. The TPS5401 reduces the external component count by integrating
the boot recharge diode. The bias voltage for the integrated high-side MOSFET is supplied by a capacitor from
the BOOT pin to the PH pin. The boot capacitor voltage is monitored by a UVLO circuit and turns the high-side
MOSFET off when the boot voltage falls below a preset threshold. The TPS5401 can operate at high duty cycles
because of the boot UVLO. The output voltage can be stepped down to as low as the 0.8-V reference.
The TPS5401 has a power-good comparator (PWRGD) which asserts when the regulated output voltage is less
than 92% or greater than 109% of the nominal output voltage. The PWRGD pin is an open-drain output which
deasserts when the VSENSE pin voltage is between 94% and 107% of the nominal output voltage, allowing the
pin to transition high when a pullup resistor is used.
The TPS5401 minimizes excessive output overvoltage (OV) transients by taking advantage of the OV
power-good comparator. When the OV comparator is activated, the high-side MOSFET is turned off and masked
from turning on until the output voltage is lower than 107%.
The SS/TR (slow-start/tracking) pin is used to minimize inrush currents or provide power-supply sequencing
during power up. A small-value capacitor should be coupled to the pin to adjust the slow-start time. A resistor
divider can be coupled to the pin for critical power-supply sequencing requirements. The SS/TR pin is discharged
before the output powers up. This discharging ensures a repeatable restart after an overtemperature fault, UVLO
fault, or a disabled condition.
The TPS5401 also discharges the slow-start capacitor during overload conditions with an overload recovery
circuit. The overload recovery circuit slow-starts the output from the fault voltage to the nominal regulation
voltage once a fault condition is removed. A frequency-foldback circuit reduces the switching frequency during
start-up and overcurrent fault conditions to help control the inductor current.
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DETAILED DESCRIPTION
Fixed Frequency PWM Control
The TPS5401 uses an adjustable fixed-frequency, peak current mode control. The output voltage is compared
through external resistors on the VSENSE pin to an internal voltage reference by an error amplifier which drives
the COMP pin. An internal oscillator initiates the turnon of the high-side power switch. The error amplifier output
is compared to the high-side power-switch current. When the power-switch current reaches the level set by the
COMP voltage, the power switch is turned off. The COMP pin voltage increases and decreases as the output
current increases and decreases. The device implements a current limit by clamping the COMP pin voltage to a
maximum level. The Eco-mode control scheme is implemented with a minimum clamp on the COMP pin.
Slope Compensation Output Current
The TPS5401 adds a compensating ramp to the switch current signal. This slope compensation prevents
sub-harmonic oscillations. The available peak inductor current remains constant over the full duty-cycle range.
Pulse-Skip Eco-mode Control Scheme
The TPS5401 operates in a pulse-skip Eco-mode control scheme at light load currents to improve efficiency by
reducing switching and gate-drive losses. The TPS5401 is designed so that if the output voltage is within
regulation and the peak switch current at the end of any switching cycle is below the pulse-skipping current
threshold, the device enters the Eco-mode control scheme. This current threshold is the current level
corresponding to a nominal COMP voltage of 500 mV.
When in the Eco-mode control scheme, the COMP pin voltage is clamped at 500 mV, and the high-side
MOSFET is inhibited. Further decreases in load current or increases in output voltage cannot drive the COMP
pin below this clamp voltage level.
Because the device is not switching, the output voltage begins to decay. As the voltage control loop
compensates for the falling output voltage, the COMP pin voltage begins to rise. At this time, the high-side
MOSFET is enabled, and a switching pulse initiates on the next switching cycle. The peak current is set by the
COMP pin voltage. The output voltage recharges the regulated value (see Figure 25); then the peak switch
current starts to decrease and eventually falls below the Eco-mode control-scheme threshold, at which time the
device again enters the Eco-mode control scheme.
For Eco-mode control-scheme operation, the TPS5401 senses peak current, not average or load current, so the
load current where the device enters the Eco-mode control scheme is dependent on the output inductor value.
For example, the circuit in Figure 40 enters the Eco-mode control scheme at about 20 mA of output current.
When the load current is low and the output voltage is within regulation, the device enters a sleep mode and
draws only 116 mA of input quiescent current. The internal PLL remains operating when in sleep mode. When
operating at light load currents in the pulse-skip mode, the switching transitions occur synchronously with the
external clock signal.
12
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DETAILED DESCRIPTION (continued)
VOUTac
VPH
IL
Figure 25. Pulse-Skip Mode Operation
Low-Dropout Operation and Bootstrap Voltage (BOOT)
The TPS5401 has an integrated boot regulator, and requires a small ceramic capacitor between the BOOT and
PH pins to provide the gate-drive voltage for the high-side MOSFET. The BOOT capacitor is refreshed when the
high-side MOSFET is off and the low-side diode conducts. The value of this ceramic capacitor should be 0.1 mF.
A ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating of 10 V or higher is recommended
because of the stable characteristics over temperature and voltage.
To improve dropout, the TPS5401 is designed to operate at 100% duty cycle as long as the BOOT-to-PH pin
voltage is greater than 2.1 V. When the voltage from BOOT to PH drops below 2.1 V, the high-side MOSFET is
turned off using an UVLO circuit, which allows the low-side diode to conduct and refresh the charge on the
BOOT capacitor. Because the supply current sourced from the BOOT capacitor is low, the high-side MOSFET
can remain on for more switching cycles than are required to refresh the capacitor; thus, the effective duty cycle
of the switching regulator is high.
The effective duty cycle during dropout of the regulator is mainly influenced by the voltage drops across the
power MOSFET, inductor resistance, low-side diode, and printed circuit board resistance. During operating
conditions in which the input voltage drops and the regulator is operating in continuous conduction mode, the
high-side MOSFET can remain on for 100% of the duty cycle to maintain output regulation, until the BOOT-to-PH
voltage falls below 2.1 V.
Attention must be taken in maximum-duty-cycle applications which experience extended time periods with light
loads or no load. When the voltage across the BOOT capacitor falls below the 2.1-V UVLO threshold, the
high-side MOSFET is turned off, but there may not be enough inductor current to pull the PH pin down to
recharge the BOOT capacitor. The high-side MOSFET of the regulator stops switching because the voltage
across the BOOT capacitor is less than 2.1 V. The output capacitor then decays until the difference in the input
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DETAILED DESCRIPTION (continued)
voltage and output voltage is greater than 2.1 V, at which point the BOOT UVLO threshold is exceeded, and the
device starts switching again until the desired output voltage is reached. This operating condition persists until
the input voltage and/or the load current increases. It is recommended to adjust the VIN stop voltage greater
than the BOOT UVLO trigger condition at the minimum load of the application using the adjustable VIN UVLO
feature with resistors on the EN pin.
The start and stop voltages for typical 3.3-V and 5-V output applications are shown in Figure 26 and Figure 27.
The voltages are plotted versus load current. The start voltage is defined as the input voltage needed to regulate
the output within 3.5%. The stop voltage is defined as the input voltage at which the output drops by 5% or stops
switching.
During high-duty-cycle conditions, the inductor current ripple increases while the BOOT capacitor is being
recharged, resulting in an increase in ripple voltage on the output. This is due to the recharge time of the boot
capacitor being longer than the typical high-side off time, when switching occurs every cycle.
5.6
VOUT = 3.3 V
VOUT = 5 V
VIN – Input Voltage – V
VIN – Input Voltage – V
5.4
5.2
Start
5
Stop
4.8
4.6
0
0.05
IOUT – Output Current – A
Figure 26. 3.3-V Start/Stop Voltage
0.10
0.15
IOUT – Output Current – A
0.20
Figure 27. 5-V Start/Stop Voltage
Error Amplifier
The TPS5401 has a transconductance amplifier as the error amplifier. The error amplifier compares the VSENSE
voltage to the lower of the SS/TR pin voltage or the internal 0.8-V voltage reference. The transconductance (gm)
of the error amplifier is 97 mA/V during normal operation. During the slow-start operation, the transconductance is
a fraction of the normal operating gm. When the voltage of the VSENSE pin is below 0.8 V and the device is
regulating using the SS/TR voltage, the gm is 26 mA/V.
The frequency compensation components (series resistor and capacitor) are connected between the COMP pin
and ground.
Voltage Reference
The voltage reference system produces a precise ±3.5% voltage reference over temperature by scaling the
output of a temperature-stable bandgap circuit.
Adjusting the Output Voltage
The output voltage is set with a resistor divider from the output node to the VSENSE pin. It is recommended to
use 1% tolerance or better divider resistors. Start with 10 kΩ for the R2 resistor and use Equation 1 to calculate
R1. To improve efficiency at light loads, consider using larger-value resistors. If the values are too high, the
regulator becomes more susceptible to noise, and voltage errors from the VSENSE input current become
noticeable.
- 0.8 V ö
æV
R1 = R2 ´ ç OUT
÷
0.8
V
è
ø
(1)
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DETAILED DESCRIPTION (continued)
Enabling and Adjusting Undervoltage Lockout
The TPS5401 is disabled when the VIN pin voltage falls below 2.5 V. If an application requires a higher
undervoltage lockout (UVLO), use the EN pin as shown in Figure 28 to adjust the input voltage UVLO by using
the two external resistors. Though it is not necessary to use the UVLO adjust resistors, for operation it is highly
recommended to provide consistent power-up behavior. The EN pin has an internal pullup current source, I1, of
0.9 mA that provides the default condition of the TPS5401 operating when the EN pin floats. Once the EN pin
voltage exceeds the enable threshold voltage (VENA) of 1.25 V, an additional 2.9 mA of hysteresis, IHYS, is added.
This additional current facilitates input-voltage hysteresis. Use Equation 2 to set the external hysteresis for the
input voltage. Use Equation 3 to set the input start voltage.
TPS5401
VIN
I1
0.9 mA
R1
R2
IHYS
2.9 mA
+
EN
1.25 V
–
Figure 28. Adjustable Undervoltage Lockout (UVLO)
V
- VSTOP
R1 = START
IHYS
R2 =
(2)
VENA
VSTART - VENA
+ I1
R1
(3)
Slow-Start/Tracking Pin (SS/TR)
The TPS5401 effectively uses the lower voltage of the internal voltage reference or the SS/TR pin voltage as the
power-supply reference voltage and regulates the output accordingly. A capacitor on the SS/TR pin to ground
implements a slow-start time. The TPS5401 has an internal pullup current source of 2 mA that charges the
external slow-start capacitor. The calculations for the slow-start time (10% to 90%) are shown in Equation 4. The
voltage reference (Vref) is 0.8 V and the slow-start current (ISS) is 2 mA. The slow-start capacitor should remain
lower than 0.47 mF and greater than 0.47 nF.
t (ms) ´ Iss (m A)
Css (nF) = ss
Vref (V) ´ 0.8
(4)
At power up, the TPS5401 does not start switching until the slow-start pin is discharged to less than 40 mV to
ensure a proper power up; see Figure 29.
Also, during normal operation, the TPS5401 stops switching and SS/TR must be discharged to 40 mV when the
VIN UVLO is exceeded, the EN pin is pulled below 1.25 V, or a thermal shutdown event occurs.
The VSENSE voltage follows the SS/TR pin voltage with a 45-mV offset up to 85% of the internal voltage
reference. When the SS/TR voltage is greater than 85% of the internal reference voltage, the offset increases as
the effective system reference transitions from the SS/TR voltage to the internal voltage reference (see
Figure 23). The SS/TR voltage ramps linearly until clamped at 1.7 V.
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DETAILED DESCRIPTION (continued)
VEN
VSS/TR
VVSENSE
VOUT
Figure 29. Operation of SS/TR Pin When Starting
Overload Recovery Circuit
The TPS5401 has an overload recovery (OLR) circuit. The OLR circuit slow-starts the output from the overload
voltage to the nominal regulation voltage once the fault condition is removed. The OLR circuit discharges the
SS/TR pin to a voltage slightly greater than the VSENSE pin voltage using an internal pulldown of 100 mA when
the error amplifier is changed to a high voltage from a fault condition. When the fault condition is removed, the
output slow-starts from the fault voltage to nominal output voltage.
Sequencing
Many of the common power-supply sequencing methods can be implemented using the SS/TR, EN and PWRGD
pins. The sequential method can be implemented using an open-drain output of a power-on-reset pin of another
device. The sequential method is illustrated in Figure 30 using two TPS5401 devices. The power good is coupled
to the EN pin on the TPS5401, which enables the second power supply once the primary supply reaches
regulation. If needed, a 1-nF ceramic capacitor on the EN pin of the second power supply provides a 1-ms
start-up delay. Figure 31 shows the results of Figure 30.
16
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DETAILED DESCRIPTION (continued)
TPS5401
EN
PWRGD
EN
VEN1
SS /TR
SS /TR
VPWRGD1
PWRGD
VOUT1
VOUT2
Figure 30. Schematic for Sequential Start-Up
Sequence
Figure 31. Sequential Start-Up Using EN and
PWRGD
TPS54160
TPS5401
3
EN
4
SS/TR
6
PWRGD
VEN1, VEN2
VOUT1
TPS5401
TPS54160
3
EN
4
SS/TR
6
PWRGD
VOUT2
Figure 32. Schematic for Ratiometric Start-Up
Sequence
Figure 33. Ratiometric Start-Up Using Coupled
SS/TR Pins
Figure 32 shows a method for ratiometric start-up sequence by connecting the SS/TR pins together. The
regulator outputs ramp up and reach regulation at the same time. When calculating the slow-start time, the pullup
current source must be doubled in Equation 4. Figure 33 shows the results of Figure 32.
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DETAILED DESCRIPTION (continued)
TPS5401
EN
VOUT1
SS /TR
PWRGD
TPS5401
VOUT2
EN
R1
SS / TR
R2
PWRGD
VSENSE
R3
R4
Figure 34. Schematic for Ratiometric and Simultaneous Start-Up Sequence
Ratiometric and simultaneous power supply sequencing can be implemented by connecting the resistor network
of R1 and R2 shown in Figure 34 to the output of the power supply that must be tracked or to another voltage
reference source. Using Equation 5 and Equation 6, the tracking resistors can be calculated to initiate the VOUT2
slightly before, after, or at the same time as VOUT1. Equation 7 is the voltage difference between VOUT1 and VOUT2
at the 95% of nominal output regulation.
The ΔV variable is zero volts for simultaneous sequencing. To minimize the effect of the inherent SS/TR to
VSENSE offset (Vssoffset) in the slow-start circuit and the offset created by the pullup current source (Iss) and
tracking resistors, the Vssoffset and Iss are included as variables in the equations. Figure 37 shows the result when
ΔV = 0 V.
To design a ratiometric start-up in which the VOUT2 voltage is slightly greater than the VOUT1 voltage when VOUT2
reaches regulation, use a negative number in Equation 5 through Equation 7 for ΔV. Equation 7 results in a
positive number for applications in which VOUT2 is slightly lower than VOUT1 when VOUT2 regulation is achieved.
Figure 35 and Figure 36 show the start-up waveforms for negative and positive ΔV, respectively.
Because the SS/TR pin must be pulled below 40 mV before starting after an EN, UVLO, or thermal shutdown
fault, careful selection of the tracking resistors is needed to ensure the device can restart after a fault. Make sure
the calculated R1 value from Equation 5 is greater than the value calculated in Equation 8 to ensure the device
can recover from a fault.
As the SS/TR voltage becomes more than 85% of the nominal reference voltage, the Vssoffset becomes larger as
the slow-start circuits gradually hand off the regulation reference to the internal voltage reference. The SS/TR pin
voltage must be greater than 1.3 V for a complete handoff to the internal voltage reference as shown in
Figure 23.
V
+ DV
V
R1 = OUT2
× ssoffset
Vref
Iss
(5)
R2 =
18
Vref ´ R1
VOUT2 + DV - Vref
(6)
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DETAILED DESCRIPTION (continued)
DV = VOUT1 - VOUT2
(7)
R1 > 2800 ´ VOUT1 - 180 ´ DV
(8)
VEN
VEN
VOUT1
VOUT1
VOUT2
Figure 35. Ratiometric Start-Up With Vout2 Leading
Vout1
VOUT2
Figure 36. Ratiometric Start-Up With Vout1 Leading
Vout2
VEN
VOUT1
VOUT2
Figure 37. Simultaneous Start-Up
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DETAILED DESCRIPTION (continued)
Constant Switching Frequency and Timing Resistor (RT/CLK Pin)
The switching frequency of the TPS5401 is adjustable over a wide range from approximately 100 kHz to 2500
kHz by placing a resistor on the RT/CLK pin. The RT/CLK pin voltage is typically 0.5 V and must have a resistor
to ground to set the switching frequency. To determine the timing resistance for a given switching frequency, use
Equation 9 or the curves in Figure 5 or Figure 6. To reduce the solution size one would typically set the switching
frequency as high as possible, but tradeoffs of the supply efficiency, maximum input voltage, and minimum
controllable on-time should be considered.
The minimum controllable on-time is typically 130 ns and limits the maximum operating input voltage.
The maximum switching frequency is also limited by the frequency-shift circuit. More discussion on the details of
the maximum switching frequency is located as follows.
206,003
RT (kΩ) =
fSW (kHz)1.0888
(9)
Overcurrent Protection and Frequency Shift
The TPS5401 implements current-mode control, which uses the COMP pin voltage to turn off the high-side
MOSFET on a cycle-by-cycle basis. Each cycle, the switch current and COMP pin voltage are compared; when
the peak switch current intersects the COMP voltage, the high-side switch is turned off. During overcurrent
conditions that pull the output voltage low, the error amplifier responds by driving the COMP pin high, increasing
the switch current. The error amplifier output is clamped internally, which functions as a switch-current limit.
To increase the maximum operating switching frequency at high input voltages, the TPS5401 implements a
frequency shift. The switching frequency is divided by 8, 4, 2, and 1 as the voltage ramps from 0 to 0.8 volts on
the VSENSE pin.
The device implements a digital frequency shift to enable synchronizing to an external clock during normal
start-up and fault conditions. Because the device can only divide the switching frequency by 8, there is a
maximum input voltage limit in which the device operates and still has frequency-shift protection.
During short-circuit events (particularly with high-input-voltage applications), the control loop has a finite minimum
controllable on-time, and the output has a low voltage. During the switch on-time, the inductor current ramps to
the peak current limit because of the high input voltage and minimum on-time. During the switch off-time, the
inductor current would normally not have enough off-time and output voltage to ramp down by the ramp-up
amount. The frequency shift effectively increases the off-time, allowing the current to ramp down.
Selecting the Switching Frequency
The switching frequency that is selected should be the lower value of the two equations, Equation 10 and
Equation 11. Equation 10 is the maximum switching frequency limitation set by the minimum controllable on-time.
Setting the switching frequency above this value causes the regulator to skip switching pulses.
Equation 11 is the maximum switching frequency limit set by the frequency-shift protection. To have adequate
output short-circuit protection at high input voltages, the switching frequency should be set to be less than the
fSW(maxshift) frequency. In Equation 11, to calculate the maximum switching frequency, one must take into account
that the output voltage decreases from the nominal voltage to 0 volts and the fDIV integer increases from 1 to 8,
corresponding to the frequency shift.
In Figure 38, the solid line illustrates a typical safe operating area regarding frequency shift and assumes an
output voltage of zero volts, an inductor resistance of 0.13 Ω, FET on-resistance of 0.2 Ω, and a diode voltage
drop of 0.5 V. The dashed line is the maximum switching frequency to avoid pulse skipping. Enter these
equations in a spreadsheet or other software or use the SwitcherPro design software to determine the switching
frequency.
æ 1 ö æ IL ´ Rdc + VOUT + Vd ö
fSW(maxskip) = ç
÷
÷´ç
è t ON ø çè VIN - IL ´ rDS(on) + Vd ÷ø
(10)
æf
ö æ I ´ Rdc + VOUTSC + Vd
fSW(shift) = ç DIV ÷ ´ ç L
è t ON ø çè VIN - IL ´ rDS(on) + Vd
20
ö
÷
÷
ø
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DETAILED DESCRIPTION (continued)
IL
Inductor current
Rdc
Inductor resistance
VIN
Maximum input voltage
VOUT
Output voltage
VOUTSC
Output voltage during short
Vd
Diode voltage drop
rDS(on)
Switch on resistance
tON
Controllable on-time
ƒDIV
Frequency divide equals (1, 2, 4, or 8)
fSW – Switching Frequency – kHz
2500
Skip
2000
1500
Shift
1000
500
VOUT = 5 V
0
10
20
30
VIN – Input Voltage – V
40
Figure 38. Maximum Switching Frequency vs. Input Voltage
How to Interface to RT/CLK Pin
The RT/CLK pin can be used to synchronize the regulator to an external system clock. To implement the
synchronization feature connect a square wave to the RT/CLK pin as shown in Figure 39. The square-wave
amplitude must transition lower than 0.5 V and higher than 2.2 V on the RT/CLK pin and have an on-time greater
than 40 ns and an off-time greater than 40 ns. The synchronization frequency range is 300 kHz to 2200 kHz. The
rising edge of the PH signal is synchronized to the falling edge of the RT/CLK pin signal.
TPS5401
RT/CLK
PLL
Clock
Source
RT
Figure 39. Synchronizing to a System Clock
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DETAILED DESCRIPTION (continued)
Power Good (PWRGD Pin)
The PWRGD pin is an open-drain output. Once the VSENSE pin is between 94% and 107% of the internal
voltage reference, the PWRGD pin is de-asserted and the pin floats. It is recommended to use a pullup resistor
between the values of 10 kΩ and 100 kΩ to a voltage source that is 5.5 V or less. PWRGD is in a defined state
once the VIN input voltage is greater than 1.5 V, but with reduced current-sinking capability. PWRGD achieves
full current-sinking capability as the VIN input voltage approaches 3 V.
The PWRGD pin is pulled low when VSENSE is lower than 92% or greater than 109% of the nominal internal
reference voltage. Also, PWRGD is pulled low if the UVLO or thermal shutdown is asserted or the EN pin is
pulled low.
Overvoltage Transient Protection
The TPS5401 incorporates an overvoltage transient protection (OVTP) circuit to minimize voltage overshoot
when recovering from output fault conditions or strong unload transients on power-supply designs with low-value
output capacitance. For example, when the power-supply output is overloaded, the error amplifier compares the
actual output voltage to the internal reference voltage. If the VSENSE pin voltage is lower than the internal
reference voltage for a considerable time, the output of the error amplifier responds by clamping the
error-amplifier output to a high voltage, thus requesting the maximum output current. Once the condition is
removed, the regulator output rises and the error amplifier output transitions to the steady-state duty cycle. In
some applications, the power-supply output voltage can respond faster than the error-amplifier output can
respond; this actuality leads to the possibility of an output overshoot. The OVTP feature minimizes the output
overshoot, when using a low-value output capacitor, by implementing a circuit to compare the VSENSE pin
voltage to OVTP threshold, which is 109% of the internal voltage reference. If the VSENSE pin voltage is greater
than the OVTP threshold, the high-side MOSFET is disabled, preventing current from flowing to the output and
minimizing output overshoot. When the VSENSE voltage drops lower than the OVTP threshold, the high-side
MOSFET is allowed to turn on at the next clock cycle.
Thermal Shutdown
The device implements an internal thermal shutdown to protect itself if the junction temperature exceeds 182°C.
The thermal shutdown forces the device to stop switching when the junction temperature exceeds the thermal
trip threshold. Once the die temperature decreases below 182°C, the device reinitiates the power-up sequence
by discharging the SS/TR pin.
Current-Mode Compensation Design
To simplify design efforts using the TPS5401, the typical designs for common applications are listed in Table 2.
For designs using ceramic output capacitors, proper derating of ceramic output capacitance is recommended
when doing the stability analysis. This is because the actual ceramic capacitance drops considerably from the
nominal value when the applied voltage increases. Advanced users may see the Step-by-Step Design Procedure
in the Application Information section for detailed guidelines or use the SwitcherPro software tool
(http://focus.ti.com/docs/toolsw/folders/switcherpro.html).
Table 2. Typical Designs (Referring to Simplifed Schematic on Page 1)
VIN
(V)
VOUT
(V)
fSW
(kHz)
LOUT
(µH)
7.5 V–35 V
5
700
47
7.5 V–35 V
5
700
12 V–42 V
5
700
12 V–42 V
3.3
8 V–14 V
5
22
R1
(kΩ)
R2
(kΩ)
C2
(pF)
C1
(pF)
R3
(kΩ)
Aluminum, 220 µF/260 mΩ
52.3
10
82
3300
698
47
Ceramic, 47 µF/10V
52.3
10
5.6
3300
75
47
Aluminum, 100 µF/300 mΩ
52.3
10
100
3300
316
700
33
Ceramic, 33 µF/10 V
30.9
10
10
3300
47
700
33
Ceramic, 47 µF/10 V
52.3
10
5.6
3300
75
COUT
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APPLICATION INFORMATION
Typical Application
The TPS5401 is good for the E-meter application. The power supply design is a challenge work to E-meter
designers because it must get enough output current from a wide input voltage range with a limited input power.
For example, in China, a single-phase or three-phase E-meter must work properly with an input AC electricity
voltage range from 0.7 Un to 1.9 Un. Here Un is the phase voltage of 220 V. The input active power limit is 1.5
W. A typical power supply design uses a transformer followed by a rectifier bridge to get a low unregulated dc
voltage. Then a voltage regulator generates 5 V for the whole system. Considering the ac transformer has large
internal dc resistance and the following rectifier has a voltage drop, the output voltage of the rectifier circuit is
different between empty load and heavy load. So the input voltage for the voltage regulator within the whole ac
input voltage range may have a maximum-to-minimum ratio of 4. In this situation, a linear regulator is not suitable
due to its very low efficiency at high input voltage.
Figure 40 shows a typical TPS5401 application schematic for this requirement. The input range is set to 7.5 V to
35 V. With such a wide input voltage range, an inexpensive transformer with high dc resistance can be used to
save total cost.
+
Figure 40. Typical Application Schematic
Step-by-Step Design Procedure
The following design procedure can be used to select component values for the TPS5401. Alternately, the
SwitcherPro software may be used to generate a complete design. The SwitcherPro software uses an iterative
design procedure and accesses a comprehensive database of components when generating a design. This
section presents a simplified discussion of the design process. To begin the design process, a few parameters
must be decided upon. Table 3 lists the parameters the designer needs to know.
Table 3.
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage range
7.5 V to 35 V
Output voltage
5V
Input ripple voltage
300 mV
Output ripple voltage
1% of VOUT
Output transient response for 0 to 500-mA load step 4% of VOUT
Maximum output current
500 mA
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Switching Frequency
The first step is to decide on a switching frequency for the regulator. Typically, the user may want to choose the
highest switching frequency possible, because this produces the smallest solution size. The high switching
frequency allows for lower-valued inductors and smaller output capacitors compared to a power supply that
switches at a lower frequency. Alternatively, the user may choose a lower switching frequency to improve
efficiency. At lower switching frequencies, switching losses are minimized. The switching frequency that can be
selected is limited by the minimum on-time of the internal power switch, the input voltage, the output voltage, and
the frequency shift limitation.
Equation 10 and Equation 11 must be used to find the maximum switching frequency for the regulator; choose
the lower value of the two equations. Switching frequencies higher than these values result in pulse skipping or
the lack of overcurrent protection during a short circuit.
The typical minimum on-time, ton(min), is 130 ns for the TPS5401. For this example, the output voltage is 5 V and
the maximum input voltage is 35 V, which allows for a maximum switch frequency up to 1213 kHz when including
the inductor resistance, on-resistance, and diode voltage in Equation 10. To ensure overcurrent runaway is not a
concern during short circuits in your design, use Equation 11 or the solid curve in Figure 38 to determine the
maximum switching frequency. With a maximum input voltage of 35 V, assuming a diode voltage of 0.5 V,
inductor resistance of 130 mΩ, switch resistance of 400 mΩ, a current-limit value of 0.94 A, and a short-circuit
output voltage of 0.1 V, the maximum switching frequency is approximately 1265 kHz. Choosing high frequency
can reduce external component size but results in higher switching loss. To achieve a balanced design, a
switching frequency of 700 kHz is used. To determine the timing resistance for a given switching frequency, use
Equation 9 to get a nearest standard resistance of 165 kΩ. The switching frequency is set to 698 kHz by the
resistor R3 shown in Figure 40.
Output Voltage Setpoint
The output voltage of the TPS5401 is externally adjustable using a resistor divider network. In the application
circuit of Figure 40, this divider network is comprised of R6 and R7. The relationship of the output voltage to the
resistor divider is given by Equation 12:
æV
ö
R6 = ç OUT - 1÷ x R7
è Vref
ø
(12)
Choosing R7 = 10 kΩ, R6 is calculated to be 52.3 kΩ for an output voltage of 5 V.
Due to current leakage of the VSENSE pin, the current flowing through the feedback network should be greater
than 1 mA in order to maintain the output voltage accuracy. This requirement makes the maximum value of R7
equal to 800 kΩ. Choosing higher resistor values decreases quiescent current and improve efficiency at low
output currents but may introduce noise immunity problems.
Input Capacitor
The TPS5401 requires a high-quality ceramic, type X5R or X7R, input decoupling capacitor with at least 3 mF of
effective capacitance, and in some applications a bulk capacitance. The effective capacitance includes any dc
bias effects. The voltage rating of the input capacitor must be greater than the maximum input voltage. The
capacitor must also have a ripple-current rating greater than the maximum input-current ripple of the TPS5401.
The input ripple current can be calculated using Equation 13.
The value of a ceramic capacitor varies significantly over temperature and the amount of dc bias applied to the
capacitor. The capacitance variations due to temperature can be minimized by selecting a dielectric material that
is stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors
because they have a high capacitance-to-volume ratio and are fairly stable over temperature. The output
capacitor must also be selected with the dc bias taken into account. The capacitance value of a capacitor
decreases as the dc bias across a capacitor increases.
For this example design, a ceramic capacitor with at least a 60-V voltage rating is required to support the
maximum input voltage. Common standard ceramic capacitor voltage ratings include 4 V, 6.3 V, 10 V, 16 V, 25
V, 50 V, and 100 V, so a 100-V capacitor should be selected. For this example, two 2.2-mF, 100-V capacitors in
parallel have been selected. Table 4 shows a selection of high-voltage capacitors. The input capacitance value
determines the input ripple voltage of the regulator. The input voltage ripple can be calculated using Equation 14.
24
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Using the design example values, IOUT(max) = 0.5 A, CIN = 4.4 mF,and fSW = 700 kHz, yields an input voltage ripple
of 40.6 mV and a maximum rms input ripple current of 0.25 A when VIN is equal to 10 V, which is 2 times the
output voltage of 5 V.
IINRMS = IOUT ×
VINRIPPLE =
(VIN -VOUT )
VOUT
×
VIN
VIN
(13)
IOUT(max) × 0.25
CIN × fSW
(14)
Table 4. Capacitor Types
VENDOR
VALUE (mF)
1 to 2.2
Murata
1 to 4.7
1
1 to 2.2
1 10 1.8
Vishay
1 to 1.2
1 to 3.9
1 to 1.8
1 to 2.2
TDK
1.5 to 6.8
1. to 2.2
1 to 3.3
1 to 4.7
AVX
1
1 to 4.7
1 to 2.2
EIA Size
1210
1206
2220
2225
1812
1210
1210
1812
VOLTAGE
DIELECTRIC
100 V
COMMENTS
GRM32 series
50 V
100 V
GRM31 series
50 V
50 V
100 V
VJ X7R series
50 V
100 V
100 V
50 V
100 V
50 V
X7R
C series C4532
C series C3225
50 V
100 V
50 V
X7R dielectric series
100 V
Output Inductor Selection
To calculate the minimum value of the output inductor, use Equation 15.
VIN(max) - VOUT
VOUT
´
LOUT(min) =
IOUT ´ KIND
VIN(max) ´ fSW
(15)
KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output current.
The inductor ripple current is filtered by the output capacitor. Therefore, choosing high inductor ripple currents
impacts the selection of the output capacitor, because the output capacitor must have a ripple-current rating
equal to or greater than the inductor ripple current. In general, the inductor ripple value is at the discretion of the
designer; however, the following guidelines may be used.
For designs using low-ESR output capacitors such as ceramics, a value as high as KIND = 0.4 may be used.
When using higher-ESR output capacitors, KIND = 0.2 yields better results. Because the inductor ripple current is
part of the PWM control system, the TPS5401 requires ripple current that is always greater than 30 mA for
dependable operation. In a wide-input voltage regulator, it is best to choose an inductor ripple current on the
larger side. This allows the inductor to still have a measurable ripple current with the input voltage at its
minimum. The maximum value of the input inductance is limited by Equation 16.
VIN(min) - VOUT
VOUT
´
LOUT(max) =
30mA
VIN(min) ´ fSW
(16)
For this design example, use KIND = 0.3, and the minimum inductor value is calculated to be 42 mH. The nearest
standard value was chosen: 47 mH. For the output-filter inductor, it is important that the root-mean-square (rms)
current and saturation current ratings not be exceeded. The rms and peak inductor current can be found from
Equation 17 to Equation 19.
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ILRIPPLE =
IL(RMS) =
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( VIN(max) - VOUT ) ´ VOUT
VIN(max) ´ LOUT ´ fSW
(IOUT )2 +
IL(peak) = IOUT
(17)
1 æ ( VIN(max) - VOUT ) ´ VOUT
´ç
12 çè VIN(max) ´ LOUT ´ fSW
ö
÷
÷
ø
2
I
+ RIPPLE
2
(18)
(19)
As the equation set demonstrates, lower ripple currents reduce the output voltage ripple of the regulator but
require a larger value of inductance. Selecting higher ripple currents increases the output voltage ripple of the
regulator but allows for a lower inductance value. The current flowing through the inductor is the inductor ripple
current plus the output current. During power up, faults or transient load conditions, the inductor current can
increase above the calculated peak inductor current level. In transient conditions, the inductor current can
increase up to the switch current limit of the device. For this reason, the most conservative approach is to specify
an inductor with a saturation current rating equal to or greater than the switch current limit rather than the peak
inductor current. For this design, ILRIPPLE = 0.1303 A, IL(RMS) = 0.501 A and IL(peak) = 0.565 A. The inductor used is
a Coilcraft MSS1048-473ML type, with a saturation current rating of 1.44 A and an rms current rating of 1.83 A.
Output Capacitor
There are three primary considerations for selecting the value of the output capacitor. The output capacitor
determines the modulator pole, the output voltage ripple, and how the regulator responds to a large change in
load current. The output capacitance must be selected based on the most-stringent of these three criteria.
The desired response to a large change in the load current is the first criterion. The output capacitor must supply
the load with current when the regulator cannot. This situation occurs if there are desired hold-up times for the
regulator where the output capacitor must hold the output voltage above a certain level for a specified amount of
time after the input power is removed. The regulator also is temporarily unable to supply sufficient output current
if there is a large, fast increase in the current needs of the load, such as when transitioning from no load to a full
load. The regulator usually requires two or more clock cycles for the control loop to see the change in load
current and output voltage and adjust the duty cycle to react to the change. The output capacitor must be sized
to supply the extra current to the load until the control loop responds to the load change. The output capacitance
must be large enough to supply the difference in current for two clock cycles while only allowing a tolerable
amount of droop in the output voltage. Equation 20 shows the minimum output capacitance necessary to
accomplish this.
2 ´ DIOUT
COUT >
fSW ´ (DVOUT - DIOUT ´ RESR )
(20)
Where:
• ΔIOUT is the change in output current
• fSW is the regulator switching frequency
• ΔVOUT is the allowable change in the output voltage
• RESR is the equivalent series resistance (ESR) of the output capacitor.
Equation 20 indicates the ESR must be less than ΔVOUT/ΔIOUT. For this example, the transient load response is
specified as a 4% change in VOUT for a load step from 0 A (no load) to 0.5 A (full load). For this example, ΔIOUT =
0.5A and ΔVOUT = 0.04 × 5 V = 0.2 V. For ceramic capacitors, the ESR is usually small enough to ignore in this
calculation. Aluminum electrolytic and tantalum capacitors have higher ESR that should be taken into account.
Using these numbers gives a minimum capacitance of 7.14 mF for ceramic capacitor and 20.4 µF for electrolytic
capacitor with 260 mΩ ESR.
The catch diode of the regulator cannot sink current, so any stored energy in the inductor produces an output
voltage overshoot when the load current rapidly decreases. The output capacitor must also be sized to absorb
energy stored in the inductor when transitioning from a high load current to a lower load current. The excess
energy that gets stored in the output capacitor increases the voltage on the capacitor. The capacitor must be
sized to maintain the desired output voltage during these transient periods. Equation 21 is used to calculate the
minimum capacitance to keep the output voltage overshoot to a desired value, where LOUT is the value of the
26
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inductor, IOH is the output current under heavy load, IOL is the output under light load, VFIN is the final peak output
voltage, and VINI is the initial capacitor voltage. For this example, the worst-case load step is from 0.5 A to 0 A.
The output voltage increases during this load transition, and the stated maximum in our specification is 4% of the
output voltage. This makes VFIN = 1.04 × 5 V = 5.2 V. VINI is the initial capacitor voltage, which is the nominal
output voltage of 5 V. Using these numbers in Equation 21 yields a minimum capacitance of 5.76 mF.
COUT > LOUT ´
IOH2 - IOL 2
VFIN2 - VINI2
(21)
Equation 22 calculates the minimum output capacitance needed to meet the output-voltage ripple specification,
where fSW is the switching frequency, VORIPPLE is the maximum allowable output voltage ripple, and ILRIPPLE is the
inductor ripple current. Equation 22 shows the ESR of the output capacitor must be less than VORIPPLE/ILRIPPLE to
meet the output-voltage ripple requirement. Low-ESR capacitors are preferred to keep the output-voltage ripple
low. If a high-ESR electrolytic capacitor is used, a small ESR ceramic capacitor is recommended to be in parallel
with the electrolytic capacitor to minimize the output voltage ripple. In this application, an aluminum electrolytic
capacitor is chosen as the output capacitor. It has 260 mΩ ESR. Equation 22 yields 1.44 µF.
1
1
´
COUT >
8 ´ fSW VORIPPLE
- RESR
ILRIPPLE
(22)
The most stringent criterion for the output capacitor is 20.5 µF of capacitance to keep the output voltage in
regulation during a load transient in this example.
Additional capacitance de-ratings for aging, temperature and dc bias should be factored in, which increases this
minimum value. For this example, a 220 µF electrolytic capacitor with 260 mΩ of ESR can be used for low cost
target.
Capacitors generally have limits to the amount of ripple current they can handle without failing or producing
excess heat. An output capacitor that can support the inductor ripple current must be specified. Some capacitor
data sheets specify the root-mean-square (rms) value of the maximum ripple current. Equation 23 can be used to
calculate the rms ripple current the output capacitor must support. For this application, Equation 23 yields 37.6
mA.
ICO(RMS) =
(
VOUT ´ VIN(max) - VOUT
)
12 ´ VIN(max) ´ LOUT ´ fSW
(23)
Catch Diode
The TPS5401 requires an external catch diode between the PH pin and GND. The selected diode must have a
reverse voltage rating equal to or greater than VIN(MAX). The peak current rating of the diode must be greater than
the maximum inductor current. The diode should also have a low forward voltage. Schottky diodes are typically a
good choice for the catch diode, due to their low forward voltage. The lower the forward voltage of the diode, the
higher the efficiency of the regulator.
Typically, when the voltage and current ratings for the diode are higher, then the forward voltage is higher.
Because the design example has an input voltage up to 42 V, a diode with a minimum of 42-V reverse voltage is
selected.
For the example design, the B160A Schottky diode is selected for its lower forward voltage, and it comes in a
larger package size which has good thermal characteristics over small devices. The typical forward voltage of the
B160A is 0.5 V.
The diode must also be selected with an appropriate power rating. The diode conducts the output current during
the off-time of the internal power switch. The off-time of the internal switch is a function of the maximum input
voltage, the output voltage, and the switching frequency. The output current during the off-time is multiplied by
the forward voltage of the diode, which equals the conduction losses of the diode. At higher switch frequencies,
the ac losses of the diode must be taken into account. The ac losses of the diode are due to the charging and
discharging of the junction capacitance and reverse recovery. Equation 24 is used to calculate the total power
dissipation, conduction losses plus ac losses, of the diode.
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The B160A has a junction capacitance (CJ) of 110 pF. Using Equation 24, the selected diode dissipates 0.29 W.
This power dissipation, depending on mounting techniques, should produce a 5.9°C temperature rise in the diode
when the input voltage is 42 V and the load current is 0.5 A.
If the power supply spends a significant amount of time at light load currents or in sleep mode, consider using a
diode which has a low leakage current and slightly higher forward voltage drop.
2
PD
(VIN(max) - VOUT ) ´ IOUT ´ VFD + CJ ´ fSW ´ (VIN(max) + VFD )
=
VIN(max)
2
(24)
Slow-Start Capacitor
The slow-start capacitor determines the minimum amount of time it takes for the output voltage to reach its
nominal programmed value during power up. This is useful if a load requires a controlled voltage-slew rate. This
is also used if the output capacitance is large and would require large amounts of current to quickly charge the
capacitor to the output voltage level. The large currents necessary to charge the capacitor may make the
TPS5401 reach the current limit, or excessive current draw from the input power supply may cause the input
voltage rail to sag. Limiting the output voltage-slew rate solves both of these problems.
The slow-start time must be long enough to allow the regulator to charge the output capacitor up to the output
voltage without drawing excessive current. Equation 25 can be used to find the minimum slow-start time, tSS,
necessary to charge the output capacitor, COUT, from 10% to 90% of the output voltage, VOUT, with an average
slow-start current of ISSAVG. In the example, to charge the 220 mF output capacitor up to 5 V while only allowing
the average input current ISSAVG to be 0.2 A would require a 4.4-ms slow-start time.
C
´ VOUT ´ 0.8
t SS > OUT
ISSAVG
(25)
Once the slow-start time is known, the slow-start capacitor value can be calculated using Equation 4. For the
example circuit, the slow-start time is set to a value of 3.2 ms, which requires a 0.01-µF capacitor.
Bootstrap Capacitor Selection
A 0.1-mF ceramic capacitor must be connected between the BOOT and PH pins for proper operation. It is
recommended to use a ceramic capacitor with X5R or better grade dielectric. The capacitor should have a 10-V
or higher voltage rating.
Undervoltage Lockout Set Point
The undervoltage lockout (UVLO) can be adjusted using an external voltage divider on the EN pin of the
TPS5401. The UVLO has two thresholds, one for power up when the input voltage is rising and one for power
down or brown outs when the input voltage is falling. The supply should turn on and start switching once the
input voltage increases above power-up threshold (enabled). After the regulator starts switching, it should
continue to do so until the input voltage falls below the power-down threshold (UVLO stop).
The programmable UVLO and enable voltages are set using a resistor divider between Vin and ground to the EN
pin. Equation 2 through Equation 3 can be used to calculate the resistance values necessary. For the example
application, because the minimum input voltage is 7.5 V, when the maximum input voltage is 35 V, the voltage at
the EN pin exceeds the absolute voltage rating of the EN pin. So the UVLO is not set externally in this design.
Compensation
The external compensation used with the TPS5401 allows for a wide range of output filter configurations. A large
range of capacitor values and types of dielectric is supported. This design example uses an aluminum electrolytic
output capacitor. A design example with the ceramic dielectric output capacitors can be found in the TPS54040
data sheet (SLVS918). More accurate designs can be found in the SwitcherPro software.
The peak-current mode PWM modulator and the output filter generate a pair of power stage pole and zero which
are determined using Equation 26 and Equation 27.
IOUT
fPmod =
2p ´ VOUT ´ COUT
(26)
28
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fZmod =
SLVSAB0 – DECEMBER 2010
1
2p ´ RESR ´ COUT
(27)
A type 2A compensation scheme is recommended for the TPS5401. As RC, CC, CP shown in Figure 41, the
compensation components are chosen to set the desired loop crossover frequency and phase margin for output
filter components. The type 2A compensation has the following characteristics: a dc gain, a low-frequency pole,
and a mid-frequency zero / pole pair.
VOUT
VSENSE
Type2A
gmEA
V ref
COMP
RO
CO
RC
CP
CC
Figure 41. The Compensation Recommended for the TPS5401
The dc gain is determined by Equation 28:
V
GDCEA = ADCEA × ref
VOUT
(28)
where:
ADCEA is the error amplifier open-loop dc gain. It is 10,000 V/V.
The low-frequency pole is determined by Equation 29:
1
fP0 =
2p ´ RO ´ CC
(29)
where:
RO is the output resistance of the error amplifier. It can be calculated by Equation 30.
A
R0 = DCEA
gmEA
(30)
The mid-frequency zero is determined by Equation 31.
1
fZ1 =
2p ´ RC ´ CC
(31)
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And, the mid-frequency pole is given by Equation 32.
1
fP1 =
2p ´ RC ´ CP
(32)
The first step is to choose the closed-loop crossover frequency fCO. In general, the closed-loop crossover
frequency could be less than 1/10 of the minimum operating frequency. But for the TPS5401, it is recommended
that the maximum closed-loop crossover frequency be not greater than 40 kHz. For this example, we choose
15-kHz crossover frequency. Next, by definition, the magnitude of the loop gain at the crossover frequency is 0
dB. By placing the compensation zero at the power stage pole, and the mid-compensation pole at the power
stage zero, the RC, CC and CP can be approximately calculated by Equation 33 through Equation 35.
2p ´ fCO ´ COUT
VOUT
RC =
×
gmPS
Vref ´ gmEA
(33)
1
2p ´ fPmod
(34)
1
CP =
2p ´ fZmod
(35)
CC =
Where:
• gmPS is the power-stage transconductance. It is 1.9 A/V
• gmEA is the error-amplifier transconductance. It is 97 mA/V
For this design example, R4, C7, and C8 make the compensation network. R4 = 698 kΩ, C7 = 3300 pF, C8 = 82
pF.
Discontinuous Mode and Eco-mode Control-Scheme Boundary
With an input voltage of 34 V, the power supply enters discontinuous mode when the output current is less than
50 mA. The power supply enters Eco-mode control scheme when the output current is lower than 30 mA.
Application Curves
VIN = 20 V / div
VOUT = 50 mV / div (ac coupled)
VOUT = 2 V / div
IOUT = 200 mA / div
EN = 2 V / div
30
Time = 200 µsec / div
Time = 5 msec / div
Figure 42. Load Transient (100 mA to 350 mA)
Figure 43. Start-Up With Input Voltage
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PH = 20 V div
PH = 20 V div
VOUT = 20 mV / div (ac coupled)
VOUT = 20 mV / div (ac coupled)
Inductor Current = 100 mA div
Inductor Current = 100 mA div
Time = 1 usec / div
Time = 1 usec / div
Figure 44. Output Ripple, CCM
Figure 45. Output Ripple, DCM
PH = 20 V / div
PH = 20 V div
VIN = 100 mV / div (ac coupled)
VOUT = 20 mV / div (ac coupled)
Inductor Current = 100 mA div
Inductor Current = 100 mA / div
Time = 1 µsec / div
Time = 20 usec / div
Figure 46. Output Ripple, PSM
Figure 47. Input Ripple, CCM
100
PH = 20 V div
90
80
VIN = 15 V
70
Efficiency - %
VOUT = 100 mV / div (ac coupled)
VIN = 7.5 V
VIN = 24 V
60
VIN = 35 V
50
40
30
Inductor Current = 100 mA div
20
10
0
Time = 1 usec / div
0
0.1
0.2
0.3
0.4
0.5
IOUT - Output Current - A
Figure 48. Input Ripple, DCM
Figure 49. Efficiency vs Load Current
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60
100
180
90
Phase
40
80
20
Gain - dB
VIN = 15 V
60
50
VIN = 24 V
40
30
60
0
Phase - °
70
Efficiency - %
120
VIN = 7.5 V
0
Gain
−20
−60
−40
−120
VIN = 35 V
20
10
0
0.001
0.01
0.1
1
−60
100
1000
IOUT - Output Current - A
Figure 50. Light-Load Efficiency
0.10
0.08
0.08
VIN = 24 V
0.06
IOUT = 0.25 A
0.06
0.04
Regulation - %
0.04
Regulation - %
−180
1000000
100000
Figure 51. Overall Loop-Frequency Response
0.10
0.02
0.00
−0.02
0.02
0.00
−0.02
−0.04
−0.04
−0.06
−0.06
−0.08
−0.08
−0.10
−0.10
0
0.1
0.2
0.3
IOUT - Output Current - A
0.4
Figure 52. Regulation vs Load Current
32
10000
Frequency - Hz
0.5
5
10
15
20
25
VIN - Input Voltage - V
30
35
Figure 53. Regulation vs Input Voltage
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Power Dissipation Estimate
The following formulas show how to estimate the IC power dissipation under continuous-conduction mode (CCM)
operation. These equations should not be used if the device is working in discontinuous-conduction mode
(DCM).
The power dissipation of the IC includes conduction loss (Pcon), switching loss (Psw), gate-drive loss (PGD), and
supply current (PSUP).
V
Pcon = IOUT 2 ´ r DS(on) ´ OUT
VIN
(36)
Psw = VIN2 ´ lOUT ´ 0.25 ´ 10-9 ´ fSW
PGD = VIN ´ 3 ´ 10
PSUP = 116 ´ 10
-6
-9
(37)
´ fSW
(38)
´ VIN
(39)
So total power dissipation is:
PTOT = PCON + PSW + PGD + PSUP
(40)
For a given ambient temperature TA, the junction temperature TJ can be estimated by Equation 41.
TJ = TA - q JA ´ PTOT
(41)
where:
• qJA is the thermal resistance of the package (°C/W)
For given TJ(max) = 150°C, the maximum allowed ambient temperature can be estimated by Equation 42.
TA(max) = TJ(max) - q JA ´ PTOT
(42)
There are additional power losses in the regulator circuit due to the inductor ac and dc losses, the catch diode
and trace resistance that impact the overall efficiency of the regulator.
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Layout
Layout is a critical portion of good power-supply design. There are several signals paths that conduct
fast-changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate
noise or degrade the power-supply performance. To help eliminate these problems, the VIN pin should be
bypassed to ground with a low-ESR ceramic bypass capacitor with X5R or X7R dielectric. Care should be taken
to minimize the loop area formed by the bypass capacitor connections, the VIN pin, and the anode of the catch
diode. See Figure 54 for a PCB layout example. The GND pin should be tied directly to the thermal pad under
the IC.
The thermal pad should be connected to any internal PCB ground planes using multiple vias directly under the
IC. The PH pin should be routed to the cathode of the catch diode and to the output inductor. Because the PH
connection is the switching node, the catch diode and output inductor should be located close to the PH pins,
and the area of the PCB conductor minimized to prevent excessive capacitive coupling. For operation at full rated
load, the top-side ground area must provide adequate heat dissipating area. The RT/CLK pin is sensitive to
noise, so the RT resistor should be located as close as possible to the IC and routed with minimal lengths of
trace. The additional external components can be placed approximately as shown. It may be possible to obtain
acceptable performance with alternate PCB layouts; however, this layout has been shown to produce good
results and is meant as a guideline.
Vout
Output
Capacitor
Topside
Ground
Area
Input
Bypass
Capacitor
Vin
UVLO
Adjust
Resistors
Slow Start
Capacitor
Output
Inductor
Route Boot Capacitor
Trace on another layer to
provide wide path for
topside ground
BOOT
Catch
Diode
PH
VIN
GND
EN
COMP
SS/TR
VSENSE
RT/CLK
PWRGD
Frequency
Set Resistor
Compensation
Network
Resistor
Divider
Thermal VIA
Signal VIA
Figure 54. PCB Layout Example
Estimated Circuit Area
The estimated printed circuit board area for the components used in the design of Figure 40 is 0.55 in2 (3.55
cm2). This area does not include test points or connectors.
34
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PACKAGE OPTION ADDENDUM
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18-Dec-2010
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package
Drawing
Pins
Package Qty
Eco Plan
(2)
Lead/
Ball Finish
MSL Peak Temp
(3)
Samples
(Requires Login)
TPS5401DGQ
ACTIVE
MSOPPowerPAD
DGQ
10
80
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-1-260C-UNLIM
Purchase Samples
TPS5401DGQR
ACTIVE
MSOPPowerPAD
DGQ
10
2500
Green (RoHS
& no Sb/Br)
CU NIPDAU Level-1-260C-UNLIM
Purchase Samples
TPS5401DGQT
ACTIVE
MSOPPowerPAD
DGQ
10
250
TBD
Call TI
Call TI
Purchase Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
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TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
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