LINER LTC3614

LTC3614
4A, 4MHz Monolithic
Synchronous Step-Down
DC/DC Converter
DESCRIPTION
FEATURES
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4A Output Current
2.25V to 5.5V Input Voltage Range
Low Output Ripple Burst Mode® Operation: IQ = 75μA
±1% Output Voltage Accuracy
Output Voltage Down to 0.6V
High Efficiency: Up to 95%
Low Dropout Operation: 100% Duty Cycle
Programmable Slew Rate on SW Node Reduces
Noise and EMI
Adjustable Switching Frequency: Up to 4MHz
Optional Active Voltage Positioning (AVP) with
Internal Compensation
Selectable Pulse-Skipping/Forced Continuous/Burst
Mode Operation with Adjustable Burst Clamp
Programmable Soft-Start
Inputs for Start-Up Tracking or External Reference
DDR Memory Mode, IOUT = ±3A
Available in a 24-Pin 3mm × 5mm QFN
Thermally Enhanced Package
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The operating frequency is externally programmable up to
4MHz, allowing the use of small surface mount inductors.
For switching-noise-sensitive applications, the LTC3614
can be synchronized to an external clock at up to 4MHz.
Forced continuous mode operation in the LTC3614 reduces
noise and RF interference. Adjustable compensation allows
the transient response to be optimized over a wide range
of loads and output capacitors.
The internal synchronous switch increases efficiency and
eliminates the need for an external catch diode, saving
external components and board space. The LTC3614 is
offered in a leadless 24-pin 3mm × 5mm thermally enhanced QFN package.
APPLICATIONS
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The LTC®3614 is a low quiescent current monolithic synchronous buck regulator using a current mode, constant
frequency architecture. The no-load DC supply current
in sleep mode is only 75μA while maintaining the output
voltage (Burst Mode operation) at no load, dropping to zero
current in shutdown. The 2.25V to 5.5V input supply voltage
range makes the LTC3614 ideally suited for single Li-Ion
as well as fixed low voltage input applications. 100% duty
cycle capability provides low dropout operation, extending
the operating time in battery-powered systems.
Point-of-Load Supplies
Distributed Power Supplies
Portable Computer Systems
DDR Memory Termination
Handheld Devices
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks of
Linear Technology Corporation. All other trademarks are the property of their respective owners.
Protected by U.S. Patents, including 6580258, 5481178, 5994885, 6304066, 6498466, 6611131.
Efficiency and Power Loss
vs Load Current
TYPICAL APPLICATION
100
SVIN
PVIN
80
330nH
VOUT
2.5V
4A
47μF
s2
665k
3614 TA01a
210k
1
70
60
0.1
50
40
0.01
30
20
10
VOUT = 2.5V
0
1
VIN = 2.8V
VIN = 3.3V
VIN = 5V
10
100
1000
OUTPUT CURRENT (mA)
POWER LOSS (W)
SRLIM/DDR
RUN
TRACK/SS
RT/SYNC
LTC3614
SW
PGOOD
SGND
ITH
PGND
MODE
VFB
90
10μF
s4
EFFICIENCY (%)
VIN
2.7V TO 5.5V
0
10000
3614 TA01b
3614f
1
LTC3614
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
MODE
VFB
ITH
TRACK/SS
TOP VIEW
24 23 22 21
SRLIM/DDR 1
20 PGOOD
RT/SYNC 2
19 RUN
SGND 3
18 SVIN
PVIN 4
17 PVIN
25
PGND
SW 5
16 SW
15 SW
SW 7
14 SW
SW 8
13 SW
NC
10 11 12
PVIN
9
PVIN
SW 6
NC
PVIN, SVIN Voltages...................................... –0.3V to 6V
SW Voltage ..................................–0.3V to (PVIN + 0.3V)
ITH, RT/SYNC Voltages ............... –0.3V to (SVIN + 0.3V)
SRLIM, TRACK/SS Voltages ....... –0.3V to (SVIN + 0.3V)
MODE, RUN, VFB Voltages .......... –0.3V to (SVIN + 0.3V)
PGOOD Voltage ............................................ –0.3V to 6V
Operating Junction Temperature Range
(Notes 2, 11) .......................................... –40°C to 125°C
Storage Temperature.............................. –65°C to 150°C
UDD PACKAGE
24-LEAD (3mm s 5mm) PLASTIC QFN
TJMAX = 125°C, θJA = 38°C/W
EXPOSED PAD (PIN 25) IS PGND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3614EUDD#PBF
LTC3614EUDD#TRPBF
LFVM
24-Lead (3mm × 5mm) Plastic QFN
–40°C to 125°C
LTC3614IUDD#PBF
LTC3614IUDD#TRPBF
LFVM
24-Lead (3mm × 5mm) Plastic QFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3614f
2
LTC3614
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating junction
temperature range, otherwise specifications are at TA ≈ TJ = 25°C. VIN = 3.3V, RT/SYNC = SVIN unless otherwise specified (Notes 1, 2, 11).
SYMBOL
PARAMETER
VIN
Operating Voltage Range
VUVLO
Undervoltage Lockout Threshold
VFB
Feedback Voltage Internal Reference
Feedback Voltage External Reference
(Note 7)
CONDITIONS
MIN
l
2.25
SVIN Ramping Down
SVIN Ramping Up
l
l
1.7
(Note 3) VTRACK = SVIN, VDDR = 0V
0°C < TJ < 85°C
–40°C < TJ < 125°C
l
TYP
MAX
UNITS
5.5
V
2.25
V
V
0.594
0.591
0.6
0.606
0.609
V
V
(Note 3) VTRACK = 0.3V, VDDR = SVIN
0.288
0.300
0.312
V
(Note 3) VTRACK = 0.5V, VDDR = SVIN
0.488
0.500
0.512
V
IFB
Feedback Input Current
VFB = 0.6V
l
±30
nA
ΔVLINEREG
Line Regulation
SVIN = PVIN = 2.25V to 5.5V
(Notes 3, 4) TRACK/SS = SVIN
l
0.2
%/V
ΔVLOADREG
Load Regulation
ITH from 0.5V to 0.9V (Notes 3, 4)
VITH = SVIN (Note 5)
0.25
2.6
%
%
IS
RDS(ON)
ILIM
Active Mode Supply Current
VFB = 0.5V, VMODE = SVIN (Note 6)
1100
Sleep Mode Supply Current
VFB = 0.7V, VMODE = 0V, ITH = SVIN
(Note 5)
75
100
μA
μA
VFB = 0.7V, VMODE = 0V (Note 4)
130
175
μA
Shutdown Current
SVIN = PVIN = 5.5V, VRUN = 0V
0.1
1
μA
Top Switch On-Resistance
PVIN = 3.3V (Note 10)
35
mΩ
Bottom Switch On-Resistance
PVIN = 3.3V (Note 10)
25
mΩ
Top Switch Current Limit
Sourcing (Note 8), VFB = 0.5V
Duty Cycle <35%
Duty Cycle = 100%
Bottom Switch Current Limit
Sinking (Note 8), VFB = 0.7V,
Forced Continuous Mode
7.5
5.3
9
10.5
A
A
–6
–8
–11
A
gm(EA)
Error Amplifier Transconductance
–5μA < IITH < 5μA (Note 4)
200
μS
IEAO
Error Amplifier Maximum Output
Current
(Note 4)
±30
μA
tSS
Internal Soft-Start Time
VFB from 0.06V to 0.54V,
TRACK/SS = SVIN
0.65
VTRACK/SS
Enable Internal Soft-Start
(Note 7 )
0.62
V
tTRACK/SS_DIS
Soft-Start Discharge Time at Start-Up
60
μs
1.2
RON(TRACK/SS_DIS) TRACK/SS Pull-Down Resistor at
Start-Up
fOSC
1.9
200
ms
Ω
Oscillator Frequency
RT/SYNC = 370k
l
0.8
1
1.2
MHz
Internal Oscillator Frequency
VRT/SYNC = SVIN
l
1.8
2.25
2.7
MHz
4
MHz
fSYNC
Synchronization Frequency Range
0.3
VRT/SYNC
SYNC Input Threshold High
1.2
SYNC Input Threshold Low
.
ISW(LKG)
Switch Leakage Current
VDDR
DDR Option Enable Voltage
VMODE
(Note 9)
Internal Burst Mode Operation
Pulse-Skipping Mode
SVIN = PVIN = 5.5V, VRUN = 0V
V
0.1
0.3
V
1
μA
SVIN – 0.3
V
0.3
SVIN – 0.3
V
V
Forced Continuous Mode
1.1
SVIN • 0.58
V
External Burst Mode Operation
0.45
0.8
V
3614f
3
LTC3614
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating junction
temperature range, otherwise specifications are at TA ≈ TJ = 25°C. VIN = 3.3V, RT/SYNC = SVIN unless otherwise specified (Notes 1, 2, 11).
SYMBOL
PARAMETER
CONDITIONS
PGOOD
Power Good Voltage Windows
TRACK/SS = SVIN, Entering Window
VFB Ramping Up
VFB Ramping Down
MIN
TYP
–3
3
–6
6
TRACK/SS = SVIN, Leaving Window
VFB Ramping Up
VFB Ramping Down
tPGOOD
Power Good Blanking Time
Entering and Leaving Window
RPGOOD
Power Good Pull-Down On-Resistance
VRUN
RUN voltage
l
l
Input High
Input Low
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3614E is guaranteed to meet performance specifications
over the 0°C to 85°C operating junction temperature range. Specifications
over the –40°C to 125°C operating junction temperature range are
assured by design, characterization and correlation with statistical process
controls. The LTC3614I is guaranteed to meet specifications over the
full –40°C to 125°C operating junction temperature range. Note that
the maximum ambient temperature is determined by specific operating
conditions in conjunction with board layout, the rated package thermal
resistance and other environmental factors.
Note 3: This parameter is tested in a feedback loop which servos VFB to
the midpoint for the error amplifier (VITH = 0.75V).
Note 4: External compensation on ITH pin.
MAX
UNITS
%
%
9
–9
11
–11
%
%
70
105
140
μs
8
17
33
Ω
0.4
V
V
1
Note 5: Tying the ITH pin to SVIN enables the internal compensation and
AVP mode.
Note 6: Dynamic supply current is higher due to the internal gate charge
being delivered at the switching frequency.
Note 7: See description of the TRACK/SS pin in the Pin Functions section.
Note 8: In sourcing mode the average output current is flowing out of the
SW pin. In sinking mode the average output current is flowing into the SW
Pin.
Note 9: See description of the MODE pin in the Pin Functions section.
Note 10: Guaranteed by correlation and design to wafer level
measurements for QFN packages.
Note 11: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
TYPICAL PERFORMANCE CHARACTERISTICS VIN = 3.3V, RT/SYNC = SVIN unless otherwise noted.
Efficiency vs Load Current
Burst Mode Operation (VMODE = 0V)
100
100
VOUT = 1.8V
90
Efficiency vs Load Current
100
VOUT = 1.2V
90
80
70
70
50
40
60
50
40
30
30
20
20
VIN = 2.5V
VIN = 3.3V
VIN = 5V
10
0
1
10
100
1000
OUTPUT CURRENT (mA)
10000
3614 G01
EFFICIENCY (%)
80
70
60
VOUT = 1.8V
90
80
EFFICIENCY (%)
EFFICIENCY (%)
Efficiency vs Load Current
Burst Mode Operation (VMODE = 0V)
60
50
40
30
VIN = 2.5V
VIN = 3.3V
VIN = 5V
10
0
1
10
100
1000
OUTPUT CURRENT (mA)
10000
3614 G02
20
Burst Mode OPERATION
PULSE-SKIPPING
FORCED CONTINUOUS
10
0
1
10
100
1000
OUTPUT CURRENT (mA)
10000
3614 G03
3614f
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LTC3614
TYPICAL PERFORMANCE CHARACTERISTICS VIN = 3.3V, RT/SYNC = SVIN unless otherwise noted.
Efficiency vs Frequency
Burst Mode Operation
(VMODE = 0V), IOUT = 2A
100
VOUT = 1.8V
90
EFFICIENCY (%)
EFFICIENCY (%)
80
70
60
50
IOUT = 6mA
IOUT = 600mA
IOUT = 2A
40
30
2.5
3
3.5
4
4.5
INPUT VOLTAGE (V)
5
5.5
95
94
93
92
91
90
89
88
87
86
85
84
83
82
0.5
Load Regulation
(VOUT = 1.8V)
1.5
VIN = 3.3V
VOUT = 1.8V
FORCED CONTINUOUS MODE
PULSE-SKIPPING MODE
INTERNAL Burst Mode OPERATION
1.3
1.1
VOUT ERROR (%)
Efficiency vs Input Voltage
Burst Mode Operation
(VMODE = 0V)
150nH
330nH
470nH
1
1.5
2 2.5 3 3.5
FREQUENCY (MHz)
4
4.5
0.9
0.7
0.5
0.3
0.1
–0.1
–0.3
0
3614 G05
3614 G04
Line Regulation
4000
2000
3000
1000
OUTPUT CURRENT (mA)
3614 G06
Burst Mode Operation
Pulse-Skipping Mode Operation
0.3
VOUT ERROR (%)
0.2
0.1
VOUT
20mV/DIV
VOUT
20mV/DIV
IL
1A/DIV
IL
1A/DIV
0
–0.1
–0.2
–0.3
2.20
2.75
3.30 3.85 4.40
INPUT VOLTAGE (V)
4.95
VOUT = 1.8V
IOUT = 150mA
VMODE = 0V
5.50
20μs/DIV
3614 G08
VOUT = 1.8V
IOUT = 150mA
VMODE = 3.3V
20μs/DIV
3614 G09
3614 G07
Load Step Transient in
Pulse-Skipping Mode
Forced Continuous Mode Operation
VOUT
20mV/DIV
IL
500mA/DIV
VOUT = 1.8V
IOUT = 100mA
VMODE = 1.5V
1μs/DIV
3614 G10
Load Step Transient in
Burst Mode Operation
VOUT
100mV/DIV
VOUT
100mV/DIV
ILOAD
2A/DIV
ILOAD
2A/DIV
100μs/DIV
VOUT = 1.8V
ILOAD = 100mA TO 4A
VMODE = 3.3V
COMPENSATION FIGURE 1
3614 G11
100μs/DIV
VOUT = 1.8V
ILOAD = 100mA TO 4A
VMODE = 0V
COMPENSATION FIGURE 1
3614 G12
3614f
5
LTC3614
TYPICAL PERFORMANCE CHARACTERISTICS VIN = 3.3V, RT/SYNC = SVIN unless otherwise noted.
Load Step Transient in Forced
Continuous Mode without AVP Mode
Load Step Transient in Forced
Continuous Mode Sourcing and
Sinking Current
Load Step Transient in Forced
Continuous Mode with AVP Mode
VOUT
100mV/DIV
VOUT
100mV/DIV
VOUT
200mV/DIV
ILOAD
2A/DIV
ILOAD
2A/DIV
ILOAD
2A/DIV
100μs/DIV
VOUT = 1.8V
ILOAD = 100mA TO 4A, VMODE = 1.5V
COMPENSATION FIGURE 1
3614 G13
100μs/DIV
VOUT = 1.8V
ILOAD = 100mA TO 4A, VMODE = 1.5V
3614 G14
Tracking Up/Down in
Forced Continuous Mode,
Non DDR Mode
Internal Start-Up in Forced
Continuous Mode
Sinking Current
3614 G15
100μs/DIV
VOUT = 1.8V
ILOAD = –3A TO 3A, VMODE = 1.5V
COMPENSATION FIGURE 1
RUN
10V/DIV
VOUT
100mV/DIV
VOUT
1V/DIV
PGOOD
10V/DIV
SW
2V/DIV
VOUT
500mV/DIV
VTRACK/SS
500mV/DIV
IL
2A/DIV
IL
2A/DIV
PGOOD
2V/DIV
1μs/DIV
VOUT = 1.8V
IOUT = –3A, VMODE = 1.5V
3614 G16
500μs/DIV
VOUT = 1.8V
IOUT = 0A, VMODE = 1.5V
Tracking Up/Down in Forced
Continuous Mode, DDR Pin Tied
to SVIN
3614 G17
Reference Voltage
vs Temperature
Switch On-Resistance
vs Input Voltage
0.05
0.606
VTRACK/SS
200mV/DIV
PGOOD
2V/DIV
0.04
MAIN SWITCH
0.602
RDS(0N) (Ω)
REFERENCE VOLTAGE (V)
0.604
VOUT
500mV/DIV
0.600
3614 G19
0.03
SYNCHRONOUS SWITCH
0.02
0.598
0.01
0.596
2ms/DIV
VOUT = 0V TO 1.2V
IOUT = 3A, VTRACK/SS = 0V TO 0.4V
VMODE = 1.5V, VSRLIM/DDR = 3.3V
3614 G18
2ms/DIV
VOUT = 0V TO 1.8V
IOUT = 3A, VTRACK/SS = 0V TO 0.7V
VMODE = 1.5V, VSRLIM/DDR = 0V
0.594
–50 –30 –10 10 30 50 70 90 110 130
TEMPERATURE (°C)
3614 G20
0
2.5
3.0
4.0
4.5
3.5
INPUT VOLTAGE (V)
5.0
5.5
3614 G21
3614f
6
LTC3614
TYPICAL PERFORMANCE CHARACTERISTICS VIN = 3.3V, RT/SYNC = SVIN unless otherwise noted.
Switch On-Resistance
vs Temperature
Frequency vs Resistor on
RT/SYNC Pin
0.040
MAIN SWITCH
0.035
FREQUENCY (kHz)
RDS(ON) (Ω)
0.8
4000
0.6
3500
0.030
SYNCHRONOUS SWITCH
0.025
0.020
0.015
3000
2500
2000
1500
0.010
1000
0.005
500
0
–50 –30 –10 10 30 50 70 90 110 130
TEMPERATURE (°C)
0
7000
200 400 600 800 1000 1200 1400
RESISTOR ON RT/SYNC PIN (kΩ)
0
–0.5
–1.0
–1.5
–2.0
7000
5000
4000
3000
2000
DYNAMIC SUPPLY CURRENT (mA)
PULSE-SKIPPING MODE
Burst Mode OPERATION
0.1
3.25 3.75 4.25 4.75
INPUT VOLTAGE (V)
5.25
3614 G28
5000
4000
3000
2000
0
–50 –30 –10 10 30 50 70 90 110 130
TEMPERATURE (°C)
3614 G27
Dynamic Supply Current vs
Temperature without AVP Mode
100
FORCED CONTINUOUS MODE
6000
3614 G26
FREQ = 2.25MHz
10
VIN = 2.25V
VIN = 3.3V
VIN = 5.5V
1000
0
–50 –30 –10 10 30 50 70 90 110 130
TEMPERATURE (°C)
5.25
Dynamic Supply Current vs Input
Voltage without AVP Mode
DYNAMIC SUPPLY CURRENT (mA)
8000
6000
3614 G25
2.75
–0.8
Switch Leakage vs Temperature,
Synchronous Switch
VIN = 2.25V
VIN = 3.3V
VIN = 5.5V
1000
0.01
2.25
–0.6
3614 G24
SWITCH LEAKAGE (nA)
SWITCH LEAKAGE (nA)
FREQUENCY VARIATION (%)
0.5
1
–0.4
Switch Leakage vs Temperature,
Main Switch
8000
3.25 3.75 4.25 4.75
INPUT VOLTAGE (V)
0
–0.2
3614 G23
1.0
2.75
0.2
–1.2
–50 –30 –10 10 30 50 70 90 100 130
TEMPERATURE (°C)
0
Frequency vs Input Voltage
–2.5
2.25
0.4
–1.0
3614 G22
100
Frequency vs Temperature
4500
FREQUENCY VARIATION (%)
0.045
10
1
VOUT Short to GND,
Forced Continuous Mode
FREQ = 2.25MHz
FORCED CONTINUOUS MODE
VOUT
500mV/DIV
PULSE-SKIPPING MODE
IL
5A/DIV
Burst Mode OPERATION
0.1
0.01
–50 –30 –10 10 30 50 70 90 110 130
TEMPERATURE (°C)
3614 G29
VOUT = 1.8V
IOUT = 0A
VMODE = 1.5V
100μs/DIV
3614 G30
3614f
7
LTC3614
TYPICAL PERFORMANCE CHARACTERISTICS VIN = 3.3V, RT/SYNC = SVIN unless otherwise noted.
Output Voltage During Sinking
vs Input Voltage (VOUT = 1.8V,
0.47μH Inductor)
Start-Up from Shutdown with
Prebiased Output (Overvoltage)
(Forced Continuous Mode)
1.88
PGOOD
5V/DIV
1.86
1.84
VOUT (V)
VOUT
500mV/DIV
1.82
–3A, 2MHz, 120°C
1.80
–3A, 2MHz, 25°C
1.78
IL
5A/DIV
1.76
50μs/DIV
PREBIASED VOUT = 2.2V
VOUT = 1.2V, IOUT = 0A
VMODE = 1.5V
3614 G31
1.74
2.25
2.75
3.25
4
4.5
INPUT VOLTAGE (V)
5.25
3614 G32
PIN FUNCTIONS
SRLIM/DDR (Pin 1): Slew Rate Limit. Tying this pin to
ground selects maximum slew rate. Minimum slew rate
is selected when the pin is open. Connecting a resistor
from SRLIM/DDR to ground allows the slew rate to be
continuously adjusted. If SRLIM/DDR is tied to SVIN, DDR
mode is selected. In DDR mode the slew rate limit is set
to maximum.
RT/SYNC (Pin 2): Oscillator Frequency. This pin provides
three ways of setting the constant switching frequency:
1. Connecting a resistor from RT/SYNC to ground will set
the switching frequency based on the resistor value.
2. Driving the RT/SYNC pin with an external clock signal
will synchronize the LTC3614 to the applied frequency.
The slope compensation is automatically adapted to the
external clock frequency.
SGND (Pin 3): Signal Ground. All small-signal and compensation components should connect to this ground, which
in turn should connect to PGND at a single point.
PVIN (Pins 4, 10, 11, 17): Power Input Supply. PVIN connects to the source of the internal P-channel power MOSFET.
This pin is independent of SVIN and may be connected to
the same voltage or to a lower voltage supply.
SW (Pins 5, 6, 7, 8, 13, 14, 15, 16): Switch Node. Connection to the inductor. These pins connect to the drains
of the internal power MOSFET switches.
NC (Pins 9, 12): Can be connected to ground or left
open.
SVIN (Pin 18): Signal Input Supply. This pin powers
the internal control circuitry and is monitored by the
undervoltage lockout comparator.
3. Tying the RT/SYNC pin to SVIN enables the internal
2.25MHz oscillator frequency.
3614f
8
LTC3614
PIN FUNCTIONS
RUN (Pin 19): Enable Pin. Forcing this pin to ground shuts
down the LTC3614. In shutdown, all functions are disabled
and the chip draws <1μA of supply current.
VFB (Pin 22): Voltage Feedback Input Pin. Senses the
feedback voltage from the external resistive divider across
the output.
PGOOD (Pin 20): Power Good. This open-drain output is
pulled down to SGND on start-up and while the FB voltage
is outside the power good voltage window. If the FB voltage increases and stays inside the power good window
for more than 100μs the PGOOD pin is released. If the
FB voltage leaves the power good window for more than
100μs the PGOOD pin is pulled down.
ITH (Pin 23): Error Amplifier Compensation. The current
comparator’s threshold increases with this control voltage. Tying this pin to SVIN enables internal compensation
and AVP mode.
In DDR mode (DDR = VIN), the power good window moves
in relation to the actual TRACK/SS pin voltage. During up/
down tracking the PGOOD pin is always pulled down.
1. Tying this pin to SVIN selects the internal soft-start
circuit.
In shutdown the PGOOD output will actively pull down
and may be used to discharge the output capacitors via
an external resistor.
MODE (Pin 21): Mode Selection. Tying the MODE pin
to SVIN or SGND enables pulse-skipping mode or Burst
Mode operation (with an internal Burst Mode clamp), respectively. If this pin is held at slightly higher than half of
SVIN, forced continuous mode is selected. Connecting this
pin to an external voltage between 0.45V and 0.8V selects
Burst Mode operation with the burst clamp set to the pin
voltage. See the Operation section for more details.
TRACK/SS (Pin 24): Track/External Soft-Start/External
Reference. Start-up behavior is programmable with the
TRACK/SS pin:
2. External soft-start timing can be programmed with a
capacitor to ground and a resistor to SVIN.
3. TRACK/SS can be used to force the LTC3614 to track
the start-up behavior of another supply.
The pin can also be used as external reference input. See the
Applications Information section for more information.
PGND (Exposed Pad Pin 25): Power Ground. This pin
connects to the source of the internal N-channel power
MOSFET. This pin should be connected close to the (–)
terminal of CIN and COUT.
3614f
9
LTC3614
FUNCTIONAL BLOCK DIAGRAM
SVIN
ITH
PVIN PVIN PVIN PVIN
ITH SENSE
COMPARATOR
+
BANDGAP
AND
BIAS
RUN
RT/SYNC
SGND
INTERNAL
COMPENSATION
OSCILLATOR
CURRENT
SENSE
–
SVIN – 0.3V
R
–
PMOS CURRENT
COMPARATOR
ITH
LIMIT
+
0.3V
–
FOLDBACK
AMPLIFIER
–
SLOPE
COMPENSATION
+
0.6V
+
VFB
+
ERROR
AMPLIFIER
–
BURST
COMPARATOR
SW
SLEEP
–
+
DRIVER
+
SW
SW
MODE
TRACK/SS
SW
SOFT-START
SW
0.555V
+
SW
SW
–
LOGIC
REVERSE
COMPARATOR
+
0.645V
SW
IREV
–
+
–
PGOOD
PGND
EXPOSED PAD
SRLIM/DDR
MODE
3614 BD
3614f
10
LTC3614
OPERATION
Main Control Loop
Mode Selection
The LTC3614 is a monolithic, constant frequency, current
mode step-down DC/DC converter. During normal operation, the internal top power switch (P-channel MOSFET) is
turned on at the beginning of each clock cycle. Current in
the inductor increases until the current comparator trips
and turns off the top power switch. The peak inductor current at which the current comparator trips is controlled by
the voltage on the ITH pin. The error amplifier adjusts the
voltage on the ITH pin by comparing the feedback signal
from a resistor divider on the VFB pin with an internal 0.6V
reference. When the load current increases, it causes a
reduction in the feedback voltage relative to the reference.
The error amplifier raises the ITH voltage until the average
inductor current matches the new load current. Typical
voltage range for the ITH pin is from 0.1V to 0.9V with
0.45V corresponding to zero current.
The MODE pin is used to select one of four different
operating modes:
When the top power switch shuts off, the synchronous
power switch (N-channel MOSFET) turns on until either
the bottom current limit is reached or the next clock cycle
begins. The bottom current limit is typically set at –8A for
forced continuous mode and 0A for Burst Mode operation
and pulse-skipping mode.
The operating frequency defaults to 2.25MHz when
RT/SYNC is connected to SVIN, or can be set by an external
resistor connected between the RT/SYNC pin and ground,
or by a clock signal applied to the RT/SYNC pin. The switching frequency can be set from 300kHz to 4MHz.
Overvoltage and undervoltage comparators pull the
PGOOD output low if the output voltage varies more than
±7.5% (typical) from the set point.
Mode Selection Voltage
SVIN
SVIN – 0.3V
SVIN • 0.58
1.1V
0.8V
0.45V
0.3V
SGND
PS
PULSE-SKIPPING MODE ENABLE
FC
FORCED CONTINUOUS MODE ENABLE
BM
EXT
Burst Mode ENABLE—EXTERNAL CLAMP,
CONTROLLED BY VOLTAGE APPLIED AT
MODE PIN
BM
Burst Mode ENABLE—INTERNAL CLAMP
3614 OP01
Burst Mode Operation—Internal Clamp
Connecting the MODE pin to SGND enables Burst Mode
operation with an internal clamp. In Burst Mode operation
the internal power switches operate intermittently at light
loads. This increases efficiency by minimizing switching
losses. During the intervals when the switches are idle,
the LTC3614 enters sleep state where many of the internal
circuits are disabled to save power. During Burst Mode
operation, the minimum peak inductor current is internally
clamped and the voltage on the ITH pin is monitored by
the burst comparator to determine when sleep mode is
enabled and disabled. When the average inductor current
is greater than the load current, the voltage on the ITH pin
drops. As the ITH voltage falls below the internal clamp,
the burst comparator trips and enables sleep mode. During sleep mode, both power MOSFETs are held off and
the load current is solely supplied by the output capacitor.
When the output voltage drops, the top power switch is
turned back on and the internal circuits are re-enabled.
This process repeats at a rate that is dependent on the
load current.
3614f
11
LTC3614
OPERATION
Burst Mode Operation—External Clamp
Dropout Operation
Connecting the MODE pin to a voltage in the range of 0.45V
to 0.8V enables Burst Mode operation with external clamp.
During this mode of operation the minimum voltage on the
ITH pin is externally set by the voltage on the MODE pin. It
is recommended to use Burst Mode operation with internal
burst clamp for temperatures above 85°C ambient.
As the input supply voltage approaches the output voltage,
the duty cycle increases toward the maximum on-time.
Further reduction of the supply voltage forces the main
switch to remain on for more than one cycle, eventually
reaching 100% duty cycle. The output voltage will then be
determined by the input voltage minus the voltage drop
across the internal P-channel MOSFET and the inductor.
Pulse-Skipping Mode Operation
Pulse-skipping mode is similar to Burst Mode operation,
but the LTC3614 does not disable power to the internal
circuitry during sleep mode. This improves output voltage
ripple but uses more quiescent current, compromising
light load efficiency.
Tying the MODE pin to SVIN enables pulse-skipping mode.
As the load current decreases, the peak inductor current
will be determined by the voltage on the ITH pin until the
ITH voltage drops below the voltage level corresponding to
0A. At this point, the peak inductor current is determined
by the minimum on-time of the current comparator. If the
load demand is less than the average of the minimum ontime inductor current, switching cycles will be skipped to
keep the output voltage in regulation.
Forced Continuous Mode
In forced continuous mode the inductor current is constantly cycled which creates a minimum output voltage
ripple at all output current levels.
Connecting the MODE pin to a voltage in the range of
1.1V to SVIN • 0.58 will enable forced continuous mode
operation.
At light loads, forced continuous mode operation is less
efficient than Burst Mode or pulse-skipping operation, but
may be desirable in some applications where it is necessary
to keep switching harmonics out of the signal band.
Low Supply Operation
The LTC3614 is designed to operate down to an input
supply voltage of 2.25V. An important consideration at low
input supply voltages is that the RDS(ON) of the P-channel
and N-channel power switches increases. The user should
calculate the power dissipation when the LTC3614 is used
at 100% duty cycle with low input voltages to ensure that
thermal limits are not exceeded. See the Typical Performance Characteristics graphs.
Short-Circuit Protection
The peak inductor current at which the current comparator
shuts off the top power switch is controlled by the voltage
on the ITH pin.
If the output current increases, the error amplifier raises
the ITH pin voltage until the average inductor current
matches the new load current. In normal operation the
LTC3614 clamps the maximum ITH pin voltage at approximately 0.9V which corresponds typically to 9A peak
inductor current.
When the output is shorted to ground, the inductor current
decays very slowly during a single switching cycle. The
LTC3614 uses two techniques to prevent current runaway
from occurring.
Forced continuous mode must be used if the output is
required to sink current.
3614f
12
LTC3614
OPERATION
If the output voltage drops below 50% of its nominal value,
the clamp voltage at ITH pin is lowered causing the maximum peak inductor current to decrease gradually with the
output voltage. When the output voltage reaches 0V the
clamp voltage at the ITH pin drops to 40% of the clamp
voltage during normal operation. The short-circuit peak
inductor current is determined by the minimum on-time
of the LTC3614, the input voltage and the inductor value.
This foldback behavior helps in limiting the peak inductor
current when the output is shorted to ground. It is disabled
during internal or external soft-start and tracking up/down
operation (see the Applications Information section).
A secondary limit is also imposed on the valley inductor
current. If the inductor current measured through the
bottom MOSFET increases beyond 12A typical, the top
power MOSFET will be held off and switching cycles will
be skipped until the inductor current is reduced.
APPLICATIONS INFORMATION
The basic LTC3614 application circuit is shown in Figure 1.
Operating Frequency
Selection of the operating frequency is a trade-off between
efficiency and component size. High frequency operation
allows the use of smaller inductor and capacitor values.
Operation at lower frequencies improves efficiency by
reducing internal gate charge losses but requires larger
inductance values and/or capacitance to maintain low
output ripple voltage.
The operating frequency of the LTC3614 is determined
by an external resistor that is connected between the
RT/SYNC pin and ground. The value of the resistor sets
the ramp current that is used to charge and discharge an
internal timing capacitor within the oscillator and can be
calculated by using the following equation:
RT =
3.82 • 1011Hz
Ω – 16kΩ
fOSC (Hz )
Although frequencies as high as 4MHz are possible, the
minimum on-time of the LTC3614 imposes a minimum
limit on the operating duty cycle. The minimum on-time
is typically 60ns; therefore, the minimum duty cycle is
equal to 60ns • fOSC(Hz)•100%.
Tying the RT/SYNC pin to SVIN sets the default internal
operating frequency to 2.25MHz ±20%.
VIN
2.25V TO 5.5V
RSS
2M
CSS
22nF
RC
15k
CC
470pF
RT
130k
CC1
10pF
(OPT)
PVIN
SVIN
RUN
TRACK/SS SRLIM/DDR
RT/SYNC
LTC3614
SW
PGOOD
SGND
ITH
PGND
MODE
VFB
CIN1
10μF
s4
L1
330nH
VOUT
1.8V
COUT2 4A
100μF
R1
392k
3614 F01
R2
196k
Figure 1. 1.8V, 4A Step-Down Regulator
3614f
13
LTC3614
APPLICATIONS INFORMATION
Frequency Synchronization
Inductor Selection
The LTC3614’s internal oscillator can be synchronized to
an external frequency by applying a square wave clock
signal to the RT/SYNC pin. During synchronization, the top
switch turn-on is locked to the falling edge of the external
frequency source. The synchronization frequency range
is 300kHz to 4MHz. During synchronization all operation
modes can be selected.
For a given input and output voltage, the inductor value
and operating frequency determine the ripple current. The
ripple current ΔIL increases with higher VIN and decreases
with higher inductance:
It is recommended that the regulator is powered down
(RUN pin to ground) before removing the clock signal
on the RT/SYNC pin in order to reduce inductor current
ripple.
AC coupling should be used if the external clock generator cannot provide a continuous clock signal throughout
start-up, operation and shutdown of the LTC3614. The
size of capacitor CSYNC depends on parasitic capacitance
on the RT/SYNC pin and is typically in the range of 10pF
to 22pF.
VIN
LTC3614
SVIN
RT/SYNC
VIN
LTC3614
SVIN
0.4V
RT/SYNC
SGND
RT
VIN
LTC3614
SVIN
RT/SYNC
SGND
fOSC
2.25MHz
LTC3614
SVIN
RT/SYNC
SGND
⎛
⎞ ⎛
⎞
V
VOUT
• ⎜ 1– OUT ⎟
L=⎜
⎟
⎝ fSW • ΔIL(MAX) ⎠ ⎝ VIN(MAX) ⎠
fOSC t1/RT
fOSC
1/TP
Inductor Core Selection
fOSC
1/TP
Once the value for L is known, the type of inductor must be
selected. Actual core loss is independent of core size for
fixed inductor value, but it is very dependent on the inductance selected. As the inductance increases, core losses decrease. Unfortunately, increased inductance requires more
turns of wire and therefore, copper losses will increase.
TP
VIN
Having a lower ripple current reduces the core losses
in the inductor, the ESR losses in the output capacitors
and the output voltage ripple. A reasonable starting point
for selecting the ripple current is ΔIL = 0.3 • IOUT(MAX).
The largest ripple current occurs at the highest VIN. To
guarantee that the ripple current stays below a specified
maximum, the inductor value should be chosen according
to the following equation:
The inductor value will also have an effect on Burst Mode
operation. The transition to low current operation begins
when the peak inductor current falls below a level set by the
burst clamp. Lower inductor values result in higher ripple
current which causes this to occur at lower load currents.
This causes a dip in efficiency in the upper range of low current operation. In Burst Mode operation, lower inductance
values will cause the burst frequency to increase.
1.2V
0.3V
CSYNC
⎛ V
⎞ ⎛ V ⎞
ΔIL = ⎜ OUT ⎟ • ⎜ 1– OUT ⎟
VIN ⎠
⎝ fSW • L ⎠ ⎝
RT
3614 F02
Figure 2. Setting the Switching Frequency
3614f
14
LTC3614
APPLICATIONS INFORMATION
Ferrite designs have very low core losses and are preferred
at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” meaning that inductance
collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor
ripple current and consequently output voltage ripple. Do
not allow a ferrite core to saturate!
Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid
or shielded pot cores in ferrite or permalloy materials are
small and don’t radiate much energy, but generally cost
more than powdered iron core inductors with similar
characteristics. The choice of which style inductor to use
mainly depends on the price versus size requirements
and any radiated field/EMI requirements. Table 1 shows
some typical surface mount inductors that work well in
LTC3614 applications.
Input Capacitor (CIN) Selection
In continuous mode, the source current of the top P-channel
MOSFET is a square wave of duty cycle VOUT/VIN. To prevent
large input voltage transients, a low ESR capacitor sized
for the maximum RMS current must be used at VIN.
The maximum RMS capacitor current is given by:
⎛ V
⎞
V
IRMS =IOUT(MAX) • OUT • ⎜ IN – 1⎟
VIN
⎝ VOUT ⎠
Table 1. Representative Surface Mount Inductors
INDUCTANCE
(μH)
DCR
(mΩ)
SATURATION
CURRENT (A)
DIMENSIONS
(mm)
HEIGHT
(mm)
Vishay IHLP-2525CZ-01
0.10
1.5
60
6.5 × 6.9
3
0.15
1.9
52
6.5 × 6.9
3
0.20
2.4
41
6.5 × 6.9
3
0.22
2.5
40
6.5 × 6.9
3
0.33
3.5
30
6.5 × 6.9
3
0.47
4
26
6.5 × 6.9
3
Sumida CDMC6D28 Series
0.2
2.5
21.7
7.25 × 4.4
3
0.3
3.2
15.4
7.25 × 4.4
3
0.47
4.2
13.6
7.25 × 4.4
3
Cooper HCP0703 Series
0.22
2.8
23
7 × 7.3
3.0
0.47
4.2
17
7 × 7.3
3.0
0.68
5.5
15
7 × 7.3
3.0
Wurth Electronik WE-HC744312 Series
0.25
2.5
18
7 × 7.7
3.8
0.47
3.4
16
7 × 7.7
3.8
Coilcraft SLC7530 Series
0.100
0.123
20
7.5 × 6.7
3
0.188
0.100
21
7.5 × 6.7
3
0.272
0.100
14
7.5 × 6.7
3
0.350
0.100
11
7.5 × 6.7
3
0.400
0.100
8
7.5 × 6.7
3
This formula has a maximum at VIN = 2VOUT , where IRMS =
IOUT/2. This simple worst-case condition is commonly used
for design because even significant deviations do not offer
much relief. Note that ripple current ratings from capacitor
manufacturers are often based on only 2000 hours of life
which makes it advisable to further derate the capacitor,
or choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to meet
size or height requirements in the design.
3614f
15
LTC3614
APPLICATIONS INFORMATION
Output Capacitor (COUT ) Selection
The selection of COUT is typically driven by the required
ESR to minimize voltage ripple and load step transients
(low ESR ceramic capacitors are discussed in the next
section). Typically, once the ESR requirement is satisfied,
the capacitance is adequate for filtering. The output ripple
ΔVOUT is determined by:
⎛
⎞
1
ΔVOUT ≤ ΔIL • ⎜ ESR +
8 • fSW • COUT ⎟⎠
⎝
where fOSC = operating frequency, COUT = output capacitance and ΔIL = ripple current in the inductor. The output
ripple is highest at maximum input voltage since ΔIL
increases with input voltage.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the capacitance, ESR or RMS
current handling requirement of the application. Aluminum
electrolytic, special polymer, ceramic and dry tantalum
capacitors are all available in surface mount packages.
Tantalum capacitors have the highest capacitance density,
but can have higher ESR and must be surge tested for
use in switching power supplies. Aluminum electrolytic
capacitors have significantly higher ESR, but can often
be used in extremely cost-sensitive applications provided
that consideration is given to ripple current ratings and
long-term reliability.
Ceramic Input and Output Capacitors
Ceramic capacitors have the lowest ESR and can be cost
effective, but also have the lowest capacitance density,
high voltage and temperature coefficients, and exhibit
audible piezoelectric effects. In addition, the high Q of
ceramic capacitors along with trace inductance can lead
to significant ringing.
Ceramic capacitors are prone to temperature effects
which require the designer to check loop stability over
the operating temperature range. To minimize their large
temperature and voltage coefficients, only X5R or X7R
ceramic capacitors should be used.
When a ceramic capacitor is used at the input and the power
is being supplied through long wires, such as from a wall
adapter, a load step at the output can induce ringing at the
VIN pin. At best, this ringing can couple to the output and
be mistaken as loop instability. At worst, the ringing at the
input can be large enough to damage the part.
Since the ESR of a ceramic capacitor is so low, the input
and output capacitor must instead fulfill a charge storage
requirement. During a load step, the output capacitor must
instantaneously supply the current until the feedback loop
raises the switch current enough to support the load. The
time required for the feedback loop to respond is dependent
on the compensation components and the output capacitor size. Typically, 3 to 4 cycles are required to respond
to a load step, but only in the first cycle does the output
drop linearly. The output droop, VDROOP , is usually about
2 to 4 times the linear drop of the first cycle; however,
this behavior can vary depending on the compensation
component values. Thus, a good place to start is with the
output capacitor size of approximately:
COUT ≈
3.5 • ΔIOUT
fSW • VDROOP
This is only an approximation; more capacitance may
be needed depending on the duty cycle and load step
requirements.
In most applications, the input capacitor is merely required
to supply high frequency bypassing, since the impedance
to the supply is very low.
They are attractive for switching regulator use because
of their very low ESR, but great care must be taken when
using only ceramic input and output capacitors.
3614f
16
LTC3614
APPLICATIONS INFORMATION
Output Voltage Programming
The resistive divider allows pin VFB to sense a fraction of
the output voltage as shown in Figure 1.
Pulse-skipping mode, which is a compromise between low
output voltage ripple and efficiency, can be implemented
by connecting MODE to SVIN. This sets IBURST to 0A. In
this condition, the peak inductor current is limited by the
minimum on-time of the current comparator. The lowest
output voltage ripple is achieved while still operating
discontinuously. During very light output loads, pulseskipping allows only a few switching cycles to skip while
maintaining the output voltage in regulation.
Burst Clamp Programming
Internal and External Compensation
If the voltage on the MODE pin is less than 0.8V, Burst
Mode operation is enabled.
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC load current.
When a load step occurs, VOUT shifts by an amount equal
to ΔILOAD(ESR), where ESR is the effective series resistance
of COUT . ΔILOAD also begins to charge or discharge COUT ,
generating the feedback error signal that forces the regulator to adapt to the current change and return VOUT to its
steady-state value. During this recovery time VOUT can
be monitored for excessive overshoot or ringing, which
would indicate a stability problem. The availability of the
ITH pin allows the transient response to be optimized over
a wide range of output capacitance.
The output voltage is set by an external resistive divider
according to the following equation:
⎛ R1⎞
VOUT = 0.6 • ⎜ 1+ ⎟ V
⎝ R2 ⎠
If the voltage on the MODE pin is less than 0.3V, the internal
default burst clamp level is selected. The minimum voltage
on the ITH pin is typically 525mV (internal clamp).
If the voltage is between 0.45V and 0.8V, the voltage on
the MODE pin (VBURST) is equal to the minimum voltage
on the ITH pin (external clamp) and determines the burst
clamp level IBURST (typically from 0A to 7A).
When the ITH voltage falls below the internal (or external)
clamp voltage, the sleep state is enabled.
As the output load current drops, the peak inductor current
decreases to keep the output voltage in regulation. When
the output load current demands a peak inductor current
that is less than IBURST , the burst clamp will force the peak
inductor current to remain equal to IBURST regardless of
further reductions in the load current.
Since the average inductor current is greater than the output
load current, the voltage on the ITH pin will decrease. When
the ITH voltage drops, sleep mode is enabled in which
both power switches are shut off along with most of the
circuitry to minimize power consumption. All circuitry is
turned back on and the power switches resume operation when the output voltage drops out of regulation. The
value for IBURST is determined by the desired amount of
output voltage ripple. As the value of IBURST increases, the
sleep period between pulses and the output voltage ripple
increase. Note that for very high VBURST voltage settings,
the power good comparator may trip, since the output
ripple may get bigger than the power good window.
The ITH external components (RC and CC) shown in Figure 1 provide adequate compensation as a starting point
for most applications. The values can be modified slightly
to optimize transient response once the final PCB layout
is done and the particular output capacitor type and value
have been determined. The output capacitors need to be
selected because the various types and values determine
the loop gain and phase. The gain of the loop will be increased by increasing RC and the bandwidth of the loop
will be increased by decreasing CC. If RC is increased by
the same factor that CC is decreased, the zero frequency
will be kept the same, thereby keeping the phase shift the
same in the most critical frequency range of the feedback
loop. The output voltage settling behavior is related to the
stability of the closed-loop system. The external capacitor, CC1, (Figure 1) is not needed for loop stability, but it
helps filter out any high frequency noise that may couple
onto that node.
The first circuit in the Typical Applications section uses
faster compensation to improve step response.
3614f
17
LTC3614
APPLICATIONS INFORMATION
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT , causing a rapid drop in VOUT . No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. More output
capacitance may be required depending on the duty cycle
and load step requirements.
AVP Mode
Fast load transient response, limited board space and low
cost are typical requirements of microprocessor power
supplies. A microprocessor will typically exhibit full load
steps with very fast slew rate. The voltage at the microprocessor must be held to about ±0.1V of nominal in spite
of these load current steps. Since the control loop cannot
respond this fast, the output capacitors must supply the
load current until the control loop can respond.
Normally, several capacitors in parallel are required to
meet microprocessor transient requirements. Capacitor
ESR and ESL primarily determine the amount of droop or
overshoot in the output voltage.
Consider the LTC3614 without AVP with a bank of tantalum
output capacitors. If a load step with very fast slew rate
occurs, the voltage excursion will be seen in both directions, for full load to minimum load transient and for the
minimum load to full load transient.
If the ITH pin is tied to SVIN, the active voltage positioning
(AVP) mode and internal compensation are selected.
AVP mode intentionally compromises load regulation by
reducing the gain of the feedback circuit, resulting in an
output voltage that slightly varies with load current. When
the load current suddenly increases, the output voltage
starts from a level slightly higher than nominal so the output
voltage can droop more and stay within the specified voltage range. When the load current suddenly decreases the
output voltage starts at a level lower than nominal so the
output voltage can have more overshoot and stay within
the specified voltage range (see Figures 3 and 4).
The benefit is a lower peak-to-peak output voltage deviation
for a given load step without having to increase the output
filter capacitance. Alternatively, the output voltage filter capacitance can be reduced while maintaining the same peak
to peak transient response. Due to the reduced loop gain
in AVP mode, no external compensation is required.
VOUT
100mV/DIV
VOUT
200mV/DIV
IL
1A/DIV
IL
1A/DIV
50μs/DIV
VIN = 3.3V
VOUT = 1.8V
ILOAD = 100mA TO 3A
VMODE = 1.5V
COMPENSATION FIGURE 1
3614 F03
Figure 3. Load Step Transient Forced
Continuous Mode (AVP Inactive)
50μs/DIV
VIN = 3.3V
VOUT = 1.8V
ILOAD = 100mA TO 3A
VMODE = 1.5V
VITH = 3.3V
OUTPUT CAPACITOR VALUE FIGURE 1
3614 F04
Figure 4. Load Step Transient Forced
Continuous Mode with AVP Mode
3614f
18
LTC3614
APPLICATIONS INFORMATION
The LTC3614 can both source and sink current if the MODE
pin is configured to forced continuous mode.
Current sinking is typically limited to 3A for 1MHz frequency
and a 0.47μH inductor, but can be lower at higher frequencies and low output voltages. If higher ripple current can
be tolerated, smaller inductor values can increase the sink
current limit. See the Typical Performance Characteristics
curves for more information.
In addition, by tying the SRLIM/DDR pin to SVIN, lower
external reference voltage and tracking output voltage are
possible. See the Output Voltage Tracking and External
Reference Input sections.
Soft-Start
The RUN pin provides a means to shut down the LTC3614.
Tying the RUN pin to SGND places the LTC3614 in a low
quiescent current shutdown state (IQ < 1μA).
When the LTC3614 is enabled by pulling the RUN pin high,
the chip enters a soft start-up state. The type of soft startup behavior is set by the TRACK/SS pin:
1. Tying TRACK/SS to SVIN selects the internal soft-start
circuit. This circuit ramps the output voltage to the final
value within 1ms.
2. If a longer soft-start period is desired, it can be set externally with a resistor and capacitor on the TRACK/SS
pin as shown in Figure 1. The TRACK/SS pin reduces the
SW PIN
10k
100k
OPEN
value of the internal reference at VFB until TRACK/SS is
pulled above 0.6V. The external soft-start duration can
be calculated by using the following formula:
⎛
⎞
SVIN
t SS = RSS • CSS • ln⎜
⎝ SVIN – 0.6V ⎟⎠
3. The TRACK/SS pin can be used to track the output
voltage of another supply.
Each time the RUN pin is tied high and the LTC3614 is
turned on, the TRACK/SS pin is internally pulled down
for ten microseconds in order to discharge the external
capacitor. This discharging time is typically adequate
for capacitors up to about 33nF. If a larger capacitor is
required, connect the external soft-start resistor to the
RUN pin.
During either internal or external soft-start, the MODE pin
is ignored and soft-start will always be in pulse-skipping
mode. In addition, the PGOOD pin is kept low and foldback
of the switching frequency is disabled.
Programmable Switch Pin Slew Rate
As switching frequencies rise, it is desirable to minimize the
transition time required when switching to minimize power
losses and blanking time for the switch to settle. However,
fast slewing of the switch node results in relatively high
external radiated EMI and high on chip supply transients,
which can cause problems for some applications.
SW PIN
DDR Mode
OPEN
100k
10k
VIN = 3.3V
VOUT = 1.8V
fSW = 2.25MHz
2ns/DIV
VIN = 3.3V
VOUT = 1.8V
fSW = 2.25MHz
2ns/DIV
3614 F05
Figure 5. Slew Rate at SW Pin vs SRLIM/DDR Resistor: Open, 100k, 10k
3614f
19
LTC3614
APPLICATIONS INFORMATION
The LTC3614 allows the user to control the slew rate of the
switching node SW by using the SRLIM/DDR pin. Tying
this pin to ground selects the fastest slew rate. The slowest slew rate is selected when the pin is open. Connecting
a resistor (between 10k and 100k) from SRLIM pin to
ground adjusts the slew rate between the maximum and
minimum values. The reduced dV/dt of the switch node
results in a significant reduction of the supply and ground
ringing, as well as lower radiated EMI.
OUTPUT VOLTAGE
VOUT1
VOUT2
TIME
(6a) Coincident Tracking
Particular attention should be used with very high switching
frequencies. Using the slowest slew rate (SRLIM open)
can reduce the minimum duty cycle capability.
OUTPUT VOLTAGE
VOUT1
Output Voltage Tracking Input
If the DDR pin is not tied to SVIN, once VTRACK/SS exceeds
0.6V, the run state is entered and the MODE selection, power
good and current foldback circuits are enabled.
In the run state, the TRACK/SS pin can be used for tracking down/up the output voltage of another supply. If the
VTRACK/SS drops below 0.6V, the LTC3614 enters the down
tracking state and VOUT is referenced to the TRACK/SS voltage. If the TRACK/SS pin drops below 0.2V, the switching
frequency is reduced to ensure that the minimum duty
cycle limit does not prevent the output from following
the TRACK/SS pin. The run state will resume if VTRACK/SS
again exceeds 0.6V and VOUT is referenced to the internal
precision reference (see Figure 8).
3614 F06
TIME
(6b) Ratiometric Tracking
Figure 6. Two Different Modes of Output Voltage Tracking
VOUT1
VOUT2
R4
R4
R3
VFB2
R2
LTC3614
Through the TRACK/SS pin, the output voltage can be set
up for either coincident or ratiometric tracking, as shown
in Figure 6.
To implement the coincident tracking behavior in Figure 6a, connect an extra resistive divider to the output
of the master channel and connect its midpoint to the
TRACK/SS pin for the slave channel. The ratio of this
divider should be selected to be the same as that of the
slave channel’s feedback divider (Figure 7a). In this tracking mode, the master channel’s output must be set higher
than slave channel’s output. To implement the ratiometric
tracking behavior in Figure 6b, different resistor divider
values must be used as specified in Figure 7b.
VOUT2
VFB1
R2
R2
TRACK/SS2
VIN
LTC3614
TRACK/SS1
R4 ≤ R3
LTC3614 CHANNEL 2
SLAVE
LTC3614 CHANNEL 1
MASTER
3614 F07a
Figure 7a. Setup for Coincident Tracking
VOUT1
VOUT2
R1
R5
R3 R1/R2 < R5/R6
R6
R4
VFB2
R2
LTC3614
TRACK/SS2
LTC3614 CHANNEL 2
SLAVE
VFB1
LTC3614
TRACK/SS1
VIN
LTC3614 CHANNEL 1
3614 F07b
MASTER
Figure 7b. Setup for Ratiometric Tracking
3614f
20
LTC3614
APPLICATIONS INFORMATION
For coincident start-up, the voltage value at the TRACK/SS
pin for the slave channel needs to reach the final reference
value after the internal soft-start time (around 1ms). The
master start-up time needs to be adjusted with an external
capacitor and resistor to ensure this.
External Reference Input (DDR Mode)
If the DDR pin is tied to SVIN (DDR mode), the run state
is entered when VTRACK/SS exceeds 0.3V and tracking
down behavior is possible if the VTRACK/SS voltage is
below 0.6V.
This allows TRACK/SS to be used as an external reference
between 0.3V and 0.6V if desired. During the run state in
DDR mode, the power good window moves in relation
to the actual TRACK/SS pin voltage if the voltage value
is between 0.3V and 0.6V. Note: if TRACK/SS voltage is
0.6V, either the tracking circuit or the internal reference
can be used.
During up/down tracking the output current foldback is
disabled and the PGOOD pin is always pulled down (see
Figure 9).
Efficiency Considerations
The efficiency of a switching regulator is equal to the output
power divided by the input power times 100%. It is often
useful to analyze individual losses to determine what is
limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of
the losses: VIN quiescent current and I2R losses. The VIN
quiescent current loss dominates the efficiency loss at
very low load currents whereas the I2R loss dominates
the efficiency loss at medium to high load currents. In a
typical efficiency plot, the efficiency curve at very low load
currents can be misleading since the actual power lost is
usually of no consequence.
1. The VIN quiescent current is due to two components: the
DC bias current as given in the Electrical Characteristics
and the internal main switch and synchronous switch
gate charge currents. The gate charge current results
from switching the gate capacitance of the internal power
MOSFET switches. Each time the gate is switched from
low to high to low again, a packet of charge dQ moves
from VIN to ground. The resulting dQ/dt is the current
out of VIN due to gate charge, and it is typically larger
than the DC bias current. Both the DC bias and gate
charge losses are proportional to VIN; thus, their effects
will be more pronounced at higher supply voltages.
2. I2R losses are calculated from the resistances of the
internal switches, RSW , and external inductor, RL. In
continuous mode the average output current flowing
through inductor L is “chopped” between the main
switch and the synchronous switch. Thus, the series
resistance looking into the SW pin is a function of both
top and bottom MOSFET RDS(ON) and the duty cycle
(DC) as follows:
RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Characteristics curves. To obtain I2R losses, simply add RSW to
RL and multiply the result by the square of the average
output current.
Other losses including CIN and COUT ESR dissipative
losses and inductor core losses generally account for
less than 2% of the total loss.
3614f
21
LTC3614
APPLICATIONS INFORMATION
VFB PIN 0.6V
VOLTAGE
0V
0.6V
TRACK/SS
PIN VOLTAGE 0.2V
0V
RUN PIN
VOLTAGE
SVIN PIN
VOLTAGE
VIN
0V
VIN
0V
TIME
SHUTDOWN SOFT-START
STATE
STATE
tSS > 1ms
RUN STATE
REDUCED
SWITCHING
FREQUENCY
DOWN
TRACKING
STATE
RUN STATE
3614 F08
UP
TRACKING
STATE
Figure 8. DDR Pin Not Tied to SVIN
0.45V
VFB PIN 0.3V
VOLTAGE 0V
EXTERNAL
VOLTAGE
REFERENCE 0.45V
0.45V
TRACK/SS 0.3V
PIN VOLTAGE 0.2V
0V
VIN
RUN PIN
VOLTAGE
SVIN PIN
VOLTAGE
0V
VIN
0V
TIME
SHUTDOWN SOFT-START
STATE
STATE
tSS > 1ms
RUN STATE
REDUCED
SWITCHING
FREQUENCY
DOWN
TRACKING
STATE
RUN STATE
3614 F09
UP
TRACKING
STATE
Figure 9. DDR Pin Tied to SVIN. Example DDR Application
3614f
22
LTC3614
APPLICATIONS INFORMATION
Thermal Considerations
In most applications, the LTC3614 does not dissipate much
heat due to its high efficiency.
However, in applications where the LTC3614 is running at
high ambient temperature with low supply voltage and high
duty cycles, such as in dropout, the heat dissipated may
exceed the maximum junction temperature of the part. If
the junction temperature reaches approximately 160°C,
both power switches will be turned off and the SW node
will become high impedance.
To prevent the LTC3614 from exceeding the maximum
junction temperature, some thermal analysis is required.
The temperature rise is given by:
TRISE = (PD)(θJA)
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to
the ambient temperature. The junction temperature, TJ,
is given by:
TJ = TA + TRISE
where TA is the ambient temperature.
As an example, consider the case when the LTC3614 is in
dropout at an input voltage of 3.3V with a load current of
4A at an ambient temperature of 85°C. From the Typical
Performance Characteristics graph of Switch Resistance,
the RDS(ON) resistance of the P-channel switch is 0.038Ω.
Therefore, power dissipated by the part is:
PD = (IOUT)2 • RDS(ON) = 0.61W
For the QFN package, the θJA is 38°C/W.
Therefore, the junction temperature of the regulator operating at 85°C ambient temperature is approximately:
TJ = 0.61W • 38°C/W + 85°C = 108°C
We can safely assume that the actual junction temperature
will not exceed the absolute maximum junction temperature of 125°C.
Note that for very low input voltage, the junction temperature will be higher due to increased switch resistance,
RDS(ON). It is not recommended to use full load current
with high ambient temperature and low input voltage.
To maximize the thermal performance of the LTC3614 the
exposed pad should be soldered to a ground plane. See
the PCB Layout Board Checklist.
3614f
23
LTC3614
PACKAGE DESCRIPTION
Design Example
As a design example, consider using the LTC3614 in an
application with the following specifications:
VIN = 2.25V to 5.5V, VOUT = 1.8V, IOUT(MAX) = 4A, IOUT(MIN)
= 200mA, f = 2.6MHz.
Efficiency is important at both high and low load current,
so Burst Mode operation will be utilized.
Finally, define the soft start-up time choosing the proper
value for the capacitor and the resistor connected to
TRACK/SS. If we set minimum tSS = 5ms and a resistor
of 2MΩ, the following equation can be solved with the
maximum SVIN = 5.5V :
CSS =
5ms
= 21.6nF
5.5V ⎞
⎛
2MΩ •In ⎜
⎝ 5.5V – 0.6V ⎟⎠
First, calculate the timing resistor:
3.8211Hz
RT =
k – 16k = 130kΩ
2.6MHz
Next, calculate the inductor value for about 33% ripple
current at maximum VIN:
1.8V
⎛
⎞ ⎛ 1.8V ⎞
L=⎜
• 1–
= 0.35µH
⎝ 2.6MHz • 1.3A ⎟⎠ ⎜⎝ 5.5V ⎟⎠
Using a standard value of 0.33μH inductor results in a
maximum ripple current of:
1.8V
⎛
⎞ ⎛ 1.8V ⎞
ΔIL = ⎜
• ⎜ 1–
⎟ = 1.41A
⎝ 2.6MHz • 0.33µH⎟⎠ ⎝ 5.5V ⎠
The standard value of 22nF guarantees the minimum
soft-start up time of 5ms.
Figure 1 shows the schematic for this design example.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3614:
1. A ground plane is recommended. If a ground plane layer
is not used, the signal and power grounds should be
segregated with all small-signal components returning
to the SGND pin at one point which is then connected
to the PGND pin close to the LTC3614.
COUT will be selected based on the ESR that is required
to satisfy the output voltage ripple requirement and the
bulk capacitance needed for loop stability. For this design,
a 100μF ceramic capacitor is used with a X5R or X7R
dielectric.
2. Connect the (+) terminal of the input capacitor(s), CIN,
as close as possible to the PVIN pin, and the (–) terminal
as close as possible to the exposed pad, PGND. This
capacitor provides the AC current into the internal power
MOSFETs.
Assuming worst-case conditions of VIN = 2VOUT, CIN should
be selected for a maximum current rating of:
3. Keep the switching node, SW, away from all sensitive
small-signal nodes.
IRMS = 4A •
1.8V
⎛ 3.6V ⎞
• ⎜
– 1 = 2ARMS
⎝ 1.8V ⎟⎠
3.6V
Decoupling PVIN with four 10μF to 22μF capacitors is
adequate for most applications.
If we set R2 = 196k, the value of R1 can now be determined
by solving the following equation.
4. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of
power components. Connect the copper areas to PGND
(exposed pad) for best performance.
5. Connect the VFB pin directly to the feedback resistors.
The resistor divider must be connected between VOUT
and SGND.
⎛ 1.8V ⎞
R1 = 196k • ⎜
−1
⎝ 0.6V ⎟⎠
A value of 392k will be selected for R1.
3614f
24
LTC3614
TYPICAL APPLICATIONS
General Purpose Buck Regulator with Fast Compensation
and Improved Step Response, 2.25MHz
VIN
2.25V TO 5.5V
10μF
s4
RF
24Ω
CF
1μF
RSS
4.7M
CSS
10nF
R4
100k
RC
43k
CC
220pF
PGOOD
CC1
10pF
R5A
1M
SVIN
PVIN
RUN
TRACK/SS SRLIM/DDR
RT/SYNC
LTC3614
SW
PGOOD
SGND
ITH
PGND
MODE
VFB
R2
196k
R5B
1M
L1
0.33μH
CO2
100μF
VOUT
1.8V
4A
R1
392k
C3
22pF
3614 TA02a
L1: VISHAY IHLP-2525CZ-01 330nH
Load Step Response in
Forced Continuous Mode
Efficiency vs Output Current
100
VOUT = 1.8V
90
EFFICIENCY (%)
80
VOUT
100mV/DIV
70
60
50
40
IOUT
2A/DIV
30
VIN = 2.5V
VIN = 3.3V
VIN = 4V
VIN = 5.5V
20
10
0
1
10
100
1000
OUTPUT CURRENT (mA)
10000
50μs/DIV
VIN = 3.3V
VOUT = 1.8V
IOUT = 100mA TO 4A
VMODE = 1.5V
3614 TA02c
3614 TA02b
3614f
25
LTC3614
TYPICAL APPLICATIONS
Master and Slave for Coincident Tracking Outputs Using a 1MHz External Clock
VIN
2.25V TO 5.5V
22μF
s4
4.7M
10nF
RF1
24Ω
CF1
1μF
1MHz
CLOCK
R5
100k
RC1
15k
CC1
470pF
PGOOD
CC2
10pF
1M
SVIN
PVIN
RUN
TRACK/SS SRLIM/DDR
RT/SYNC
LTC3614
SW
PGOOD
SGND
ITH
PGND
MODE
VFB
R2
357k
1M
L1
0.68μH
CHANNEL 1
MASTER
CO12
100μF
R1
715k
VOUT1
1.8V
4A
R3
464k
C3
22pF
R4
464k
RF2
24Ω
22μF
s4
CF2
1μF
R7
100k
RC2
15k
CC3
470pF
PGOOD
CC4
10pF
SVIN
PVIN
RUN
TRACK/SS SRLIM/DDR
RT/SYNC
LTC3614
SW
PGOOD
SGND
ITH
PGND
MODE
VFB
L1, L2: VISHAY IHLP-2525CZ-01 680nH
R6
301k
CHANNEL 2
SLAVE
L2
0.68μH
VOUT2
1.2V
CO22 4A
100μF
R5
301k
C7
22pF
3614 TA03a
Coincident Start-Up
Coincident Tracking Up/Down
VOUT1
VOUT1
VOUT2
500mV/DIV
500mV/DIV
2ms/DIV
3614 TA03b
VOUT2
200ms/DIV
3614 TA03c
3614f
26
LTC3614
PACKAGE DESCRIPTION
UDD Package
24-Lead Plastic QFN (3mm × 5mm)
(Reference LTC DWG # 05-08-1833)
0.70 ±0.05
3.50 ± 0.05
2.10 ± 0.05
3.65 ± 0.05
1.50 REF
1.65 ± 0.05
PACKAGE OUTLINE
0.25 ±0.05
0.50 BSC
3.50 REF
4.10 ± 0.05
5.50 ± 0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
0.75 ± 0.05
3.00 ± 0.10
1.50 REF
23
R = 0.05 TYP
PIN 1 NOTCH
R = 0.20 OR 0.25
s 45° CHAMFER
24
0.40 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
3.65 ± 0.10
5.00 ± 0.10
3.50 REF
1.65 ± 0.10
(UDD24) QFN 0808 REV Ø
0.200 REF
0.00 – 0.05
R = 0.115
TYP
0.25 ± 0.05
0.50 BSC
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3614f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC3614
TYPICAL APPLICATION
DDR Termination With Ratiometric Tracking of VDD, 1MHz
VIN
3.3V
VDD
1.8V
VDD
C1
22μF
s4
R6
562k
R7
187k
Ratiometric Start-Up
R3
100k
R8
365k
PGOOD
SVIN
RUN
TRACK/SS
RT/SYNC
CC
2.2nF
R5
1M
L1
0.33μH
PGOOD
CC1
10pF
ITH
MODE
L1: COILCRAFT DO3316T
VTT
500mV/DIV
SRLIM/DDR
LTC3614
RC
6k
R4
1M
PVIN
SW
C4
100μF
SGND
PGND
VTT
0.9V
C5 ±3A
47μF
500μs/DIV
3614 TA04b
R1
200k
VFB
R2
200k
C3
22pF
3614 TA04a
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC3616
5.5V, 6A (IOUT) 4MHz Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN(MIN) = 2.25V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V,
IQ = 70μA, ISD < 1μA, 3mm × 5mm QFN24 Package
LTC3612
5.5V, 3A (IOUT), 4MHz, Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN(MIN) = 2.25V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V,
IQ = 70μA, ISD <1μA, 3mm × 4mm QFN-20 TSSOP20E Package
LTC3418
5.5V, 8A (IOUT), 4MHz, Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN(MIN) = 2.25V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V,
IQ = 380μA, ISD <1μA, 5mm × 7mm QFN-38 Package
LTC3415
5.5V, 7A (IOUT), 1.5MHz, Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V,
IQ = 450μA, ISD <1μA, 5mm × 7mm QFN-38 Package
LTC3416
5.5V, 4A (IOUT), 4MHz, Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN(MIN) = 2.25V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V,
IQ = 64μA, ISD <1μA, TSSOP20E Package
LTC3413
5.5V, 3A (IOUT Sink/Source), 2MHz, Monolithic Synchronous
Regulator for DDR/QDR Memory Termination
90% Efficiency, VIN(MIN) = 2.25V, VIN(MAX) = 5.5V, VOUT(MIN) = VREF /2,
IQ = 280μA, ISD <1μA, TSSOP16E Package
LTC3412A
5.5V, 2.5A (IOUT), 4MHz, Synchronous Step-Down DC/DC
Converter
95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V,
IQ = 60μA, ISD <1μA, 4mm × 4mm QFN-16 TSSOP16E Package
3614f
28 Linear Technology Corporation
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