Evaluation board available. NX2415 TWO PHASE SYNCHRONOUS PWM CONTROLLER WITH INTEGRATED FET DRIVER AND DIFFERENTIAL CURRENT SENSE PRELIMINARY DATA SHEET Pb Free Product FEATURES DESCRIPTION The NX2415 is a two-phase PWM controller with inte- n Differential inductor DCR sensing eliminates the problem with layout parasitic grated FET driver designed for low voltage high current n External programmable voltage droop application. The two phase synchronous buck converter n Low Impedance On-board Drivers offers ripple cancelation for both input and output. The n Hiccup current limit NX2415 uses differential remote sensing using either current sense resistor or inductor DCR sensing to achieve n Power Good for power sequencing accurate current matching between the two channels. n Enable Signal allows external shutdown as well as programming the BUS voltage start up threshold Differential sensing eliminates the error caused by PCB n Programmable frequency board trace resistance that is otherwise is present when n Prebias start up using a single ended voltage sensing. In addition the n Over voltage protection without negative spike at NX2415 offers high drive current capability especially for output keeping the synchronous MOSFET off during SW node transition, accurate programmable droop allowing to re- n Pb-free and RoHS compliant duce number of output capacitors, accurate enable circuit provides programmable start up point for Bus volt- n Graphic card High Current Vcore Supply age, PGOOD output, programmable switching frequency n High Current +40A on board DC to DC converter and hiccup current limiting circuitry. applications APPLICATIONS TYPICAL APPLICATION 10 31 +5V 1uF 10k 30 PVCC1 VCC BST1 EN 6.49k HDRV1 23 +5V 1uF 24 op 7 45.3k 2 ENBUS SW1 DROOP LDRV1 RT NX2415 29 10k +5V 28 11 VOUT 430 3 PGSEN 10k 3.92k 6.8nF 5.62k 10k 20k SW2 150pF 10nF 1k 180k 100k 10nF 1nF PVCC2 HDRV2 5 FB 6 VCOMP LDRV2 4 VP 8 OCP 1 VREF 14 IOUT 0.68uH 2.15 26 100uF M1 M2 22 VOUT +1.2V/50A 2 x (1000uF,7mohm ESR) 620 1uF 21 620 CS+1 9 CS-1 10 BST2 1nF 20k 2N3906 CSCOMP 220nF 2.2nF 1.8nF PGOOD PGND1 VIN1 +12V 180uF 0.22uF 25 +12V 1.65k 1uH 2 x 10uF PGND2 18 1uF +5V 17 16 15 19 10uF 0.22uF M3 0.68uH 2.15 M4 620 1uF 20 620 12 CS+2 CS-2 13 AGND 32 Figure1 - Typical application of NX2415 ORDERING INFORMATION Device NX2415CMTR Rev.4.8 05/06/08 Temperature 0 to 70oC Package MLPQ-32L Frequency 200kHz to 1MHz Pb-Free Yes 1 NX2415 ABSOLUTE MAXIMUM RATINGS Vcc to PGND & BST to SW voltage .................... -0.3V to 6.5V BST to PGND Voltage ...................................... -0.3V to 35V SW to PGND .................................................... -2V to 35V All other pins .................................................... -0.3V to 6.5V Storage Temperature Range ............................... -65oC To 150oC Operating Junction Temperature Range ............... -40oC To 125oC Lead temperature(Soldering 5s) ........................... 260oC CAUTION: Stresses above those listed in "ABSOLUTE MAXIMUM RATINGS", may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. PACKAGE INFORMATION HDRV1 NC SW1 ENBUS PGOOD EN AGND VCC 32-LEAD PLASTIC MLPQ 5 x 5 32 31 30 29 28 27 26 25 VREF 1 24 BST1 RT 2 23 PVCC1 PGSEN 3 22 LDRV1 VP 4 FB 5 21 PGnd1 NX2415 20 PGnd2 COMP 6 19 LDRV2 DROOP 7 18 PVCC2 OCP 8 θ JA ≈ 35o C /W 17 BST2 HDRV2 SW2 IOUT CS-2 CS+2 CSCOMP CS-1 CS+1 9 10 11 12 13 14 15 16 ELECTRICAL SPECIFICATIONS Unless otherwise specified, these specifications apply over Vcc = 5V, V BST-VSW =5V, EN=HIGH, and TA = 0 to 70oC. Typical values refer to TA = 25oC. Low duty cycle pulse testing is used which keeps junction and case temperatures equal to the ambient temperature. PARAMETER Supply Voltage(Vcc) VCC ,PVCC Voltage Range SYM TEST CONDITION VCC VCC Supply Current (static) ICC (Static) EN=LOW PVCC Supply Current (Dynamic) EN&ENBUS HIGH, ICC Freq=200Khz per phase (Dynamic) CLOAD=2200PF VBST Voltage Range VBST to VSW VBST Supply Current ((Dynamic)) EN&ENBUS HIGH, VBST Freq=200Khz per phase (Dynamic) CLOAD=2200PF Rev.4.8 05/06/08 MIN TYP MAX UNITS 4.5 5 5.5 V - 6.6 mA 4 mA 4.5 5 4 5.5 V mA 2 NX2415 PARAMETER Under Voltage, Vcc , Enable(EN) & ENBUS SYM VCC-Threshold VCC_UVLO VCC-Hysteresis EN Threshold EN Hysteresis ENBUS Threshold ENBUS Hysteresis Reference Voltage Ref Voltage VCC_Hyst Ref Voltage line regulation Oscillator (Rt) Frequency for each phase Ramp-Amplitude Voltage MIN VCC Rising Vcc Rising VBUS Rising VREF Fs VRAMP Ramp Peak Ramp Valley Max Duty Cycle Min Duty Cycle Transconductance Amplifiers(CSCOMP) Open Loop Gain Transconductance Voltage Mode Error Amplifier Open Loop Gain Input Offset Voltage Output Current Source Output Current Sink Output HI Voltage Output LOW Voltage SS (Internal ) Soft Start time Power Good(Pgood) Threshold TEST CONDITION 4.5V<Vcc<5.5V Rt=45kohm 200Khz/Phase TYP MAX 4 V 0.2 0.6 0.1 1.6 0.16 V V V V V 0.8 V 1 % 400 1 KHz V 2.5 1.5 95 V V % % 0 50 65 1600 dB umoh 50 Vio_v 0 5 5 Vcc-1.5 0.5 Tss Hysteresis PGood Voltage Low UNITS dB mV mA mA V V 200Khz/Phase 20 mS VSEN Falling 74 %VID IPGood=-5mA 5 0.5 % V High Side Driver(CL=4700pF) Output Impedance , Sourcing Current Output Impedance , Sinking Current Rise Time Rsource(Hdrv) I=200mA 1.1 ohm Rsink(Hdrv) I=200mA 0.8 ohm THdrv(Rise) VBST-VSW=4.5V 24 ns Fall Time THdrv(Fall) VBST-VSW=4.5V 24 ns Deadband Time Tdead(L to Ldrv going Low to Hdrv going H) High, 10%-10% 30 ns Rev.4.8 05/06/08 3 NX2415 PARAMETER Low Side Driver (CL=4700pF) Output Impedance, Sourcing Current Output Impedance, Sinking Current Rise Time Fall Time Deadband Time Current Sense Amplifier(CS+, CS-) Current Sense Amplifier Mismatch Voltage Gain Droop Voltage Current Source(Droop) Droop Voltage Current Source OCP Adjust Blank time before activating OCP Vref Reference Voltage Driving current ability OVP Threshold OVP Threshold Rev.4.8 05/06/08 SYM TEST CONDITION MIN Rsource(Ldrv) I=200mA 1.1 ohm Rsink(Ldrv) I=200mA 0.5 ohm 40 36 30 ns ns ns 0 mV TLdrv(Rise) 10% to 90% TLdrv(Fall) 90% to 10% Tdead(H to SW going Low to Ldrv going L) High, 10% to 10% K 29.7 TYP 30 MAX 30.3 UNITS V/V V(IOUT)=0.6V,feedback resistor=10kohm,Rdroop=60 kohm 100 uA 200Khz/Phase 15 uS 1.6 5 V mA 0.96 V 4 NX2415 PIN DESCRIPTIONS PIN # SYMBOL PIN DESCRIPTION 31 VCC 25, 16 HDRV1, HDRV2 High side gate driver outputs. 22, 19 LDRV1, LDRV2 Low side gate driver outputs. 30 EN IC’s supply voltage. This pin biases the internal logic circuits. A minimum 1uF ceramic capacitor is recommended to connect from this pin to ground plane. This pin is used to remotely turn off the controller. The pin has a threshold voltage of 0.6 volts. 24, 17 BST1,BST2 These pins supplies voltage to high side FET drivers. 26,15 SW1,SW2 23, 18 PVCC1, PVCC2 These pins provide the supply voltage for the lower MOSFET drivers. 28 PGOOD This pin is an open collector output. If used, it should be pulled to 5V with a resistor greater than or equal to 10k, otherwise it my be left open. Any fault or under voltage on the enable pins will cause the signal to be pulled low. 4 VP Input to the positive pin of the error amplifier. A resistor is connected from the output of the DAC to this pin. Place a small capacitor from this pin to GND to filter any noise. 5 FB This pin is the error amplifier inverting input. It is connected to the output voltage via a voltage divider. 2 RT This pin programs the internal oscillator frequency using a resistor from this pin to ground. The frequency of each phase is 1/2 of this frequency. 9,12 These pins are connected to the source pins of the upper fets. CS+1,CS+2 Positive input of the differential current sense amplifiers. It is connected directly to the RC junction of the respective phase’s output inductor. 10,13 CS-1,CS-2 Negative input of the differential current sense amplifiers. It is connected directly to the negative side of the respective phase’s output inductor. 11 CSCOMP The output of the transconductance op amp for current balance circuit. An external RC is connected from this pin to GND to stabilize the current loop. 6 VCOMP This is the output pin of the error amplifier. The compensation network connection. 7 DROOP A resistor from this pin to ground programs an internal current source that is fed into the FB pin. This current source is proportional to the output current of the regulator. The product of this current times the external resistor RFB provides a droop voltage. Rev.4.8 05/06/08 5 NX2415 PIN # SYMBOL 8 OCP A resistor divider connected from this pin to Vref programs the current limit threshold. The outputs of the internal current sense differential amplifiers are summed together to represent the output current. This voltage is then compared to this threshold. 1 VREF A 1.6V buffered reference is brought out. 29 ENBUS This pin is used to program the under voltage lockout of the bus supply. A resistor divider from the bus voltage to this pin programs the under voltage lockout. When the voltage of this pin is greater than 1.6V, the bus voltage is assumed in operation. The pin has a 10% hysterisis. 21, 20 PGND1, PGND2 This is the ground connection for the power stage of the controller. 32 AGND 14 IOUT Input of OCP amplifier. Place a 10nF to 100nF capacitor from this pin to GND to filter any noise. 3 PGSEN Output over voltage and Pgood sensing pin. A resistor divider plus a small capacitor should be connected to the this pin to set the OVP and Pgood. Rev.4.8 05/06/08 PIN DESCRIPTION Controller analog ground pin. 6 NX2415 BLOCK DIAGRAM VCC 0.8V 1.6V Bias generator UVLO PVCC1 UVLO 1.25V PVCC2 Enbus BST1 1.6/1.44 start Hiccup BST2 EN FET driver 0.64 /0.53V Digital start Vp DrvH1 DrvH2 DrvL2 OVP 0.8V DrvL1 SS_finish Dis_EA SW1 SW2 FB R S Droop current ramp1 PGND2 Set1 VCOMP K=30 Two phase OSC Rt PGND1 Q KR V1.25 R set2 CS-1 KR CS02 ramp2 Vref CS+1 CS01 R PWM control logic and driver 1.6V KR V1.25 R CS+2 CS-2 0.8*120% R OVP KR Slave channel control PGsen V1.25 Σ 0.64/0.6 Pgood CScomp Σ gm*Ri=0.6 gm IOUT SS_finished Ri Hiccup 32 cycles filter ÷ 2 Hiccup Logic 6 Cycles filter OCP Current Mirror AGND Droop FB Rev.4.8 05/06/08 7 NX2415 APPLICATION INFORMATION Symbol Used In Application Information: VIN - Input voltage VOUT - Output voltage IOUT - Output current Choose inductor from Vishay IHLP_5050FD-01 with L=0.68uH DCR=1.4mΩ. Current Ripple is recalculated as ∆IRIPPLE = DVRIPPLE - Output voltage ripple FS L OUT =0.54uH - Operation frequency for each channel = DIRIPPLE - Inductor current ripple VIN -VOUT VOUT 1 × × LOUT VIN FS ...(2) 12V-1.2V 1.2V 1 × × = 3.97A 0.68uH 12V 400kHz Output Capacitor Selection Design Example The following is typical application for NX2415. Output capacitor value is basically decided by the VIN = 12V output voltage ripple, capacitor RMS current rating and VOUT=1.2V IOUT_max=60A load transient. Based on Voltage Ripple For electrolytic, POSCAP bulk capacitor, the ESR DVRIPPLE <=12mV (equivalent series resistance) and inductor current typi- DVDROOP<=120mV @30A step cally determines the output voltage ripple. IOUT=50A FS=400kHz ESRdesire = Phase number N=2 ∆VRIPPLE 12mV = = 3.022mΩ ∆IRIPPLE 3.97A ...(3) If low ESR is required, for most applications, mul- Output Inductor Selection tiple capacitors in parallel are better than a big capaci- The selection of inductor value is based on induc- tor. For example, for 12mV output ripple, SANYO OS- tor ripple current, power rating, working frequency and CON capacitors 2R5SEPC1000MX(1000uF 7mΩ) are efficiency. Larger inductor value normally means smaller chosen. ripple current. However if the inductance is chosen too large, it brings slow response and lower efficiency. Usu- N = ally the ripple current ranges from 20% to 40% of the output current. This is a design freedom which can be decided by design engineer according to various application requirements. The inductor value can be calculated by using the following equations: L OUT = VIN -VOUT VOUT 1 × × ∆IRIPPLE VIN FS ∆IRIPPLE =k × IOUTPUT N where k is between 0.2 to 0.4. Select k=0.2, then ...(1) E S R E × ∆ IR I P P L E ∆ VR IPPLE ...(4) Number of Capacitor is calculated as 7m Ω × 3.97A 12mV N =2.3 For ceramic capacitor, the current ripple is determined by the number of capacitor instead of ESR N= COUT = ∆IRIPPLE 8 × FS × ∆VRIPPLE ...(5) Typically, the calculated capacitance is so small that the output voltage droop during the transient can not meet the spec although ripple is small. 12V-1.2V 1.2V 1 L OUT = × × 50A 12V 400kHz 0.2 × 2 Rev.4.8 05/06/08 8 NX2415 Based On Transient Requirement Typically, the output voltage droop during transient is specified as: ∆VDROOP <∆VTRAN @ step load DISTEP During the transient, the voltage droop during the transient is composed of two sections. One Section is dependent on the ESR of capacitor, the other section is a function of the inductor, output capacitance as well as input, output voltage. For example, overshoot caused by DISTEP transient load which is from high load to low load, can be estimated as the following equation,if assuming the bandwidth of system is high enough. ∆Vovershoot = ESR × ∆Istep + VOUT × τ2 2 × L × COUT ...(6) where τ is the a function of capacitor, etc. 0 if LEFF ≤ Lcrit τ = LEFF ×∆Istep − ESR × COUT V OUT ...(7) If the OS-CON capacitors (1000uF, 7mΩ ) is used, the critical inductance is given as Lcrit = The effective inductor value is 0.34uH which is big- number of capacitors is in parallel. The above equation shows that if the selected output inductor is smaller than the critical inductance, the voltage droop or overshoot is only dependent on the ESR of output capacitor. For low frequency capacitor such as electrolytic capacitor, the product of ESR and capacitance is high and L ≤ L crit is true. In that case, the transient spec is dependent on the ESR of capacitor. In most cases, the output capacitors are multiple capacitors in parallel. The number of capacitors can be calculated by the following + VOUT × τ2 2 × L × C E × ∆Vtran LEFF × ∆Istep VOUT − ESR E × CE 0.34µH × 30A − 7mΩ × 1000µF = 1.5us 1.2V ...(8) where ESRE and CE represents ESR and capaci- ∆Vtran ESR E × C E × VOUT = ∆Istep 7mΩ × 1000µF × 1.2V = 0.28µH 30A = tance of each capacitor if multiple capacitors are used Rev.4.8 05/06/08 is 120mV for 30A load step. capacitance. if LEFF ≥ Lcrit LOUT 0.68uH = = 0.34uH N 2 ESR × COUT × VOUT ESR E × C E × VOUT = = ∆Istep ∆Istep where For example, assume voltage droop during transient τ= ESR E × ∆Istep ...(10) age transient not only dependent on the ESR, but also L EFF = N= if LEFF ≥ Lcrit ger than critical inductance. In that case, the output volt- where L crit 0 if LEFF ≤ Lcrit τ = LEFF × ∆Istep − ESR E × CE V OUT ...(9) N= ESR E ×∆Istep ∆Vtran + VOUT ×τ2 2 × LEFF × CE ×∆Vtran 7mΩ× 30A + 120mV 1.2V × (1.5us)2 2 × 0.34µH×1000µF ×120mV = 1.78 = The number of capacitors has to satisfied both ripple and transient requirement. Overall, we can choose N=2. It should be considered that the proposed equation is based on ideal case, in reality, the droop or overshoot is typically more than the calculation. The equation gives a good start. For more margin, more capacitors have to be chosen after the test. Typically, for high frequency capacitor such as high quality POSCAP especially ceramic capacitor, 20% to 100% (for ceramic) more capacitors have to be chosen since the ESR of capacitors is so low that the PCB parasitic can affect the results tremendously. More capacitors have to be selected to compensate these parasitic parameters. 9 NX2415 Control Loop Compensator Design NX2415 can control and drive two channel synchro- nous bucks with 180o phase shift between each other. One of two channels is called master, the other is called slave. They are connected together by sharing the same output capacitors. Voltage loop is designed to regulate output voltage. In order to achieve the current balance in these two synchronous buck converters, current loop FZ1 = 1 2 × π × R 4 × C2 ...(11) FZ2 = 1 2 × π × (R 2 + R3 ) × C3 ...(12) FP1 = 1 2 × π × R3 × C3 ...(13) FP2 = compensation network is employed to to make sure the 1 2 × π × R4 × ...(14) C1 × C2 C1 + C2 currents in slave is following the master. where FZ1,FZ2,FP1 and FP2 are poles and zeros in Voltage Loop Compensator Design the compensator. Due to the double pole generated by LC filter of the power stage, the power system has 180o phase shift , and therefore, is unstable by itself. In order to achieve Zf C1 Vout Zin accurate output voltage and fast transient response,compensator is employed to provide highest R3 the Bode plot of the closed loop system has crossover C2 R2 possible bandwidth and enough phase margin. Ideally, C3 R4 Fb frequency between 1/10 and 1/5 of the switching fre- Ve o quency, phase margin greater than 50 and the gain cross- R1 Vref ing 0dB with -20dB/decade. Power stage output capacitors usually decide the compensator type. If electrolytic capacitors are chosen as output capacitors, type II compensator can be used to compensate the system, Figure 2 - Type III compensator because the zero caused by output capacitor ESR is pensator should be chosen. A. Type III compensator design For low ESR output capacitors, typically such as Sanyo OSCON and POSCAP, the frequency of ESR zero Gain(db) lower than crossover frequency. Otherwise type III compower stage FLC 40dB/decade loop gain FESR caused by output capacitors is higher than the crossover frequency. In this case, it is necessary to compen- 20dB/decade sate the system with type III compensator. In design example, six electrolytic capacitors are compensator used as output capacitors. The system is compensated with type III compensator. The following figures and equations show how to realize the this type III compensator with electrolytic capacitors. FZ1 FZ2 FP1 FO FP2 Figure 3 - Bode plot of Type III compensator Rev.4.8 05/06/08 10 NX2415 The transfer function of type III compensator 6. Calculate R4 by choosing FO=40kHz. is given by: Ve VOUT (1+ sR4 × C2 ) × [1+ s(R2 + R3 ) × C3 ] 1 = × sR2 × (C2 + C1) (1+ sR × C2 × C1 ) × 1+ sR × C ( 4 3 3) C2 + C1 Use the same power stage requirement as demo board. The crossover frequency has to be selected as FLC<FESR<FO, and usually FO<=1/10~1/5FS. 1.Calculate the location of LC double pole F LC and ESR zero FESR. FLC = 1 2 × π × LEFF × COUT 1 = 2 × π × 0.34uH × 2000uF = 6.1kHz R4 = VOSC 2 × π × FO × LEFF R2 × R3 × × Vin ESR R2 + R3 1V 2 × π × 40kHz × 0.34uH 10kΩ × 3.92kΩ × × 12V 3.5mΩ 10kΩ + 3.92kΩ =5.73kΩ = Choose R4=5.62kΩ. 7. Calculate C2 with zero Fz1 at 75% of the LC double pole by equation (11). 1 2 × π × FZ1 × R 4 C2 = 1 2 × π × 0.75 × 6.1kHz × 5.62k Ω = 6.2nF = Choose C2=6.8nF. 8. Calculate C 1 by equation (14) with pole F p2 at half the switching frequency. FESR 1 = 2 × π × ESR × COUT 1 2 × π × 3.5mΩ × 2000uF = 22.7kHz = 2.Set R2 equal to10kΩ. R × VREF 10k Ω × 0.8V R1= 2 = = 20k Ω VOUT -VREF 1.2V-0.8V Choose R1= 20kΩ. 3. Calculate C3 by setting FZ2 = FLC and Fp1 =FESR. 1 2 × π × R 4 × FP2 C1 = 1 2 × π × 5.62kΩ × 200kHz = 141pF = Choose C1=150pF. B. Type II compensator design If the electrolytic capacitors are chosen as power stage output capacitors, usually the Type II compensa- 1 1 1 C3 = ) ×( 2 × π × R2 Fz2 Fp1 1 1 1 = ) ×( 2 × π × 10k Ω 6.1kHz 22.7kHz =1.9nF tor can be used to compensate the system. Type II compensator can be realized by simple RC circuit without feedback as shown in figure 4. R3 and C1 introduce a zero to cancel the double pole effect. C2 introduces a pole to suppress the switching noise. The Choose C3=1.8nF. following equations show the compensator pole zero lo- 5. Calculate R 3 by equation (13). cation and constant gain. R3 = 1 2 × π × FP1 × C3 1 2 × π × 22.7kHz × 1.8nF = 3.89k Ω = Choose R3=3.92kΩ. Rev.4.8 05/06/08 Gain= Fz = R3 R2 1 2 × π × R3 × C1 Fp ≈ 1 2 × π × R 3 × C2 ... (15) ... (16) ... (17) 11 NX2415 FLC = C2 Vout 2 ×π× LEFF × COUT 1 = C1 R3 1 2 ×π× 0.75uH×10800uF = 1.768kHz R2 Fb Ve FESR = R1 Vref 1 2 × π × ESR × COUT 1 2 × π × 13m Ω × 1800uF = 6.801kHz = Figure 4 - Type II compensator 2.Set R2 equal to10kΩ and calculate R1. R1= power stage R 2 × VREF 10k Ω × 0.8V = = 20k Ω VOUT -VREF 1.2V-0.8V Gain(db) 3. Set crossover frequency FO=15kHz. 40dB/decade 4.Calculate R3 value by the following equation. V O S C 2 × π × FO × L E F F × × R2 V in ESR R3= loop gain 1V 2 × π × 15kHz × 0.75uH × × 10kΩ 12V 2.16m Ω =27.3kΩ = 20dB/decade Choose R 3 =27.4kΩ. 5. Calculate C1 by setting compensator zero FZ compensator Gain at 75% of the LC double pole. 1 2 × π × R3 × Fz C1= FZ FLC FESR FO FP Figure 5 - Bode plot of Type II compensator 1 2 × π × 27.4kΩ × 0.75 × 1.768kHz =4.4nF = Choose C1=4.7nF. 6. Calculate C 2 by setting compensator pole Fp For this type of compensator, FO has to satisfy at half the swithing frequency. FLC<FESR<< FO and FO <=1/10~1/5Fs. Here a type II compensator is designed for the case C2= which has six electrolytic capacitors(1800uF, 13mΩ) and two 1.5uH inductors. 1.Calculate the location of LC double pole F LC 1 π × R 3 × Fs 1 π × 2 7 .4k Ω × 1 0 0 k H z =116pF = and ESR zero FESR. Choose C2=100pF. Rev.4.8 05/06/08 12 NX2415 Current Loop Compensator Design Power stage Compensation D(s) Master channel 1 d Vosc Current Sensing Amplifier Gain Vin s*L+Req iL s*L+DCR Rs*Cs*s+1 Inductor Current sense Figure 6 - Current loop control diagram VIN master channel DCR L Rs Vbias Cs Rs VIN Slave channel 1 DCR L PWM control logic and driver Ramp for slave channel VOUT Rs Cs Vbias Rs Icomp1 Rcc C1 Slave channel control C2 Slave channel control Slave channel Figure 7 - Function diagram of current loop Rev.4.8 05/06/08 13 NX2415 Inductor Current Sensing racy during the transient if droop function is required. The illustration is shown in the following figure. VIN iL L Control & Driver VOUT Rs Current Sensing Amplifier DCR Rs Cs VS_IL VS_IL----Voltage accross the sensing capacitor Cs Overshoot caused by inductor nonlinearity iL--- inductor current Figure 8 - Inductor current sensing using RC network. The inductor current can be sensed through a RC Output voltage with droop function network as shown above. The advantage of the RC network is the lossless comparing with a resistor in series Droop misbehavoir caused by overshoot of VS_IL with output inductor. The selection of the resistor sensing network is chosen by the following equation: R S × CS = L DCR ...(18) If the above equation is satisfied, the voltage across Figure 9 - Droop accuracy affected by the nonlinearity of inductor. In this case, the sensing resistor has to be chosen the sensing capacitor Cs will be equal to the inductor current times DCR of inductor for all frequency domain. VS _ IL = DCR × iL If the sensing capacitor is chosen CS = 1µF CS must be X7R or COG ceramic capacitor. The sensing resistor is calculated as RS = L DCR × CS For example, for 0.68uH inductor with 1.4mΩ DCR, we have RS = 0.68µH = 486Ω 1.4mΩ × 1µF In most of cases, the selection of sensing resistor based on the above equation will be sufficient. However, for some inductor such as toroid coiled inductor with micrometal, even the product of sensing resistor and capacitor is perfectly match with L/DCR, the voltage across the capacitor still has overshoot due to the nonlinearity of inductor. This will affect the droop accu- Rev.4.8 05/06/08 RS ≥ L DCR × CS to compensate the overshoot. This selection only affects the small signal mode of current loop. For DC accuracy, there is no effect since the DC voltage across the sensing capacitor will equal to the DCR times inductor current at DC load no matter what Rs is. In this example, Rs=620Ω. RS value is preferred to be less than 400Ω in NX2415's application, therefore we need to reiterate the calculation, choose CS 2.2uF instead. RS value is finally chosen as 301Ω . Powe dissipation of Rs resistor is calculated as followed: PD (RS ) = (VIN − VOUT )2 V 2 × D + OUT × (1 − D) RS RS (12 V − 1.2V)2 (1.2 V)2 × 0. 1 + × (1 − 0.1) 301Ω 301Ω = 0.04 W = The power rating of Rs should be over 0.04W. 14 NX2415 Current Loop Compensation FP1 = Req 2× π ×L = 7.4mΩ = 1.7kHz 2 × π × 0.68µH The current compensation transfer function is Slave channel power stage -20 dB given as D(s) = Current loop compensation gm × s × ( C1 + C2 ) 1 + s × Rcc × C1 R × C1 × C2 1 + s × cc C1 + C2 It has one zero and one pole. The ideal is to Loop gain for slave channel choose resistor Rcc to achieve desired loop gain such -20dB 0 DB as 50kHz. Rcc can be calculated as -40dB Fp1 Fzc Fo Rcc = Fpc 2 × π × Fo × L × Vosc gm × VIN × K C × DCR ...(19) where Figure 10 - Bode plot of current loop The diagram and bode plot for current loop of KC ≈ 60 ⋅ kΩ = 22.9 2kΩ+ RS through inductor sensing is amplified by current sensing 60kΩ and 2kΩ is the internal resistance for the current sensing amplifier. For fast response, we can set the current loop differential amplifier. The amplified slave current signal cross-over frequency one and half times of voltage loop is compared with the amplified inductor current from cross-over frequency. Since the voltage loop cross-over master channel (channel 1 for NX2415) through a frequency is typically selected as 1/10 of switching fre- transconductance amplifier, the difference between chan- quency, we choose FO=50kHz. NX2415 is shown in above figures. The current signal nel current will change the output of transconductance 2 × π × 50kHz × 0.68µH × 1V = 442Ω 1.6mA / V × 12 V × 22.9 ×1.4mΩ amplifier, which will compare with a internal ramp signal Rcc = and changes the duty cycle of slave channel buck con- Select verter. If the inductor are perfectly matched and the PWM Rcc = 430Ω . controller has no offset, the DC current in slave channel will equal to the DC current of master channel (channel 1) due to the gain of current loop. From the bode plot, the power stage has one pole The selection of capacitor C1 is such that the zero of compensation will cancel the pole of power stage, therefore, C1 = located at FP1 = Req L 0.68µH = = 214nF Req × Rcc 7.4mΩ × 430Ω Typically, the capacitor C1 is so big that the cur- 2×π×L where Req is the equivalent resistor and it is given by rent loop may start slowly during the start up. There- VOUT V + Rdson _ syn × 1 − OUT VIN VIN selected capacitor can not reduce too much to cause R eq ≈ DCR + R dson _ con × R dson _ con is the Rdson of control FET and R dson _ syn is the Rdson of synchronous FET. For this example, Req = 7.4mΩ fore, smaller capacitor can be selected. However, the phase droop. Select C1=220nF. The capacitor C2 is an option and it is used to filter out the switching noise. C2 can be calculated as The pole is located as Rev.4.8 05/06/08 15 NX2415 VREF 1 1 C2 = = = 1.85nF π × Rcc × FS π × 430Ω × 400kHz 100k Select C2=2.2nF. OCP ROCP Frequency Selection The frequency can be set by external Rt resistor. The relationship between frequency per phase and RT Figure 12 - Over current protection pin is shown as follows. RT ≈ Output Voltage Droop Operation 18600000 FS ...(20) The effective output impedance of the controller must be adjusted to maximize the output voltage fluctuation range. A program resistor attached to the Droop pin RDROOP will program this value. The function works by an frequency(kHz) FREQUENCY(kHz) vs RT(kohm ) 800 internal current source connected to the FB pin. This 700 current flows output of the FB pin and through the Rin 600 resistance from the FB pin to the output. 500 This current source is a function of the sensed 400 output current. As the output current increases, the droop current will increase and causes the output voltage 300 todroop proportionately. The droop current is programmed 200 by a resistor attached to the Droop pin. The value of the 100 resistor is chosen as follows. 0 0 50 100 150 200 R t(kohm ) Figure 11 - Frequency vs Rt chart Over Current/Short Circuit Protection VOUT RIN The converter will go into hiccup mode if the output current reaches a programmed limit V OCP determined by the voltage at pin OCP. VOCP = 0.6 ROCP = VP ...(21) IOCP pin. RS is the current sensing matching resistor ...(22) Where RLL is desired load impedance. For example, if we want Vout droops 60mV @ 20A, RLL = level,100kΩ is the resistor connecting VREF pin and 60mV = 3mΩ 20A IDROOP = V(IOUT) RDROOP 0.6 × = Rev.4.8 05/06/08 COMP Error Amplifer ∆VOUT = IDROOP × RIN = ∆ILOAD × RLL Where Iocp is the desired over current protection when using DCR sensing method. IDROOP Figure 13 - Output voltage droop funciton 60kΩ DCR IOCP 2kΩ + R S 2 VOCP × 100kΩ VREF − VOCP FB 60kΩ DCR × × ILOAD 2kΩ + RS 2 RDROOP ...(23) 16 NX2415 Combine equation 22 and 23, RDROOP = 0.6 60kΩ DCRILOADRIN 2 2kΩ + RS ∆VOUT be calculated. From this figure, it is obvious that a multiphase converter can have a much smaller input RMS ...(24) Where DCR is the sense resistor or the DCR of the output inductor. RS is the current sensing matching resistor when using DCR sensing method. ILOAD is the load current. RIN is the input DC resistor of the master phase compensator which connect FB pin and PGSEN pin. For example, to have the DVOUT=60mV when the load current is 20A, DCR is 1.4mΩ, RIN is 10kΩ, RS is 620Ω. RDROOP = 32kΩ 0.6 60kΩ 1.4mΩ × 20A × 10kΩ = × × 2 2kΩ + 0.62kΩ 60mV current, which results in a lower amount of input capacitors that are required. For example, Vin=12V, Vout=1.2V. The duty cycle is D=Vout/Vin=1.2/12=10%. From the figure, for two phase, the RMS current is 0.2*Iout=0.2*50A=10A. A combination of ceramic and electrolytic(SANYO WG or WF series) or OSCON type capacitors can achieve both ripple current capability together with having enough capacitance such that input voltage will not sag too much. In this application, one OSCON SVPC180M(180uF, 16V, 2.8A) and three 10uF(4A rms current, X5R) ceramic capacitors are selected. Choose RDROOP= 32kΩ. A 1uH input inductor is recommended to slow down the input current transient. Suppose power stage effi- Over Voltage Protection ciency is 0.8, then input current can estimated by Over voltage protection is achieved by sensing the output voltage through resistor divider. The sensed volt- IINPUT = age on PGSEN pin is compared with 120%*0.8V to generate the OVP signal. A small value capacitor is re- IO U T × VOUT 60 A × 1 . 2 V = = 7 .5 A 0 . 8 × 12 V η × VIN In this application, Coilcraft DO3316P_102HC with RMS rating 10A is chosen. quired to connect to PGSEN pin also. VOUT 1.2V 0.5 0.8V*120% 10k PGSEN 20k normlized OVP 1nF Singlephase 0.4 I RMS (IN ) 0.3 Iout Two phase 0.2 Figure 14 - Over voltage protection Input Filter Selection The selection criteria of input capacitor are voltage 0.1 0 Three phase 0 0.1 0.2 0.3 0.4 0.5 D rating and the RMS current rating. For conservative consideration, the capacitor voltage rating should be 1.5 times higher than the maximum input voltage. The RMS Figure 15 - Normalized input RMS current vs. duty cycle. current rating of the input capacitor for multi-phase converter can be estimated from the above Figure 15. First, determine the duty cycle of the converter (VO/ VIN). The ratio of input RMS current over output current can be obtained. Then the total input RMS current can Rev.4.8 05/06/08 17 NX2415 Power MOSFETs Selection Soft Start and Enable Signal Operation The NX2415 requires two N-Channel power MOSFETs for each channels. The selection of MOSFETs is based on maximum drain source voltage, gate source voltage, maximum current rating, MOSFET on resistance and power dissipation. The main consideration is the power loss contribution of MOSFETs to the overall converter efficiency. In this design example, eight NTD60N02 are used. They have the following parameters: VDS=25V, ID =62A,RDSON =12mΩ,QGATE =9nC. There are three factors causing the MOSFET power loss:conduction loss, switching loss and gate driver loss. The NX2415 will start operation only after Vcc and PVcc have reached their threshold voltages and EN and ENBUS have been enabled. The ENBUS pin can be programmed to turn on the converter at any input voltage. The ENBUS pin has a threshold voltage of 1.6V. Once the converter starts, there is a soft start sequence of 4082 steps between 0 and Vp. The ramp rate is determined by the switching frequency. dVO VO = ...(27) dt 4082 × FS The softstart time is calculated as followed: Tstartup = Gate driver loss is the loss generated by discharging the gate capacitor and is dissipated in driver circuits. 4082 FS ...(28) It is proportional to frequency and is defined as: Pgate = (QHGATE × VHGS + QLGATE × VLGS ) × FS Layout Considerations ...(24) where QHGATE is the high side MOSFETs gate charge,QLGATE is the low side MOSFETs gate charge,VHGS is the high side gate source voltage, and VLGS is the low side gate source voltage. This power dissipation should not exceed maximum power dissipation of the driver device. The layout is very important when designing high frequency switching converters. Layout will affect noise pickup and can cause a good design to perform with less than expected results. There are two sets of components considered in the layout which are power components and small signal components. Power components usually consist of Conduction loss is simply defined as: input capacitors, high-side MOSFET, low-side MOSFET, PHCON =IOUT × D × RDS(ON) × K 2 PLCON =IOUT 2 × (1 − D) × RDS(ON) × K PTOTAL =PHCON + PLCON inductor and output capacitors. A noisy environment is ...(25) generated by the power components due to the switching power. Small signal components are connected to sensitive pins or nodes. A multilayer layout which in- Where the RDS(ON) will increases as MOSFET junction temperature increases, K is RDS(ON) temperature dependency and should be selected for the worst case. Conduction loss should not exceed package rating or overall system thermal budget. cludes power plane, ground plane and signal plane is recommended . Layout guidelines: 1. First put all the power components in the top layer connected by wide, copper filled areas. The input Switching loss is mainly caused by crossover con- capacitor, inductor, output capacitor and the MOSFETs duction at the switching transition. The total switching should be close to each other as possible. This helps to loss can be approximated. 1 PSW = × VIN × IOUT × TSW × FS 2 reduce the EMI radiated by the power loop due to the ...(26) high switching currents through them. 2. Low ESR capacitor which can handle input RMS TSW is the sum of TR and TF which can be found in ripple current and a high frequency decoupling ceramic mosfet datasheet, IOUT is output current, and FS is switch- cap which usually is 1uF need to be practically touching ing frequency. Swithing loss PSW is frequency depen- the drain pin of the upper MOSFET, a plane connection dent. is a must. 3. The output capacitors should be placed as close Rev.4.8 05/06/08 18 NX2415 as to the load as possible and plane connection is required. 4. Drain of the low-side MOSFET and source of the high-side MOSFET need to be connected thru a plane ans as close as possible. A snubber nedds to be placed as close to this junction as possible. 5. Source of the lower MOSFET needs to be connected to the GND plane with multiple vias. One is not enough. This is very important. The same applies to the output capacitors and input capacitors. 6. Hdrv and Ldrv pins should be as close to MOSFET gate as possible. The gate traces should be wide and short. A place for gate drv resistors is needed to fine tune noise if needed. 7. Vcc capacitor, BST capacitor or any other bypassing capacitor needs to be placed first around the IC and as close as possible. The capacitor on comp to GND or comp back to FB needs to be place as close to the pin as well as resistor divider. 8. The output sense line which is sensing output back to the resistor divider should not go through high frequency signals. 9. All GNDs need to go directly thru via to GND plane. 10. The feedback part of the system should be kept away from the inductor and other noise sources, and be placed close to the IC. 11. In multilayer PCB, separate power ground and analog ground. These two grounds must be connected together on the PC board layout at a single point. The goal is to localize the high current path to a separate loop that does not interfere with the more sensitive analog control function. 12. Inductor current sense line should be connected directly to the inductor solder pad. Rev.4.8 05/06/08 19 NX2415 MLPQ 32 PIN 5 x 5 PACKAGE OUTLINE DIMENSIONS NOTE: ALL DIMENSIONS ARE DISPLAYED IN MILLIMETERS. Rev.4.8 05/06/08 20 NX2415 MLPQ 32 PIN 5 x 5 TAPE AND REEL INFORMATION NOTE: 1. R7 = 7 INCH LOCK REEL, R13 = 13 INCH LOCK REEL. 2. ALL DIMENSIONS ARE DISPLAYED IN MILLIMETERS. Rev.4.8 05/06/08 21