TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 INTEGRATED DUAL USB SWITCHES, BOOST CONVERTER AND LDO Check for Samples: TPS2505 FEATURES 1 • • • • • • • • • • Integrated Synchronous Boost Converter, LDO and Dual USB Current-Limited Switches 1.8-V to 5.25-V Input Voltage (2.2-V Minimum Start-Up Voltage) Adjustable Independent USB Current Limit – 100 mA to 1100 mA Auxiliary 5.1-V Output 3.3-V Linear Regulator Output Inrush Current < 100 mA Minimal External Components Required Deglitched Independent Fault Reporting Small 5-mm x 5-mm QFN-20 Package Industrial Temperature Range Typical Application 2.2µH Power Supply 1.8V – 5.25V SW IN 10µF EN RESET 3.3V Power LDOOUT 1µF EN_LDO LDOIN HUB CONTROLLER RFAULT 1 5V USB Power USB1 5V Output AUX 120µF 22µF TPS2505 RFAULT 2 5V USB Power USB2 FAULT 1 120µF FAULT 2 ILIM1 RLIM1 ENUSB1 ILIM2 ENUSB2 GND PAD PGND RLIM2 APPLICATIONS • • • • Portable Applications Using Single Li+ Cell Bus Powered USB Hosts USB Hosts Without Native 5-V Supplies Computer Peripherals DESCRIPTION The TPS2505 provides an integrated solution to meet USB 5-V power requirements from a 1.8-V to 5.25-V input supply. The features include a 5.1-V, 1100-mA boost converter, a 200-mA, 3.3-V LDO linear regulator and dual USB 2.0 compliant power outputs with independent output switch enable, current limit, and over-current fault reporting. The 1.8-V to 5.25-V input can be supplied by sources including DC/DC regulated supplies (e.g. 3.3 V), or batteries such as single cell Li+, two-cell or three-cell NiCd, NiMH or alkaline. The output trip current for the dual USB switches can be programmed via external resistors from as low as 100 mA to as high as 1100 mA. An auxiliary 5.1-V output is provided, where the total current supplied by the USB outputs and the auxiliary by cannot exceed 1100 mA. 1 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2009–2010, Texas Instruments Incorporated TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. Table 1. ORDERING INFORMATION (1) (1) (2) (3) TA (2) PACKAGE (3) ORDERABLE PART NUMBER TOP-SIDE MARKING –40ºC to 85ºC RGW (QFN) TPS2505B1RGWR TPS2505 For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. Maximum ambient temperature is a function of device junction temperature and system-level considerations, such as power dissipation and board layout. See Dissipation Ratings and Recommended Operating Conditions for specific information related to these devices. Package drawings, thermal data, and symbolization are available at www.ti.com/packaging. ABSOLUTE MAXIMUM RATINGS (1) (2) Over operating free-air temperature range (unless otherwise noted). VALUE Input voltage range on SW, AUX, IN, USB, ENUSB, EN, FAULT, ILIM V 25 mA 1 mA –40 to 125 °C FAULT sink current ILIM source current TJ Operating junction temperature ESD (1) (2) UNIT -0.3 to 7 HBM (Human Body Model) 2000 CMD (Charged Device Model) 500 V Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. Voltages are referenced to GND and PGND tied together. DISSIPATION RATINGS PACKAGE RGW THERMAL RESISTANCE RqJA RqJB 42°C/W 14°C/W TA ≤ 25°C 2.4 W RECOMMENDED OPERATING CONDITIONS over operating free-air temperature range (unless otherwise noted) MIN VIN Supply voltage at IN 1.8 VSTART Supply voltage at IN for start-up 2.2 Enable voltage at EN, ENUSB1, ENUSB2, ENLDO NOM MAX UNIT 5.25 V V 0 5.25 V TA Operating free air temperature range –40 85 °C TJ Operating junction temperature range –40 125 °C RECOMMENDED EXTERNAL COMPONENTS over operating free-air temperature range (unless otherwise noted) MIN Inductor RILIM 2 NOM 2.2 MAX 4.7 UNIT µH Boost input capacitance (ceramic capacitor, X5R, 10V, 0805) 10 µF Boost output capacitance (ceramic capacitor, X5R, 10V, 1210) 22 µF LDO input capacitance (ceramic capacitor, X5R) 4.7 µF 1 µF LDO output capacitance (ceramic capacitor, X5R) 0.7 Current-limit set resistor from ILIM to GND 20 Submit Documentation Feedback 220 kΩ Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 ELECTRICAL CHARACTERISTICS (SHARED BOOST, LDO AND USB) over recommended operating conditions (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT BIAS Bias Current Shutdown current VIN VIN = 3.3 V, VAUX = 5.2 V, VEN = VIN, VENUSB1 = VAUX, IAUX = IUSB = 0 A VAUX VIN = 3.3 V, VEN = VENUSB = 0 V, AUX and USB OPEN, -40°C ≤ TJ ≤ 85°C VIN 15 25 500 600 µA 5 µA UVLO VIN rising Undervoltage lockout threshold on IN for boost converter 2.08 VIN falling, VAUX = 5.2 V Threshold VIN falling, VAUX = OPEN Threshold Hysteresis VAUX falling 1.85 0.4 1.93 Hysteresis V 2.05 0.15 VAUX rising Undervoltage lockout threshold on AUX for USB switches 2.20 1.69 Threshold 4.18 4.45 4.1 4.37 Hysteresis V 0.09 THERMAL SHUTDOWN Full thermal shutdown thereshold 150 Hysteresis °C 10 USB only thermal shutdown °C 130 Hysteresis °C 10 °C ELECTRICAL CHARACTERISTICS (BOOST ONLY) over recommended operating conditions (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT 4.75 5.1 5.25 V APPLICATION SPECIFICATIONS VAUX AUX regulation voltage VRIPPLE AUX ripple voltage Load regulation (1) Line regulation VREF Includes ripple and line/load regulation PFM, IO = 100 mA 250 PWM, IO = 1100 mA 75 IO = 0 mA to 1100 mA (PWM operation only) 50 IO = 1100 mA (PWM operation) 50 300 (2) IO = 1100 mA, VIN = 3.6 V to 5.25 V Internal reference voltage 1.35 mV mV mV V OSCILLATOR freq VLFM Switching frequency, normal mode VIN < VLFM 850 1000 1150 kHz Switching frequency, low-frequency mode VIN > VLFM 225 250 275 kHz 4.25 4.35 4.45 V Low-frequency mode input voltage threshold Hysteresis VNFM (1) (2) No-frequency mode input voltage threshold (Boost SYNC MOSFET always on) 200 VIN rising 4.9 5.05 mV 5.17 V Load regulation in No Frequency or Pass-Through is given by IR drop across SWP switch resistance.. Includes voltage drop when transitioning to No Frequency or Pass-Through Mode, where VAUX is no longer a closed loop regulated voltage and drops to VNFM – ILOADRSWP. For No Frequency or Pass-Through, ΔVAUX/ ΔVIN = 1. Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 3 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com ELECTRICAL CHARACTERISTICS (BOOST ONLY) (continued) over recommended operating conditions (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT Hysteresis 75 mV Maximum duty cycle 85 % Minimum controllable on-time 85 ns 420 mA PULSE FREQUENCY MODE (PFM) IINDLOW Demanded peak current to enter PFM mode Peak inductor current, falling AUXLOW AUX too low comparator threshold Resume switching due to AUX, falling 0.98 * VAUX V POWERSTAGE Switch on resistance (SWN) 120 Peak switch current limit (SWN MOSFET) ISW 3 Switch on resistance (SWP) Vsg = VMAX Switch on resistance (SWP + USB) VIN > VNFM 4.5 125 6 mΩ A 125 mΩ 185 mΩ 0.1 A START UP (3) ISTART Constant current VEXIT Constant current exit threshold (VIN –VAUX) tstartup Boost startup time 700 VIN = 5.1 V, COUT = 150 µF 25 mV 40 ms BOOST ENABLE (EN) Enable threshold, boost converter IEN (3) Input current VEN = 0 V or 5.5 V 0.7 1 V -0.5 0.5 µA VAUX pin must be unloaded during startup. ELECTRICAL CHARACTERISTICS (USB1/2 ONLY) over recommended operating conditions (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT USB1, USB2 rDS(on) USB switch resistance 80 mΩ 2 3 ms 2.5 3.5 ms 5.25 mV tr Rise time, output VAUX = 5.1 V, CL = 100 µF, RL = 10 Ω tf Fall time, output VAUX = 5.1 V, CL = 100 µF, RL = 10 Ω VUSB1/2 USB1/2 output voltage Including ripple 4.75 0.7 1 V Input current VENUSB = 0 V or 5.5 V -0.5 0.5 µA Turnon time CL = 100 µF, RL = 10 Ω 5 ms Turnoff time CL = 100 µF, RL = 10 Ω 10 ms 150 mV 1 µA 10 ms USB ENABLE (ENUSB1, ENUSB2) Enables threshold, USB switch IENUSB /FAULT1, /FAULT2 tDEG 4 Output low voltage I/FAULT = 1 mA Off-state current V/FAULT = 5.5 V /FAULT deglitch /FAULT assertion or deassertion due to over-current condition Submit Documentation Feedback 6 8 Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 ELECTRICAL CHARACTERISTICS (USB1/2 ONLY) (continued) over recommended operating conditions (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT ILIM1, ILIM2 IOS Short-circuit output current RILIM = 100 kΩ 190 RILIM = 40 kΩ 550 380 875 RILIM = 20 kΩ 1140 1700 mA ELECTRICAL CHARACTERISTICS (LDO AND RESET ONLY) over recommended operating conditions (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT 3.8 5.1 5.25 V 3.2 3.3 3.4 V LDO SPECIFICATIONS Input voltage Output voltage Including line/load regulation DC accuracy Line regualtion ILOAD: 200 mA Line transient 500 mV step at 50 mV/ms Load regulation Load transient ILOAD: 0 mA - 200 mA in 1 ms Dropout voltage Output overshoot tr Rise time, output VAUX = 5.1 V, CL = 1 µF tf Fall time, output VAUX = 5.1 V, CL = 1 µF IOS Short-circuit output current PSRR 350 20 Hz < f < 20 kHz, IL = 100 mA ±3 % 5 mV 15 mV 20 mV 120 mV 300 mV 3 % 200 µs 1 ms 800 mA 40 dB RESET SPECIFICATIONS Threshold voltage Deglitch timing VLDOOUT rising 3.09 3.1 3.11 VLDOOUT falling 2.91 2.975 3.03 Low to high transition 150 175 200 ms 8 10 12 kΩ Internal pull-up resistance Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 V 5 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com TERMINAL FUNCTIONS TERMINAL TYPE (1) DESCRIPTION NAME NO. PGND 1 P Source connection for the internal low-side boost converter power switch. Connect to GND with a low impedance connection to the input and output capacitors. IN 2 I Input supply voltage for boost converter. EN 3 I Enable input for boost converter. Tie to IN to enable. GND 4 P Control / logic ground. Must be tied to PGND close to the IC externally. ILIM1 5 I Program the nominal USB1 switch current-limit threshold with a resistor to GND. ILIM2 6 I Program the nominal USB2 switch current-limit threshold with a resistor to GND. /FAULT1 7 O Active low USB1 fault indicator (open drain). ENUSB1 8 I Enable input for the USB1 switch. Tie to IN or AUX to enable. RESET 9 O Active low LDO output good indicator (open drain). LDOOUT 10 O Fixed 3.3-V LDO output. Connect a low-ESR ceramic capacitor from LDOOUT to GND. LDOIN 11 I Input supply voltage for LDO. Connect to AUX. ENLDO 12 I Enable input for the LDO. Tie to AUX to enable. /FAULT2 13 O Active low USB2 fault indicator (open drain). ENUSB2 14 I Enable input for the USB2 switch. Tie to IN or AUX to enable. USB2 15 O Output of the USB2 power switch. Connect to the USB2 port. USB1 16 O Output of the USB1 power switch. Connect to the USB1 port. USB1 17 O Output of the USB1 power switch. Connect to the USB1 port. AUX 18 O Fixed 5.1-V boost converter output. Connect a low-ESR ceramic capacitor from AUX to PGND. SW 19 P Boost and rectifying switch input. This node is switched between PGND and AUX. Connect the boost inductor from IN to SW. SW 20 P Boost and rectifying switch input. This node is switched between PGND and AUX. Connect the boost inductor from IN to SW. THERMAL PAD — — Internally connected to PGND. Must be soldered to board ground for thermal dissipation. (1) I = Input; O = Output; P = Power RGW PACKAGE (TOP VIEW) SW USB1 USB1 AUX SW PGND IN 20 16 1 15 PAD EN /FAULT2 ENLDO GND ILIM1 5 11 6 LDOIN 10 LDOOUT RESET ENUSB1 /FAULT1 ILIM2 6 USB2 ENUSB2 Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 PGND PGND is the internal ground connection for the source of the low-side N-channel MOSFET in the boost converter. Connect PGND to an external plane near the ground connection of the input and output capacitors to minimize parasitic effects due to high switching currents of the boost converter. Connect PGND to GND and the thermal pad externally at a single location to provide a star-point ground. See LAYOUT RECOMMENDATIONS for further details. IN IN is the input voltage supply for the boost converter. Connect a 10-µF ceramic capacitor (minimum) from IN to PGND. See COMPONENT RECOMMENDATIONS for further details on selecting the input capacitor. EN EN is a logic-level input that enables the boost converter. Pull EN above 1 V to enable the device and below 0.7 V to disable the device. EN also disables the USB switches and LDO. GND Signal and logic circuits of the TPS2505 are referenced to GND. Connect GND to a quiet ground plane near the device. An optional 0.1-µF capacitor can be connected from VIN to GND close the device to provide local decoupling. Connect GND and PGND to the thermal pad externally at a single location to provide a star-point ground. See LAYOUT RECOMMENDATIONS for further details. ILIM1/2 Connect a resistor from ILIM1/2 to GND to program the current-limit threshold of the USB switches. Place this resistor as close to the device as possible to prevent noise from coupling into the internal circuitry. Do not drive ILIM1/2 with an external source. The current-limit threshold is proportional to the current through the RILIM resistor. See Programming the Current-Limit Threshold Resistor RILIM for details on selecting the current-limit resistor. RESET The RESET output indicates when the LDO output reaches 3.1 V. It has a 175-ms delay for deglitch in the low to high transition. The output has in internal 10-kΩ pull-up resistor to the LDO output. LDOOUT LDOOUT is the LDO output. Internal feedback regulates LDOOUT to 3.3 V. Connect a 1-µF ceramic capacitor from LDOOUT to PGND for compensation. See COMPONENT RECOMMENDATIONS for further details. LDOIN LDOIN is the input voltage supply for the LDO. Connect a 4.7-µF ceramic capacitor from LDOIN to PGND when not powered by AUX. See COMPONENT RECOMMENDATIONS for further details on selecting the input capacitor. ENLDO ENLDO is a logic-level input that enables the 3.3-V LDO. Pull EN above 1 V to enable the device and below 0.7 V to disable the device. The boost converter must be enabled in order for the LDO to be enabled. The boost converter is independent of ENLDO and continues to operate even when ENLDO disables the LDO. Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 7 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com FAULT1/2 FAULT1/2 are open-drain outputs that indicate when the USB switches are in an overcurrent or over-temperature condition. FAULT1/2 have a fixed internal deglitch of tDEG to prevent false triggering from noise or transient conditions. FAULT1/2 assert low if the USB switches remain in an overcurrent condition for longer than tDEG. FAULT1/2 de-assert when the overcurrent condition is removed after waiting for the same tDEG period. Over-temperature conditions bypass the internal delay period and assert/de-assert the FAULT1/2 output immediately upon entering or leaving an over-temperature condition. FAULT1/2 are asserted low when VAUX falls below VTRIP (4.6 V, typical). ENUSB1/2 ENUSB1/2 are logic-level inputs that enable the USB switches. Pull ENUSB1/2 above 1 V to enable the USB switches and below 0.7 V to disable the USB switches. ENUSB1/2 only enables the USB switches. The boost converter is independent of ENUSB1/2 and continues to operate even when ENUSB disables the USB switch. USB1/2 USB1/2 are the outputs of the USB switches and should be connected to the USB connectors to provide USB power. Although the device does not require it for operation, a bulk capacitor may be connected from USB to PGND to meet USB standard requirements. See the latest USB 2.0 specification for further details. AUX AUX is the boost converter output and provides power to the USB switches and to any additional load connected to AUX. Internal feedback regulates AUX to 5.1 V. Connect a 22-µF ceramic capacitor from AUX to PGND to filter the boost converter output. See COMPONENT RECOMMENDATIONS for further details. Additional external load can be connected to AUX as long as the total current drawn by the USB switches and external load does not overload the boost converter. See Determining the Maximum Allowable AUX and USB1/2 Current for details. SW SW is the internal boost converter connection of the low-side N-channel MOSFET drain and the high-side P-channel drain. Connect the boost inductor from IN to SW close to the device to minimize parasitic effects on the device operation. THERMAL PAD The thermal pad connection is used to heat-sink the device to the printed-circuit board (PCB). The thermal pad may not be connected externally to a potential other than ground because it is connected to GND internally. The thermal pad must be soldered to the PCB to remove sufficient thermal energy in order to stay within the recommended operating range of the device. 8 Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 SW FUNCTIONAL BLOCK DIAGRAM STARTUP 60Ω GATE CONTROL AUX SWP SWN CURRENT SENSE CURRENT SENSE BYPASS CURENT LIMIT UVLO LATCH IN DC VREF OSCILLATOR EN CHARGE PUMP ENLDO ENUSB1 CURRENT SENSE DRIVER LDOIN BANDGAP REFERENCE (VREF ) USB1 FAULT 1 CURRENT LIMIT LDOOUT DEGLITCH OVER TEMP. DRIVER ENUSB2 CURRENT LIMIT RESET 175ms Delay Falling Edge CURRENT SENSE USB2 FAULT 2 DEGLITCH 3.1V (VREF ILIM2 ILIM1 referenced) Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 9 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com PARAMETER MEASUREMENT INFORMATION OUT tf tr RL 90% CL 90% VOUT 10% TEST CIRCUIT 50% VEN 50% ton toff 90% VOUT 10% VOLTAGE WAVEFORMS Figure 1. Test Circuit and Voltage Waveforms TYPICAL CHARACTERISTICS Maximum Total DC/DC Current vs Input Voltage Boost Efficiency vs Output Current 100.00 2500 VIN = 2.7V 80.00 2000 VIN =3.3V Typical VIN = 5.0V 70.00 1500 Boost Efficiency - % I O –Total Boost Output Current - mA 90.00 1000 Conservative 500 60.00 50.00 40.00 30.00 20.00 10.00 0 1.75 0.00 2.25 2.75 3.25 3.75 4.25 4.75 5.25 0 0.2 0.4 0.6 0.8 1 1.2 VIN – Input Voltage - V Figure 2. 10 Figure 3. Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 TYPICAL CHARACTERISTICS (continued) Boost Output Voltage vs Input Voltage TPS2505 Typical On Input Current With No Load 14 5.30 Temp . = 25°C ENUSB 1/2 = Hi ENLDO = Hi Boost Switching No load 5.25 VAUX – B oost Output Voltage - V 12 IIN – Input Curre nt - mA 10 8 6 4 5.20 5.15 IAUX = 500 mA IAUX = 200 mA 5.10 5.05 I AUX = 0mA 5.00 IAUX = 1000 mA 4.95 4.90 2 4.85 4.80 0 1.8 1.8 2.2 2.6 3 3.4 3.8 4.2 4.6 5 2.2 2.6 3.0 3.4 3.8 4.2 4.6 5.0 VIN – Input Voltage - V VIN –Input Voltage - V Figure 4. Figure 5. Boost Output Voltage vs Load Current Boost Startup after Enable – No Load 5.3 V IN VAUX – Boost Output Voltage - V 5.25 VIN = 5 .25V V IN = 4.95V CAUX = 22uF No load 5.2 VAUX 5.15 VIN = 2.8V 5.1 IIN 5.05 VIN = 5.0V VIN =3.0 V 5 0 200 400 600 800 1000 IAUX –Boost Output Current - A Figure 6. Figure 7. Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 11 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com TYPICAL CHARACTERISTICS (continued) USB1 Startup after Enable – No Load Boost Output Ripple in PFM Operation V IN VIN = 3.0V CAUX = 22uF CUSB1 = 100uF CUSB2 = 100uF IO = 100mA VAUX 100mV/div VIN = 4.95V CAUX= 22uF CUSB1 = 220uF RILIM1 = 39 kO No load V AUX VSW 2V/div I IN V USB1 200us/div Figure 8. Figure 9. Boost Load Transient Response, 500 mA - 1 A, USB1/2 Enabled LDO RESET Deglitch Time VAUX VLDOIN VUSB1 VIN = 4.95V CAUX = 22uF CUSB1 = 100uF CUSB2 = 100uF IUSB1 = 500mA IAUX = 0mA V USB2 VLDOOUT VRESET IUSB2 Figure 10. 12 Figure 11. Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 TYPICAL CHARACTERISTICS (continued) LDO Output Voltage vs Output Current LDO Load Transient Response 0 mA - 200 mA 4 VAUX VIN = VAUX VLDOOUT VIN = 4.95V CAUX = 22uF LDOIN connected to AUX VRESET ILDOOUT VO(L DOOU T) – LDO Output V oltag e - V 3.5 3 2.5 2 1.5 1 0.5 0 0 100 200 300 400 500 I O( LDOOUT) – LDO Output Current - mA Figure 12. Figure 13. THEORY OF OPERATION DESCRIPTION This device targets applications for host-side USB devices where a 5-V power rail, required for USB operation, is unavailable. The TPS2505 integrates the functionality of a synchronous boost converter, 3.3-V LDO with power good RESET signal and dual USB switches into a monolithic integrated circuit so that lower-voltage rails can be used directly to provide USB power. The TPS2505 can also be powered by an upstream USB port as it limits the inrush current during power up to less than 100 mA to meet USB 2.0 specifications. The boost converter is highly integrated, including the switching MOSFETs (low-side N-channel, high-side synchronous P-channel), gate-drive and analog-control circuitry, and control-loop compensation. Additional features include high-efficiency light-load operation, overload and short-circuit protection, and controlled monotonic soft start. The USB switch integrates all necessary functions, including back-to-back series N-channel MOSFETs, charge-pump gate driver, and analog control circuitry. The current-limit protection is user-adjustable by selecting the RILIM1/2 resistors from ILIM1/2 to GND. BOOST CONVERTER Start-Up Input power to the TPS2505 is provided from IN to GND. The device has an undervoltage lockout (UVLO) circuit that disables the device until the voltage on IN exceeds 2.15 V (typical). The TPS2505 goes through its normal start-up process and attempts to regulate the AUX voltage to 5.1 V (typical). The boost converter has a two-step start-up sequence. During the initial startup, the output of the boost is connected to VIN through a resistive switch that limits the startup current, ISTART, to be below 100mA. This allows the TPS2505 to be USB 2.0 compliant when powered by an upstream USB port. The boost output must be unloaded during startup. ISTART charges the output capacitance on VAUX until VAUX reaches VIN – VEXIT. The converter begins to switch once VAUX exceeds VIN – VEXIT. The initial duty cycle of the device is limited by a closed-loop soft start that ramps the reference voltage to the internal error amplifier to provide a controlled, monotonic start-up on VAUX. The boost converter goes through this cycle any time the voltage on VAUX drops below VIN – VEXIT due to overload conditions or the boost converter re-enables after normal shutdown. Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 13 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com The USB switches are powered directly from VAUX and turns on once the UVLO of the USB switches is met (4.3 V typical). The turn-on is controlled internally to provide a monotonic start-up on VUSB1/2. Normal Operation The boost converter runs at a 1-MHz fixed frequency and regulates the output voltage VAUX using a pulse-width modulating (PWM) topology that adjusts the duty cycle of the low-side N-channel MOSFET on a cycle-by-cycle basis. The PWM latch is set at the beginning of each clock cycle and commands the gate driver to turn on the low-side MOSFET. The low-side MOSFET remains on until the PWM latch is reset. Voltage regulation is controlled by a peak-current-mode control architecture. The voltage loop senses the voltage on VAUX and provides negative feedback into an internal, transconductance-error amplifier with internal compensation and resistor divider. The output of the transconductance-error amplifier is summed with the output of the slope-compensation block and provides the error signal that is fed into the inverting input of the PWM comparator. Slope compensation is necessary to prevent subharmonic oscillations that may occur in peak-current mode control architectures that exceed 50% duty cycle. The PWM ramp fed into the non-inverting input of the PWM comparator is provided by sensing the inductor current through the low-side N-channel MOSFET. The PWM latch is reset when the PWM ramp intersects the error signal and terminates the pulse width for that clock period. The TPS2505 stops switching if the peak-demanded current signal from the error amplifier falls below the zero-duty-cycle threshold of the device. Low-Frequency Mode The TPS2505 enters low-frequency mode above VIN = VLFM (4.35 V typical) by reducing the dc/dc converter frequency from 1 MHz (typical) to 250 kHz (typical). Current-mode control topologies require internal leading-edge blanking of the current-sense signal to prevent nuisance trips of the PWM control MOSFET. The consequence of leading-edge blanking is that the PWM controller has a minimum controllable on-time (85 ns typical) that results in a minimum controllable duty cycle. In a boost converter, the demanded duty cycle decreases as the input voltage increases. The boost converter pulse-skips if the demanded duty cycle is less than what the minimum controllable on-time allows, which is undesirable due to the excessive increase in switching ripple. When the TPS2505 enters low-frequency mode above VIN = VLFM, the minimum controllable duty cycle is increased because the minimum controllable on-time is a smaller percentage of the entire switching period. Low-frequency mode prevents pulse skipping at voltages larger than VLFM. The TPS2505 resumes normal 1-MHz switching operation when VIN decreases below VLFM. One effect of reducing the switching frequency is that the ripple current in the inductor and output AUX capacitors is increased. It is important to verify that the peak inductor current does not exceed the peak switch current limit ISW (4.5 A typical) and that the increase in AUX ripple is acceptable during low-frequency mode. No-Frequency Mode The TPS2505 enters no-frequency mode above VIN = VNFM (5.05 V typical) by disabling the oscillator and turning on the high-side synchronous PMOS 100% of the time. The input voltage is now directly connected to the AUX output through the inductor and high-side PMOS. Power dissipation in the device is reduced in no-frequency mode because there is no longer any switching loss and no RMS current flows through the low-side control NMOS, which results in higher system-level efficiency. The boost converter resumes switching when VIN falls below VNFM. Pulsed Frequency Mode (PFM) Light-Load Operation The TPS2505 enters the PFM control scheme at light loads to increase efficiency. The device reduces power dissipation while in the PFM control scheme by disabling the gate drivers and power MOSFETs and entering a pulsed-frequency mode (PFM). PFM works by disabling the gate driver when the PFM latch is set. During this time period there is no switching, and the load current is provided solely by the output capacitor. There are two comparators that determine when the device enters or leaves the PFM control scheme. The first comparator is the PFM-enter comparator. The PFM-enter comparator monitors the peak demanded current in the inductor and allows the device to enter the PFM control scheme when the inductor current falls below IINDLOW (420 mA typical). The second comparator is the AUX-low comparator. The AUX-low comparator monitors AUX and forces the converter out of the PFM control scheme and resumes normal operation when the voltage on AUX falls below AUXLOW (5 V typical). The PFM control scheme is disabled during low-frequency mode when VIN > VLFM (4.35 V typical). 14 Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 Overvoltage Protection The TPS2505 provides overvoltage protection on VAUX to protect downstream devices. Overvoltage protection is provided by disabling the gate drivers and power MOSFETs when an overvoltage condition is detected. The TPS2505 uses a single AUX-high comparator to monitor the AUX voltage by sensing the voltage on the internal feedback node fed into the error amplifier. The AUX-high comparator disables the gate driver whenever the voltage on AUX exceeds the regulation point by 5% (typical). The gate driver remains disabled until the AUX voltage falls below the 5% high OVP threshold. The overvoltage protection feature is disabled when VIN > VNFM (5.05 V typical) to prevent unwanted shutdown. Overload Conditions The TPS2505 boost converter uses multiple overcurrent protection features to limit current in the event of an overload or short-circuit condition. The first feature is the lower current-limit comparator that works on a cycle-by-cycle basis. This comparator turns off the low-side MOSFET by resetting the PWM latch whenever the current through the low-side MOSFET exceeds 4.5 A (typical). The low-side MOSFET remains off until the next switching cycle. The second feature is the upper current-limit comparator that disables switching for eight switching cycles whenever the current in the low-side MOSFET exceeds 6.7 A (typical). After eight switching cycles, the boost converter resumes normal operation. The third feature is the constant-current start-up ISTART comparator that disables switching and regulates the current through the high-side MOSFET whenever the voltage on VAUX drops below the input voltage by VEXIT (700 mV typ). This feature protects the boost converter in the event of an output short circuit on VAUX. ISTART also current-limit protects the synchronous MOSFET in no-frequency mode when VIN > VNFM (5.05 V typical). The converter goes through normal start-up operation once the short-circuit condition is removed. A fourth feature is the 85% (typical) maximum-duty-cycle clamp that prevents excessive current from building in the inductor. Determining the Maximum Allowable AUX and USB1/2 Current The maximum output current of the boost converter out of AUX depends on several system-level factors including input voltage, inductor value, switching frequency, and ambient temperature. The limiting factor for the TPS2505 is the peak inductor current, which cannot exceed ISW (3 A minimum). The cycle-by-cycle current-limit turns off the low-side NMOS as a protection mechanism whenever the inductor current exceeds ISW. Figure 2 can be used as a guideline for determining the maximum total current at different input voltages. The typical plot assumes nominal conditions: 2.2-µH inductor, 1-MHz/250-kHz switching frequency, nominal MOSFET on-resistances. The conservative plot assumes more pessimistic conditions: 1.7-µH inductor, 925-kHz/230-kHz switching frequency, and maximum MOSFET on-resistances. The graph accounts for the frequency change from 1-MHz to 250-kHz when VIN > VLFM (4.35 V typical) and for the no-frequency mode when VIN > VNFM (5.05 V typical), which explains the discontinuities of the graph. Table 2. Maximum Total DC/DC Current (IAUX + IUSB1+ IUSB2) at Common Input Voltages INPUT VOLTAGE (V) MAXIMUM TOTAL OUTPUT CURRENT (IAUX + IUSB1 + IUSB2) CONSERVATIVE (mA) TYPICAL (mA) 599 757 2.5 916 1113 2.7 1008 1216 3 1148 1374 3.3 1308 1536 3.6 1445 1704 4.35 1241 1730 4.5 1364 1858 4.75 1593 2093 5.05 2300 2300 5.25 2300 2300 1.8 Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 15 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com USB SWITCHES Overview The TPS2505 integrates a current-limited, power-distribution switches using N-channel MOSFETs for applications where short circuits or heavy capacitive loads are encountered. The current-limit threshold is user-programmable between 130 mA and 1.4 A (typical) by selecting an external resistor. The device incorporates an internal charge pump and gate-drive circuitry necessary to fully enhance the N-channel MOSFETs. The internal gate drivers controls the MOSFETs turn-on to limit large current and voltage surges by providing built-in soft-start functionality. The power switches have an independent undervoltage lockout (UVLO) circuit that disables them until the voltage on AUX reaches 4.3 V (typical). Built-in hysteresis prevents unwanted on/off cycling due to input voltage drop on AUX from current surges on the output of the power switch. The power switches have an independent logic-level enable control (ENUSB1/2) that gates power-switch turn-on and bias for the charge pump, driver, and miscellaneous control circuitry. A logic-high input on ENUSB1/2 enables the drivers, control circuits, and power switches. The enable input are compatible with CMOS, TTL, LVTTL, 2.5-V, and 1.8-V logic levels. Overcurrent Conditions The TPS2505 power switches respond to overcurrent conditions by limiting its output current to the IOS level. The device maintains a constant output current and reduces the output voltage accordingly during an overcurrent condition. Two possible overload conditions can occur. The first condition is when a short circuit or partial short circuit is present on the output of the switch prior to device turn-on and the device is powered up or enabled. The output voltage is held near zero potential with respect to ground, and the TPS2505 ramps the output current to IOS. The TPS2505 power switches limit the current to IOS until the overload condition is removed or the device begins to cycle thermally. The second condition is when a short circuit, partial short circuit, or transient overload occurs while the device is already enabled and powered on. The current-sense amplifier is overdriven during this time and momentarily disables the power switch. The current-sense amplifier recovers and limits the output current to IOS. The power switches thermally cycle if an overload condition is present long enough to activate thermal limiting in any of the foregoing cases. The power switches turns off when the junction temperature exceeds 130°C while in current-limit. The power switches remains off until the junction temperature cools 10°C and then restarts. The TPS2505 power switches cycles on/off until the overload is removed. The boost converter is independent of the power-switch thermal sense and continues to operate as long as the temperature of the boost converter remains less than 150°C and does not trigger the boost-converter thermal sense. FAULT1/2 Response The FAULT1/2 open-drain outputs are asserted low during an overcurrent condition that causes VUSB to fall below VTRIP (4.6 V typical) or causes the junction temperature to exceed the shutdown threshold (130°C). The TPS2505 asserts the FAULT1/2 signals until the fault condition is removed and the power switches resume normal operation. The FAULT1/2 signals are independent of the boost converter or each other. The FAULT1/2 signals use an internal delay deglitch circuit (8-ms typical) to delay asserting the FAULT1/2 signals during an overcurrent condition. The power switches must remain in an overcurrent condition for the entire deglitch period or the deglitch timer is restarted. This ensures that FAULT1/2 are not accidentally asserted due to normal operation such as starting into a heavy capacitive load. The deglitch circuitry delays entering and leaving fault conditions. Overtemperature conditions are not deglitched and assert the FAULT1/2 signals immediately. Undervoltage Lockout The undervoltage lockout (UVLO) circuit disables the TPS2505 power switch until the input voltage on AUX reaches the power switch UVLO turn-on threshold of 4.3 V (typical). Built-in hysteresis prevents unwanted on/off cycling due to input-voltage drop from large current surges. Programming the Current-Limit Threshold Resistor RILIM The overcurrent thresholds are user programmable via external resistors. The TPS2505 uses an internal regulation loop to provide a regulated voltage on the ILIM1/2 pins. The current-limit thresholds are proportional to the current sourced out of ILIM1/2. The recommended 1% resistor range for RILIM1/2 is 16.1 kΩ ≤ RILIM ≤ 200 kΩ to ensure stability of the internal regulation loop. Many applications require that the minimum current limit is above a certain current level or that the maximum current limit is below a certain current level, so it is important to consider the tolerance of the overcurrent threshold when selecting a value for 16 Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 RILIM1/2. The following equations and Figure 14 can be used to calculate the resulting overcurrent threshold for a given external resistor value (RILIM1/2). Figure 14 includes current-limit tolerance due to variations caused by temperature and process. However, the equations do not account for tolerance due to external resistor variation, so it is important to account for this tolerance when selecting RILIM1/2. The traces routing the RILIM1/2 resistors to the TPS2505 should be as short as possible to reduce parasitic effects on the current-limit accuracy. RILIM1/2 can be selected to provide a current-limit threshold that occurs 1) above a minimum load current or 2) below a maximum load current. To design above a minimum current-limit threshold, find the intersection of RILIM and the maximum desired load current on the IOS(min) curve and choose a value of RILIM below this value. Programming the current limit above a minimum threshold is important to ensure start up into full load or heavy capacitive loads. The resulting maximum current-limit threshold is the intersection of the selected value of RILIM1/2 and the IOS(max) curve. To design below a maximum current-limit threshold, find the intersection of RILIM and the maximum desired load current on the IOS(max) curve and choose a value of RILIM1/2 above this value. Programming the current limit below a maximum threshold is important to avoid current-limiting upstream power supplies, causing the input voltage bus to droop. The resulting minimum current-limit threshold is the intersection of the selected value of RILIM1/2 and the IOS(min) curve. Current-limit threshold equations (IOS): 27,570V RILIM 1/ 2 0.93 k W (1) 28, 235V RILIM 1/ 2 0.998 k W (2) 32,114V I OS (min) (mA) = RILIM 1/ 21.114 k W (3) I OS (max) (mA) = I OS (typ ) (mA) = USB Current-Limit Threshold vs RILIM Over Temperature and Process (VIN = 3.3 V, IAUX = 0 A, Secondary USB Switch Disabled) 1800 IAUX = 0 mA VIN = 3 .3 V No load on secondary USB switch 1600 I OS – US B Curre nt LImi t - mA 1400 IOS(max ) 1200 1000 800 IOS(typ) 600 400 IOS(min) 200 0 20 30 40 50 60 70 80 90 100 R ILIM1/2 – USB Current Limit Resistance - kO Figure 14. Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 17 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com 3.3-V LDO The TPS2505 integrates a 3.3-V LDO with a maximum load capacity of 200 mA. The LDO can be powered by the AUX boost output to allow operation when there is only a low voltage supply such as an alkaline battery. The LDO will only turn on once VAUX reaches the UVLO threshold. The LDO can also be connected to be powered to an external supply if no additional load to AUX is desired or to reduce power dissipation (in case the supply is lower than the 5.1-V boost output). However, the boost must be enabled to allow the LDO to operate, even if connected to a separate supply. RESET COMPARATOR The Reset Comparator integrated in the TPS2505 provides a power-good signal that indicates when the LDO output has reached a 3.1-V threshold. The comparator has a 175-ms deglitch delay for the low-to-high transition to prevent any glitches when the LDO is powering up. Hysteresis has been added to the comparator to increase noise immunity and avoid unwanted glitches in the output during LDO transients. THERMAL SHUTDOWN The TPS2505 self-protects using two independent thermal sensing circuits that monitor the operating temperatures of the boost converter and power switch independently and disable operation if the temperature exceeds recommended operating conditions. The boost converter and power switches each have an ambient thermal sensor that disables operation if the measured junction temperature in that part of the circuit exceeds 150°C. The boost converter continues to operate even if the power switch is disabled due to an over-temperature condition. COMPONENT RECOMMENDATIONS The main functions of the TPS2505 are integrated and meet recommended operating conditions with a wide range of external components. The following sections give guidelines and trade-offs for external component selection. The recommended values given are conservative and intended over the full range of recommended operating conditions. Boost Inductor Connect the boost inductor from IN to SW. The inductance controls the ripple current through the inductor. A 2.2-µH inductor is recommended, and the minimum and maximum inductor values are constrained by the integrated features of the TPS2505. The minimum inductance is limited by the peak inductor-current value. The ripple current in the inductor is inversely proportional to the inductance value, so the output voltage may fall out of regulation if the peak inductor current exceeds the cycle-by-cycle current-limit comparator (3 A minimum). Using a nominal 2.2-µH inductor allows full recommended current operation even if the inductance is 20% low (1.76 µH) due to component variation. The maximum inductance value is limited by the internal compensation of the boost-converter control loop. A maximum 4.7-µH (typical) inductor value is recommended to maintain adequate phase margin over the full range of recommended operating conditions. IN Capacitance Connect the input capacitance from IN to the reference ground plane (see LAYOUT RECOMMENDATIONS for connecting PGND and GND to the ground plane). Input capacitance reduces the AC voltage ripple on the input rail by providing a low-impedance path for the switching current of the boost converter. The TPS2505 does not have a minimum or maximum input capacitance requirement for operation, but a 10-µF, X7R or X5R ceramic capacitor is recommended for most applications for reasonable input-voltage ripple performance. There are several scenarios where it is recommended to use additional input capacitance: • The output impedance of the upstream power supply is high, or the power supply is located far from the TPS2505. • The TPS2505 is tested in a lab environment with long, inductive cables connected to the input, and transient voltage spikes could exceed the absolute maximum voltage rating of the device. • The device is operating in PFM control scheme near VIN = 1.8 V, where insufficient input capacitance may cause the input ripple voltage to fall below the minimum 1.75-V (typical) UVLO circuit, causing device turnoff. Additionally, it is good engineering practice to use an additional 0.1-µF ceramic decoupling capacitor close to the IC to prevent unwanted high-frequency noise from coupling into the device. 18 Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 AUX Capacitance Connect the boost-converter output capacitance from AUX to the reference ground plane. The AUX capacitance controls the ripple voltage on the AUX rail and provides a low-impedance path for the switching and transient-load currents of the boost converter. It also sets the location of the output pole in the control loop of the boost converter. There are limitations to the minimum and maximum capacitance on AUX. The recommended minimum capacitance on AUX is a 22-µF, X5R or X7R ceramic capacitor. A 10-V rated ceramic capacitor is recommended to minimize the capacitance derating loss due to dc bias applied to the capacitor. The low ESR of the ceramic capacitor minimizes ripple voltage and power dissipation from the large, pulsating currents of the boost converter and provides adequate phase margin across all recommended operating conditions. In some applications, it is desirable to add additional AUX capacitance. Additional AUX capacitance reduces transient undershoot/overshoot voltages due to load steps and reduces AUX ripple in the PFM control scheme. Adding AUX capacitance changes the control loop, resulting in reduced phase margin, so it is recommended that no more than 220 µF of additional capacitance be added in parallel to the 22-µF ceramic capacitor. The combined output capacitance on AUX and USB should not exceed 500 µF. USB Capacitance Connect the USB1/2 capacitances from USB1/2 to the reference ground plane. The USB1/2 capacitances are on the outputs of the power switches and provide energy for transient load steps. The TPS2505 does not require any USB capacitance for operation. Additional capacitance can be added on USB1/2 outputs, but it is recommended to not exceed 220 µF to maintain adequate phase margin for the boost converter control loop. The combined output capacitance on AUX and USB should not exceed 500 µF. USB applications require a minimum of 120 µF on downstream facing ports. ILIM1/2 and FAULT1/2 Resistors Connect the ILIM1/2 resistors from ILIM1/2 to the reference ground plane. The ILIM1/2 resistors programs the current-limit threshold of the USB power switches (see Programming the Current-Limit Threshold Resistor RILIM). The ILIM1/2 pins are the output of internal linear regulators that provide a fixed 400-mV output. The recommended nominal resistor value using 1% resistors on ILIM1/2 is 16.1 kΩ ≤ RILIM ≤ 200 kΩ. This range should be adjusted accordingly if 1% resistors are not used. Do not overdrive ILIM1/2 with an external voltage or connect directly to GND. Connect the ILIM1/2 resistors as close to the TPS2505 as possible to minimize the effects of parasitics on device operation. Do not add external capacitance on the ILIM1/2 pins. The ILIM1/2 pins should not be left floating. Connect the FAULT1/2 resistors from the FAULT1/2 pins to an external voltage source such as VAUX or VIN. The FAULT1/2 pins are open-drain outputs capable of sinking a maximum current of 10 mA continuously. The FAULT1/2 resistors should be sized large enough to limit current to under 10 mA continuously. Do not tie FAULT1/2 directly to an external voltage source. The maximum recommended voltage on FAULT1/2 is 6.5 V. The FAULT1/2 pin can be left floating if not used. LAYOUT RECOMMENDATIONS Layout is an important design step due to the high switching frequency of the boost converter. Careful attention must be applied to the PCB layout to ensure proper function of the device and to obtain the specified performance. Potential issues resulting from poor layout techniques include wider line and load regulation tolerances, EMI noise issues, stability problems, and USB current-limit shifts. It is critical to provide a low-impedance ground path that minimizes parasitic inductance. Wide and short traces should be used in the high-current paths, and components should be placed as close to the device as possible. Grounding is an important part of the layout. The device has a PGND and a GND pin. The GND pin is the quiet analog ground of the device and should have its own separate ground pour; connect the quiet signals to GND including the RILIM1/2 resistors and any input decoupling capacitors to the GND pour. It is important that the RILIM1/2 resistors be tied to a quiet ground to avoid unwanted shifts in the current-limit threshold. The PGND pin is the high-current power-stage ground; the ground pours of the output (AUX) and bulk input capacitors should be tied to PGND. PGND and GND should to be tied together in one location at the IC thermal pad, creating a star-point ground. The output filter of the boost converter is also critical for layout. The inductor and AUX capacitors should be placed to minimize the area of current loop through AUX–PGND–SW.The layout for the TPS2505EVM evaluation board is shown in Figure 15 and should be followed as closely as possible for best performance. Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 19 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 AUX www.ti.com SW VIN PGND PGND HSPORT1 PGND LDOOUT HSPORT2 AUX PGND AGND LDOIN PGND AGND PGND VIN HS PO R T2 HSPORT1 U OO LD T VIN Figure 15. Layout Example for TPS2505 Application – 4 Layer Board 20 Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 APPLICATION INFORMATION STEP-BY-STEP DESIGN PROCEDURE The following design procedure provides an example for selecting component values for the TPS2505. The following design parameters are needed as inputs to the design process. • Input voltage range • Output voltage on AUX • Input ripple voltage • Output ripple voltage on AUX • Output current rating of AUX rail • Output current rating of USB rail • Nominal efficiency target • Operating frequency A power inductor, input and output filter capacitors, and current-limit threshold resistor are the only external components required to complete the TPS2505 boost-converter design. The input ripple voltage, AUX ripple voltage, and total output current affect the selection of these components. This design example assumes the following input specifications. PARAMETER EXAMPLE VALUE Input voltage range (VIN) 2.7 V to 4.2 V AUX voltage (VAUX) 5.1 V (internally fixed) Input ripple voltage (ΔVIN) 15 mV AUX ripple voltage (ΔVAUX) 50 mV AUX current (IAUX) 0.3 A LDO current (ILDO) (powered from AUX) 0.1 A USB1 current (IUSB1 ) 0.5 A USB2 current (IUSB2 ) 0.1 A Total current (ITOTAL = IAUX + ILDO + IUSB1 + IUSB2) 1A Efficiency target, nominal 90% Switching frequency (fSW) 1 MHz SWITCHING FREQUENCY The switching frequency of the TPS2505 is internally fixed at 1 MHz for the specified VIN range. AUX VOLTAGE The AUX voltage of the TPS2505 is internally fixed at 5.1 V. DETERMINE MAXIMUM TOTAL CURRENT (IAUX + ILDO + IUSB1 + IUSB2 ) Using Figure 2, the maximum total current at VIN = 2.7 V is 1 A using the conservative line. The design requirements are met for this application. POWER INDUCTOR The inductor ripple current, Δi, should be at least 20% of the average inductor current to avoid erratic operation of the peak-current-mode PWM controller. Assume an inductor ripple current, Δi, which is 30% of the average inductor current and a power-converter efficiency, h, of 90%. Using the minimum input voltage, the average inductor current at VIN = 2.7 V is: I IN = VAUX ´ ITOTAL 5.1V ´1A = = 2.1A VIN ´h 2.7V ´ 0.9 (4) Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 21 TPS2505 SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 www.ti.com IL Di I L_pk IIN Time Figure 16. Waveform of Current in Boost Inductor The corresponding inductor ripple current is: Di = 0.3 ´ I IN = 0.3 ´ 2.1A = 630mA (5) Verify that the peak inductor current is less than the 3-A peak switch current: I L _ pk = I IN + Di = 2.42 A < 3 A 2 (6) The following equation estimates the duty cycle of the low-side SWN MOSFET: D= ton V - VIN + I IN ´ ( RSWP + RL ) 5.1V - 2.7V + 2.1A ´ (0.1W + 0.07W) = AUX = = 0.54 5.1V + 2.1A ´ (0.1W + 0.1W) ton + toff VAUX + VIN ´ ( RSWP + RSWN ) (7) where RSWN is the low-side control MOSFET on-resistance, RSWP is the high-side synchronous MOSFET on-resistance, and RL is an estimate of the inductor dc resistance. The following equation calculates the recommended inductance for this design. L= VIN ´ D 2, 7V ´ 0.54 = = 2.31m H f ´ Di 1´106 Hz ´ 0.63 A (8) The rms inductor current is: 2 I L _ RMS = I IN 2 2 æ Di ö æ 0.63 A ö 2 +ç ÷ = (2.1A) + ç ÷ = 2.11A è2 3ø è 2 3 ø (9) Select a Coilcraft LPS4018-222ML inductor. This 2.2-mH inductor has a saturation current rating of 2.7 A and an rms current rating of 2.3 A. See COMPONENT RECOMMENDATIONS for specific additional information. OUTPUT AUX CAPACITOR SELECTION The AUX output capacitor, CAUX, discharges during the PWM MOSFET on-time, resulting in an output ripple voltage of ΔVAUX. ΔVAUX is largest at maximum load current. C AUX = D ´ ITOTAL f ´ DVAUX C AUX _ min (10) 0.54 ´1A = = 10.8m F 1´106 Hz ´ 50mV (11) Ceramic capacitors exhibit a dc bias effect, whereby the capacitance falls with increasing bias voltage. The effect is worse for capacitors in smaller case sizes and lower voltage ratings. X5R and X7R capacitors exhibit less DC bias effect than Y5V and Z5U capacitors. 22 Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 TPS2505 www.ti.com SPAS093A – DECEMBER 2009 – REVISED FEBRUARY 2010 Select a TDK C3225X5R1A226M 22-mF, 10-V X5R ceramic capacitor to allow for a 50% drop in capacitance due to the dc bias effect. See COMPONENT RECOMMENDATIONS for specific additional information. OUTPUT USB1/2 CAPACITOR SELECTION The USB1/2 output capacitors provide energy during a load step on the USB outputs. The TPS2505 does not require a USB output capacitor, but many USB applications require that downstream-facing ports be bypassed with a minimum of 120-mF, low-ESR capacitance. Select a Panasonic EEVFK1A151P 150-mF, 10-V capacitor. INPUT CAPACITOR SELECTION The ripple current through the input filter capacitor is equal to the ripple current through the inductor. If the ESL and ESR of the input filter capacitor are ignored, then the required input filter capacitance is: CIN = 630mA Di = = 5.25m F 8 ´ f ´ DVIN 8 ´1´106 Hz ´15mV (12) Select a TDK C2012X5R1A106K 10-µF, 10-V, X5R, size 805 ceramic capacitor. The capacitance drops 20% at 3.3-V bias, resulting in an effective capacitance of 8 µF. An additional 0.1-µF ceramic capacitor should be placed locally from IN to GND to prevent noise from coupling into the device if the input capacitor cannot be located physically near to the device. In applications where long, inductive cables connect the input power supply to the device, additional bulk input capacitance may be necessary to minimize voltage overshoot. See COMPONENT RECOMMENDATIONS for specific additional information. CURRENT-LIMIT THRESHOLD RESISTOR RILIM The current-limit threshold IOS of the power switches are externally adjustable by selecting the RILIM1/2 resistors. To eliminate the possibility of false tripping, RILIM1/2 should be selected so that the minimum tolerance of the current-limit threshold is greater than the maximum specified USB load, IUSB. It is also important to account for IOS shifts due to variation in VIN and IAUX. This shift due to the additional loading in AUX can add up to ±75 mA of variation to the IOS as calculated according to Programming the Current-Limit Threshold Resistor RILIM. Select RILIM1 so that the minimum current-limit threshold equals 600 mA to ensure a minimum IUSB1 current-limit threshold of 525 mA. In the same way, select RILIM2 so that the minimum current-limit threshold equals 200 mA to ensure a minimum IUSB2 current-limit threshold of 125 mA. 1 RILIM 1/ 2 æ 32.114 ö1.114 =ç ÷ è I OS min ø æ 32.114 ö RILIM 1 = ç ÷ è 600mA ø 1 1.114 æ 32.114 ö RILIM 2 = ç ÷ è 200mA ø (13) = 35.62k W (14) 1 1.114 = 95.49k W (15) Choose the next smaller 1% resistor, which are 34.8 kΩ for RLIM1 and 95.3 kΩ for RLIM2. Submit Documentation Feedback Copyright © 2009–2010, Texas Instruments Incorporated Product Folder Link(s): TPS2505 23 PACKAGE OPTION ADDENDUM www.ti.com 15-Feb-2010 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing TPS2505B1RGWR ACTIVE VQFN RGW Pins Package Eco Plan (2) Qty 20 3000 Green (RoHS & no Sb/Br) Lead/Ball Finish CU NIPDAU MSL Peak Temp (3) Level-2-260C-1 YEAR (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. 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