AAT2512 Dual 400mA High Frequency Buck Converter General Description Features The AAT2512 is a member of AnalogicTech's Total Power Management IC™ (TPMIC™) product family. It is a dual channel synchronous buck converter operating with an input voltage range of 2.7V to 5.5V, making it ideal for applications with singlecell lithium-ion/polymer batteries. • • Both regulators have independent input and enable pins. Offered with fixed or adjustable output voltages, each channel is designed to operate with 27µA (typical) of quiescent current, allowing for high efficiency under light load conditions. The AAT2512 requires only three external components (CIN, COUT, and LX) for each converter, minimizing cost and real estate. Both channels are designed to deliver 400mA of load current and operate with a switching frequency of 1.4MHz, reducing the size of external components. The AAT2512 is available in a Pb-free, 12-pin TDFN33 package and is rated over the -40°C to +85°C temperature range. • • • • • • • • • • SystemPower™ VIN Range: 2.7V to 5.5V Output Current: — Channel 1: 400mA — Channel 2: 400mA 98% Efficient Step-Down Converter Integrated Power Switches 100% Duty Cycle 1.4MHz Switching Frequency Internal Soft Start 150µs Typical Turn-On Time Over-Temperature Protection Current Limit Protection Available in TDFN33-12 Package -40°C to +85°C Temperature Range Applications • • • • • Cellular Phones Digital Cameras Handheld Instruments Microprocessor / DSP Core/ IO Power PDAs and Handheld Computers Typical Application V BAT C IN VIN1 LX1 VIN2 FB1 L1 4.7µH AAT2512 EN1 LX2 EN2 FB2 GND 2512.2006.06.1.4 V OUT1 VOUT2 L2 4.7µH COUT 4.7µF 4.7µF 1 AAT2512 Dual 400mA High Frequency Buck Converter Pin Descriptions Pin # Symbol 1 EN1 2 FB1 3, 6, 7, 10 4 GND EN2 5 FB2 8 LX2 9 11 VIN2 LX1 12 VIN1 Function Enable pin for Channel 1. When connected low, it disables the channel and consumes less than 1µA of current. When connected high, normal operation. Feedback input pin for Channel 1. This pin is connected to the converter output. It is used to see the output of the converter to regulate to the desired value via an external resistor divider. Ground. Enable pin for Channel 2. When connected low, it disables the channel and consumes less than 1µA of current. When connected high, normal operation. Feedback input pin for Channel 2. This pin is connected to the converter output. It is used to see the output of the converter to regulate to the desired value via an external resistor divider. Power switching node for Channel 2. Output switching node that connects to the output inductor. Input supply voltage for Channel 2. Must be closely decoupled. Power switching node for Channel 2. Output switching node that connects to the output inductor. Input supply voltage for Channel 1. Must be closely decoupled. Pin Configuration TDFN33-12 (Top View) EN1 FB1 GND EN2 FB2 GND 2 1 12 2 11 3 10 4 9 5 8 6 7 VIN1 LX1 GND VIN2 LX2 GND 2512.2006.06.1.4 AAT2512 Dual 400mA High Frequency Buck Converter Absolute Maximum Ratings1 Symbol VIN VLX VFB VEN TJ TLEAD Description Input Voltages to GND LX to GND FB1 and FB2 to GND EN1 and EN2 to GND Operating Junction Temperature Range Maximum Soldering Temperature (at leads, 10 sec) Value Units 6.0 -0.3 to VP + 0.3 -0.3 to VP + 0.3 -0.3 to 6.0 -40 to 150 300 V V V V °C °C Value Units 2.0 50 W °C/W Thermal Information Symbol PD θJA Description Maximum Power Dissipation Thermal Resistance2 1. Stresses above those listed in Absolute Maximum Ratings may cause permanent damage to the device. Functional operation at conditions other than the operating conditions specified is not implied. Only one Absolute Maximum Rating should be applied at any one time. 2. Mounted on an FR4 board. 2512.2006.06.1.4 3 AAT2512 Dual 400mA High Frequency Buck Converter Electrical Characteristics1 VIN = 3.6V; TA = -40°C to +85°C, unless otherwise noted. Typical values are TA = 25°C. Symbol Description VIN Input Voltage VOUT Output Voltage Tolerance VOUT IQ ISHDN Output Voltage Range Quiescent Current Shutdown Current LX Leakage Current Feedback Leakage P-Channel Current Limit High Side Switch On Resistance Low Side Switch On Resistance Line Regulation Oscillator Frequency ILX_LEAK IFB ILIM RDS(ON)H RDS(ON)L ΔVLINE FOSC TS TSD THYS VEN(L) VEN(H) IEN Start-Up Time Over-Temperature Shutdown Threshold Over-Temperature Shutdown Hysteresis Enable Threshold Low Enable Threshold High Input Low Current Conditions IOUT = 0 to 400mA; VIN = 2.7V to 5.5V Min Typ Max Units 2.7 5.5 V -3.0 3.0 % VIN 70 1.0 1.0 0.2 1.2 0.45 0.40 0.2 1.4 V µA µA µA µA A Ω Ω % MHz 150 µs 140 °C 15 °C 0.6 Per Channel EN1 = EN2 = GND VIN = 5.5V, VLX = 0 to VIN VFB = 1.0V Both Channels 27 VIN = 2.7V to 5.5V From Enable to Output Regulation; Both Channels 0.6 VIN = VFB = 5.5V 1.4 -1.0 1.0 V V µA 1. The AAT2512 is guaranteed to meet performance specifications over the -40°C to +85°C operating temperature range and is assured by design, characterization, and correlation with statistical process controls. 4 2512.2006.06.1.4 AAT2512 Dual 400mA High Frequency Buck Converter Typical Characteristics EN1 = VIN; EN2 = GND. Efficiency vs. Load DC Regulation (VOUT = 1.8V; L = 4.7μ μH) (VOUT = 1.8V) 1.0 100 Efficiency (%) 90 80 Output Error (%) VIN = 2.7V VIN = 4.2V VIN = 3.6V 70 60 50 0.1 1 10 100 0.5 VIN = 4.2V 0.0 VIN = 3.6V -0.5 -1.0 0.1 1000 VIN = 2.7V 1 Output Current (mA) Efficiency vs. Load DC Regulation 1.0 Output Error (%) Efficiency (%) 90 VIN = 5.0V 80 VIN = 4.2V VIN = 3.6V 60 VIN = 4.2V 0.5 VIN = 5.0V 0.0 VIN = 3.6V -0.5 VIN = 3.0V 50 0.1 1 10 100 -1.0 1000 0.1 1 Output Current (mA) 1.0 Output Error (%) Efficiency (%) 90 VIN = 4.2V 80 VIN = 5.0V 60 1 10 Output Current (mA) 2512.2006.06.1.4 1000 (VOUT = 3.3V; L = 6.8µH) VIN = 3.6V 50 0.1 100 DC Regulation (VOUT = 3.3V; L = 6.8μ μH) 70 10 Output Current (mA) Efficiency vs. Load 100 1000 (VOUT = 2.5V) VIN = 2.7V 70 100 Output Current (mA) (VOUT = 2.5V; L = 6.8μ μH) 100 10 100 1000 VIN = 5.0V 0.5 VIN = 4.2V 0.0 -0.5 -1.0 VIN = 3.6V 0.1 1 10 100 1000 Output Current (mA) 5 AAT2512 Dual 400mA High Frequency Buck Converter Typical Characteristics EN1 = VIN; EN2 = GND. Line Regulation (VOUT = 1.8V) 0.40 1.6 0.30 1.2 0.20 2.0 1.0 1.0 0.8 0.0 0.6 -1.0 0.4 -2.0 0.2 VEN VO IL -3.0 0.0 -4.0 -0.2 -5.0 -0.4 Accuracy (%) 1.4 3.0 4.0 Inductor Current (bottom) (A) Enable and Output Voltage (top) (V) 5.0 Soft Start (VIN = 3.6V; VOUT = 1.8V; IOUT = 400mA) IOUT = 10mA 0.10 0.00 -0.10 IOUT = 1mA IOUT = 400mA -0.20 -0.30 -0.40 2.5 3.0 3.5 Time (100μ μs/div) 4.0 4.5 5.0 5.5 6.0 Input Voltage (V) Output Voltage Error vs. Temperature Switching Frequency vs. Temperature (VIN = 3.6V; VO = 1.8V; IOUT = 400mA) (VIN = 3.6V; VOUT = 1.8V) 2.0 15.0 9.0 1.0 Variation (%) Output Error (%) 12.0 0.0 -1.0 6.0 3.0 0.0 -3.0 -6.0 -9.0 -12.0 -2.0 -40 -20 0 20 40 60 80 -15.0 -40 100 -20 0 Temperature (°°C) 80 100 50 VOUT = 1.8V 1.0 Supply Current (μ μA) Frequency Variation (%) 60 No Load Quiescent Current vs. Input Voltage 2.0 0.0 -1.0 VOUT = 2.5V -2.0 VOUT = 3.3V -3.0 2.7 3.1 3.5 3.9 4.3 Input Voltage (V) 6 40 Temperature (°°C) Frequency vs. Input Voltage -4.0 20 4.7 5.1 5.5 45 40 35 25°C 85°C 30 25 20 -40°C 15 10 2.7 3.1 3.5 3.9 4.3 4.7 5.1 5.5 Input Voltage (V) 2512.2006.06.1.4 AAT2512 Dual 400mA High Frequency Buck Converter Typical Characteristics EN1 = VIN; EN2 = GND. P-Channel RDS(ON) vs. Input Voltage N-Channel RDS(ON) vs. Input Voltage 750 750 700 700 120°C 650 100°C RDS(ON) (mΩ Ω) RDS(ON) (mΩ Ω) 650 600 550 85°C 500 450 25°C 400 120°C 600 550 500 85°C 450 400 25°C 350 350 300 300 2.5 3.0 3.5 4.0 4.5 5.0 5.5 2.5 6.0 3.0 Input Voltage (V) (300mA to 400mA; VIN = 3.6V; VOUT = 1.8V; C1 = 4.7μ μF) 300mA 1mA 1.90 1.85 Output Voltage (top) (V) Output Voltage (top) (V) IO 1.80 1.75 VO IO 400mA 300mA 0.4 0.3 IL 0.2 0.1 Time (50μs/div) Load Transient Response Load Transient Response (300mA to 400mA; VIN = 3.6V; VOUT = 1.8V; C1 = 10μ μF) (300mA to 400mA; VIN = 3.6V; VOUT = 1.8V; C1 = 10μ μF; C4 = 100pF) 0.4 0.3 IL 0.2 0.1 Time (50μs/div) 2512.2006.06.1.4 1.825 Output Voltage (top) (V) 300mA 1.850 1.800 1.775 VO IO 400mA 300mA 0.4 0.3 IL 0.2 0.1 Load and Inductor Current (200mA/div) (bottom) 400mA Load and Inductor Current (200mA/div) (bottom) VO IO 6.0 Time (50μs/div) 1.90 1.75 5.5 Load and Inductor Current (200mA/div) (bottom) VO 0 1.80 5.0 Load Transient Response IL 1.85 4.5 Load Transient Response 1.8 1.7 4.0 Input Voltage (V) Load and Inductor Current (200mA/div) (bottom) 1.9 3.5 (1mA to 300mA; VIN = 3.6V; VOUT = 1.8V; C1 = 10μ μF; CFF = 100pF) 2.0 Output Voltage (top) (V) 100°C Time (50μs/div) 7 AAT2512 Dual 400mA High Frequency Buck Converter Typical Characteristics EN1 = VIN; EN2 = GND. Output Ripple (VIN = 3.6V; VOUT = 1.8V; IOUT = 1mA) 6.0 5.5 1.80 5.0 1.79 4.5 1.78 4.0 1.77 3.5 1.76 3.0 Time (25μ μs/div) 40 20 0.30 0.25 VO 0 0.20 -20 0.15 -40 0.10 -60 -80 0.05 IL 0.00 -100 -0.05 -120 -0.10 Inductor Current (bottom) (A) 1.81 Input Voltage (bottom) (V) Output Voltage (top) (V) 1.82 Output Voltage (AC coupled) (top) (mV) Line Response (VOUT = 1.8V @ 400mA) Time (10µs/div) Output Ripple 0.9 40 20 0.8 VO 0 0.7 -20 0.6 -40 0.5 -60 0.4 0.3 -80 -100 IL Inductor Current (bottom) (A) Output Voltage (AC coupled) (top) (mV) (VIN = 3.6V; VOUT = 1.8V; IOUT = 400mA) 0.2 0.1 -120 Time (500ns/div) 8 2512.2006.06.1.4 AAT2512 Dual 400mA High Frequency Buck Converter Functional Block Diagram FB1 VIN1 DH Comp. Err. Amp. LX1 Logic Voltage Reference SGND1 DL Control Logic EN1 GND1 VIN2 See Note FB2 DH Comp. Err. Amp. LX2 Logic Voltage Reference Control Logic EN2 DL GND2 See Note SGND2 Note: Internal resistor divider included for ≥1.2V versions. For low voltage versions, the feedback pin is tied directly to the error amplifier input. Functional Description The AAT2512 is a high performance power management IC comprised of two buck converters. Each channel has independent input voltages and enable/disable pins. Designed to operate at 1.4MHz of switching frequency, the converters require only three external components (CIN, COUT, and LX), minimizing cost and size of external components. The AAT2512 also features soft-start control to limit inrush current. Soft start increases the inductor current limit point in discrete steps when power is applied to the input or when the enable pins are pulled high. It limits the current surge seen at the input and eliminates output voltage overshoot. The enable input, when pulled low, forces the converter into a low power, non-switching state consuming less than 1µA of current. Both converters are designed to operate with an input voltage range of 2.7V to 5.5V. Typical values of the output filter are 4.7µH and 4.7µF ceramic capacitor. The output voltage operates to as low as 0.6V and is offered as both fixed and adjustable. Power devices are sized for 400mA current capability while maintaining over 90% efficiency at full load. Light load efficiency is maintained at greater than 80% down to 500µA of load current. Both channels have excellent transient response, load, and line regulation. Transient response time is typically less than 20µs. For overload conditions, the peak input current is limited. As load impedance decreases and the output voltage falls closer to zero, more power is dissipated internally, raising the device temperature. Thermal protection completely disables switching when internal dissipation becomes excessive, protecting the device from damage. The junction overtemperature threshold is 140°C with 15°C of hysteresis. The under-voltage lockout guarantees sufficient VIN bias and proper operation of all internal circuits prior to activation. 2512.2006.06.1.4 9 AAT2512 Dual 400mA High Frequency Buck Converter Applications Information Inductor Selection The step-down converter uses peak current mode control with slope compensation to maintain stability for duty cycles greater than 50%. The output inductor value must be selected so the inductor current down slope meets the internal slope compensation requirements. The internal slope compensation for the adjustable and low-voltage fixed versions of the AAT2512 is 0.24A/µsec. This equates to a slope compensation that is 75% of the inductor current down slope for a 1.5V output and 4.7µH inductor. m= 0.75 ⋅ VO 0.75 ⋅ 1.5V A = = 0.24 L 4.7μH μsec The 4.7µH CDRH3D16 series inductor selected from Sumida has a 105mΩ DCR and a 900mA DC current rating. At full load, the inductor DC loss is 17mW which gives a 2.8% loss in efficiency for a 400mA 1.5V output. Input Capacitor This is the internal slope compensation for the adjustable (0.6V) version or low-voltage fixed version. When externally programming the 0.6V version to a 2.5V output, the calculated inductance would be 7.5µH. L= Manufacturer's specifications list both the inductor DC current rating, which is a thermal limitation, and the peak current rating, which is determined by the saturation characteristics. The inductor should not show any appreciable saturation under normal load conditions. Some inductors may meet the peak and average current ratings yet result in excessive losses due to a high DCR. Always consider the losses associated with the DCR and its effect on the total converter efficiency when selecting an inductor. Select a 4.7µF to 10µF X7R or X5R ceramic capacitor for the input. To estimate the required input capacitor size, determine the acceptable input ripple level (VPP) and solve for C. The calculated value varies with input voltage and is a maximum when VIN is double the output voltage. 0.75V 0.75 ⋅ VO μsec ≈ 3 A ⋅ VO = m 0.24A /μsec CIN = μsec =3 ⋅ 2.5V = 7.5μH A In this case, a standard 6.8µH value is selected. For high-voltage fixed versions (2.5V and above), m = 0.48A/µsec. Table 1 displays inductor values for the AAT2512 fixed and adjustable options. Configuration 0.6V Adjustable With External Feedback Fixed Output V ⎞ VO ⎛ ⋅ 1- O VIN ⎝ VIN ⎠ ⎛ VPP ⎞ - ESR ⋅ FS ⎝ IO ⎠ This equation provides an estimate for the input capacitor required for a single channel. Output Voltage Inductor 1V, 1.2V 2.2µH 1.5V, 1.8V 4.7µH 2.5V, 3.3V 6.8µH 0.6V to 3.3V 4.7µH Table 1: Inductor Values. 10 2512.2006.06.1.4 AAT2512 Dual 400mA High Frequency Buck Converter The equation below solves for input capacitor size for both channels. It makes the worst-case assumptions that both converters are operating at 50% duty cycle and are synchronized. 1 CIN = ⎛ VPP ⎞ - ESR · 4 · FS ⎝ IO1 + IO2 ⎠ Because the AAT2512 channels will generally operate at different duty cycles and are not synchronized, the actual ripple will vary and be less than the ripple (VPP) used to solve for the input capacitor in the equation above. Always examine the ceramic capacitor DC voltage coefficient characteristics when selecting the proper value. For example, the capacitance of a 10µF 6.3V X5R ceramic capacitor with 5V DC applied is actually about 6µF. The maximum input capacitor RMS current is: IRMS = IO1 · ⎛ ⎝ VO1 ⎛ V ⎞ · 1 - O1 ⎞ + IO2 · ⎛ VIN ⎝ VIN ⎠ ⎠ ⎝ VO2 ⎛ V ⎞ · 1 - O2 ⎞ VIN ⎝ VIN ⎠ ⎠ The input capacitor RMS ripple current varies with the input and output voltage and will always be less than or equal to half of the total DC load current of both converters combined. I +I IRMS(MAX) = O1(MAX) O2(MAX) 2 This equation also makes the worst-case assumption that both converters are operating at 50% duty cycle and are synchronized. Since the converters are not synchronized and are not both operating at 50% duty cycle, the actual RMS current will always be less than this. Losses associated with the input ceramic capacitor are typically minimal. VO ⎛ VO ⎞ The term VIN · ⎝1 - VIN ⎠ appears in both the input voltage ripple and input capacitor RMS current equations. It is a maximum when VO is twice VIN. This is why the input voltage ripple and the input capacitor RMS current ripple are a maximum at 50% duty cycle. 2512.2006.06.1.4 The input capacitor provides a low impedance loop for the edges of pulsed current drawn by the AAT2512. Low ESR/ESL X7R and X5R ceramic capacitors are ideal for this function. To minimize the stray inductance, the capacitor should be placed as closely as possible to the IC. This keeps the high frequency content of the input current localized, minimizing EMI and input voltage ripple. The proper placement of the input capacitor (C3 and C8) can be seen in the evaluation board layout in Figure 4. Since decoupling must be as close to the input pins as possible, it is necessary to use two decoupling capacitors. C3 provides the bulk capacitance required for both converters, while C8 is a high frequency bypass capacitor for the second channel (see C3 and C8 placement in Figure 4). A laboratory test set-up typically consists of two long wires running from the bench power supply to the evaluation board input voltage pins. The inductance of these wires, along with the low ESR ceramic input capacitor, can create a high Q network that may affect converter performance. This problem often becomes apparent in the form of excessive ringing in the output voltage during load transients. Errors in the loop phase and gain measurements can also result. Since the inductance of a short printed circuit board trace feeding the input voltage is significantly lower than the power leads from the bench power supply, most applications do not exhibit this problem. In applications where the input power source lead inductance cannot be reduced to a level that does not affect converter performance, a high ESR tantalum or aluminum electrolytic capacitor should be placed in parallel with the low ESR, ESL bypass ceramic capacitor. This dampens the high Q network and stabilizes the system. Output Capacitor The output capacitor limits the output ripple and provides holdup during large load transitions. A 4.7µF to 10µF X5R or X7R ceramic capacitor typically provides sufficient bulk capacitance to stabilize the output during large load transitions and has the ESR and ESL characteristics necessary for low output ripple. 11 AAT2512 Dual 400mA High Frequency Buck Converter The output voltage droop due to a load transient is dominated by the capacitance of the ceramic output capacitor. During a step increase in load current the ceramic output capacitor alone supplies the load current until the loop responds. As the loop responds, the inductor current increases to match the load current demand. This typically takes two to three switching cycles and can be estimated by: COUT = 3 · ΔILOAD VDROOP · FS Once the average inductor current increases to the DC load level, the output voltage recovers. The above equation establishes a limit on the minimum value for the output capacitor with respect to load transients. The internal voltage loop compensation also limits the minimum output capacitor value to 4.7µF. This is due to its effect on the loop crossover frequency (bandwidth), phase margin, and gain margin. Increased output capacitance will reduce the crossover frequency with greater phase margin. The maximum output capacitor RMS ripple current is given by: IRMS(MAX) = 1 2· 3 · VOUT · (VIN(MAX) - VOUT) L · F · VIN(MAX) Dissipation due to the RMS current in the ceramic output capacitor ESR is typically minimal, resulting in less than a few degrees rise in hot spot temperature. Adjustable Output Resistor Selection For applications requiring an adjustable output voltage, the 0.6V version can be programmed externally. Resistors R1 through R4 of Figure 2 program the output to regulate at a voltage higher than 0.6V. 12 To limit the bias current required for the external feedback resistor string, the minimum suggested value for R2 and R4 is 59kΩ. Although a larger value will reduce the quiescent current, it will also increase the impedance of the feedback node, making it more sensitive to external noise and interference. Table 2 summarizes the resistor values for various output voltages with R2 and R4 set to either 59kΩ for good noise immunity or 221kΩ for reduced no load input current. ⎛ VOUT ⎞ ⎛ 1.5V ⎞ R1 = V -1 · R2 = 0.6V - 1 · 59kΩ = 88.5kΩ ⎝ REF ⎠ ⎝ ⎠ The adjustable version of the AAT2512 in combination with an external feedforward capacitor (C4 and C5 of Figure 2) delivers enhanced transient response for extreme pulsed load applications. The addition of the feedforward capacitor typically requires a larger output capacitor (C1 and C2) for stability. Ω R2, R4 = 59kΩ Ω R2, R4 = 221kΩ VOUT (V) Ω) R1, R3 (kΩ R1, R3 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.8 1.85 2.0 2.5 3.3 19.6 29.4 39.2 49.9 59.0 68.1 78.7 88.7 118 124 137 187 267 75K 113K 150K 187K 221K 261K 301K 332K 442K 464K 523K 715K 1.00M Table 2: Adjustable Resistor Values For Use With 0.6V Version. 2512.2006.06.1.4 AAT2512 Dual 400mA High Frequency Buck Converter Thermal Calculations There are three types of losses associated with the AAT2512 converter: switching losses, conduction losses, and quiescent current losses. Conduction losses are associated with the RDS(ON) characteristics of the power output switching devices. Switching losses are dominated by the gate charge of the power output switching devices. At full load, assuming continuous conduction mode (CCM), a simplified form of the dual converter losses is given by: Given the total losses, the maximum junction temperature can be derived from the θJA for the TDFN33-12 package which is 50°C/W. TJ(MAX) = PTOTAL · ΘJA + TAMB PCB Layout The following guidelines should be used to insure a proper layout. PTOTAL = + IO12 · (RDSON(HS) · VO1 + RDSON(LS) · [VIN -VO1]) VIN IO22 · (RDSON(HS) · VO2 + RDSON(LS) · [VIN -VO2]) VIN + (tsw · F · [IO1 + IO2] + 2 · IQ) · VIN IQ is the AAT2512 quiescent current for one channel and tsw is used to estimate the full load switching losses. For the condition where channel one is in dropout at 100% duty cycle, the total device dissipation reduces to: PTOTAL = IO12 · RDSON(HS) + IO22 · (RDSON(HS) · VO2 + RDSON(LS) · [VIN -VO2]) VIN + (tsw · F · IO2 + 2 · IQ) · VIN Since RDS(ON), quiescent current, and switching losses all vary with input voltage, the total losses should be investigated over the complete input voltage range. 2512.2006.06.1.4 1. Due to the pin placement of VIN for both converters, proper decoupling is not possible with just one input capacitor. The large input capacitor C3 should connect as closely as possible to VP and GND, as shown in Figure 4. The additional input bypass capacitor C8 is necessary for proper high frequency decoupling of the second converter. 2. The output capacitor and inductor should be connected as closely as possible. The connection of the inductor to the LX pin should also be as short as possible. 3. The feedback trace should be separate from any power trace and connect as closely as possible to the load point. Sensing along a high-current load trace will degrade DC load regulation. If external feedback resistors are used, they should be placed as closely as possible to the FB pin. This prevents noise from being coupled into the high impedance feedback node. 4. The resistance of the trace from the load return to GND should be kept to a minimum. This will help to minimize any error in DC regulation due to differences in the potential of the internal signal ground and the power ground. 5. For good thermal coupling, PCB vias are required from the pad for the TDFN paddle to the ground plane. The via diameter should be 0.3mm to 0.33mm and positioned on a 1.2 mm grid. 13 AAT2512 Dual 400mA High Frequency Buck Converter Design Example Specifications VO1 = 2.5V @ 400mA (adjustable using 0.6V version), pulsed load ΔILOAD = 300mA VO2 = 1.8V @ 400mA (adjustable using 0.6V version), pulsed load ΔILOAD = 300mA VIN = 2.7V to 4.2V (3.6V nominal) FS = 1.4 MHz TAMB = 85°C 2.5V VO1 Output Inductor L1 = 3 μsec μsec ⋅ VO1 = 3 ⋅ 2.5V = 7.5μH A A (see Table 1) For Sumida inductor CDRH3D16, 10µH, DCR = 210mΩ. ΔI1 = ⎛ 2.5V⎞ VO ⎛ V ⎞ 2.5V ⋅ 1 - O1 = ⋅ ⎝1 = 72.3mA VIN ⎠ 10μH ⋅ 1.4MHz 4.2V⎠ L1 ⋅ F ⎝ IPK1 = IO1 + ΔI1 = 0.4A + 0.036A = 0.44A 2 PL1 = IO12 ⋅ DCR = 0.4A2 ⋅ 210mΩ = 34mW 1.8V VO2 Output Inductor L2 = 3 μsec μsec ⋅ VO2 = 3 ⋅ 1.8V = 5.4μH (see Table 1) A A For Sumida inductor CDRH3D16, 4.7µH, DCR = 105mΩ. ΔI2 = ⎛ 1.8V ⎞ VO2 ⎛ V ⎞ 1.8V ⋅ 1 - O2 = ⋅ 1= 156mA VIN ⎠ 4.7μH ⋅ 1.4MHz ⎝ 4.2V⎠ L⋅F ⎝ IPK2 = IO2 + ΔI2 = 0.4A + 0.078A = 0.48A 2 PL2 = IO22 ⋅ DCR = 0.4A2 ⋅ 105mΩ = 17mW 14 2512.2006.06.1.4 AAT2512 Dual 400mA High Frequency Buck Converter 2.5V Output Capacitor COUT = 3 · ΔILOAD 3 · 0.3A = = 3.2μF VDROOP · FS 0.2V · 1.4MHz IRMS(MAX) = (VOUT) · (VIN(MAX) - VOUT) 1 2.5V · (4.2V - 2.5V) · = 21mArms = L · F · VIN(MAX) 2 · 3 10μH · 1.4MHz · 4.2V 2· 3 1 · Pesr = esr · IRMS2 = 5mΩ · (21mA)2 = 2.2μW 1.8V Output Capacitor COUT = 3 · ΔILOAD 3 · 0.3A = = 3.2μF 0.2V · 1.4MHz VDROOP · FS IRMS(MAX) = (VOUT) · (VIN(MAX) - VOUT) 1 1.8V · (4.2V - 1.8V) · = 45mArms = L · F · VIN(MAX) 2 · 3 4.7μH · 1.4MHz · 4.2V 2· 3 1 · Pesr = esr · IRMS2 = 5mΩ · (45mA)2 = 10μW Input Capacitor Input Ripple VPP = 25mV. CIN = 1 ⎛ VPP ⎞ - ESR · 4 · FS ⎝ IO1 + IO2 ⎠ IRMS(MAX) = = 1 = 6.8μF ⎛ 25mV ⎞ - 5mΩ · 4 · 1.4MHz ⎝ 0.8A ⎠ IO1 + IO2 = 0.4Arms 2 P = esr · IRMS2 = 5mΩ · (0.4A)2 = 0.8mW 2512.2006.06.1.4 15 AAT2512 Dual 400mA High Frequency Buck Converter AAT2512 Losses The maximum dissipation occurs at dropout where VIN = 2.7V. All values assume an ambient temperature of 85°C and a junction temperature of 120°C. PTOTAL = IO12 · (RDSON(HS) · VO1 + RDSON(LS) · (VIN -VO1)) + IO22 · (RDSON(HS) · VO2 + RDSON(LS) · (VIN -VO2)) VIN + (tsw · F · IO2 + 2 · IQ) · VIN = 0.42 · (0.725Ω · 2.5V + 0.7Ω · (2.7V - 2.5V)) + 0.42 · (0.725Ω · 1.8V + 0.7Ω · (2.7V - 1.8V)) 2.7V + 5ns · 1.4MHz · 0.4A + 60μA) · 2.7V = 239mW TJ(MAX) = TAMB + ΘJA · PLOSS = 85°C + (50°C/W) · 239mW = 97°C Output 1 Enable VIN 1 2 3 R1 see Table 3 C41 U1 AAT2512 1 2 1 C5 3 R3 see Table 3 4 5 6 R4 59.0k EN1 VIN1 FB1 LX1 SGND1 GND1 EN2 VIN2 FB2 LX2 SGND2 GND2 LX1 12 L1 see Table 3 11 VO1 C3 10 10μF 9 8 LX2 VO2 L2 see Table 3 C11 4.7μF 7 R2 59.0k C8 C7 0.01μF C6 0.01μF C21 4.7μF 0.1μF GND GND 3 2 1 Output 2 Enable Figure 3: AAT2512 Evaluation Board Schematic. 1. For enhanced transient configuration C5, C4 = 100pF and C1, C2 = 10µF. 16 2512.2006.06.1.4 AAT2512 Dual 400mA High Frequency Buck Converter Adjustable Version (0.6V device) Ω R2, R4 = 59kΩ Ω1 R2, R4 = 221kΩ VOUT (V) Ω) R1, R3 (kΩ Ω) R1, R3 (kΩ L1, L2 (µH) 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.8 1.85 2.0 2.5 3.3 19.6 29.4 39.2 49.9 59.0 68.1 78.7 88.7 118 124 137 187 267 75.0 113 150 187 221 261 301 332 442 464 523 715 1000 2.2 2.2 2.2 2.2 2.2 2.2 4.7 4.7 4.7 4.7 6.8 6.8 6.8 Fixed Version R2, R4 Not Used VOUT (V) Ω) R1, R3 (kΩ L1, L2 (µH) 0.6-3.3V 0 4.7 Table 3: Evaluation Board Component Values. Figure 4: AAT2512 Evaluation Board Top Side. Figure 5: AAT2512 Evaluation Board Bottom Side. 1. For reduced quiescent current, R2 and R4 = 221kΩ. 2512.2006.06.1.4 17 AAT2512 Dual 400mA High Frequency Buck Converter Manufacturer Sumida Sumida Sumida MuRata MuRata Coilcraft Coiltronics Coiltronics Coiltronics Part Number Inductance (µH) Max DC Current (A) DCR Ω) (Ω Size (mm) LxWxH Type CDRH3D16-2R2 CDRH3D16-4R7 CDRH3D16-6R8 LQH2MCN4R7M02 LQH32CN4R7M23 LPO3310-472 SD3118-4R7 SD3118-6R8 SDRC10-4R7 2.2 4.7 6.8 4.7 4.7 4.7 4.7 6.8 4.7 1.20 0.90 0.73 0.40 0.45 0.80 0.98 0.82 1.30 0.072 0.105 0.170 0.80 0.20 0.27 0.122 0.175 0.122 3.8x3.8x1.8 3.8x3.8x1.8 3.8x3.8x1.8 2.0x1.6x0.95 2.5x3.2x2.0 3.2x3.2x1.0 3.1x3.1x1.85 3.1x3.1x1.85 5.7x4.4x1.0 Shielded Shielded Shielded Non-Shielded Non-Shielded 1mm Shielded Shielded 1mm Shielded Table 4: Typical Surface Mount Inductors. Manufacturer MuRata MuRata MuRata Part Number Value Voltage Temp. Co. Case GRM219R61A475KE19 GRM21BR60J106KE19 GRM21BR60J226ME39 4.7µF 10uF 22uF 10V 6.3V 6.3V X5R X5R X5R 0805 0805 0805 Table 5: Surface Mount Capacitors. 18 2512.2006.06.1.4 AAT2512 Dual 400mA High Frequency Buck Converter Ordering Information Voltage Package Channel 1 Channel 2 Marking1 Part Number (Tape and Reel)2 TDFN33-12 TDFN33-12 0.6V 1.8V 0.6V 1.6V QKXYY QYXYY AAT2512IWP-AA-T1 AAT2512IWP-IH-T1 All AnalogicTech products are offered in Pb-free packaging. The term “Pb-free” means semiconductor products that are in compliance with current RoHS standards, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. For more information, please visit our website at http://www.analogictech.com/pbfree. Legend Voltage Adjustable (0.6V) 0.9 1.2 1.5 1.8 1.9 2.5 2.6 2.7 2.8 2.85 2.9 3.0 3.3 4.2 Code A B E G I Y N O P Q R S T W C 1. XYY = assembly and date code. 2. Sample stock is generally held on part numbers listed in BOLD. 2512.2006.06.1.4 19 AAT2512 Dual 400mA High Frequency Buck Converter TDFN33-12 2.40 ± 0.05 Detail "B" 3.00 ± 0.05 Index Area (D/2 x E/2) 0.3 ± 0.10 0.16 0.375 ± 0.125 0.075 ± 0.075 3.00 ± 0.05 1.70 ± 0.05 Top View Bottom View Pin 1 Indicator (optional) 0.23 ± 0.05 Detail "A" 0.45 ± 0.05 0.1 REF 0.05 ± 0.05 0.229 ± 0.051 + 0.05 0.8 -0.20 7.5° ± 7.5° Option A: C0.30 (4x) max Chamfered corner Side View Option B: R0.30 (4x) max Round corner Detail "B" Detail "A" All dimensions in millimeters. © Advanced Analogic Technologies, Inc. AnalogicTech cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in an AnalogicTech product. No circuit patent licenses, copyrights, mask work rights, or other intellectual property rights are implied. AnalogicTech reserves the right to make changes to their products or specifications or to discontinue any product or service without notice. Customers are advised to obtain the latest version of relevant information to verify, before placing orders, that information being relied on is current and complete. All products are sold subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those pertaining to warranty, patent infringement, and limitation of liability. AnalogicTech warrants performance of its semiconductor products to the specifications applicable at the time of sale in accordance with AnalogicTech’s standard warranty. Testing and other quality control techniques are utilized to the extent AnalogicTech deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily performed. AnalogicTech and the AnalogicTech logo are trademarks of Advanced Analogic Technologies Incorporated. All other brand and product names appearing in this document are registered trademarks or trademarks of their respective holders. Advanced Analogic Technologies, Inc. 830 E. Arques Avenue, Sunnyvale, CA 94085 Phone (408) 737-4600 Fax (408) 737-4611 20 2512.2006.06.1.4