AOSMD AOZ1010

AOZ1010
EZBuck™ 2A Simple Regulator
General Description
Features
The AOZ1010 is a high efficiency, simple to use, 2A buck
regulator. The AOZ1010 works from a 4.5V to 16V input
voltage range, and provides up to 2A of continuous
output current with an output voltage adjustable down to
0.8V.
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The AOZ1010 comes in an SO-8 package and is rated
over a -40°C to +85°C ambient temperature range.
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4.5V to 16V operating input voltage range
130mΩ internal PFET switch for high efficiency:
up to 95%
Internal Schottky diode
Internal soft start
Output voltage adjustable to 0.8V
2A continuous output current
Fixed 500kHz PWM operation
Cycle-by-cycle current limit
Short-circuit protection
Thermal shutdown
Small size SO-8 package
Applications
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Point of load DC/DC conversion
PCIe graphics cards
Set top boxes
DVD drives and HDD
LCD panels
Cable modems
Telecom/networking/datacom equipment
Typical Application
Cd
C1
22µF Ceramic
VIN
L1
3.3µH
EN
VOUT
AOZ1010
LX
R1
COMP
RC
CC
C2, C3
22µF Ceramic
FB
AGND
PGND
R2
Figure 1. 3.3V/2A Buck Regulator
Rev. 1.0 November 2006
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Page 1 of 14
AOZ1010
Ordering Information
Part Number
Ambient Temperature Range
Package
Environmental
AOZ1010AI
-40°C to +85°C
SO-8
RoHS
S nt
RoH plia
Com
All AOS Products are offering in packaging with Pb-free plating and compliant to RoHS standards.
Please visit wwww.aosmd.com/web/rohs_compliant.jsp for additional information.
Pin Configuration
PGND
1
8
LX
VIN
2
7
LX
AGND
3
6
EN
FB
4
5
COMP
SO-8
(Top View)
Pin Description
Pin Number
Pin Name
Pin Function
1
PGND
2
VIN
3
AGND
Reference connection for controller section. Also used as thermal connection for controller
section. Electrically needs to be connected to PGND.
4
FB
The FB pin is used to determine the output voltage via a resistor divider between the output
and GND.
5
COMP
6
EN
The enable pin is active high. Connect EN pin to VIN if not used. Do not leave the EN pin floating.
7, 8
LX
PWM output connection to inductor. Thermal connection for output stage.
Power ground. Electrically needs to be connected to AGND.
Supply voltage input. When VIN rises above the UVLO threshold the device starts up.
External loop compensation pin.
Block Diagram
VIN
UVLO
& POR
EN
Internal
+5V
5V LDO
Regulator
OTP
+
ISen
–
Reference
& Bias
Softstart
Q1
ILimit
+
0.8V
+
EAmp
FB
–
–
PWM
Comp
PWM
Control
Logic
+
Level
Shifter
+
FET
Driver
LX
LX
D1
COMP
500kHz
Oscillator
AGND
Rev. 1.0 November 2006
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PGND
Page 2 of 14
11
µ
2
AOZ1010
Absolute Maximum Ratings
Recommend Operating Ratings
Exceeding the Absolute Maximum ratings may damage the
device.
The device is not guaranteed to operate beyond the Maximum
Operating Ratings.
Parameter
Parameter
Rating
Rating
Supply Voltage (VIN)
18V
Supply Voltage (VIN)
4.5V to 16V
LX to AGND
-0.7V to VIN+0.3V
Output Voltage Range
0.8V to VIN
EN to AGND
-0.3V to VIN+0.3V
Ambient Temperature (TA)
-40°C to +85°C
FB to AGND
-0.3V to 6V
87°C/W
COMP to AGND
-0.3V to 6V
Package Thermal Resistance SO-8
(ΘJA)(1)
PGND to AGND
-0.3V to +0.3V
Junction Temperature (TJ)
+150°C
Storage Temperature (TS)
-65°C to +150°C
Note:
1. The value of ΘJA is measured with the device mounted on 1-in2 FR-4
board with 2oz. Copper, in a still air environment with TA = 25°C. The
value in any given application depends on the user's specific board
design.
Electrical Characteristics
TA = 25°C, VIN = VEN = 12V, VOUT = 3.3V unless otherwise specified(2)
Symbol
VIN
VUVLO
Parameter
Conditions
Supply Voltage
Input Under-Voltage Lockout Threshold
Min.
Typ.
4.5
VIN Rising
VIN Falling
Max.
Units
16
V
4.00
3.70
V
Supply Current (Quiescent)
IOUT = 0, VFB = 1.2V, VEN > 1.2V
2
3
mA
IOFF
Shutdown Supply Current
VEN = 0V
3
20
µA
VFB
Feedback Voltage
0.8
0.818
IIN
V
0.5
%
Line Regulation
1
%
IFB
Feedback Voltage Input Current
VEN
EN Input Threshold
VHYS
0.782
Load Regulation
200
Off Threshold
On Threshold
0.6
2.0
EN Input Hysteresis
100
nA
V
mV
MODULATOR
fO
Frequency
350
DMAX
Maximum Duty Cycle
100
DMIN
Minimum Duty Cycle
500
600
kHz
%
6
%
Error Amplifier Voltage Gain
500
V/ V
Error Amplifier Transconductance
200
µA / V
PROTECTION
ILIM
Current Limit
Over-Temperature Shutdown Limit
tSS
2.5
TJ Rising
TJ Falling
Soft Start Interval
3.6
A
145
100
°C
4
ms
OUTPUT STAGE
High-Side Switch On-Resistance
VIN = 12V
VIN = 5V
97
166
130
200
mΩ
Note:
2. Specification in BOLD indicate an ambient temperature range of -40°C to +85°C. These specifications are guaranteed by design.
Rev. 1.0 November 2006
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Page 3 of 14
AOZ1010
Typical Performance Characteristics
Circuit of Figure 1. TA = 25°C, VIN = VEN = 12V, VOUT = 3.3V unless otherwise specified.
Light Load (DCM) Operation
Full Load (CCM) Operation
Vin
ripple
Vin
ripple
0.1V/div
0.1V/div
Vo
ripple
Vo
ripple
20mV/div
20mV/div
IL
1A/div
IL
1A/div
VLX
10V/div
VLX
10V/div
1µs/div
1µs/div
Startup to Full Load
Full Load to Turnoff
Vin
5V/div
Vin
5V/div
Vo
1V/div
Vo
1V/div
Iin
0.5A/div
Iin
0.5A/div
1ms/div
1ms/div
50% to 100% Load Transient
Light Load to Turnoff
Vo
Ripple
Vin
5V/div
50mV/div
Vo
1V/div
Io
1A/div
100µs/div
Rev. 1.0 November 2006
Iin
0.5A/div
1s/div
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Page 4 of 14
AOZ1010
Typical Performance Characteristics (Continued)
Circuit of Figure 1. TA = 25°C, VIN = VEN = 12V, VOUT = 3.3V unless otherwise specified.
Short Circuit Protection
Short Circuit Recovery
Vo
2V/div
Vo
2V/div
IL
1A/div
IL
1A/div
100µs/div
1ms/div
AOZ1010AI Efficiency
Efficiency (VIN = 12V) vs. Load Current
100
8.0V OUTPUT
Efficieny (%)
95
5.0V OUTPUT
90
3.3V OUTPUT
85
80
75
0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
Load Current (A)
Thermal de-rating curves for SO-8 package part under typical input and output condition based on the evaluation board.
25°C ambient temperature and natural convection (air speed < 50LFM) unless otherwise specified.
Derating Curve at 5V Input
Derating Curve at 12V Input
2.5
2.5
Output Current (IO)
Output Current (IO)
3.3V, 5.0V OUTPUT
2.0
1.8V OUTPUT
1.5
1.0
0.5
0
25
35
45
55
65
75
85
5.0V OUTPUT
1.5
3.3V OUTPUT
1.8V OUTPUT
1.0
0.5
0
25
Ambient Temperature (TA)
Rev. 1.0 November 2006
8.0V OUTPUT
2.0
35
45
55
65
75
85
Ambient Temperature (TA)
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Page 5 of 14
AOZ1010
Detailed Description
The AOZ1010 is a current-mode step down regulator
with integrated high side PMOS switch and a low side
freewheeling Schottky diode. It operates from a 4.5V to
16V input voltage range and supplies up to 2A of load
current. The duty cycle can be adjusted from 6% to 100%
allowing a wide range of output voltage. Features include
enable control, Power-On Reset, input under voltage
lockout, fixed internal soft-start and thermal shut down.
The AOZ1010 uses a P-Channel MOSFET as the high
side switch. It saves the bootstrap capacitor normally
seen in a circuit which is using an NMOS switch. It allows
100% turn-on of the upper switch to achieve linear regulation mode of operation. The minimum voltage drop from
VIN to VO is the load current times DC resistance of
MOSFET plus DC resistance of buck inductor. It can be
calculated by equation below:
The AOZ1010 is available in SO-8 package.
V O _MAX = V IN – I O × ( R DS ( ON ) + R inductor )
Enable and Soft Start
where;
The AOZ1010 has internal soft start feature to limit
in-rush current and ensure the output voltage ramps up
smoothly to regulation voltage. A soft start process
begins when the input voltage rises to 4.0V and voltage
on EN pin is HIGH. In soft start process, the output
voltage is ramped to regulation voltage in typically 4ms.
The 4ms soft start time is set internally.
The EN pin of the AOZ1010 is active high. Connect the
EN pin to VIN if enable function is not used. Pulling EN to
ground will disable the AOZ1010. Do not leave it open.
The voltage on EN pin must be above 2.0V to enable the
AOZ1010. When voltage on EN pin falls below 0.6V, the
AOZ1010 is disabled. If an application circuit requires the
AOZ1010 to be disabled, an open drain or open collector
circuit should be used to interface to EN pin.
Steady-State Operation
Under steady-state conditions, the converter operates
in fixed frequency and Continuous-Conduction Mode
(CCM).
The AOZ1010 integrates an internal P-MOSFET as the
high-side switch. Inductor current is sensed by amplifying
the voltage drop across the drain to source of the high
side power MOSFET. Output voltage is divided down by
the external voltage divider at the FB pin. The difference
of the FB pin voltage and reference is amplified by the
internal transconductance error amplifier. The error voltage, which shows on the COMP pin, is compared against
the current signal, which is sum of inductor current signal
and ramp compensation signal, at PWM comparator
input. If the current signal is less than the error voltage,
the internal high-side switch is on. The inductor current
flows from the input through the inductor to the output.
When the current signal exceeds the error voltage, the
high-side switch is off. The inductor current is freewheeling through the internal Schottky diode to output.
Rev. 1.0 November 2006
VO_MAX is the maximum output voltage,
VIN is the input voltage from 4.5V to 16V,
IO is the output current from 0A to 2A,
RDS(ON) is the on resistance of internal MOSFET, the value is
between 97mΩ and 200mΩ depending on input voltage and
junction temperature, and
Rinductor is the inductor DC resistance.
Switching Frequency
The AOZ1010 switching frequency is fixed and set by an
internal oscillator. The actual switching frequency could
range from 350kHz to 600kHz due to device variation.
Output Voltage Programming
Output voltage can be set by feeding back the output
to the FB pin with a resistor divider network. In the
application circuit shown in Figure 1. The resistor divider
network includes R2 and R3. Usually, a design is started
by picking a fixed R3 value and calculating the required
R2 with equation below:
R 

V O = 0.8 ×  1 + ------1-
R 2

Some standard values of R1 and R2 for the most commonly used output voltage values are listed in Table 1.
Table 1.
VO (V)
R1 (kΩ)
R2 (kΩ)
0.8
1.0
1.2
4.99
10
1.5
10
11.5
1.8
12.7
10.2
2.5
21.5
10
3.3
31.6
10
5.0
52.3
10
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Open
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AOZ1010
The combination of R1 and R2 should be large enough to
avoid drawing excessive current from the output, which
will cause power loss.
Thermal Protection
Since the switch duty cycle can be as high as 100%, the
maximum output voltage can be set as high as the input
voltage minus the voltage drop on upper PMOS and
inductor.
An internal temperature sensor monitors the junction
temperature. It shuts down the internal control circuit and
high side PMOS if the junction temperature exceeds
145°C. The regulator will restart automatically under the
control of soft-start circuit when the junction temperature
decreases to 100°C.
Protection Features
Application Information
The AOZ1010 has multiple protection features to prevent
system circuit damage under abnormal conditions.
The basic AOZ1010 application circuit is shown in
Figure 1. Component selection is explained below.
Over Current Protection (OCP)
The sensed inductor current signal is also used for
over current protection. Since the AOZ1010 employs
peak current mode control, the COMP pin voltage is
proportional to the peak inductor current. The COMP pin
voltage is limited to be between 0.4V and 2.5V internally.
The peak inductor current is automatically limited cycle
by cycle.
The cycle by cycle current limit threshold is set between
2.5A and 3.6A. When the load current reaches the current limit threshold, the cycle by cycle current limit circuit
turns off the high side switch immediately to terminate
the current duty cycle. The inductor current stops rising.
The cycle by cycle current limit protection directly limits
inductor peak current. The average inductor current is
also limited due to the limitation on peak inductor current.
When the cycle by cycle current limit circuit is triggered,
the output voltage drops as the duty cycle is decreasing.
The AOZ1010 has internal short circuit protection to
protect itself from catastrophic failure under output short
circuit conditions. The FB pin voltage is proportional to
the output voltage. Whenever FB pin voltage is below
0.2V, the short circuit protection circuit is triggered.
As a result, the converter is shut down and hiccups at a
frequency equal to 1/8 of normal switching frequency.
The converter will start up via a soft start once the short
circuit condition disappears. In short circuit protection
mode, the inductor average current is greatly reduced
because of the low hiccup frequency.
Power-On Reset (POR)
A power-on reset circuit monitors the input voltage.
When the input voltage exceeds 4V, the converter starts
operation. When input voltage falls below 3.7V, the
converter will stop switching.
Rev. 1.0 November 2006
Input Capacitor
The input capacitor (C1 in Figure 1), must be connected
to the VIN pin and PGND pin of the AOZ1010 to maintain
steady input voltage and filter out the pulsing input
current. A small decoupling capacitor (Cd in Figure 1),
usually 1µF, should be connected to the VIN pin and
AGND pin for stable operation of the AOZ1010. The
voltage rating of input capacitor must be greater than
maximum input voltage plus ripple voltage.
The input ripple voltage can be approximated by the
equation below:
IO
VO VO

∆V IN = ------------------ ×  1 – ---------- × ---------f × C IN 
V IN  V IN
Since the input current is discontinuous in a buck
converter, the current stress on the input capacitor is
another concern when selecting the capacitor. For a
buck circuit, the RMS value of input capacitor current can
be calculated by:
VO 
VO
I CIN _RMS = I O × ---------  1 – ---------
V IN 
V IN 
If let m equal the conversion ratio:
VO
---------- = m
V IN
The relationship between the input capacitor RMS current and voltage conversion ratio is calculated and shown
in Figure 2 on the next page. It can be seen that when VO
is half of VIN, CIN is under the worst current stress. The
worst current stress on CIN is 0.5 x IO .
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Page 7 of 14
AOZ1010
frequency together decide the inductor ripple current,
which is:
0.5
VO 
VO
∆I L = ----------- ×  1 – ---------
f ×L 
V IN 
0.4
ICIN_RMS(m) 0.3
IO
0.2
The peak inductor current is:
∆I
I Lpeak = I O + --------L2
0.1
0
0
0.5
m
1
Figure 2. ICIN vs. Voltage Conversion Ratio
For reliable operation and best performance, the input
capacitors must have current rating higher than ICIN_RMS
at the worst operating conditions. Ceramic capacitors are
preferred for input capacitors because of their low
ESR and high ripple current rating. Depending on the
application circuits, other low ESR tantalum capacitors
or aluminum electrolytic capacitors may also be used.
When selecting ceramic capacitors, X5R or X7R type
dielectric ceramic capacitors are preferred for their better
temperature and voltage characteristics. Note that the
ripple current rating from capacitor manufacturers is
based on certain amount of life time. Further de-rating
may be necessary for practical design requirement.
Inductor
The inductor is used to supply constant current to the
output when it is driven by a switching voltage. For a
given input and output voltage, inductance and switching
High inductance gives low inductor ripple current but
requires a larger size inductor to avoid saturation. Low
ripple current reduces inductor core losses. Low ripple
current also reduces RMS current through the inductor
and switches, which results in less conduction loss.
When selecting the inductor, make sure it is able to
handle the peak current at the highest operating temperature without saturation.
The inductor takes the highest current in a buck circuit.
The conduction loss on the inductor needs to be checked
for thermal and efficiency requirements.
Surface mount inductors in different shape and styles are
available from Coilcraft, Elytone and Murata. Shielded
inductors are small and radiate less EMI noise, but they
cost more than unshielded inductors. The choice
depends on EMI requirement, price and size.
Table 2 lists some inductors for typical output voltage
design.
Table 2. Typical Inductors
VOUT
5.0V
3.3V
1.8V
L1
Unshielded, 4.7µH, LQH55DN4R7M03
Manufacture
MURATA
Shielded, 4.7µH, LQH66SN4R7M03
MURATA
Shielded, 5.8µH, ET553-5R8
ELYTONE
Unshielded, 6.7µH, DO3316P-682MLD
Coilcraft
Unshielded, 4.7µH, LQH55DN3R3M03
MURATA
Shielded, 4.7µH, LQH66SN3R3M03
MURATA
Shielded, 3.3µH, ET553-3R3
ELYTONE
Unshielded, 4.7µH, DO3316P-472MLD
Coilcraft
Unshielded, 4.7µH, DO1813P-472HC
Coilcraft
Unshielded, 2.2µH, LQH55DN1R5M03
MURATA
Shielded, 2.2µH, LQH66SN1R5M03
MURATA
Shielded, 2.2µH, ET553-2R2
ELYTONE
Unshielded, 2.2µH, DO3316P-222MLD
Coilcraft
Unshielded, 2.2µH, DO1813P-222HC
Coilcraft
Rev. 1.0 November 2006
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Page 8 of 14
AOZ1010
Output Capacitor
The output capacitor is selected based on the DC output
voltage rating, output ripple voltage specification, and
ripple current rating.
The selected output capacitor must have a higher rated
voltage specification than the maximum desired output
voltage including ripple. De-rating needs to be considered for long term reliability.
Output ripple voltage specification is another important
factor for selecting the output capacitor. In a buck
converter circuit, output ripple voltage is determined by
inductor value, switching frequency, output capacitor
value and ESR. It can be calculated by the equation
below:
1
∆V O = ∆I L ×  ES R CO + ---------------------------

8 × f × C O
Usually, the ripple current rating of the output capacitor
is a smaller issue because of the low current stress.
When the buck inductor is selected to be very small
and inductor ripple current is high, output capacitor could
be overstressed.
Loop Compensation
The AOZ1010 employs peak current mode control for
easy use and fast transient response. Peak current mode
control eliminates the double pole effect of the output
L&C filter. It greatly simplifies the compensation loop
design.
With peak current mode control, the buck power stage
can be simplified to be a one-pole and one-zero system
in frequency domain. The pole is dominant pole and can
be calculated by:
1
f P 1 = -----------------------------------2π × C O × R L
where,
CO is output capacitor value, and
The zero is a ESR zero due to output capacitor and its
ESR. It is can be calculated by:
ESRCO is the equivalent series resistance of the output
capacitor.
When low ESR ceramic capacitor is used as output
capacitor, the impedance of the capacitor at the switching
frequency dominates. Output ripple is mainly caused by
capacitor value and inductor ripple current. The output
ripple voltage calculation can be simplified to:
1
f Z 1 = -------------------------------------------------2π × C O × ESR CO
where;
CO is the output filter capacitor,
RL is load resistor value, and
1
∆V O = ∆I L × --------------------------8×f ×C
ESRCO is the equivalent series resistance of output capacitor.
O
∆V O = ∆I L × ES R CO
The compensation design is actually to shape the
converter close loop transfer function to get desired gain
and phase. Several different types of compensation
networks can be used for AOZ1010. For most cases, a
series capacitor and resistor network connected to the
COMP pin sets the pole-zero and is adequate for a stable
high-bandwidth control loop.
For lower output ripple voltage across the entire operating temperature range, X5R or X7R dielectric type of
ceramic, or other low ESR tantalum capacitor or aluminum electrolytic capacitor may also be used as output
capacitors.
In the AOZ1010, FB pin and COMP pin are the inverting
input and the output of internal transconductance error
amplifier. A series R and C compensation network
connected to COMP provides one pole and one zero.
The pole is:
In a buck converter, output capacitor current is continuous. The RMS current of output capacitor is decided
by the peak to peak inductor ripple current. It can be
calculated by:
G EA
f P 2 = ------------------------------------------2π × C C × G VEA
If the impedance of ESR at switching frequency
dominates, the output ripple voltage is mainly decided
by capacitor ESR and inductor ripple current. The output
ripple voltage calculation can be further simplified to:
∆I L
I CO _RMS = ---------12
where;
GEA is the error amplifier transconductance, which is 200 x 10-6
A/V,
GVEA is the error amplifier voltage gain, which is 500 V/V, and
CC is compensation capacitor.
Rev. 1.0 November 2006
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Page 9 of 14
AOZ1010
The zero given by the external compensation network,
capacitor CC (C5 in Figure 1), and resistor RC (R1 in
Figure 1), is located at:
1
f Z 2 = ------------------------------------2π × C C × R C
To design the compensation circuit, a target crossover
frequency fC for close loop must be selected. The system
crossover frequency is where control loop has unity gain.
The crossover frequency is also called the converter
bandwidth. Generally a higher bandwidth means faster
response to load transient. However, the bandwidth
should not be too high because of system stability
concerns. When designing the compensation loop,
converter stability under all line and load condition must
be considered.
Usually, it is recommended to set the bandwidth to be
less than 1/10 of switching frequency. The AOZ1010
operates at a fixed switching frequency range from
350kHz to 600kHz. The recommended crossover
frequency is less than 30kHz.
f C = 30kHz
The strategy for choosing RC and CC is to set the cross
over frequency with RC and set the compensator zero
with CC. Using selected crossover frequency, fC, to
calculate RC:
VO
2π × C O
R C = f C × ----------× ----------------------------G ×G
V
FB
EA
CS
where;
fC is the desired crossover frequency,
VFB is 0.8V,
GEA is the error amplifier transconductance, which is 200 x 10-6
A/V, and
GCS is the current sense circuit transconductance, which is
5.64 A/V.
The compensation capacitor CC and resistor RC together
make a zero. This zero is put somewhere close to the
dominate pole, fP1, but lower than 1/5 of the selected
crossover frequency. CC can is selected by:
1.5
C C = ------------------------------------2π × R C × f P 1
Rev. 1.0 November 2006
The previous equation can also be simplified to:
CO × RL
C C = ---------------------RC
An easy-to-use application software which helps to
design and simulate the compensation loop can be found
at www.aosmd.com.
Table 3 lists the values for a typical output voltage design
when output is 44µF ceramics capacitor.
Table 3.
VOUT
L1
RC
CC
1.8V
2.2µH
20kΩ
1.5nF
3.3V
3.3µH
31.6kΩ
1.0nF
5V
4.7µH
49.9kΩ
1.0nF
8V
10µH
80.6kΩ
0.82nF
Thermal Management and Layout
Consideration
In the AOZ1010 buck regulator circuit, high pulsing current flows through two circuit loops. The first loop starts
from the input capacitors, to the VIN pin, to the LX pins, to
the filter inductor, to the output capacitor and load, and
then returns to the input capacitor through ground.
Current flows in the first loop when the high side switch is
on. The second loop starts from the inductor, to the
output capacitors and load, to the PGND pin of the
AOZ1010, and to the LX pins of the AZO1010. Current
flows in the second loop when the low side diode is on.
In PCB layout, minimizing the two loops area reduces the
noise of this circuit and improves efficiency. A ground
plane is recommended to connect input capacitor, output
capacitor, and PGND pin of the AOZ1010.
In the AOZ1010 buck regulator circuit, the two major
power dissipating components are the AOZ1010 and
output inductor. The total power dissipation of converter
circuit can be measured by input power minus output
power.
P total _loss = V IN × I IN – V O × I O
The power dissipation of inductor can be approximately
calculated by output current and DCR of inductor.
P inductor _loss = IO2 × R inductor × 1.1
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Page 10 of 14
AOZ1010
The junction to ambient temperature can be got from
power dissipation in the AOZ1010 and thermal impedance from junction to ambient.
2. Input capacitor should be connected to the VIN pin
and the PGND pin as close as possible.
T (jun-amb) = ( P totalloss – P inductorloss ) × Θ JA
The maximum junction temperature of AOZ1010 is
145°C, which limits the maximum load current capability.
Please see the thermal de-rating curves for the maximum
load current of the AOZ1010 under different ambient
temperatures.
The thermal performance of the AOZ1010 is strongly
affected by the PCB layout. Extra care should be taken
by users during the design process to ensure that the IC
will operate under the recommended environmental
conditions.
Several layout tips are listed below for the best electric
and thermal performance. Figure 3 below illustrates a
single layer PCB layout example as a reference.
1. Do not use thermal relief connection to the VIN and
the PGND pin. Pour a maximized copper area to
the PGND pin and the VIN pin to help thermal
dissipation.
Cin
PGND
1
VIN
2
AGND
8
3. A ground plane is preferred. If a ground plane is not
used, separate PGND from AGND and connect them
only at one point to avoid the PGND pin noise
coupling to the AGND pin. In this case, a decoupling
capacitor should be connected between VIN pin and
AGND pin.
4. Make the current trace from LX pins to L to Co to the
PGND as short as possible.
5. Pour copper plane on all unused board area and
connect it to stable DC nodes, like VIN, GND, or VOUT.
6. The two LX pins are connected to the internal PFET
drain. They are low resistance thermal conduction
path and most noisy switching node. Connect a
copper plane to the LX pin to help thermal
dissipation. This copper plane should not be too
large otherwise switching noise may be coupled to
other parts of the circuit.
7. Keep sensitive signal traces such as trace connecting FB pin and COMP pin away from the LX pins.
LX
7
LX
3
6
EN
4
5
COMP
SO-8
Cd
R2
FB
Cout
L
R1
Cc
Rc
Figure 3. AOZ1010 PCB Layout
Rev. 1.0 November 2006
www.aosmd.com
Page 11 of 14
AOZ1010
Package Dimensions, SO-8L
D
Gauge Plane
Seating Plane
e
0.25
8
L
E
E1
h x 45°
1
C
θ
7° (4x)
A2 A
0.1
b
A1
Dimensions in millimeters
2.20
5.74
1.27
0.80
Unit: mm
Symbols
A
A1
A2
b
c
D
E1
e
E
h
L
θ
Min.
1.35
0.10
1.25
0.31
0.17
4.80
3.80
Nom.
1.65
—
1.50
—
—
4.90
3.90
1.27 BSC
5.80
6.00
0.25
—
0.40
—
0°
—
Max.
1.75
0.25
1.65
0.51
0.25
5.00
4.00
6.20
0.50
1.27
8°
Dimensions in inches
Symbols
A
A1
A2
b
c
D
E1
e
E
h
L
θ
Min.
0.053
0.004
0.049
0.012
0.007
0.189
0.150
Nom. Max.
0.065 0.069
—
0.010
0.059 0.065
—
0.020
—
0.010
0.193 0.197
0.154 0.157
0.050 BSC
0.228 0.236 0.244
0.010
—
0.020
0.016
—
0.050
0°
—
8°
Notes:
1. All dimensions are in millimeters.
2. Dimensions are inclusive of plating
3. Package body sizes exclude mold flash and gate burrs. Mold flash at the non-lead sides should be less than 6 mils.
4. Dimension L is measured in gauge plane.
5. Controlling dimension is millimeter, converted inch dimensions are not necessarily exact.
Rev. 1.0 November 2006
www.aosmd.com
Page 12 of 14
AOZ1010
Tape and Reel Dimensions
SO-8 Carrier Tape
P1
D1
See Note 3
P2
T
See Note 5
E1
E2
E
See Note 3
B0
K0
A0
D0
P0
Feeding Direction
Unit: mm
Package
SO-8
(12mm)
A0
6.40
±0.10
B0
5.20
±0.10
K0
2.10
±0.10
D0
1.60
±0.10
D1
1.50
±0.10
E
12.00
±0.10
SO-8 Reel
E1
1.75
±0.10
E2
5.50
±0.10
P0
8.00
±0.10
P2
2.00
±0.10
P1
4.00
±0.10
T
0.25
±0.10
W1
S
G
N
M
K
V
R
H
W
W
N
Tape Size Reel Size
M
12mm
ø330
ø330.00 ø97.00 13.00
±0.10 ±0.30
±0.50
W1
17.40
±1.00
K
H
10.60
ø13.00
+0.50/-0.20
S
2.00
±0.50
G
—
R
—
V
—
SO-8 Tape
Leader/Trailer
& Orientation
Trailer Tape
300mm min. or
75 empty pockets
Rev. 1.0 November 2006
Components Tape
Orientation in Pocket
www.aosmd.com
Leader Tape
500mm min. or
125 empty pockets
Page 13 of 14
AOZ1010
AOZ1010 Package Marking
Z1010AI
FAYWLT
Part Number Code
Assembly Lot Code
Fab & Assembly Location
Year & Week Code
Rev. 1.0 November 2006
www.aosmd.com
Page 14 of 14