VCA2617 SBOS326 −AUGUST 2005 Dual, Variable Gain Amplifier FEATURES DESCRIPTION D INDEPENDENT CHANNEL CONTROLS: The VCA2617 is a dual-channel, continuously variable, voltage-controlled gain amplifier well-suited for a variety of ultrasound systems as well as applications in proximity detectors and test equipment. The VCA2617 uses a true variable-gain amplifier architecture, achieving very good noise performance at low gains, while not sacrificing high gain distortion performance. − Gain Range: 48dB − Clamping Levels − Post-Gain: 0, +6dB D D D D D D D D LOW NOISE: 4.1nV//Hz LOW POWER: 52mW/Channel BANDWIDTH: 50MHz HARMONIC DISTORTION: −53dBc at 5MHz CROSSTALK: −61dB at 5MHz 5V SINGLE SUPPLY POWER-DOWN MODE Following a linear-in-dB response, the VCA2617 gain can be varied over a 48dB range with a 0.2V to 2.3V control voltage. Two separate high-impedance control inputs allow for a channel independent variation of the gains. Each channel of the VCA2617 can be configured to provide a gain range of −10dB to 38dB, or −16dB to 32dB, depending on the gain select pin (HG). This post-gain feature allows the user to optimize the output swing of VCA2617 for a variety of high-speed analog-to-digital converters (ADC). As a means to improve system overload recovery time, the VCA2617 also provides an internal clamping circuitry where an externally applied voltage sets the desired output clamping level. SMALL QFN-32 PACKAGE (5x5mm) APPLICATIONS D MEDICAL AND INDUSTRIAL ULTRASOUND D D SYSTEMS − Suitable for 10-Bit and 12-Bit Systems TEST EQUIPMENT SONAR The VCA2617 operates on a single +5V supply while consuming only 52mW per channel. It is available in a small QFN-32 package (5x5mm). RELATED PRODUCTS V CNTLA PART NUMBER DESCRIPTION VCA2615 Dual, Low-Noise, Variable-Gain Amplifier with Preamp V CLMPA Gain Select A C EXTA Out A+ In A+ V−I Clamping Circuitry I−V Out A− In A− Out B+ In B+ V−I Clamping Circuitry I−V Out B− In B− V CNTLB V CLMPB Gain Select B C EXTB Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. Copyright 2005, Texas Instruments Incorporated ! ! www.ti.com " #$%& www.ti.com SBOS326 −AUGUST 2005 ABSOLUTE MAXIMUM RATINGS(1) Power Supply (VDD) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +6V VCA2617 Analog Input . . . . . . . . . . . . . . . . . −0.3V to (+VS + 0.3V) Logic Input . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.3V to (+VS + 0.3V) Case Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +100°C Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . −40°C to +150°C (1) Stresses above those listed under Absolute Maximum Ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may affect device reliability. This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. PACKAGE/ORDERING INFORMATION(1) PRODUCT PACKAGE-LEAD PACKAGE DESIGNATOR SPECIFIED TEMPERATURE RANGE PACKAGE MARKING VCA2617 QFN−32 RHB −40°C to +85°C VCA2617 ORDERING NUMBER TRANSPORT MEDIA, QUANTITY VCA2617RHBT Tape and Reel, 250 VCA2617RHBR Tape and Reel, 3000 (1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI website at www.ti.com. 2 " #$%& www.ti.com SBOS326 −AUGUST 2005 ELECTRICAL CHARACTERISTICS All specifications at TA = +25°C, VDD = 5V, load resistance = 500Ω on each output to ground, differential output (1VPP), CA, CB = 3.9µF, single-ended input configuration, fIN = 5MHz, HG = Low (High-Gain Mode), VCNTL = 2.3V, unless otherwise noted. VCA2617 PARAMETER CONDITIONS MIN TYP MAX UNIT VARIABLE GAIN AMPLIFIER Input Resistance Input Capacitance Maximum Input Voltage Differential Single-Ended Input Voltage Noise Input Current Noise Noise Figure Input Common-Mode Voltage Bandwidth Clipping Voltage Range (VCLMP) Clipping Voltage Variation Maximum Capacitive Output Loading Slew Rate Maximum Output Signal(1) Output Common-Mode Voltage Output Impedance Output Short-Circuit Current 2nd-Harmonic Distortion 3rd-Harmonic Distortion Overload Distortion (2nd-Harmonic) Crosstalk Delay Matching Single-Ended 300 8 kΩ pF 2.0 1.0 4.1 1 13.3 2.5 50 0.25 to 2.6 ±75 80 100 6 2.5 3 60 −53 −62 −40 −61 ±1 VPP VPP nV/√Hz pA/√Hz dB V MHz V mV pF V/µs VPP V Ω mA dBc dBc dBc dB ns Linear Operation(1); Each Input Gain = 45dB, RS = 0Ω Independent of Gain Internal VCLMP = 0.5V, VCAOUT = 1.0VPP 50Ω in Series at 5MHz, Single-Ended, Either Output VOUT = 1VPP, VCNTL = 1.5V VOUT = 1VPP, VCNTL = 1.5V Input Signal = 1VPP, VCNTL = 2V −43 −43 ACCURACY Gain Slope Gain Error(2) Gain Range Gain Range (HG) Output Offset Voltage, Differential Channel-to-Channel Gain Matching VCNTL = 0.4V to 2.0V VCNTL = 0.4V to 2.0V VCNTL = 0.2V to 2.3V VCNTL = 0.4V to 2.0V HG = 0 (+6dB); VGA High Gain; VCNTL = 0.2V to 2.3V HG = 1 (0dB); VGA Low Gain; VCNTL = 0.2V to 2.3V VCNTL = 0.4V to 2.0V 22 ±0.9 48 35.5 −10 to +38 −16 to +32 ±50 ±0.8 42dB Gain Change; to 90% Signal Level 0.2 to 2.3 1 0.5 −2 +2 dB/V dB dB dB dB dB mV dB GAIN CONTROL INTERFACE (VCNTL) Input Voltage Range Input Resistance Response Time V MΩ µs DIGITAL INPUTS(3), (4) (HGA, HGB, PD) VIH, High-Level Input Voltage VIL, Low-Level Input Voltage Input Resistance Input Capacitance 2.0 0.8 1 5 V V MΩ pF POWER SUPPLY Supply Voltage Power Dissipation Power-Down Power-Up Response Time Power-Down Response Time 4.75 5.0 105 22 25 2 5.25 125 V mW mW µs µs +85 °C °C/W °C/W THERMAL CHARACTERISTICS Temperature Range Thermal Resistance, qJA qJC Ambient, Operating Soldered Pad; Four-Layer PCB with Thermal Vias −40 36.7 4.0 (1) 2nd, 3rd-harmonic distortion less than or equal to −30dBc. (2) Referenced to best fit dB-linear curve. (3) Parameters ensured by design; not production tested. (4) Do not leave inputs floating; no internal pull-up/pull-down resistors. 3 " #$%& www.ti.com SBOS326 −AUGUST 2005 INA− NC CA2 CA1 VDDA GNDA Out A− Out A+ 32 31 30 29 28 27 26 25 PIN CONFIGURATION INA + 1 24 VCNTLA NC 2 23 VCLMPA VDDR 3 22 HGA 21 NC 20 PD VCA2617 (Thermal Pad tied to Ground Potential) 16 VCNTLB OutB+ 17 15 8 OutB− INB+ 14 VCLMPB GNDB 18 13 7 VDDB NC CB1 HGB 12 19 11 6 CB2 GNDR 10 5 NC VCM 9 4 INB− VB PIN DESCRIPTIONS 4 PIN DESIGNATOR 1 INA+ PIN DESIGNATOR Channel A +Input DESCRIPTION 17 VCNTLB Channel B Gain Control Voltage Input No Internal Connection 18 VCLMPB Channel B Clamp Voltage Reference Supply 19 HGB Bias Voltage, 0.1µF Bypass Capacitor 20 PD Power Down (Active High) Common-mode Voltage, 0.1µF Bypass Capacitor 21 NC No Internal Connection Internal Reference Ground 22 HGA No Internal Connection 23 2 NC 3 4 VDDR VB 5 VCM 6 GNDR 7 NC 8 INB+ Channel B+ Input 24 VCLMPA VCNTLA DESCRIPTION Channel B High/Low Output Gain (High = Low Gain) Channel A High/Low Output Gain (High = Low Gain) Channel A Clamp Voltage Channel A Gain Control Voltage Input 9 INB− Channel B− Input 25 OutA+ Channel A+ Output 10 NC No Internal Connection 26 OutA− Channel A− Output 11 CB2 Channel B, External Coupling Capacitor 27 GNDA Channel A Ground 12 CB1 Channel B, External Coupling Capacitor 28 VDDA Channel A Supply 13 VDDB Channel B Supply 29 CA1 Channel A; External Coupling Capacitor 14 GNDB Channel B Ground 30 CA2 Channel A; External Coupling Capacitor 15 OutB− Channel B− Output 31 NC No Internal Connection 16 OutB+ Channel B+ Output 32 INA− Channel A− Input " #$%& www.ti.com SBOS326 −AUGUST 2005 TYPICAL CHARACTERISTICS GAIN vs VCNTL GAIN ERROR vs VCNTL 2.0 1.5 HG = 0 1.0 HG = 0 HG = 1 Gain Error (dB) 0.5 0 HG = 1 −0.5 −1.0 VCNTL (V) 40 1.5 30 1.0 GAIN vs VCNTL vs TEMPERATURE 2.0 1.9 1.8 1.7 1.6 1.5 1.4 2.0 1.9 1.8 1.7 1.6 1.4 GAIN ERROR vs VCNTL vs TEMPERATURE 2.0 1.5 −40_ C 20 +25_ C +85_ C −40_C 1.0 Gain Error (dB) 30 25 +25_C 0.5 0 +85_C −0.5 0 −5 −10 −1.0 −15 −2.0 VCNTL (V) Figure 5 2.0 1.9 1.8 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 0.9 0.8 0.7 0.6 0.5 0.4 −1.5 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 2.1 2.2 2.3 Gain (dB) 1.3 Figure 4 35 10 5 1.2 VCNTL (V) Figure 3 15 1.1 1.0 0.9 0.8 0.7 −2.0 0.6 −1.5 VCNTL (V) 45 40 1.3 1MHz 2MHz 5MHz 10MHz −1.0 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 2.1 2.2 2.3 −10 −0.5 1.5 1MHz 2MHz 5MHz 10MHz 0 0.5 10 0.5 0.4 20 −20 GAIN ERROR vs VCNTL vs FREQUENCY 2.0 Gain Error (dB) Gain (dB) Figure 2 GAIN vs VCNTL vs FREQUENCY 0 1.2 VCNTL (V) Figure 1 50 1.1 1.0 0.9 0.8 0.7 0.6 −2.0 0.5 −1.5 0.4 60 55 50 45 40 35 30 25 20 15 10 5 0 −5 −10 −15 −20 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 2.1 2.2 2.3 Gain (dB) All specifications at TA = +25°C, VDD = 5V, load resistance = 500Ω on each output to ground, differential output (1VPP), CA, CB = 3.9µF, single-ended input configuration, fIN = 5MHz, HG = Low (High-Gain Mode), VCNTL = 2.3V, unless otherwise noted. VCNTL (V) Figure 6 5 " #$%& www.ti.com SBOS326 −AUGUST 2005 TYPICAL CHARACTERISTICS (continued) All specifications at TA = +25°C, VDD = 5V, load resistance = 500Ω on each output to ground, differential output (1VPP), CA, CB = 3.9µF, single-ended input configuration, fIN = 5MHz, HG = Low (High-Gain Mode), VCNTL = 2.3V, unless otherwise noted. GAIN MATCHING, CHA to CHB GAIN MATCHING, CHA to CHB 70 100 VCNTL = 0.4V VCNTL = 2.0V 90 60 80 70 60 40 Units Units 50 30 50 40 30 20 20 10 10 0 −1.92 −1.76 −1.60 −1.44 −1.28 −1.12 −0.96 −0.80 −0.64 −0.48 −0.32 −0.16 0.00 0.16 0.32 0.48 0.64 0.80 0.96 1.12 1.28 1.44 1.60 1.76 1.92 −1.80 −1.65 −1.50 −1.35 −1.20 −1.05 −0.90 −0.75 −0.60 −0.45 −0.30 −0.15 0.00 0.15 0.30 0.45 0.60 0.75 0.90 1.05 1.20 1.35 1.50 1.65 1.80 0 Delta Gain (dB) Delta Gain (dB) Figure 7 Figure 8 GAIN vs FREQUENCY GAIN vs FREQUENCY 40 45 3.9µF HG = 0, VCNTL = 2.3V 40 39 0.1µF 35 38 HG = 1, VCNTL = 2.3V 37 25 Gain (dB) Gain (dB) 30 20 15 10 HG = 0, VCNTL = 0.7V 36 4700pF 0.022µF 35 34 5 33 0 32 −5 HG = 1, VCNTL = 0.7V −10 0.1 1 10 31 0.1 100 1 OUTPUT−REFERRED NOISE vs VCNTL INPUT−REFERRED NOISE vs VCNTL 1000 1000 100 HG = 1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 2.1 2.2 2.3 2.4 2.5 10 VCNTL (V) Figure 11 HG = 1 100 HG = 0 10 1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 2.1 2.2 2.3 2.4 2.5 HG = 0 Noise (nV/√Hz) Noise (nV/√Hz) 100 Figure 10 Figure 9 6 10 Frequency (MHz) Frequency (MHz) VCNTL (V) Figure 12 " #$%& www.ti.com SBOS326 −AUGUST 2005 TYPICAL CHARACTERISTICS (continued) All specifications at TA = +25°C, VDD = 5V, load resistance = 500Ω on each output to ground, differential output (1VPP), CA, CB = 3.9µF, single-ended input configuration, fIN = 5MHz, HG = Low (High-Gain Mode), VCNTL = 2.3V, unless otherwise noted. INPUT−REFERRED NOISE vs FREQUENCY (Hi VGA Gain) NOISE FIGURE vs VCNTL 100 10 Noise (nV/√Hz) Noise nV/√Hz HG = 1 HG = 0 10 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 1.9 2.0 2.1 2.2 2.3 2.4 2.5 1 −50 1 Frequency (MHz) Figure 13 Figure 14 DISTORTION vs FREQUENCY (2nd−Harmonic, VCNTL = 2.3V) DISTORTION vs FREQUENCY (3rd−Harmonic, VCNTL = 2.3V) −50 −52 −52 −54 −54 HG = 1 −56 Distortion (dB) Distortion (dB) 10 VCNTL (V) −58 −60 −62 −64 HG = 0 −66 −56 −60 −62 −64 −66 −68 −68 −70 −70 1 10 −25 HG = 1 −58 HG = 0 1 10 Frequency (MHz) Frequency (MHz) Figure 15 Figure 16 DISTORTION vs VCNTL (Lo VGA Gain) DISTORTION vs VCNTL (Hi VGA Gain) −40 −30 Distortion (dB) −40 −45 −50 −55 2nd−Harmonic −50 −55 −60 3rd−Harmonic −60 VCNTL (V) VCNTL (V) Figure 17 Figure 18 2.3 2.2 2.1 2.0 1.9 1.8 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 0.9 −65 0.8 2.3 2.2 2.1 2.0 1.9 1.8 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 3rd−Harmonic 0.9 −65 0.8 Distortion (dB) −45 2nd−Harmonic −35 7 " #$%& www.ti.com SBOS326 −AUGUST 2005 TYPICAL CHARACTERISTICS (continued) All specifications at TA = +25°C, VDD = 5V, load resistance = 500Ω on each output to ground, differential output (1VPP), CA, CB = 3.9µF, single-ended input configuration, fIN = 5MHz, HG = Low (High-Gain Mode), VCNTL = 2.3V, unless otherwise noted. DISTORTION vs RESISTIVE LOAD DISTORTION vs VCA OUTPUT −40 −40 −45 −45 Distortion (dB) Distortion (dB) −50 −55 2nd−Harmonic −60 −65 −50 2nd−Harmonic −55 −60 −70 3rd−Harmonic −75 VCNTL (V) VCNTL (V) Figure 21 Figure 22 TOTAL POWER vs TEMPERATURE 1.8 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 0.9 0.8 2.3 2.2 2.1 2.0 1.9 1.8 1.7 −75 1.6 600 550 1MHz 2MHz 5MHz 10MHz −71 −75 1.4 500 −69 −73 1.3 450 −67 −73 1.2 750 HG = 1 1.1 700 −69 −65 2.0 −67 −63 1.9 HG = 0 1.0 2.3 −61 0.9 650 −61 0.8 2.2 −59 Crosstalk (dB) −57 −59 −71 CROSSTALK vs VCNTL −55 −57 1.5 Crosstalk (dB) Figure 20 CROSSTALK vs VCNTL −65 2.1 Figure 19 −63 400 Resistance (Ω) VCA Output (VPP) −55 350 300 250 200 150 100 50 −70 2.0 1.9 1.8 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 0.9 0.8 0.7 0.6 3rd−Harmonic 0.5 −80 −65 DISTORTION vs TEMPERATURE −80 105.8 105.7 −75 3rd−Harmonic 105.5 Distortion (dB) Power (mW) 105.6 105.4 105.3 105.2 105.1 10 20 30 40 Temperature (_C) Figure 23 8 −65 −60 105.0 104.9 −40 −30 −20 −10 0 −70 50 60 70 80 2nd−Harmonic −55 −40 −30 −20 −10 0 10 20 30 40 Temperature (_C) Figure 24 50 60 70 80 " #$%& www.ti.com SBOS326 −AUGUST 2005 TYPICAL CHARACTERISTICS (continued) All specifications at TA = +25°C, VDD = 5V, load resistance = 500Ω on each output to ground, differential output (1VPP), CA, CB = 3.9µF, single-ended input configuration, fIN = 5MHz, HG = Low (High-Gain Mode), VCNTL = 2.3V, unless otherwise noted. VCLMP vs VOUT (150mVPP, S/E Input) 2ND−HARMONIC DISTORTION −20 −28 VOUT (PP) −32 −36 −40 −44 0.50 0.75 1.00 VCLMP (V) VIN (VPP) Figure 25 Figure 26 2ND−HARMONIC DISTORTION vs VCNTL (Differential Input) −45 −45 −50 −50 VOUT (PP) 2.3 2.2 2.1 2.0 1.9 1.8 1.7 1.6 HG = 1 1.3 −75 2.3 2.2 2.1 2.0 1.9 1.8 1.7 1.6 1.5 1.4 1.3 1.1 1.0 0.9 0.8 −75 −70 3rd−Harmonic 0.9 −70 −65 1.2 −65 HG = 0 −60 0.8 −60 −55 1.1 2nd−Harmonic 1.0 Distortion (dBc) −55 1.2 Distortion (dBc) DISTORTION vs VOUT (Differential Input) 1.5 −48 0.25 1.4 Distortion (dB) −24 6.2 5.8 5.4 5.0 4.6 4.2 HG = 0 3.8 3.4 3.0 2.6 2.2 1.8 1.4 HG = 1 1.0 0.6 0.2 −0.2 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCNTL (V) Figure 27 Figure 28 3RD−HARMONIC DISTORTION vs VCNTL (Differential Input) DISTORTION vs RESISTIVE LOAD (Differential) −45 −30 HG = 1 −35 −50 2nd−Harmonic Distortion (dB) −55 −60 −65 −50 −55 VCNTL (V) Figure 29 2.3 2.2 2.1 2.0 1.9 1.8 1.7 1.6 1.5 1.4 1.3 1.2 1.1 1.0 −65 0.9 −75 −45 −60 HG = 0 −70 0.8 Distortion (dBc) −40 −70 3rd−Harmonic 50 100 150 200 250 300 350 400 450 500 550 600 650 700 750 Resistance (Ω) Figure 30 9 " #$%& www.ti.com SBOS326 −AUGUST 2005 TYPICAL CHARACTERISTICS (continued) All specifications at TA = +25°C, VDD = 5V, load resistance = 500Ω on each output to ground, differential output (1VPP), CA, CB = 3.9µF, single-ended input configuration, fIN = 5MHz, HG = Low (High-Gain Mode), VCNTL = 2.3V, unless otherwise noted. OUTPUT IMPEDANCE vs FREQUENCY GROUP DELAY vs FREQUENCY 100 25 Group Delay (ns) ROUT (Ω) 23 10 21 19 17 1 15 0.1 1 10 100 1 Frequency (MHz) Figure 32 Figure 31 2V 0V 1VPP VGA Output (V) VCNTL (V) GAIN CONTROL TRANSIENT RESPONSE Time (0.4µs/div) Figure 33 10 10 Frequency (MHz) 100 " #$%& www.ti.com SBOS326 −AUGUST 2005 Table 1. Gain and Noise Performance of Various VCA Blocks THEORY OF OPERATION The VCA2617 is a dual-channel variable gain amplifier (VGA) with each channel being independant. The VGA is a true variable-gain amplifier, achieving lower noise output at lower gains. The output amplifier has two gains, allowing for further optimization with different analog-to-digital converters. Figure 34 shows a simplified block diagram of a single channel of the dual-channel system. The VGA can be powered down in order to conserve system power when necessary. BLOCK GAIN (Loss) dB NOISE nV/√Hz Buffer 0 3.9 Attenuator (VCA2619) 0 2.2 Attenuator (VCA2619) −40 2.2 VCA1 (VCA2617) 40 4.1 VCA1 (VCA2617) 0 90 VCA1 (VCA2619) 40 3.9 When the block diagram shown in Figure 34 has a gain of 40dB, the noise referred to the input (RTI) is: Total Noise (RTI) + 4.1nVń ǸHz (1) VGA Figure 34. Simplified Block Diagram of VCA2617 When the block diagram shown in Figure 35 has the combined gain of 60dB, the noise referred to the input (RTI) is given by the expression: Total Noise (RTI) + Ǹ(Buffer Noise)2 ) (ATTEN Noise)2 ) (VCA Noise)2 VGA—OVERVIEW + The VGA that is used with the VCA2617 is a true variable-gain amplifier; as the gain is reduced, the noise contribution from the VGA itself is also reduced. This design is in contrast with another popular device architecture (used by the VCA2619), where an effective VCA characteristic is obtained by a voltage-variable attenuator succeeded by a fixed-gain amplifier. At the highest gain, systems with either architecture are dominated by the noise produced at the input to either the fixed or variable gain amplifier. At low gains, however, the noise output is dominated by the contribution from the VGA. Therefore, the overall system with lower VGA gain will produce less noise. The following example will illustrate this point. Figure 34 shows a block diagram of the variable-gain amplifier; Figure 35 shows a block diagram of a variable attenuation attenuator followed by a fixed gain amplifier. For purposes of this example, let us assume the performance characteristics shown in Table 1; these values are the typical performance data of the VCA2617 and the VCA2619. Buffer ATTENUATOR Ǹ(3.9)2 ) (2.2)2 ) (3.9)2 + 5.9nVń ǸHz (2) Repeating the above measurements for both VCA configurations when the overall gain is 10dB yields the following results: For the VCA with a variable gain amplifier (Figure 34): Total Noise (RTI) + 90nVń ǸHz (3) For the VCA with a variable attenuation attenuator (Figure 35): Total Noise + Ǹ(3.9)2 ) (2.2) 2 ) (2.0ń0.10)2 + 14nVń ǸHz (4) The VGA has a continuously-variable gain range of 48dB with the ability to select either of two options. The gain of the VGA can either be varied from −10dB to 38dB, or from −16dB to 32dB. The VGA output can be programmed to clip at a predetermined voltage that is selected by the user. When the user applies a voltage to the VCLMP-pin, the output will have a peak-to-peak voltage that is given by the graph shown in Figure 26. Amplifier Figure 35. Block Diagram of Older VCA Models 11 " #$%& www.ti.com SBOS326 −AUGUST 2005 VOLTAGE-CONTROLLED AMPLIFIER (VCA)— DETAIL is fixed, will not show diminished noise in this manner. Refer to Table 2, which shows a comparison between the noise performance at different gains for the VCA2617 and the older VCA2619. Figure 36 shows a simplified schematic of the VCA. The VCA2617 is a true voltage-controlled amplifier, with the gain expressed in dB directly proportional to a control signal. This architecture compares to the older VCA products where a voltage-controlled attenuator was followed by a fixed-gain amplifier. With a variable-gain amplifier, the output noise diminishes as the gain reduces. A variable-gain amplifier, where the output amplifier gain Table 2. Noise vs Gain (RG = 0) NOISE RTI (nV/√Hz) PRODUCT GAIN (dB) VCA2617 40 4.1 VCA2617 0 100 VCA2619 40 5.9 VCA2619 0 500 Clipping Program Circuitry VDD HGA/B R1 Q1 Q5 Q7 Q9 R2 +IN Q2 D1 D2 D3 D4 A1 Q3 A2 VCM Q4 Q6 External Capacitor Q8 VCNTL Q10 Q12 Q14 Q16 Q18 Q20 Q22 Q24 C1 VCA Program Circuitry C C2 Q11 Q13 Q15 Q17 Q19 Q21 Q23 Q25 Voltage−Controlled Resistor Network Q26 Q30 Control Signal Q32 VCM Q27 D5 D6 D7 D8 A4 A3 R3 Q28 −IN R4 Q29 Q31 Q33 Q34 VDD Clipping Program Circuitry VCLMP Figure 36. Block Diagram of VCA 12 " #$%& www.ti.com SBOS326 −AUGUST 2005 The VCA accepts a differential input at the +IN and −IN terminals. Amplifier A1, along with transistors Q2 and Q3, forms a voltage follower that buffers the +IN signal to be able to drive the voltage-controlled resistor. Amplifier A3, along with transistors Q27 and Q28, plays the same role as A1. The differential signal applied to the voltage-controlled resistor network is converted to a current that flows through transistors Q1 through Q4. Through the mirror action of transistors Q1/Q5 and Q4/Q6, a copy of this same current flows through Q5 and Q6. Assuming that the signal current is less than the programmed clipping current (that is, the current flowing through transistors Q7 and Q8), the signal current will then go through the diode bridge (D1 through D4) and be sent through either R2 or R1, depending upon the state of Q9. This signal current multiplied by the feedback resistor associated with amplifier A2, determines the signal voltage that is designated −OUT. Operation of the circuitry associated with A3 and A4 is identical to the operation of the previously described function to create the signal +OUT. A1 and its circuitry form a voltage-to-current converter, while A2 and its circuitry form a current-to-voltage converter. This architecture was adapted because it has excellent signal-handling capability. A1 has been designed to handle a large voltage signal without overloading, and the various mirroring devices have also been sized to handle large currents. Good overload capability is achieved as both the input and output amplifier are not required to amplify voltage signals. Voltage amplification occurs when the input voltage is converted to a current; this current in turn is converted back to a voltage as amplifier A2 acts as a transimpedance amplifier. The overall gain of the output amplifier A2 can be altered by 6dB by the action of the HG signal. This enables more optimum performance when the VCA interfaces with either a 10-bit or 12-bit analog-to-digital converter (ADC). An external capacitor (C) is required to provide a low impedance connection to join the two sections of the resistor network. Capacitor C could be replaced by a short-circuit. By providing a DC connection, the output offset will be a function of the gain setting. Typically, the offset at this point is ±10mV; thus, if the gain varies from 1 to 100, the output offset would vary from ±10mV to ±100mV. VARIABLE GAIN CHARACTERISTICS Transistors Q10, Q12, Q14, Q16, Q18, Q20, Q22, and Q24 form a variable resistor network that is programmed in an exponential manner to control the gain. Transistors Q11, Q13, Q15, Q17, Q19, Q21, Q23, and Q25 perform the same function. These two groups of FET variable resistors are configured in this manner to balance the capacitive loading on the total variable-resistor network. This balanced configuration is used to keep the second harmonic component of the distortion low. The common source connection associated with each group of FET variable resistors is brought out to an external pin so that an external capacitor can be used to make an AC connection. This connection is necessary to achieve an adequate low-frequency bandwidth because the coupling capacitor would be too large to include within the monolithic chip. The value of this variable resistor ranges in value from 15Ω to 5000Ω to achieve a gain range of about 48dB. The low-frequency bandwidth is then given by the formula: Low Frequency BW + 1ń2pRC (5) where: R is the value of the attenuator. C is the value of the external coupling capacitor. For example, if a low-frequency bandwidth of 500kHz was desired and the value of R was 10Ω, then the value of the coupling capacitor would be 0.05µF. One of the benefits of this method of gain control is that the output offset is independent of the variable gain of the output amplifier. The DC gain of the output amplifier is extremely low; any change in the input voltage is blocked by the coupling capacitor, and no signal current flows through the variable resistor. This method also means that any offset voltage existing in the input is stored across this coupling capacitor; when the resistor value is changed, the DC output will not change. Therefore, changes in the control voltage do not appear in the output signal. Figure 37 shows the output waveform resulting from a step change in the control voltage, and Figure 38 shows the output voltage resulting when the control voltage is a full-scale ramp. Channel 1 VCNTL (2V/div) Channel 2 Output (20mV/div) Time (400ns/div) Figure 37. Response to Step Change of VCNTL 13 " #$%& www.ti.com SBOS326 −AUGUST 2005 When FETs used as variable resistors are placed in parallel, the attenuation characteristic that is created behaves according to this same exponential characteristic at discrete points as a function of the control voltage. It does not perfectly obey an ideal exponential characteristic at other points; however, an 8-section approximation yields a ±1dB error to an ideal curve. Channel 1 VCNTL (2V/div) PROGRAMMABLE CLIPPING Channel 2 Output (20mV/div) Time (400ns/div) Figure 38. Response to Ramp Change of VCNTL The exponential gain control characteristic is achieved through a piecewise approximation to a perfectly smooth exponential curve. Eight FETs, operated as variable resistors whose value is progressively 1/2 of the value of the adjacent parallel FET, are turned on progressively, or their value is lowered to create the exponential gain characteristic. This characteristic can be shown in the following way. An exponential such as y = ex increases in the y dimension by a constant ratio as the x dimension increases by a constant linear amount. In other words, for a constant (x1 − x2), the ratio ex1/ex2 remains the same. The clipping level of the VCA2617 can be programmed to a desired output. The programming feature is useful when matching the clipped level from the output of the VCA to the full-scale range of a subsequent VCA, in order to prevent the VCA from generating false spectral signals; see the circuit diagram shown in Figure 39. The signal node at the drain junction of Q5 and Q6 is sent to the diode bridge formed by diode-connected transistors, D1 through D4. The diode bridge output is determined by the current that flows through transistors Q7 and Q8. The maximum current that can then flow into the summing node of A2 is this same current; consequently, the maximum voltage output of A2 is this same current multiplied by the feedback resistor associated with A2. The maximum output voltage of A2, which would be the clipped output, can then be controlled by adjusting the current that flows through Q7 and Q8; see the circuit diagram shown in Figure 36. The circuitry of A1, R1, and Q1 converts the clamp voltage (VCLMP) to a current that controls equal and opposite currents flowing through transistors Q5 and Q6. VDD R1 Q9 Q1 A1 Q7 Q5 From Buffered Input Q2 Clip Adjust Input R2 D1 D2 D3 D4 VCM HGA/B A2 Output Amplifier Q6 Q8 Figure 39. Clipping Level Adjust Circuitry 14 " #$%& www.ti.com SBOS326 −AUGUST 2005 When HG = 0, the previously described circuitry is designed so that the value of the VCLMP signal is equal to the peak differential signal developed between +VOUT and −VOUT. When HG = 1, the differential output will be equal to the clamp voltage. This method of controlled clipping also achieves fast and clean settling waveforms at the output of the VCA, as shown in Figure 40 through Figure 43. The sequence of waveforms demonstrate the clipping performance during various stages of overload. In a typical application, the VCA2617 will drive an anti-aliasing filter, which in turn will drive an ADC. Many modern ADCs, such as the ADS5270, are well-behaved with as much as 2x overload. This means that the clipping level of the VCA should be set to overcome the loss in the filter such that the clipped input to the ADC is just slightly over the full-scale input. By setting the clipping level in this manner, the lowest harmonic distortion level will be achieved without interfering with the overload capability of the ADC. Input (0.5V/div) Input (0.5V/div) Differential Output (0.5V/div) Differential Output (0.5V/div) VCNTL = 1.0V Time (200ns/div) Figure 40. Before Overload (630mVPP Input) VCNTL = 1.0V Figure 42. Overload (1.5VPP Input) Input (0.5V/div) Input (2V/div) Differential Output (0.5V/div) Differential Output (1V/div) VCNTL = 1.0V Time (200ns/div) Figure 41. Approaching Overload (700mVPP Input) Time (200ns/div) VCNTL = 1.0V Time (200ns/div) Figure 43. Extreme Overload (3.8VPP Input) 15 PACKAGE OPTION ADDENDUM www.ti.com 27-Feb-2006 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Eco Plan (2) Qty VCA2617RHBR ACTIVE QFN RHB 32 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR VCA2617RHBRG4 ACTIVE QFN RHB 32 3000 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR VCA2617RHBT ACTIVE QFN RHB 32 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR VCA2617RHBTG4 ACTIVE QFN RHB 32 250 Green (RoHS & no Sb/Br) CU NIPDAU Level-2-260C-1 YEAR Lead/Ball Finish MSL Peak Temp (3) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. Addendum-Page 1 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. Customers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. All products are sold subject to TI’s terms and conditions of sale supplied at the time of order acknowledgment. TI warrants performance of its hardware products to the specifications applicable at the time of sale in accordance with TI’s standard warranty. Testing and other quality control techniques are used to the extent TI deems necessary to support this warranty. Except where mandated by government requirements, testing of all parameters of each product is not necessarily performed. TI assumes no liability for applications assistance or customer product design. Customers are responsible for their products and applications using TI components. To minimize the risks associated with customer products and applications, customers should provide adequate design and operating safeguards. TI does not warrant or represent that any license, either express or implied, is granted under any TI patent right, copyright, mask work right, or other TI intellectual property right relating to any combination, machine, or process in which TI products or services are used. Information published by TI regarding third-party products or services does not constitute a license from TI to use such products or services or a warranty or endorsement thereof. Use of such information may require a license from a third party under the patents or other intellectual property of the third party, or a license from TI under the patents or other intellectual property of TI. Reproduction of information in TI data books or data sheets is permissible only if reproduction is without alteration and is accompanied by all associated warranties, conditions, limitations, and notices. Reproduction of this information with alteration is an unfair and deceptive business practice. TI is not responsible or liable for such altered documentation. Resale of TI products or services with statements different from or beyond the parameters stated by TI for that product or service voids all express and any implied warranties for the associated TI product or service and is an unfair and deceptive business practice. TI is not responsible or liable for any such statements. Following are URLs where you can obtain information on other Texas Instruments products and application solutions: Products Applications Amplifiers amplifier.ti.com Audio www.ti.com/audio Data Converters dataconverter.ti.com Automotive www.ti.com/automotive DSP dsp.ti.com Broadband www.ti.com/broadband Interface interface.ti.com Digital Control www.ti.com/digitalcontrol Logic logic.ti.com Military www.ti.com/military Power Mgmt power.ti.com Optical Networking www.ti.com/opticalnetwork Microcontrollers microcontroller.ti.com Security www.ti.com/security Mailing Address: Telephony www.ti.com/telephony Video & Imaging www.ti.com/video Wireless www.ti.com/wireless Texas Instruments Post Office Box 655303 Dallas, Texas 75265 Copyright 2006, Texas Instruments Incorporated