MPS MP2355

TM
MP2355
3A, 23V, 380KHz
Step-Down Converter
The Future of Analog IC Technology
TM
DESCRIPTION
FEATURES
The MP2355 is a step-down regulator with a
built in internal Power MOSFET. It achieves 3A
continuous output current over a wide input
supply range with excellent load and line
regulation.
•
•
•
Current mode operation provides fast transient
response and eases loop stabilization. Fault
condition protection includes cycle-by-cycle
current limiting and thermal shutdown. Adjustable
soft-start reduces the stress on the input source
at turn-on. In shutdown mode the regulator draws
20µA of supply current.
The MP2355 uses a minimum number of
readily available external components to
complete a 3A step-down DC to DC converter
solution.
EVALUATION BOARD REFERENCE
•
•
•
•
•
•
•
•
Programmable Soft-Start
100mΩ Internal Power MOSFET Switch
Stable with Low ESR Output Ceramic
Capacitors
Up to 95% Efficiency
20µA Shutdown Mode
3A Output Current
Wide 4.75V to 23V Operating Input Range
Fixed 380KHz Frequency
Thermal Shutdown
Cycle-by-Cycle Over Current Protection
Under Voltage Lockout
APPLICATIONS
•
•
•
Distributed Power Systems
Battery Chargers
Pre-Regulator for Linear Regulators
“MPS” and “The Future of Analog IC Technology” are Trademarks of Monolithic
Power Systems, Inc.
Board Number
Dimensions
EV2355DN-00A
2.0”X x 1.3”Y x 0.5”Z
TYPICAL APPLICATION
INPUT
4.75V to 23V
10nF
8
1
3
2
VIN
BST
LX
RUN
MP2355
SS
FB
GND
10nF
5
95
4
6
D1
B330A
OUTPUT
3.3V / 3A
COMP
4.7nF
VOUT=5.0V
90
7
EFFICIENCY (%)
OPEN
AUTOMATIC
STARTUP
Efficiency vs
Load Current
85
80
VOUT=3.3V
VOUT=2.5V
75
70
65
60
MP2355_TAC_S01
0
500 1000 1500 2000 2500 3000 3500
LOAD CURRENT (mA)
MP2355_EC01
MP2355 Rev. 1.5
5/1/2006
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1
TM
MP2355 – 3A, 23V, 380KHz STEP-DOWN CONVERTER
ABSOLUTE MAXIMUM RATINGS (1)
PACKAGE REFERENCE
TOP VIEW
SS
1
8
RUN
BST
2
7
COMP
VIN
3
6
FB
LX
4
5
GND
EXPOSED PAD
ON BACKSIDE
CONNECT TO PIN 5
MP2355_PD01-SOIC8N
Supply Voltage VIN ....................... –0.3V to +25V
Switch Voltage VLX ....................... –0.3V to +26V
Boost Voltage VBST ..........VLX – 0.3V to VLX + 6V
All Other Pins................................. –0.3V to +6V
Junction Temperature...............................150°C
Lead Temperature ....................................260°C
Storage Temperature .............–65°C to +150°C
Recommended Operating Conditions
Input Voltage VIN ............................ 4.75V to 23V
Operating Temperature .............–40°C to +85°C
Thermal Resistance
Part Number*
MP2355DN
*
Package
SOIC8N
(Exposed Pad)
Temperature
–40°C to +85°C
For Tape & Reel, add suffix –Z (eg. MP2355DN–Z)
For RoHS compliant packaging, add suffix –LF (eg.
MP2355DN –LF–Z)
(2)
(3)
θJA
θJC
SOIC8N .................................. 50 ...... 10... °C/W
Notes:
1) Exceeding these ratings may damage the device.
2) The device is not guaranteed to function outside of its
operating conditions.
3) Measured on approximately 1” square of 1 oz copper.
ELECTRICAL CHARACTERISTICS
VIN = 12V, TA = +25°C, unless otherwise noted.
Parameter
Shutdown Supply Current
Supply Current
Symbol Condition
VRUN = 0V
VRUN = 2.8, VFB = 1.5V
Feedback Voltage
VFB
Error Amplifier Voltage Gain
Error Amplifier
Transconductance
High-Side Switch-On
Resistance
Low-Side Switch-On
Resistance
High-Side Switch Leakage
Current
Current Limit (4)
Current Sense to COMP
Transconductance
Oscillation Frequency
Short Circuit Oscillation
Frequency
Maximum Duty Cycle
Minimum Duty Cycle
EN Shutdown Threshold
Voltage
AVEA
MP2355 Rev. 1.5
5/1/2006
GEA
4.75V ≤ VIN ≤ 23V,
VCOMP < 2V
Min
Typ
20
1.0
Max
30
1.2
Units
µA
mA
1.194
1.222
1.250
V
400
∆ICOMP = ±10µA
500
800
V/V
1120
µA/V
RDS(ON)1
95
mΩ
RDS(ON)2
10
Ω
VRUN = 0V, VLX = 0V
0
3.7
GCS
fS
VFB = 0V
DMAX
DMIN
10
µA
4.3
A
3.8
A/V
330
380
430
KHz
20
35
50
KHz
0
%
%
1.5
V
VFB = 1.0V
VFB = 1.5V
90
0.9
1.2
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2
TM
MP2355 – 3A, 23V, 380KHz STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS (continued)
VIN = 12V, TA = +25°C, unless otherwise noted.
Parameter
Enable Pull Up Current
EN UVLO Threshold
EN UVLO Threshold
Hysteresis
Soft-Start Period
Symbol Condition
VRUN = 0V
VEN Rising
Min
1.1
2.37
Typ
1.8
2.54
CSS = 0.1µF
Thermal Shutdown
Max
2.5
2.71
Units
µA
V
210
mV
10
ms
150
°C
Note:
4) Equivalent output current = 1.5A ≥ 50% Duty Cycle
2.0A ≤ 50% Duty Cycle
Assumes ripple current = 30% of load current.
Slope compensation changes current limit above 40% duty cycle.
TYPICAL PERFORMANCE CHARACTERISTICS
Circuit of Figure 2, VIN = 12V, VO = 3.3V, L1 = 15µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless
otherwise noted.
Heavy Load
Operation
Light Load
Operation
3A Load
No Load
VIN, AC
200mV/div.
VIN, AC
20mV/div.
VO, AC
20mV/div.
VO, AC
20mV/div.
IL
1A/div.
IL
1A/div.
VSW
10V/div.
VSW
10V/div.
MP2355-TPC01
MP2355-TPC02
Startup from Shutdown
Startup from Shutdown
Startup from Shutdown
No C4
3A Resistive Load
C4 = 10nF
3A Resistive Load
C4 = 10nF
No Load
VEN
5V/div.
VEN
5V/div.
VOUT
1V/div.
VOUT
1V/div.
IL
1A/div.
VOUT
1V/div.
IL
1A/div.
IL
1A/div.
MP2355-TPC03
MP2355 Rev. 1.5
5/1/2006
VEN
5V/div.
MP2355-TPC04
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MP2355-TPC05
3
TM
MP2355 – 3A, 23V, 380KHz STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Circuit of Figure 2, VIN = 12V, VO = 3.3V, L1 = 15µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless
otherwise noted.
Load Transient
Short Circuit Protection
VO, AC
200mV/div.
Short Circuit Recovery
VOUT
2V/div.
VOUT
2V/div.
IL
1A/div.
ILOAD
1A/div.
IL
2A/div.
IL
2A/div.
MP2355-TPC06
MP2355-TPC07
MP2355-TPC08
PIN FUNCTIONS
Pin #
1
2
3
4
5
6
7
8
Name
Description
Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to
SS
GND to set the soft-start period. A 0.1µF capacitor sets the soft-start period to 10ms. To
disable the soft-start feature, leave SS unconnected.
High-Side Gate Drive Boost Input. BST supplies the drive for the high-side N-Channel
BST MOSFET switch. Connect a 10nF or greater capacitor from LX to BST to power the high
side switch.
Power Input. VIN supplies the power to the IC, as well as the step-down converter switches.
VIN
Drive VIN with a 4.75V to 23V power source. Bypass VIN to GND with a suitably large
capacitor to eliminate noise on the input to the IC. See Input Capacitor
Power Switching Output. LX is the switching node that supplies power to the output.
LX
Connect the output LC filter from LX to the output load. Note that a capacitor is required
from LX to BST to power the high-side switch.
GND Ground. (Note: Connect the exposed pad on backside to Pin 5.)
Feedback Input. FB senses the output voltage to regulate that voltage. Drive FB with a
FB
resistive voltage divider from the output voltage. The feedback threshold is 1.222V. See
Setting the Output Voltage
Compensation Node. COMP is used to compensate the regulation control loop. Connect a
COMP series RC network from COMP to GND to compensate the regulation control loop. In some
cases, an additional capacitor from COMP to GND is required. See Compensation
Enable/UVLO. A voltage greater than 2.71V enables operation. For complete low current
RUN
shutdown the EN pin voltage needs to be less than 900mV.
MP2355 Rev. 1.5
5/1/2006
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4
TM
MP2355 – 3A, 23V, 380KHz STEP-DOWN CONVERTER
OPERATION
is compared to the switch current measured
internally to control the output voltage.
The MP2355 is a current-mode step-down
regulator. It regulates input voltages from 4.75V to
23V down to an output voltage as low as 1.222V,
and is able to supply up to 3A of load current.
The converter uses an internal N-Channel
MOSFET switch to step-down the input voltage to
the regulated output voltage. Since the MOSFET
requires a gate voltage greater than the input
voltage, a boost capacitor connected between LX
and BST drives the gate. The capacitor is
internally charged while LX is low.
The MP2355 uses current-mode control to
regulate the output voltage. The output voltage is
measured at FB through a resistive voltage
divider and amplified through the internal error
amplifier.
The
output
current
of
the
transconductance error amplifier is presented at
COMP where a network compensates the
regulation control system. The voltage at COMP
An internal 10Ω switch from LX to GND is used to
insure that LX is pulled to GND when LX is low to
fully charge the BST.capacitor.
VIN 3
CURRENT
SENSE
AMPLIFIER
INTERNAL
REGULATORS
OSCILLATOR
42/380kHz
+
0.7V
--
RUN 8
-2.37V/
2.62V
+
FREQUENCY
FOLDBACK
COMPARATOR
+
SLOPE
COMP
5V
--
CLK
+
SHUTDOWN
COMPARATOR
--
S
Q
R
Q
CURRENT
COMPARATOR
2
BST
4
LX
5
GND
LOCKOUT
COMPARATOR
1.8V
--
+
--
0.7V 1.222V
6
FB
+
ERROR
AMPLIFIER
7
COMP
1
SS
MP2355_BD01
Figure 1—Functional Block Diagram
MP2355 Rev. 1.5
5/1/2006
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TM
MP2355 – 3A, 23V, 380KHz STEP-DOWN CONVERTER
APPLICATIONS INFORMATION
C5
10nF
INPUT
4.75V to 23V
3
OPEN
AUTOMATIC
STARTUP
8
1
2
VIN
RUN
FB
5
OUTPUT
3.3V / 3A
4
MP2355
SS
GND
C4
10nF
BST
LX
6
D1
B330A
COMP
7
C6
OPEN
C3
4.7nF
MP2355_TAC_F02
Figure 2—MP2355 with Murata 22µF, 10V Ceramic Output Capacitor
COMPONENT SELECTION
Setting the Output Voltage
The output voltage is set using a resistive
voltage divider from the output voltage to FB
pin. The voltage divider divides the output
voltage down to the feedback voltage by the
ratio:
VFB = VOUT
R2
R1 + R2
Thus the output voltage is:
VOUT = 1.22 ×
R1 + R2
R2
Where VFB is the feedback voltage and VOUT is
the output voltage.
A typical value for R2 can be as high as 100kΩ,
but a typical value is 10kΩ. Using that value, R1
is determined by:
R1 = 8.18 × ( VOUT − 1.22)(kΩ )
For example, for a 3.3V output voltage, R2 is
10kΩ, and R1 is 17kΩ.
Inductor
The inductor is required to supply constant
current to the output load while being driven by
the switched input voltage. A larger value
inductor will result in less ripple current that will
result in lower output ripple voltage. However,
the larger value inductor will have a larger
physical size, higher series resistance, and/or
lower saturation current.
MP2355 Rev. 1.5
5/1/2006
A good rule for determining the inductance to
use is to allow the peak-to-peak ripple current in
the inductor to be approximately 30% of the
maximum switch current limit. Also, make sure
that the peak inductor current is below the
maximum switch current limit. The inductance
value can be calculated by:
L=
⎛
⎞
VOUT
V
× ⎜⎜1 − OUT ⎟⎟
fS × ∆IL ⎝
VIN ⎠
Where VIN is the input voltage, fS is the 380KHz
switching frequency, and ∆IL is the peak-topeak inductor ripple current.
Choose an inductor that will not saturate under
the maximum inductor peak current. The peak
inductor current can be calculated by:
ILP = ILOAD +
⎛
VOUT
V
× ⎜1 − OUT
2 × f S × L ⎜⎝
VIN
⎞
⎟⎟
⎠
Where ILOAD is the load current.
Table 1 lists a number of suitable inductors
from various manufacturers. The choice of
which style inductor to use mainly depends on
the price vs. size requirements and any EMI
requirement.
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6
TM
MP2355 – 3A, 23V, 380KHz STEP-DOWN CONVERTER
Table 1—Inductor Selection Guide
Vendor/
Model
Package
Dimensions
(mm)
Core
Type
Core
Material
W
L
H
Open
Ferrite
7.0
7.8
5.5
Sumida
CR75
CDH74
Open
Ferrite
7.3
8.0
5.2
CDRH5D28
Shielded
Ferrite
5.5
5.7
5.5
CDRH5D28
Shielded
Ferrite
5.5
5.7
5.5
Sumida (continued)
CDRH6D28
Shielded
Ferrite
6.7
6.7
3.0
CDRH104R
Shielded
Ferrite
10.1
10.0
3.0
Input Capacitor
The input current to the step-down converter is
discontinuous, therefore a capacitor is required
to supply the AC current to the step-down
converter while maintaining the DC input
voltage. Use low ESR capacitors for the best
performance. Ceramic capacitors are preferred,
but tantalum or low-ESR electrolytic capacitors
may also suffice.
Since the input capacitor (C1) absorbs the input
switching current it requires an adequate ripple
current rating. The RMS current in the input
capacitor can be estimated by:
I C1 = ILOAD ×
Toko
D53LC
Type A
Shielded
Ferrite
5.0
5.0
3.0
D75C
Shielded
Ferrite
7.6
7.6
5.1
D104C
Shielded
Ferrite
10.0
10.0
4.3
D10FL
Open
Ferrite
9.7
1.5
4.0
DO3308
Open
Ferrite
9.4
13.0
3.0
DO3316
Open
Ferrite
9.4
13.0
5.1
Choose a diode which has a maximum reverse
voltage rating is greater than the maximum
input voltage, and who’s current rating is
greater than the maximum load current. Table 2
lists
example
Schottky
diodes
and
manufacturers.
Table 2—Diode Selection Guide
Diode
SK33
SK34
B330
B340
MBRS330
MBRS340
MP2355 Rev. 1.5
5/1/2006
⎞
⎟
⎟
⎠
The worst-case condition occurs at VIN = 2VOUT,
where:
I C1 =
Coilcraft
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is off. To
reduce losses due to the diode forward voltage
and recovery times, use a Schottky diode.
VOUT ⎛⎜ VOUT
× 1−
VIN ⎜⎝
VIN
ILOAD
2
For simplification, choose the input capacitor
whose RMS current rating greater than half of the
maximum load current.
The input capacitor can be electrolytic, tantalum
or ceramic. When using electrolytic or tantalum
capacitors, a small, high quality ceramic
capacitor, i.e. 0.1µF, should be placed as close to
the IC as possible. When using ceramic
capacitors, make sure that they have enough
capacitance to provide sufficient charge to
prevent excessive voltage ripple at input. The
input voltage ripple caused by capacitance can be
estimated by:
∆VIN =
⎛
ILOAD
V
V
× OUT × ⎜⎜ 1 − OUT
f s × C1
VIN
VIN
⎝
⎞
⎟⎟
⎠
Voltage/Current Manufacture
Rating
30V, 3A
40V, 3A
30V, 3A
40V, 3A
30V, 3A
40V, 3A
Diodes Inc.
Diodes Inc.
Diodes Inc.
Diodes Inc.
On Semiconductor
On Semiconductor
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7
TM
MP2355 – 3A, 23V, 380KHz STEP-DOWN CONVERTER
Output Capacitor
The output capacitor is required to maintain the
DC output voltage. Ceramic, tantalum, or low
ESR electrolytic capacitors are recommended.
Low ESR capacitors are preferred to keep the
output voltage ripple low. The output voltage
ripple can be estimated by:
∆VOUT =
VOUT ⎛
V
× ⎜⎜1 − OUT
fS × L ⎝
VIN
⎞
⎞ ⎛
1
⎟
⎟⎟ × ⎜ R ESR +
⎜
8 × f S × C2 ⎟⎠
⎠ ⎝
Where L is the inductor value, C2 is the output
capacitance value, and RESR is the equivalent
series resistance (ESR) value of the output
capacitor.
In the case of ceramic capacitors, the impedance
at the switching frequency is dominated by the
capacitance. The output voltage ripple is mainly
caused by the capacitance. For simplification, the
output voltage ripple can be estimated by:
∆VOUT =
⎛
V
× ⎜⎜1 − OUT
VIN
× L × C2 ⎝
VOUT
8 × fS
2
⎞
⎟⎟
⎠
In the case of tantalum or electrolytic capacitors,
the ESR dominates the impedance at the
switching frequency. For simplification, the output
ripple can be approximated to:
∆VOUT =
VOUT ⎛
V
× ⎜⎜1 − OUT
fS × L ⎝
VIN
⎞
⎟⎟ × R ESR
⎠
The characteristics of the output capacitor also
affect the stability of the regulation system. The
MP2355 can be optimized for a wide range of
capacitance and ESR values.
Compensation Components
MP2355 employs current mode control for easy
compensation and fast transient response. The
system stability and transient response are
controlled through the COMP pin. COMP pin is
the output of the internal transconductance
error amplifier. A series capacitor-resistor
combination sets a pole-zero combination to
control the characteristics of the control system.
The DC gain of the voltage feedback loop is
given by:
A VDC = R LOAD × G CS × A VEA ×
Where AVEA is the error amplifier voltage gain,
is
the
current
sense
400V/V;
GCS
transconductance, 3.8A/V; RLOAD is the load
resistor value.
The system has two poles of importance. One
is due to the compensation capacitor (C3) and
the output resistor of error amplifier, and the
other is due to the output capacitor and the load
resistor. These poles are located at:
fP1 =
GEA
2π × C3 × A VEA
fP2 =
1
2π × C2 × R LOAD
Where
GEA
is
the
transconductance, 800µA/V.
error
amplifier
The system has one zero of importance, due to
the compensation capacitor (C3) and the
compensation resistor (R3). This zero is located
at:
f Z1 =
1
2π × C3 × R3
The system may have another zero of
importance, if the output capacitor has a large
capacitance and/or a high ESR value. The zero,
due to the ESR and capacitance of the output
capacitor, is located at:
fESR =
1
2π × C2 × RESR
In this case (as shown in Figure 2), a third pole
set by the compensation capacitor (C6) and the
compensation resistor (R3) is used to
compensate the effect of the ESR zero on the
loop gain. This pole is located at:
f P3 =
MP2355 Rev. 1.5
5/1/2006
VFB
VOUT
1
2π × C6 × R3
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TM
MP2355 – 3A, 23V, 380KHz STEP-DOWN CONVERTER
The goal of compensation design is to shape
the converter transfer function to get a desired
loop gain. The system crossover frequency
where the feedback loop has the unity gain is
important.
Lower crossover frequencies result in slower
line and load transient responses, while higher
crossover frequencies could cause system
unstable. A good rule of thumb is to set the
crossover frequency to approximately one-tenth
of the switching frequency. Switching frequency
for the MP2355 is 380KHz, so the desired
crossover frequency is around 38KHz.
Table 3 lists the typical values of compensation
components for some standard output voltages
with various output capacitors and inductors.
The values of the compensation components
have been optimized for fast transient
responses and good stability at given
conditions.
Table 3—Compensation Values for Typical
Output Voltage/Capacitor Combinations
VOUT
L1
C2
R3
C3
C6
2.5V
10µH
min.
22µF
Ceramic
3.9kΩ
5.6nF
None
3.3V
15µH
min.
22µF
Ceramic
4.7kΩ
4.7nF
None
5V
15µH
min.
22µF
Ceramic
7.5kΩ
2.7nF
None
12V
22µH
min.
22µF
Ceramic
15kΩ
1.5nF
None
2.5V
10µH
min.
560µF Al.
30mΩ ESR
100kΩ
1nF
150pF
3.3V
15µH
min.
560µF Al
30mΩ ESR
120kΩ
1nF
120pF
5V
15µH
min.
470µF Al.
30mΩ ESR
150kΩ
1nF
82pF
12V
22µH
min.
220µF Al.
30mΩ ESR
169kΩ
1nF
39pF
1) Choose the compensation resistor (R3) to
set the desired crossover frequency. Determine
the R3 value by the following equation:
MP2355 Rev. 1.5
5/1/2006
C3 >
4
2π × R3 × f C
Where R3 is the compensation resistor value and
fC is the desired crossover frequency, 38KHz.
3) Determine if the second compensation
capacitor (C6) is required. It is required if the ESR
zero of the output capacitor is located at less than
half of the 380KHz switching frequency, or the
following relationship is valid:
f
1
< S
2π × C2 × R ESR
2
If this is the case, then add the second
compensation capacitor (C6) to set the pole fP3 at
the location of the ESR zero. Determine the C6
value by the equation:
C6 =
C2 × R ESR
R3
External Bootstrap Diode
It is recommended that an external bootstrap
diode be added when the system has a 5V
fixed input or the power supply generates a 5V
output. This helps improve the efficiency of the
regulator. The bootstrap diode can be a low
cost one such as IN4148 or BAT54.
5V
To optimize the compensation components for
conditions not listed in Table 2, the following
procedure can be used.
R3 =
2) Choose the compensation capacitor (C3) to
achieve the desired phase margin. For
applications with typical inductor values, setting
the compensation zero, fZ1, to less than one forth
of the crossover frequency provides sufficient
phase margin. Determine the C3 value by the
following equation:
2π × C2 × f C VOUT
×
G EA × G CS
VFB
BS
10nF
MP2355
SW
MP2355_F03
Figure 3—External Bootstrap Diode
This diode is also recommended for high duty
cycle operation (when
VOUT
>65%) and high
VIN
output voltage (VOUT>12V) applications.
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9
TM
MP2355 – 3A, 23V, 380KHz STEP-DOWN CONVERTER
PACKAGE INFORMATION
SOIC8N (EXPOSED PAD)
0.229(5.820)
0.244(6.200)
PIN 1 IDENT.
NOTE 4
0.150(3.810)
0.157(4.000)
0.0075(0.191)
0.0098(0.249)
SEE DETAIL "A"
NOTE 2
0.011(0.280) x 45o
0.020(0.508)
0.013(0.330)
0.020(0.508)
0.050(1.270)BSC
0o-8o
NOTE 3
0.189(4.800)
0.197(5.000)
0.053(1.350)
0.068(1.730)
0.016(0.410)
0.050(1.270)
.050
0.049(1.250)
0.060(1.524)
DETAIL "A"
.028
0.200 (5.07 mm)
SEATING PLANE
0.001(0.030)
0.004(0.101)
0.140 (3.55mm)
0.060
Land Pattern
NOTE:
1) Control dimension is in inches. Dimension in bracket is millimeters.
2) Exposed Pad Option (N-Package) ; 2.31mm -2.79mm x 2.79mm - 3.81mm.
Recommend Solder Board Area: 2.80mm x 3.82mm = 10.7mm 2 (16.6 mil2)
3) The length of the package does not include mold flash. Mold flash shall not exceed 0.006in. (0.15mm) per side.
With the mold flash included, over-all length of the package is 0.2087in. (5.3mm) max.
4) The width of the package does not include mold flash. Mold flash shall not exceed 0.10in. (0.25mm) per side.
With the mold flash included, over-all width of the package is 0.177in. (4.5mm) max.
NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications.
Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS
products into any application. MPS will not assume any legal responsibility for any said applications.
MP2355 Rev. 1.5
5/1/2006
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MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited.
© 2006 MPS. All Rights Reserved.
10