TM MP2361 2A, 23V, 1.4MHz Step-Down Converter The Future of Analog IC Technology TM DESCRIPTION FEATURES The MP2361 is a monolithic step-down switch mode converter with a built-in internal power MOSFET. It achieves 2A continuous output current over a wide input supply range with excellent load and line regulation. • • • Current mode operation provides fast transient response and eases loop stabilization. Fault condition protections include cycle-by-cycle current limiting and thermal shutdown. In shutdown mode the regulator draws 20µA of supply current. Programmable soft-start minimizes the inrush supply current and the output overshoot at initial startup. The MP2361 requires a minimum number of readily available standard external components. • • • • • • • • • • 2A Output Current with QFN Package 0.18Ω Internal Power MOSFET Switch Stable with Low ESR Output Ceramic Capacitors 90% Efficiency 20µA Shutdown Mode Fixed 1.4MHz Frequency Thermal Shutdown Cycle-by-Cycle Over Current Protection Wide 4.75V to 23V Operating Input Range Output Adjustable from 0.92V to 16V Programmable Under Voltage Lockout Available in 10-pin QFN (3mm x 3mm) and Tiny MSOP Packages Evaluation Board Available APPLICATIONS EVALUATION BOARD REFERENCE Board Number Dimensions EV2361DQ-00A 2.3”X x 1.5”Y x 0.5”Z • • • • Distributed Power Systems Battery Charger DSL Modems Pre-Regulator for Linear Regulators “MPS” and “The Future of Analog IC Technology” are Trademarks of Monolithic Power Systems, Inc. TYPICAL APPLICATION 9 10 4 2 IN BS EN SW FB GND 6 C4 10nF C5 10nF MP2361 SS Efficiency vs Load Current 100 5 7 D1 B220A VOUT 2.5V/2A COMP 8 C6 OPEN C3 1.8nF VOUT=5V 90 EFFICIENCY (%) INPUT 4.75V to 23V 80 VOUT=2.5V VOUT=3.3V 70 60 50 MP2361_TAC_S01 0 0.5 1.0 1.5 LOAD CURRENT (A) 2.0 MP2361-EC01 MP2361 Rev. 1.2 1/11/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 1 TM MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER PACKAGE REFERENCE TOP VIEW TOP VIEW NC 1 10 SS BS 2 9 EN NC 3 8 COMP IN 4 7 FB SW 5 6 GND EXPOSED PAD ON BACKSIDE NC 1 10 SS BS 2 9 EN NC 3 8 COMP IN 4 7 FB SW 5 6 GND MP2361_PD01-MSOP10 MP2361_PD01-QFN10 Part Number* Package Temperature Part Number** Package Temperature MP2361DQ QFN10 (3mm x 3mm) –40°C to +85°C MP2361DK MSOP10 –40°C to +85°C * For Tape & Reel, add suffix –Z (eg. MP2361DQ–Z) For Lead Free, add suffix –LF (eg. MP2361DQ –LF–Z) ** For Tape & Reel, add suffix –Z (eg. MP2361DK–Z) For Lead Free, add suffix –LF (eg. MP2361DK –LF–Z) ABSOLUTE MAXIMUM RATINGS (1) Recommended Operating Conditions Supply Voltage (VIN)..................................... 25V Switch Node Voltage (VSW) .......................... 26V Bootstrap Voltage (VBS) ....................... VSW + 6V Feedback Voltage (VFB) .................–0.3V to +6V Enable/UVLO Voltage (VEN)...........–0.3V to +6V Comp Voltage (VCOMP) ...................–0.3V to +6V Junction Temperature .............................+150°C Lead Temperature ..................................+260°C Storage Temperature.............. –65°C to +150°C Supply Voltage (VIN) ...................... 4.75V to 23V Operating Temperature .............–40°C to +85°C Thermal Resistance (3) θJA (2) θJC QFN10 (3mmx3mm) ............... 50 ...... 12... °C/W MSOP10 ................................ 150 ..... 65... °C/W Notes: 1) Exceeding these ratings may damage the device. 2) The device is not guaranteed to function outside of its operating conditions. 3) Measured on approximately 1” square of 1 oz copper. ELECTRICAL CHARACTERISTICS VIN = 12V, TA = +25°C, unless otherwise noted. Parameter Feedback Voltage Upper Switch On Resistance Lower Switch On Resistance Upper Switch Leakage Current Limit (4) Current Sense Transconductance Output Current to Comp Pin Voltage Error Amplifier Voltage Gain Error Amplifier Transconductance Oscillator Frequency Short Circuit Frequency Soft-Start Pin Equivalent Output Resistance MP2361 Rev. 1.2 1/11/2006 Symbol Condition VFB 4.75V ≤ VIN ≤ 23V RDS(ON)1 RDS(ON)2 VEN = 0V; VSW = 0V Min 0.892 2.8 Typ 0.920 0.18 10 0 3.5 Max 0.948 10 Units V Ω Ω µA A GCS 1.95 A/V AVEA GEA fS 400 930 1.4 210 V/V µA/V MHz KHz ∆IC = ±10µA VFB = 0V 630 9 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 1230 kΩ 2 TM MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER ELECTRICAL CHARACTERISTICS (continued) VIN = 12V, TA = +25°C, unless otherwise noted. Parameter Maximum Duty Cycle Minimum On Time EN Shutdown Threshold Voltage Enable Pull-Up Current EN UVLO Threshold Rising EN UVLO Threshold Hysteresis Symbol DMAX tON VEN IEN VUVLO Condition VFB = 0.8V Min ICC > 100µA VEN = 0V VEN Rising 0.7 Supply Current (Shutdown) IOFF VEN ≤ 0.4V 20 36 µA Supply Current (Quiescent) ION VEN ≥ 3V 1.2 1.4 mA Thermal Shutdown 2.37 Typ 70 100 1.0 1.0 2.50 210 160 Max 1.3 2.62 Units % ns V µA V mV °C Note: 4) Equivalent output current = 1.5A ≥ 50% Duty Cycle 2.0A ≤ 50% Duty Cycle Assumes ripple current = 30% of load current. Slope compensation changes current limit above 40% duty cycle. PIN FUNCTIONS Pin # 1 2 3 4 5 6 7 8 9 10 Name Description NC BS No Connect. Bootstrap (C5). This capacitor is needed to drive the power switch’s gate above the supply voltage. It is connected between SW and BS pins to form a floating supply across the power switch driver. The voltage across C5 is about 5V and is supplied by the internal +5V supply when the SW pin voltage is low. NC No Connect. IN Supply Voltage. The MP2361 operates from a +4.75V to +23V unregulated input. C1 is needed to prevent large voltage spikes from appearing at the input. SW Switch. This connects the inductor to either IN through M1 or to GND through M2. GND Ground. This pin is the voltage reference for the regulated output voltage. For this reason care must be taken in its layout. This node should be placed outside of the D1 to C1 ground path to prevent switching current spikes from inducing voltage noise into the part. FB Feedback. An external resistor divider from the output to GND, tapped to the FB pin sets the output voltage. To prevent current limit run away during a short circuit fault condition the frequency foldback comparator lowers the oscillator frequency when the FB voltage is below 400mV. COMP Compensation. This node is the output of the transconductance error amplifier and the input to the current comparator. Frequency compensation is done at this node by connecting a series R-C to ground. See the compensation section for exact details. EN Enable/UVLO. A voltage greater than 2.62V enables operation. Leave EN unconnected for automatic startup. An Under Voltage Lockout (UVLO) function can be implemented by the addition of a resistor divider from VIN to GND. For complete low current shutdown it’s the EN pin voltage needs to be less than 700mV. SS Soft-Start Pin. Connect SS to an external capacitor to program the soft-start. If unused, leave it open. MP2361 Rev. 1.2 1/11/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 3 TM MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER OPERATION MP2361 reverts to its initial M1 off, M2 on state. If the Current Sense Amplifier plus Slope Compensation signal does not exceed the COMP voltage, then the falling edge of the CLK resets the Flip-Flop. The MP2361 is a current mode regulator. That is, the COMP pin voltage is proportional to the peak inductor current. At the beginning of a cycle: the upper transistor M1 is off; the lower transistor M2 is on (see Figure 1); the COMP pin voltage is higher than the current sense amplifier output; and the current comparator’s output is low. The rising edge of the 1.4MHz CLK signal sets the RS Flip-Flop. Its output turns off M2 and turns on M1 thus connecting the SW pin and inductor to the input supply. The increasing inductor current is sensed and amplified by the Current Sense Amplifier. Ramp compensation is summed to Current Sense Amplifier output and compared to the Error Amplifier output by the Current Comparator. When the Current Sense Amplifier plus Slope Compensation signal exceeds the COMP pin voltage, the RS Flip-Flop is reset and the The output of the Error Amplifier integrates the voltage difference between the feedback and the 0.92V bandgap reference. The polarity is such that the FB pin voltage lower than 0.92V increases the COMP pin voltage. Since the COMP pin voltage is proportional to the peak inductor current an increase in its voltage increases current delivered to the output. The lower 10Ω switch ensures that the bootstrap capacitor voltage is charged during light load conditions. External Schottky Diode D1 carries the inductor current when M1 is off. IN 4 INTERNAL REGULATORS CURRENT SENSE AMPLIFIER 5V OSCILLATOR 210KHz/ 1.4MHz 0.7V -- EN 9 -2.29V/ 2.50V + FREQUENCY FOLDBACK COMPARATOR SLOPE COMP 5V -- CLK + + + SHUTDOWN COMPARATOR -- S Q R Q CURRENT COMPARATOR 2 BS 5 SW 6 GND LOCKOUT COMPARATOR -- + -- 0.4V 0.92V 7 FB + SS 10 1.8V ERROR AMPLIFIER 8 COMP MP2361_BD01 Figure 1—Functional Block Diagram MP2361 Rev. 1.2 1/11/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 4 TM MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER APPLICATION INFORMATION COMPONENT SELECTION Setting the Output Voltage The output voltage is set using a resistive voltage divider from the output voltage to FB pin. The voltage divider divides the output voltage down to the feedback voltage by the ratio: VFB = VOUT R2 R1 + R2 Thus the output voltage is: VOUT = 0.92 × R1 + R2 R2 Where VOUT is the output voltage and VFB is the feedback voltage. A typical value for R2 can be as high as 100kΩ, but a typical value is 10kΩ. Using that value, R1 is determined by: R1 = 10.87 × ( VOUT − 0.92) For example, for a 3.3V output voltage, R2 is 10kΩ, and R1 is 25.8kΩ. Inductor The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor will result in less ripple current that will result in lower output ripple voltage. However, the larger value inductor will have a larger physical size, higher series resistance, and/or lower saturation current. A good rule for determining the inductance to use is to allow the peak-to-peak ripple current in the inductor to be approximately 30% of the maximum switch current limit. Also, make sure that the peak inductor current is below the maximum switch current limit. The inductance value can be calculated by: L= ⎛ ⎞ VOUT V × ⎜⎜1 − OUT ⎟⎟ fS × ∆IL ⎝ VIN ⎠ Where fS is the switching frequency, ∆IL is the peak-to-peak inductor ripple current and VIN is the input voltage. MP2361 Rev. 1.2 1/11/2006 Choose an inductor that will not saturate under the maximum inductor peak current. The peak inductor current can be calculated by: ILP = ILOAD + ⎛ VOUT V × ⎜⎜1 − OUT 2 × fS × L ⎝ VIN ⎞ ⎟⎟ ⎠ Where ILOAD is the load current. Output Rectifier Diode The output rectifier diode supplies the current to the inductor when the high-side switch is off. To reduce losses due to the diode forward voltage and recovery times, use a Schottky diode. Choose a diode whose maximum reverse voltage rating is greater than the maximum input voltage, and whose current rating is greater than the maximum load current. Input Capacitor The input current to the step-down converter is discontinuous, therefore a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. Use low ESR capacitors for the best performance. Ceramic capacitors are preferred, but tantalum or low-ESR electrolytic capacitors may also suffice. Since the input capacitor (C1) absorbs the input switching current it requires an adequate ripple current rating. The RMS current in the input capacitor can be estimated by: I C1 = ILOAD × VOUT ⎛⎜ VOUT × 1− VIN ⎜⎝ VIN ⎞ ⎟ ⎟ ⎠ The worst-case condition occurs at VIN = 2VOUT, where: IC1 = ILOAD 2 For simplification, choose the input capacitor whose RMS current rating greater than half of the maximum load current. www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 5 TM MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER The input capacitor can be electrolytic, tantalum or ceramic. When using electrolytic or tantalum capacitors, a small, high quality ceramic capacitor, i.e. 0.1µF, should be placed as close to the IC as possible. When using ceramic capacitors, make sure that they have enough capacitance to provide sufficient charge to prevent excessive voltage ripple at input. The input voltage ripple caused by capacitance can be estimated by: ∆VIN = ⎛ ILOAD V V × OUT × ⎜1 − OUT fS × C1 VIN ⎜⎝ VIN ⎞ ⎟⎟ ⎠ VOUT ⎛ V × ⎜⎜1 − OUT fS × L ⎝ VIN ⎞ ⎞ ⎛ 1 ⎟ ⎟⎟ × ⎜ R ESR + ⎜ 8 × f S × C2 ⎟⎠ ⎠ ⎝ Where L is the inductor value, RESR is the equivalent series resistance (ESR) value of the output capacitor and C2 is the output capacitance value. In the case of ceramic capacitors, the impedance at the switching frequency is dominated by the capacitance. The output voltage ripple is mainly caused by the capacitance. For simplification, the output voltage ripple can be estimated by: ∆VOUT = ⎛ V × ⎜⎜1 − OUT VIN × L × C2 ⎝ VOUT 8 × fS 2 ⎞ ⎟⎟ ⎠ In the case of tantalum or electrolytic capacitors, the ESR dominates the impedance at the switching frequency. For simplification, the output ripple can be approximated to: ∆VOUT = VOUT ⎛ V × ⎜⎜1 − OUT fS × L ⎝ VIN ⎞ ⎟⎟ × R ESR ⎠ The characteristics of the output capacitor also affect the stability of the regulation system. The MP2361 can be optimized for a wide range of capacitance and ESR values. MP2361 Rev. 1.2 1/11/2006 The DC gain of the voltage feedback loop is given by: A VDC = R LOAD × G CS × A VEA × Output Capacitor The output capacitor is required to maintain the DC output voltage. Ceramic, tantalum, or low ESR electrolytic capacitors are recommended. Low ESR capacitors are preferred to keep the output voltage ripple low. The output voltage ripple can be estimated by: ∆VOUT = Compensation Components The MP2361 employs current mode control for easy compensation and fast transient response. The system stability and transient response are controlled through the COMP pin. COMP pin is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system. VFB VOUT Where RLOAD is the load resistor value, GCS is the current sense transconductance and AVEA is the error amplifier voltage gain. The system has two poles of importance. One is due to the compensation capacitor (C3) and the output resistor of error amplifier, and the other is due to the output capacitor and the load resistor. These poles are located at: fP1 = GEA 2π × C3 × A VEA fP2 = 1 2π × C2 × R LOAD Where GEA is transconductance. the error amplifier The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at: f Z1 = 1 2π × C3 × R3 The system may have another zero of importance, if the output capacitor has a large capacitance and/or a high ESR value. The zero, due to the ESR and capacitance of the output capacitor, is located at: fESR = 1 2π × C2 × RESR www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 6 TM MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER In this case, a third pole set by compensation capacitor (C6) and compensation resistor (R3) is used compensate the effect of the ESR zero on loop gain. This pole is located at: f P3 = the the to the 1 2π × C6 × R3 The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency where the feedback loop has the unity gain is important. Lower crossover frequencies result in slower line and load transient responses, while higher crossover frequencies could cause system unstable. A good rule of thumb is to set the crossover frequency to below one-tenth of the switching frequency. To optimize the compensation components, the following procedure can be used: 1. Choose the compensation resistor (R3) to set the desired crossover frequency. Determine the R3 value by the following equation: 2π × C2 × f C VOUT R3 = × G EA × G CS VFB Where fC is the desired crossover frequency, which is typically less than one tenth of the switching frequency. 2. Choose the compensation capacitor (C3) to achieve the desired phase margin. For applications with typical inductor values, setting the compensation zero, fZ1, to below one forth of the crossover frequency provides sufficient phase margin. Determine the C3 value by the following equation: C3 > 3. Determine if the second compensation capacitor (C6) is required. It is required if the ESR zero of the output capacitor is located at less than half of the switching frequency, or the following relationship is valid: f 1 < S 2π × C2 × R ESR 2 If this is the case, then add the second compensation capacitor (C6) to set the pole fP3 at the location of the ESR zero. Determine the C6 value by the equation: C6 = C2 × R ESR R3 External Boost Diode For 5V input or 5V output applications, it is recommended that an external boost diode be added when the system has a 5V fixed input or the power supply generates a 5V output. This helps improve the efficiency of the MP2361 regulator. The boost diode can be a low cost one such as IN4148 or BAT54. 5V BOOST DIODE BS 2 10nF MP2361 SW 5 MP2361_F02 Figure 2—External Boost Diode 2 π × R3 × f C Where R3 is the compensation resistor value. MP2361 Rev. 1.2 1/11/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 7 TM MP2361 – 2A, 23V, 1.4MHz STEP-DOWN CONVERTER PACKAGE INFORMATION QFN10 (3mm x 3mm) 0.35 0.45 2.95 3.05 Pin 1 Identif ication 0.20 0.30 Pin 1 Identif ication R0.200TY P 10 1 QFN10L (3 x 3mm) 2.95 3.05 2.35 2.000 2.45 Ref Exp. DAP 0.500 Bsc 6 5 1.65 1.75 Exp. DAP Top View BottomView 0.85 0.95 0.178 0.228 0.0000.050 Side View Note: 1) Dimensions arein millimeters. MSOP10 0.0197(0.500)TYP 10 6 0.004(0.100) 0.008(0.200) PIN 1 IDENT. 0.114(2.900) 0.122(3.100) 0.184(4.700) 0.200(5.100) SEE DETAIL "A" 0.014(0.350)TYP 1 GATE PLANE 0.010(0.250) 5 0.014(0.350)TYP 0o -6o 0.017(0.400) 0.025(0.600) 0.032(0.800) 0.044(1.100) 0.008(0.200)REF DETAIL "A" 0.030(0.750) 0.038(0.950) 0.002(0.050) 0.006(0.150) NOTE: 1) Control dimension is in inches. Dimension in bracket is millimeters. 2) Package length does not include mold flash, protrusions or gate burr. 3) Package width does not include interlead flash or protrusions. NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications. MP2361 Rev. 1.2 1/11/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 8