MP2363 3A, 27V, 365KHz Step-Down Converter The Future of Analog IC Technology DESCRIPTION FEATURES The MP2363 is a non-synchronous step-down regulator with an integrated Power MOSFET. It achieves 3A continuous output current over a wide input supply range with excellent load and line regulation. • Current mode operation provides fast transient response and eases loop stabilization. Fault condition protection includes cycle-bycycle current limiting and thermal shutdown. Adjustable soft-start reduces the stress on the input source at turn-on. In shutdown mode, the regulator draws 20µA of supply current. The MP2363 requires a minimum number of readily available external components to complete a 3A step-down DC to DC converter solution. The MP2363 is available in an 8-pin SOIC package. • • • • • • • • • • • 3A Continuous Output Current, 4A Peak Output Current Programmable Soft-Start 100mΩ Internal Power MOSFET Switch Stable with Low ESR Output Ceramic Capacitors Up to 95% Efficiency 20µA Shutdown Mode Fixed 365KHz frequency Thermal Shutdown Cycle-by-Cycle Over Current Protection Wide 4.75V to 27V Operating Input Range Output is Adjustable From 0.92V to 21V Under Voltage Lockout APPLICATIONS • • • Distributed Power Systems Battery Chargers Pre-Regulator for Linear Regulators “MPS” and “The Future of Analog IC Technology” are Registered Trademarks of Monolithic Power Systems, Inc. EVALUATION BOARD REFERENCE Board Number Dimensions EV2363DN-00A 2.0”X x 1.9”Y x 0.4”Z TYPICAL APPLICATION Efficiency Curve OPEN = AUTOMATIC STARTUP 7 8 10nF 1 2 IN BS SW EN MP2363 SS GND FB COMP 4 3 5 6 OPEN 100 B330A 6.8nF OUTPUT 2.5V 3A VIN = 12V VOUT=5.0V 90 EFFICIENCY (%) INPUT 4.75V to 27V VOUT=2.5V 80 VOUT=3.3V 70 60 50 MP2363 Rev. 1.0 6/15/2006 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 LOAD CURRENT (A) www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 1 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER ABSOLUTE MAXIMUM RATINGS (1) PACKAGE REFERENCE Supply Voltage VIN ....................... –0.3V to +28V Switch Voltage VSW ................. –1V to VIN + 0.3V Boost Voltage VBS ..........VSW – 0.3V to VSW + 6V All Other Pins................................. –0.3V to +6V Junction Temperature...............................150°C Lead Temperature ....................................260°C Storage Temperature .............–65°C to +150°C TOP VIEW BS 1 8 SS IN 2 7 EN SW 3 6 COMP GND 4 5 FB Recommended Operating Conditions EXPOSED PAD ON BACKSIDE CONNECT TO PIN 4 Input Voltage VIN ............................ 4.75V to 27V Ambient Operating Temp ..........–40°C to +85°C Thermal Resistance (3) θJA θJC SOIC8N .................................. 50 ...... 10... °C/W Part Number* Package Temperature MP2363DN SOIC8N –40°C to +85°C * (2) Notes: 1) Exceeding these ratings may damage the device. 2) The device is not guaranteed to function outside of its operating conditions. 3) Measured on approximately 1” square of 1 oz copper. For Tape & Reel, add suffix –Z (eg. MP2363DN–Z) For RoHS Compliant Packaging, add suffix –LF (eg. MP2363DN–LF–Z) ELECTRICAL CHARACTERISTICS VIN = 12V, TA = +25°C, unless otherwise noted. Parameters Symbol Condition Shutdown Supply Current Supply Current VEN = 0V VEN = 3V, VFB = 1.4V Feedback Voltage VFB Error Amplifier Voltage Gain Min (4) Error Amplifier Transconductance High-Side Switch On-Resistance (4) Low-Side Switch On-Resistance High-Side Switch Leakage Current Short Circuit Current Limit Current Sense to COMP Transconductance Oscillation Frequency Short Circuit Oscillation Frequency Maximum Duty Cycle Minimum On Time (4) EN Threshold Voltage Enable Pull Up Current Under Voltage Lockout Threshold Under Voltage Lockout Threshold Hysteresis Units 20 1.0 30 1.2 µA mA 0.94 V 1120 µA/V 0.90 0.92 ∆ICOMP = ±10µA 500 800 400 RDS(ON)1 RDS(ON)2 VEN = 0V, VSW = 0V 4.5 GCS fS DMAX TON Max 4.75V ≤ VIN ≤ 27V AVEA GEA Typ VFB = 0V VFB = 0.8V VEN = 0V VIN Rising 315 20 0.9 0.9 2.37 Thermal Shutdown (4) 100 6 0.1 5.7 7.0 365 35 88 120 1.2 1.4 2.54 210 160 V/V 10 415 50 1.5 2.2 2.71 mΩ Ω µA A A/V KHz KHz % ns V µA V mV °C Note: 4) Guaranteed by design. MP2363 Rev. 1.0 6/15/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 2 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER PIN FUNCTIONS Pin # 1 2 3 4 5 6 7 8 Name Description High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET BS switch. Connect a 10nF or greater capacitor from SW to BS to power the high-side switch. Power Input. IN supplies the power to the IC, as well as the step-down converter switches. Drive IN with a 4.75V to 27V power source. Bypass IN to GND with a suitably large capacitor IN to eliminate noise on the input to the IC. See Input Capacitor section of Application Information. Power Switching Output. SW is the switching node that supplies power to the output. Connect SW the output LC filter from SW to the output load. Note that a capacitor is required from SW to BS to power the high-side switch. GND Ground. Connect the exposed pad on backside to Pin 4. Feedback Input. FB senses the output voltage to regulate said voltage. Drive FB with a FB resistive voltage divider from the output voltage. The feedback threshold is 0.92V. See Setting the Output Voltage section of Application Information. Compensation Node. COMP is used to compensate the regulation control loop. Connect a series RC network from COMP to GND to compensate the regulation control loop. In some COMP cases, an additional capacitor from COMP to GND is required. See Compensation section of Application Information. Enable Input. EN is a digital input that turns the regulator on or off. Drive EN higher than 2.71V EN to turn on the regulator, lower than 0.9V to turn it off. For automatic startup, leave EN unconnected. Soft Start Control Input. SS controls the soft start period. Connect a capacitor from SS to GND SS to set the soft-start period. Soft-start cap is always recommended to eliminate the start-up inrush current and for a smooth start-up waveform. TYPICAL PERFORMANCE CHARACTERISTICS VIN = 12V, VOUT = 2.5V, L = 15µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless otherwise noted. Efficiency Curve vs Load Current 85 80 VIN=12V 75 70 VIN=24V 65 80 65 50 MP2363 Rev. 1.0 6/15/2006 VIN=24V 70 55 0.5 1.0 1.5 2.0 2.5 3.0 3.5 LOAD CURRENT (A) VIN=12V 75 60 0 6.5 85 55 7.0 VIN=9V 90 60 50 VOUT = 5V 95 VIN=9V 90 EFFICIENCY (%) 100 0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 LOAD CURRENT (A) CURRENT LIMIT (A) VOUT = 3.3V EFFICIENCY (%) 95 Limit Current vs Duty Cycle Efficiency Curve vs Load Current 6.0 5.5 5.0 4.5 4.0 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 DUTY CYCLE (%) www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 3 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER TYPICAL PERFORMANCE CHARACTERISTICS (continued) VIN = 12V, VOUT = 2.5V, L = 15µH, C1 = 10µF, C2 = 22µF, TA = +25°C, unless otherwise noted. SWITCHING FREQUENCY (KHz) Switching Frequency vs Die Temperature Steady State Test OUT = 1.5A Resistive Load 400 IL 1A/div. 390 VOUT AC Coupled 100mV/div. 380 370 VOUT 10mV/div. 360 350 VIN 200mV/div. ILOAD 1A/div. 340 VSW 10V/div. 330 320 -40 -20 0 20 40 60 80 100 120 DIE TEMPERATURE (oC) Steady State Test Startup through Enable Startup through Enable IOUT = 3A Resistive Load IOUT = 3A Resistive Load IOUT = 1.5A Resistive Load IL 2A/div. VOUT 10mV/div. VIN 200mV/div. VOUT 1V/div. VOUT 1V/div. IL 1A/div. IL 2A/div. VSW 10V/div. 2ms/div. 4ms/div. Shutdown through Enable Shutdown through Enable IOUT = 3A Resistive Load IOUT = 1.5A Resistive Load VOUT 1V/div. VOUT 1V/div. IL 1A/div. IL 2A/div. MP2363 Rev. 1.0 6/15/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 4 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER OPERATION The converter uses an internal N-Channel MOSFET switch to step-down the input voltage to the regulated output voltage. Since the MOSFET requires a gate voltage greater than the input voltage, a boost capacitor connected between SW and BS drives the gate. The capacitor is charged by an internal 5V supply while SW is low. The MP2363 is a current-mode step-down regulator. It regulates an input voltage between 4.75V to 27V down to an output voltage as low as 0.92V, and is able to supply up to 3A of load current. The MP2363 uses current-mode control to regulate the output voltage. The output voltage is measured at the FB pin through a resistive voltage divider and amplified through the internal error amplifier. The output current of the transconductance error amplifier is presented at COMP where a network compensates the regulation control system. The voltage at COMP is compared to the switch current measured internally to control the output voltage. An internal 10Ω switch from SW to GND is used to insure that SW is pulled to GND when SW is low to fully charge the boost.capacitor. IN 2 CURRENT SENSE AMPLIFIER INTERNAL REGULATORS OSCILLATOR 35KHz/ 365KHz SLOPE COMP -- EN 7 -2.54V/ 2.33V + FREQUENCY FOLDBACK COMPARATOR 5V -- CLK + 1.2V + + SHUTDOWN COMPARATOR -- S Q R Q CURRENT COMPARATOR 1 BS 3 SW 4 GND LOCKOUT COMPARATOR -- + -- 0.35V 0.92V 5 FB + SS 8 1.8V ERROR AMPLIFIER 6 COMP Figure 1—Functional Block Diagram MP2363 Rev. 1.0 6/15/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 5 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER APPLICATION INFORMATION COMPONENT SELECTION (Refer to the Typical Application Circuit on page 10) Setting the Output Voltage The output voltage is set using a resistive voltage divider from the output voltage to FB pin. The voltage divider divides the output voltage down to the feedback voltage by the ratio: VFB = VOUT R2 R1 + R2 Where VFB is the feedback voltage and VOUT is the output voltage. Thus the output voltage is: VOUT R1 = 8.18 × ( VOUT − 0.92)(kΩ) Inductor The inductor is required to supply constant current to the output load while being driven by the switched input voltage. A larger value inductor will result in less ripple current that will result in lower output ripple voltage. However, the larger value inductor will have a larger physical size, higher series resistance, and/or lower saturation current. A good rule for determining the inductance to use is to allow the peak-to-peak ripple current in the inductor to be approximately 30% of the maximum switch current limit. Also, make sure that the peak inductor current is below the maximum switch current limit. The inductance value can be calculated by: ⎛ ⎞ VOUT V × ⎜⎜1 − OUT ⎟⎟ fS × ∆IL ⎝ VIN ⎠ Where VIN is the input voltage, fS is the 365KHz switching frequency, and ∆IL is the peak-topeak inductor ripple current. MP2363 Rev. 1.0 6/15/2006 ILP = ILOAD + ⎛ VOUT V × ⎜⎜1 − OUT 2 × fS × L ⎝ VIN ⎞ ⎟⎟ ⎠ Where ILOAD is the load current and fS is the 365KHz switching frequency. Table 1 lists a number of suitable inductors from various manufacturers. The choice of which style inductor to use mainly depends on the price vs. size requirements and any EMI requirement. Table 1—Inductor Selection Guide R1 + R2 = 0.92 × R2 A typical value for R2 can be as high as 100kΩ, but a typical value is 10kΩ. Using that value, R1 is determined by: L= Choose an inductor that will not saturate under the maximum inductor peak current. The peak inductor current can be calculated by: Vendor/ Model Package Dimensions (mm) Core Type Core Material W L H CR75 Open Ferrite 7.0 7.8 5.5 CDH74 Open Sumida Ferrite 7.3 8.0 5.2 CDRH5D28 Shielded Ferrite 5.5 5.7 5.5 CDRH5D28 Shielded Ferrite 5.5 5.7 5.5 CDRH6D28 Shielded Ferrite 6.7 6.7 3.0 CDRH104R Shielded Ferrite 10.1 10.0 3.0 Toko D53LC Type A Shielded Ferrite 5.0 5.0 3.0 D75C Shielded Ferrite 7.6 7.6 5.1 D104C Shielded Ferrite 10.0 10.0 4.3 D10FL Open Ferrite 9.7 1.5 4.0 DO3308 Open Ferrite 9.4 13.0 3.0 DO3316 Open Ferrite 9.4 13.0 5.1 Coilcraft Output Rectifier Diode The output rectifier diode supplies the current to the inductor when the high-side switch is off. To reduce losses due to the diode forward voltage and recovery times, use a Schottky diode. www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 6 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER Choose a diode whose maximum reverse voltage rating is greater than the maximum input voltage, and whose current rating is greater than the maximum load current. Table 2 lists example Schottky diodes and manufacturers. Table 2—Diode Selection Guide Voltage/Current Manufacture Rating Diode SK33 SK34 B330 B340 MBRS330 MBRS340 30V, 3A 40V, 3A 30V, 3A 40V, 3A 30V, 3A 40V, 3A Diodes Inc. Diodes Inc. Diodes Inc. Diodes Inc. On Semiconductor On Semiconductor Input Capacitor The input current to the step-down converter is discontinuous, therefore a capacitor is required to supply the AC current to the step-down converter while maintaining the DC input voltage. Use low ESR capacitors for the best performance. Ceramic capacitors are preferred, but tantalum or low-ESR electrolytic capacitors may also suffice. Since the input capacitor (C1) absorbs the input switching current it requires an adequate ripple current rating. The RMS current in the input capacitor can be estimated by: VOUT ⎛⎜ VOUT × 1− VIN ⎜⎝ VIN I C1 = ILOAD × ⎞ ⎟ ⎟ ⎠ ILOAD is the load current, VOUT is the output voltage, and VIN is the input voltage. The worstcase condition occurs at VIN = 2VOUT, where: IC1 = ILOAD 2 For simplification, choose the input capacitor whose RMS current rating greater than half of the maximum load current. The input capacitor can be electrolytic, tantalum or ceramic. When using electrolytic or tantalum capacitors, a small, high quality ceramic capacitor, i.e. 0.1µF, should be placed as close to the IC as possible. When using ceramic capacitors, make sure that they have enough capacitance to provide sufficient charge to prevent excessive voltage ripple at input. The input voltage ripple caused by capacitance can be estimated by: ∆VIN = ⎛ ILOAD V V × OUT × ⎜1 − OUT fS × C1 VIN ⎜⎝ VIN ⎞ ⎟⎟ ⎠ Output Capacitor The output capacitor (C2) is required to maintain the DC output voltage. Ceramic, tantalum or low ESR electrolytic capacitors are recommended. Low ESR capacitors are preferred to keep the output voltage ripple low. The output voltage ripple can be estimated by: ∆VOUT = VOUT ⎛ V × ⎜⎜1 − OUT fS × L ⎝ VIN ⎞ ⎞ ⎛ 1 ⎟ ⎟⎟ × ⎜ R ESR + ⎜ 8 × f S × C2 ⎟⎠ ⎠ ⎝ Where L is the inductor value and RESR is the equivalent series resistance (ESR) value of the output capacitor. In the case of ceramic capacitors, the impedance at the switching frequency is dominated by the capacitance. The output voltage ripple is mainly caused by the capacitance. For simplification, the output voltage ripple can be estimated by: ∆VOUT = ⎛ V × ⎜⎜1 − OUT VIN × L × C2 ⎝ VOUT 8 × fS 2 ⎞ ⎟⎟ ⎠ In the case of tantalum or electrolytic capacitors, the ESR dominates the impedance at the switching frequency. For simplification, the output ripple can be approximated to: ∆VOUT = VOUT ⎛ V ⎞ × ⎜1 − OUT ⎟ × RESR fS × L ⎝ VIN ⎠ The characteristics of the output capacitor also affect the stability of the regulation system. The MP2363 can be optimized for a wide range of capacitance and ESR values. MP2363 Rev. 1.0 6/15/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 7 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER Compensation Components MP2363 employs current mode control for easy compensation and fast transient response. The system stability and transient response are controlled through the COMP pin. COMP pin is the output of the internal transconductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the characteristics of the control system. The DC gain of the voltage feedback loop is given by: A VDC = R LOAD × G CS × A VEA × VFB VOUT Where AVEA is the error amplifier voltage gain, 400V/V; GCS is the current sense transconductance, 7A/V, and RLOAD is the load resistor value. The system has two poles of importance. One is due to the compensation capacitor (C3) and the output resistor of error amplifier, and the other is due to the output capacitor and the load resistor. These poles are located at: fP1 = GEA 2π × C3 × A VEA fP2 = 1 2π × C2 × R LOAD Where GEA is the transconductance, 800µA/V. error amplifier The system has one zero of importance, due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at: f Z1 = 1 2π × C3 × R3 The system may have another zero of importance, if the output capacitor has a large capacitance and/or a high ESR value. The zero, due to the ESR and capacitance of the output capacitor, is located at: fESR = MP2363 Rev. 1.0 6/15/2006 In this case, a third pole set by compensation capacitor (C6) and compensation resistor (R3) is used compensate the effect of the ESR zero on loop gain. This pole is located at: f P3 = the the to the 1 2π × C6 × R3 The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency where the feedback loop has the unity gain is important. Lower crossover frequencies result in slower line and load transient responses, while higher crossover frequencies can cause system instability. A good rule of thumb is to set the crossover frequency to approximately one-tenth of the switching frequency. Switching frequency for the MP2363 is 365KHz, so the desired crossover frequency is around 36.5KHz. Table 3 lists the typical values of compensation components for some standard output voltages with various output capacitors and inductors. The values of the compensation components have been optimized for fast transient responses and good stability at given conditions. Table 3—Compensation Values for Typical Output Voltage/Capacitor Combinations VOUT L C2 R3 C3 C6 1.8V 4.7µH 100µF Ceramic 5.6kΩ 3.3nF None 2.5V 4.7–10µH 47µF 3.32kΩ Ceramic 6.8nF None 3.3V 6.8–10µH 22µFx2 4.02kΩ Ceramic 8.2nF None 5V 10–15µH 22µFx2 6.49kΩ Ceramic 10nF None 12V 15–20µH 22µFx2 Ceramic 4.7nF None 15kΩ 1 2π × C2 × R ESR www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 8 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER To optimize the compensation components for conditions not listed in Table 2, the following procedure can be used. 1. Choose the compensation resistor (R3) to set the desired crossover frequency. Determine the R3 value by the following equation: R3 = 2π × C2 × f C VOUT × G EA × G CS VFB Where fC is the desired crossover frequency (which typically has a value no higher than 37.5KHz). 2. Choose the compensation capacitor (C3) to achieve the desired phase margin. For applications with typical inductor values, setting the compensation zero, fZ1, below one forth of the crossover frequency provides sufficient phase margin. Determine the C3 value by the following equation: 4 C3 > 2π × R3 × f C 3. Determine if the second compensation capacitor (C6) is required. It is required if the ESR zero of the output capacitor is located at less than half of the 365KHz switching frequency, or the following relationship is valid: f 1 < S 2π × C2 × R ESR 2 Soft-Start Capacitor To reduce input inrush current during startup, a programmable soft-start is provided by connecting a capacitor (C4) from pin SS to GND. The soft-start time is given by: t SS (ms ) = 45 × C SS (µF) To reduce the susceptibility to noise, do not leave SS pin open. Use a capacitor with small value if you do not need soft-start function. External Bootstrap Diode It is recommended that an external bootstrap diode be added when the system has a 5V fixed input or the power supply generates a 5V output. This helps improve the efficiency of the regulator. The bootstrap diode can be a low cost one such as IN4148 or BAT54. 5V DIODE BS 1 10nF MP2363 SW 3 Figure 2—External Bootstrap Diode This diode is also recommended for high duty cycle operation (when VOUT >65%) and high VIN output voltage (VOUT>12V) applications. If this is the case, then add the second compensation capacitor (C6) to set the pole fP3 at the location of the ESR zero. Determine the C6 value by the equation: C6 = MP2363 Rev. 1.0 6/15/2006 C2 × R ESR R3 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 9 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER TYPICAL APPLICATION CIRCUITS C5 10nF INPUT 4.75V to 27V 2 OPEN = AUTOMATIC STARTUP 1 IN 7 BS SW EN OUTPUT 3.3V 3A 3 MP2363 8 SS GND FB COMP 4 5 6 C3 8.2nF C6 D1 B330A OPEN Figure 3—MP2363 for 3.3V Output with 47µF, 6.3V Ceramic Output Capacitor C5 10nF INPUT 4.75V to 27V OPEN = AUTOMATIC STARTUP 2 1 IN 7 BS SW EN OUTPUT 5V 3A 3 MP2363 8 SS GND FB COMP 4 5 6 C6 C3 10nF D1 OPEN Figure 4—MP2363 for 5V Output with 47µF, 6.3V Ceramic Output Capacitor MP2363 Rev. 1.0 6/15/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 10 MP2363 – 3A, 27V, 365KHz STEP-DOWN CONVERTER PACKAGE INFORMATION SOIC8N (EXPOSED PAD) 0.229(5.820) 0.244(6.200) PIN 1 IDENT. NOTE 4 0.150(3.810) 0.157(4.000) 0.0075(0.191) 0.0098(0.249) SEE DETAIL "A" NOTE 2 0.011(0.280) x 45o 0.020(0.508) 0.013(0.330) 0.020(0.508) 0.050(1.270)BSC 0o-8o NOTE 3 0.189(4.800) 0.197(5.000) 0.053(1.350) 0.068(1.730) DETAIL "A" 0.016(0.410) 0.050(1.270) .050 0.049(1.250) 0.060(1.524) .028 0.200 (5.07 mm) SEATING PLANE 0.001(0.030) 0.004(0.101) 0.140 (3.55mm) 0.060 Land Pattern NOTE: 1) Control dimension is in inches. Dimension in bracket is millimeters. 2) Exposed Pad Option (N-Package) ; 2.31mm -2.79mm x 2.79mm - 3.81mm. Recommend Solder Board Area: 2.80mm x 3.82mm = 10.7mm 2 (16.6 mil2) 3) The length of the package does not include mold flash. Mold flash shall not exceed 0.006in. (0.15mm) per side. With the mold flash included, over-all length of the package is 0.2087in. (5.3mm) max. 4) The width of the package does not include mold flash. Mold flash shall not exceed 0.10in. (0.25mm) per side. With the mold flash included, over-all width of the package is 0.177in. (4.5mm) max. NOTICE: The information in this document is subject to change without notice. Please contact MPS for current specifications. Users should warrant and guarantee that third party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not assume any legal responsibility for any said applications. MP2363 Rev. 1.0 6/15/2006 www.MonolithicPower.com MPS Proprietary Information. Unauthorized Photocopy and Duplication Prohibited. © 2006 MPS. All Rights Reserved. 11