LP2975 MOSFET LDO Driver/Controller General Description Features A high-current LDO regulator is simple to design with the LP2975 LDO Controller. Using an external P-FET, the LP2975 will deliver an ultra low dropout regulator with extremely low quiescent current. High open loop gain assures excellent regulation and ripple rejection performance. The trimmed internal bandgap reference provides precise output voltage over the entire operating temperature range. Dropout voltage is “user selectable” by sizing the external FET: the minimum input-output voltage required for operation is the maximum load current multiplied by the RDS(ON) of the FET. Overcurrent protection of the external FET is easily implemented by placing a sense resistor in series with VIN. The 57 mV detection threshold of the current sense circuitry minimizes dropout voltage and power dissipation in the resistor. The standard product versions available provide output voltages of 12V, 5V, or 3.3V with guaranteed 25˚C accuracy of 1.5% (“A” grade) and 2.5% (standard grade). n n n n n n n n n n n n Block Diagram Connection Diagram Simple to use, few external components Ultra-small mini SO-8 package 1.5% (A grade) precision output voltage Low-power shutdown input < 1 µA in shutdown Low operating current (180 µA typical @ VIN = 5V) Wide supply voltage range (1.8V to 24V) Built-in current limit amplifier Overtemperature protection 12V, 5V, and 3.3V standard output voltages Can be programmed using external divider −40˚C to +125˚C junction temperature range Applications n High-current 5V to 3.3V regulator n Post regulator for switching converter n Current-limited switch Surface Mount Mini SO-8 Package DS100034-2 Top View For Order Numbers See Table 1 of this Document See NS Package Number MUA08A DS100034-1 *RSET values are: 208k for 12V part, 72.8k for 5V part, and 39.9k for 3.3V part. © 1999 National Semiconductor Corporation DS100034 www.national.com LP2975 MOSFET LDO Driver/Controller September 1997 Ordering Information TABLE 1. Package Marking and Ordering Information Output Voltage Grade Order Information Package Marking Supplied As: 12 A LP2975AIMMX-12 L47A 3.5k Units on Tape and Reel 12 A LP2975AIMM-12 L47A 250 Units on Tape and Reel 12 STD LP2975IMMX-12 L47B 3.5k Units on Tape and Reel 12 STD LP2975IMM-12 L47B 250 Units on Tape and Reel 5.0 A LP2975AIMMX-5.0 L46A 3.5k Units on Tape and Reel 5.0 A LP2975AIMM-5.0 L46A 250 Units on Tape and Reel 5.0 STD LP2975IMMX-5.0 L46B 3.5k Units on Tape and Reel 5.0 STD LP2975IMM-5.0 L46B 250 Units on Tape and Reel 3.3 A LP2975AIMMX-3.3 L45A 3.5k Units on Tape and Reel 3.3 A LP2975AIMM-3.3 L45A 250 Units on Tape and Reel 3.3 STD LP2975IMMX-3.3 L45B 3.5k Units on Tape and Reel 3.3 STD LP2975IMM-3.3 L45B 250 Units on Tape and Reel www.national.com 2 Absolute Maximum Ratings (Note 1) Input Supply Voltage (Survival) −0.3V to +26V If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. Input Supply Voltage (Operating) +1.8V to +24V Storage Temperature Range −65˚C to +150˚C Operating Junction Temperature Range −40˚C to +125˚C Lead Temp. (Soldering, 5 seconds) Current Limit Pins (Survival) −0.3V to +VIN Comp Pin (Survival) −0.3V to +2V Gate Pin (Survival) −0.3V to +VIN ON/OFF Pin (Survival) −0.3V to +20V Feedback Pin (Survival) −0.3V to +24V 260˚C ESD Rating 2 kV Power Dissipation (Note 2) Internally Limited Electrical Characteristics Limits in standard typeface are for TJ = 25˚C, and limits in boldface type apply over the full operating temperature range. Unless otherwise specified; VON/OFF = 1.5V, VIN = 15V. Symbol VREG Parameter Regulation Voltage (12V Versions) Regulation Voltage (5V Versions) Regulation Voltage (3.3V Versions) VCOMP Comp Pin Voltage Quiescent Current IQ Conditions Typ 12.5 < VIN < 24V 12.0 (VIN - 0.5V) > VGATE > (VIN - 5V) 5.5 < VIN < 24V 5.0 (VIN - 0.5V) > VGATE > (VIN - 4.5V) 3.8 < VIN < 24V 3.3 (VIN - 0.5V) > VGATE > (VIN - 3.3V) VREG < VIN < 24V VIN = 5V VON/OFF Current Limit Sense Voltage VIN = 15V VFB = 0.9 X VREG ON/OFF Threshold Output = ON 0.94 Output = OFF ION/OFF IG Min Max 11.820 12.180 11.700 12.300 11.640 12.360 11.520 12.480 4.925 5.075 4.875 5.125 4.850 5.150 4.800 5.200 3.250 3.350 3.217 3.383 3.201 3.399 3.168 3.432 1.215 1.265 1.203 1.277 1.209 1.271 1.196 1.284 VON/OFF = 1.5V Gate Drive Current (Sourcing) VG = 7.5V VFB= 1.1 X VREG 3.5 VG = 7.5V VFB = 0.9 X VREG 1100 240 240 320 320 1 1 45 69 45 69 39 72 39 72 1.10 34 1.20 0.70 0.70 0.40 0.40 50 50 75 75 1.3 1.3 0.3 0.3 350 350 40 40 15 Units V V µA mV 1.10 1.20 0.87 ON/OFF Input Bias Current Gate Drive Current (Sinking) Max 0.01 57 LM2975I-X.X (Note 3) Min 180 VON/OFF = 0V VCL 1.240 LM2975AI-X.X (Note 3) 19 15 V µA mA µA VG(MIN) Gate Clamp Voltage VIN = 24V VFB = 0.9 X VREG 17 19 V R(VIN-G) Resistance from Gate to VIN VIN = 24V VON/OFF = 0 500 kΩ Open Loop Voltage Gain VIN = 15V 0.5V ≤ VGATE ≤ 13 5000 V/V Note 1: Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the device outside of its rated operating conditions. Note 2: The LP2975 has internal thermal shutdown which activates at a die temperature of about 150˚C. It should be noted that the power dissipated within the LP2975 is low enough that this protection circuit should never activate due to self-heating, even at elevated ambient temperatures. 3 www.national.com Electrical Characteristics (Continued) Note 3: Limits are 100% production tested at 25˚C. Limits over the operating temperature range are guaranteed through correlation using Statistical Quality Control (SQC) methods. The limits are used to calculate National’s Average Outgoing Quality Level (AOQL). Typical Application Circuits 5V - 3.3V @ 5A LDO Regulator DS100034-3 *See Application Hints. **If current limiting is not required, short out this resistor. Adjustable Voltage 5A LDO Regulator DS100034-4 *See Application Hints. ***If current limiting is not required, short out this resistor. www.national.com 4 Typical Performance Characteristics Unless otherwise specified: TA = 25˚C, CIN = 1 µF, ON/OFF pin is tied to 1.5V. VIN Referred Gate Clamp Voltage Minimum Operating Voltage DS100034-5 ON/OFF Threshold DS100034-6 Current Limit Sense Voltage DS100034-7 ON/OFF Pin Current DS100034-8 Supply Current DS100034-10 DS100034-9 5 www.national.com Typical Performance Characteristics Unless otherwise specified: TA = 25˚C, CIN = 1 µF, ON/OFF pin is tied to 1.5V. (Continued) ON/OFF Input Resistance Gate Current DS100034-34 DS100034-11 Gate-Ground Saturation Line Regulation DS100034-14 DS100034-13 Load Regulation Leakage Current DS100034-35 DS100034-15 www.national.com 6 Typical Performance Characteristics Unless otherwise specified: TA = 25˚C, CIN = 1 µF, ON/OFF pin is tied to 1.5V. (Continued) Controller Gain and Phase Response DS100034-36 maximum output voltage. The measured change in output voltage is divided by the nominal output voltage and expressed as a percentage. VIN = 5.6V for this test. Reference Designs The LP2975 controller can be used with virtually any P-channel MOSFET to build a wide variety of linear voltage regulators. Since it would be impossible to document all the different voltage and current combinations that could be built, a number of reference designs will be presented along with performance data for each. THE PERFORMANCE DATA SHOWN IS ACTUAL TEST DATA, BUT IS NOT GUARANTEED. 5 mA ≤ IL ≤ 5A: LOAD REGULATION = 0.012% 0 ≤ IL ≤ 5A: LOAD REGULATION = 0.135% Line Regulation Line regulation is defined as the maximum change in output voltage as the input voltage is varied. It is measured by changing the input voltage and recording the minimum/ maximum output voltage. The measured change in output voltage is divided by the nominal output voltage and expressed as a percentage. IL = 5A for this test. 5.4V ≤ VIN ≤ 10V: LINE REGULATION = 0.03% DESIGN #1: VOUT = 5V @ 5A (Refer to Typical Application Circuits) COMPONENTS: CIN = 82 µF Aluminum Electrolytic COUT = 120 µF Aluminum Electrolytic CF = 220 pF RSC = 10 mΩ P-FET = NDP6020P Heatsink: (assuming VIN ≤ 7V and TA ≤ 60˚C) if protection against a continuous short-circuit is required, a heatsink with θS-A ≤ 1.5 ˚C/W must be used. However, if continuous shortcircuit survivability is not needed, a heatsink with θS-A ≤ 6 ˚C/W is adequate. Output Noise Voltage Output noise voltage was measured by connecting a wideband AC voltmeter (HP 400E) directly across the output capacitor. VIN = 6V and IL = 5A for this test. NOISE = 75 µV (rms) Transient Response Transient response is defined as the change in output voltage which occurs after the load current is suddenly changed. VIN = 5.6V for this test. The load resistor is connected to the regulator output using a switch so that the load current increases from 0 to 5A abruptly. The change in output voltage is shown in the scope photo below (the vertical scale is 200 mV/division and the horizontal scale is 10 µs/division). The regulator nominal output (5V) is located on the center line of the photo. The output shows a maximum change of about −600 mV compared to nominal. This is due to the relatively small output capacitor chosen for this design. Increasing COUT greatly improves transient response (see Designs #2 and #3). PERFORMANCE DATA: Dropout Voltage Dropout voltage is defined as the minumum input-to-output differential voltage required by the regulator to keep the output in regulation. It is measured by reducing VIN until the output voltage drops below the nominal value (the nominal value is the output voltage measured with VIN = 5.5V). IL = 5A for this test. DROPOUT VOLTAGE = 323 mV Load Regulation Load regulation is defined as the maximum change in output voltage as the load current is varied. It is measured by changing the load resistance and recording the minimum/ 7 www.national.com Reference Designs maximum output voltage. The measured change in output voltage is divided by the nominal output voltage and expressed as a percentage. VIN = 3.5V for this test. (Continued) 0 ≤ IL ≤ 0.5A: LOAD REGULATION = 0.034% Line Regulation Line regulation is defined as the maximum change in output voltage as the input voltage is varied. It is measured by changing the input voltage and recording the minimum/ maximum output voltage. The measured change in output voltage is divided by the nominal output voltage and expressed as a percentage. IL = 0.5A for this test. 3.5V ≤ VIN ≤ 6V: LINE REGULATION = 0.017% Output Noise Voltage Output noise voltage was measured by connecting a wideband AC voltmeter (HP 400E) directly across the output capacitor. VIN = 5V and IL = 0.5A for this test. NOISE = 85 µV (rms) DS100034-37 Transient Response Transient response is defined as the change in output voltage which occurs after the load current is suddenly changed. VIN = 3.5V for this test. The load resistor is connected to the regulator output using a switch so that the load current increases from 0 to 0.5A abruptly. The change in output voltage is shown in the scope photo (the vertical scale is 20 mV/division and the horizontal scale is 50 µs/division). The regulator nominal output (3V) is located on the center line of the photo. A maximum change of about −50 mV is shown. Transient Response for 0–5A Load Step DESIGN #2: VOUT = 3V @ 0.5A (Refer to Typical Application Circuits, Adjustable Voltage Regulator) COMPONENTS: CIN = 68 µF Tantalum COUT = 2 X 68 µF Tantalum CC = 470 pF R1 = 237 kΩ, 1% R2 = NOT USED RSC = 0.1Ω Tie feedback pin to VOUT P-FET = NDT452P Heatsink: Tab of N-FET is soldered down to 0.6 in2 copper area on PC board. Output Voltage Adjustment: For this application, a 3.3V part is “trimmed” down to 3V by using a single external 237 kΩ resistor at R1, which parallels the internal 39.9 kΩ resistor (reducing the effective resistance to 34.2 kΩ). Because the tempco of the external resistor will not match the tempco of the internal resistor (which is typically 3000 ppm), this method of adjusting VOUT by using a single resistor is only recommended in cases where the output voltage is adjusted ≤ 10% away from the nominal value. PERFORMANCE DATA: DS100034-38 Dropout Voltage Transient Response for 0–0.5A Load Step Dropout voltage is defined as the minimum input-to-output differential voltage required by the regulator to keep the output in regulation. It is measured by reducing VIN until the output voltage drops below the nominal value (the nominal value is the output voltage measured with VIN = 5V). IL = 0.5A for this test. Minimizing COUT It is often desirable to decrease the value of COUT to save cost and reduce size. The design guidelines suggest selecting COUT to set the first pole ≤ 200 Hz (see later section Output Capacitor), but this is not an absolute requirement in all cases. The effect of reducing COUT is to decrease phase margin. As phase margin is decreased, the output ringing will increase when a load step is applied to the output. Eventually, if COUT is made small enough, the regulator will oscillate. To demonstrate these effects, the value of COUT in reference design #2 is halved by removing one of the two 68 µF output capacitors and the transient response test is repeated (see DROPOUT VOLTAGE = 141 mV Load Regulation Load regulation is defined as the maximum change in output voltage as the load current is varied. It is measured by changing the load resistance and recording the minimum/ www.national.com 8 Reference Designs DESIGN #3: VOUT = 1.5V @ 6A. (Refer to Typical Application Circuits, Adjustable Voltage Regulator) (Continued) photo below). The total overshoot increases from −50 mV to about −75 mV, and the second “ring” on the transient is noticeably larger. COMPONENTS: CIN = 1000 µF Aluminum Electrolytic COUT = 4 X 330 µF OSCON Aluminum Electrolytic CC = NOT USED R1 = 261Ω, 1% R2 = 1.21 kΩ, 1% RSC = 6 mΩ P-FET = NDP6020P Heatsink: (Assuming VIN ≤ 3.3V and TA ≤ 60˚C) if protection against a continuous short-circuit is required, a heatsink with θS-A < 2.5 ˚C/W must be used. However, if continuous shortcircuit survivability is not needed, a heatsink with θS-A < 7 ˚C/W is adequate. PERFORMANCE DATA: Dropout Voltage Dropout voltage is defined as the minimum input-to-output differential voltage required by the regulator to keep the output in regulation. It is measured by reducing VIN until the output voltage drops below the nominal value (the nominal value is the output voltage measured with VIN = 3.3V). IL = 6A for this test. DROPOUT VOLTAGE = 0.68V DS100034-39 Transient Response with Output Capacitor Halved The design is next tested with only a 4.7 µF output capacitor (see scope photo below). Observe that the vertical scale has been increased to 100 mV/division to accommodate the −250 mV undershoot. More important is the severe ringing as the transient decays. Most designers would recognize this immediately as the warning sign of a marginally stable design. Load Regulation Load regulation is defined as the maximum change in output voltage as the load current is varied. It is measured by changing the load resistance and recording the minimum/ maximum output voltage. The measured change in output voltage is divided by the nominal output voltage and expressed as a percentage. VIN = 3.3V for this test. 0 ≤ IL ≤ 6A: LOAD REGULATION = 0.092% Line Regulation Line regulation is defined as the maximum change in output voltage as the input voltage is varied. It is measured by changing the input voltage and recording the minimum/ maximum output voltage. The measured change in output voltage is divided by the nominal output voltage and expressed as a percentage. IL = 6A for this test. 3.3V ≤ VIN ≤ 5V: LINE REGULATION = 0.033% Output Noise Voltage Output noise voltage was measured by connecting a wideband AC voltmeter (HP 400E) directly across the output capacitor. VIN = 3.3V and IL = 6A for this test. NOISE = 60 µV (rms) DS100034-40 Transient Response with Only 4.7 µF Output Cap The reason this design is marginally stable is that the 4.7 µF output capacitor (along with the 6Ω output load) sets the pole fp at 5 kHz. Analysis shows that the unity-gain frequency of the loop is increased to about 100 kHz, allowing the FET’s gate capacitance pole fpg to cause significant phase shift before the loop gain goes below unity. Also, because of the low output voltage, the feedforward capacitor provides less than 10˚ of positive phase shift. For good stability, the output capcitor needs to be larger than 4.7 µF. Transient Response Transient response is defined as the change in output voltage which occurs after the load current is suddenly changed. VIN = 3.3V for this test. The load resistor is connected to the regulator output using a switch so that the load current increases from 0 to 6A abruptly. The change in output voltage is shown in the scope photo (the vertical scale is 50 mV/division and the horizontal scale is 20 µs/division. The regulator nominal output (1.5V) is located on the center line of the photo. A maximum change of about −80 mV is shown. For detailed information on stability and phase margin, see the Application Hints section. 9 www.national.com Reference Designs means faster changes in Gate voltage (which corresponds to faster transient response) will occur with a smaller amount of Gate capacitance. (Continued) 2) The Gate capacitance forms a pole in the loop gain which can reduce phase margin. When possible, this pole should be kept at a higher frequency than the cross-over frequency of the regulator loop (see later section CROSS-OVER FREQUENCY AND PHASE MARGIN). A high value of Gate capacitance may require that a feedforward capacitor be used to cancel some of the excess phase shift (see later section FEED-FORWARD CAPACITOR) to prevent loop instability. POWER DISSIPATION: The maximum power dissipated in the FET in any application can be calculated from: PMAX = (VIN − VOUT) x IMAX Where the term IMAX is the maximum output current. It should be noted that if the regulator is to be designed to withstand short-circuit, a current sense resistor must be used to limit IMAX to a safe value (refer to section SHORT-CIRCUIT CURRENT LIMITING). The power dissipated in the FET determines the best choice for package type. A TO-220 package device is best suited for applications where power dissipation is less than 15W. Power levels above 15W would almost certainly require a TO-3 type device. In low power applications, surface-mount package devices are size-efficient and cost-effective, but care must be taken to not exceed their power dissipation limits. DS100034-41 Transient Response for 0–6A Load Step Application Hints SELECTING THE FET The best choice of FET for a specific application will depend on a number of factors: VOLTAGE RATING: The FET must have a Drain-to-Source breakdown voltage (sometimes called BVDSS) which is greater than the input voltage. DRAIN CURRENT: On-state Drain current must be specified to be greater than the worst-case (short circuit) load current for the application. TURN-ON THRESHOLD: The Gate-to-Source voltage where the FET turns on (called the Gate Threshold Voltage) is very important. Many FET’s are intended for use with G-to-S voltages in the 5V to 10V range. These should only be used in applications where the input voltage is high enough to provide > 5V of drive to the Gate. Newer FET’s are becoming available with lower turn-on thresholds (Logic-Level FET’s) which turn on fully with a gate voltage of only 3V to 4V. Low threshold FET’s should be used in applications where the input voltage is ≤ 5V. ON RESISTANCE: FET on resistance (often called RDSON) is a critical parameter since it directly determines the minimum input-to-output voltage required for operation at a given load current (also called dropout voltage). RDSON is highly dependent on the amount of Gate-toSource voltage applied. For example, the RDSON of a FET with VG-S = 5V will typically decrease by about 25% as the VG-S is increased to 10V. RDSON is also temperature dependent, increasing at higher temperatures. POWER DISSIPATION AND HEATSINKING Since the LP2975 controller is suitable for use with almost any external P-FET, it follows that designs can be built which have very high power dissipation in the pass FET. Since the controller can not protect the FET from overtemperature damage, thermal design must be carefully done to assure a reliable design. THERMAL DESIGN METHOD: The temperature of the FET and the power dissipated is defined by the equation: TJ = (θJ-A x PD) + TA Where: TJ is the junction temperature of the FET. TA is the ambient temperature. PD is the power dissipated by the FET. θJ-A is the junction-to-ambient thermal resistance. To ensure a reliable design, the following guidelines are recommended: 1) Design for a maximum (worst-case) FET junction temperature which does not exceed 150˚C. 2) Heatsinking should be designed for worst-case (maximum) values of TA and PD. 3) In designs which must survive a short circuit on the output, the maximum power dissipation must be calculated assuming that the output is shorted to ground: PD(MAX) = VIN x ISC The dropout voltage of any LDO design is directly related to RDSON, as given by: VDROPOUT = ILOAD x (RDSON + RSC) Where RSC is the short-circuit current limit set resistor (see Application Circuit). Where ISC is the short-circuit output current. 4) If the design is not intended to be short-circuit proof, the maximum power dissipation for intended operation will be: PD(MAX) = (VIN − VOUT) x IMAX Where IMAX is the maximum output current. LOW POWER ( < 2W) APPLICATIONS: In most cases, some type of small surface-mount device will be used for the GATE CAPACITANCE: Selecting a FET with the lowest possible Gate capacitance improves LDO performance in two ways: 1) The Gate pin of the LP2975 (which drives the Gate of the FET) has a limited amount of current to source or sink. This www.national.com 10 Application Hints of 0.3–0.5˚C/W. The best source of information for this is heatsink catalogs (Wakefield, AAVID, Thermalloy) since they also sell mounting hardware. θS-A is the heatsink-to-ambient thermal resistance, which defines how well a heatsink transfers heat into the air. Once this is determined, a heatsink must be selected which has a value which is less than or equal to the computed value. The value of θS-A is usually listed in the manufacturer’s data sheet for a heatsink, but the information is sometimes given in a graph of temperature rise vs. dissipated power. (Continued) FET in low power designs. Because of the increased cell density (and tiny packages) used by modern FET’s, the current carrying capability may easily exceed the power dissipation limits of the package. It is possible to parallel two or more FET’s, which divides the power dissipation among all of the packages. It should be noted that the “heatsink” for a surface mount package is the copper of the PC board and the package itself (direct radiation). DESIGN EXAMPLE: A design is to be done which takes 3.3V in and provides 2.5V out at a load current of 7A. The power dissipation will be calculated for both normal operation and short circuit conditions. For normal operation: PD = (VIN − VOUT) x ILOAD = 5.6W Surface-mount devices have the value of θJ-A specified for a typical PC board mounting on their data sheet. In most cases it is best to start with the known data for the application (PD, TA, TJ) and calculate the required value of θJ-A needed. This value will define the type of FET and, possibly, the heatsink required for cooling. If the output is shorted to ground: PD(SC) = VIN x ISC = 3.3 x 7.7 = 25.4W (Assuming that a sense resistor is selected to set the value of ISC 10% above the nominal 7A). θJ-A will be calculated assuming a maximum TA of 70˚C and a maximum TJ of 150˚C: θJ-A = (TJ − TA)/PD(MAX) For normal operation: θJ-A = (150 − 70) / 5.6 = 14.3˚C/W For designs which must operate with the output shorted to ground: θJ-A = (150 − 70) / 25.4 = 3.2˚C/W θJ-A = (TJ − TA)/PD(MAX) DESIGN EXAMPLE: A design is to be done with VIN = 5V and VOUT = 3.3V with a maximum load current of 300 mA. Based on these conditions, power dissipation in the FET during normal operation would be: PD = (VIN − VOUT) x ILOAD Solving, we find that PD = 0.51W. Assuming that the maximum allowable value of TJ is 150˚C and the maximum TA is 70˚C, the value of θJ-A is found to be 157˚C/W. However, if this design must survive a continuous short on the output, the power dissipated in the FET is higher: PD(SC) = VIN x ISC = 5 x 0.33 = 1.65W (This assumes the current sense resistor is selected for an ISC value that is 10% higher than the required 0.3A). The value of θJ-A required to survive continuous short circuit is calculated to be 49˚C/W. Having solved for the value(s) of θJ-A, a FET can be selected. It should be noted that a FET must be used with a θJ-A value less than or equal to the calculated value. HIGH POWER (≥2W) APPLICATIONS: As power dissipation increases above 2W, a FET in a larger package must be used to obtain lower values of θJ-A. The same formulae derived in the previous section are used to calculate PD and θJ-A. Having found θJ-A, it becomes necessary to calculate the value of θS-A (the heatsink-to-ambient thermal resistance) so that a heatsink can be selected: θS-A = θJ-A − (θJ-C + θC-S) Where: θJ-C is the junction-to-case thermal resistance. This parameter is the measure of thermal resistance between the semiconductor die inside the FET and the surface of the case of the FET where it mounts to the heatsink (the value of θJ-C can be found on the data sheet for the FET). A typical FET in a TO-220 package will have a θJ-C value of approximately 2–4˚C/W, while a device in a TO-3 package will be about 0.5–2˚C/W. θC-S is the case-to-heatsink thermal resistance, which measures how much thermal resistance exists between the surface of the FET and the heatsink. θC-S is dependent on the package type and mounting method. A TO-220 package with mica insulator and thermal grease secured to a heatsink will have a θC-S value in the range of 1–1.5˚C/W. A TO-3 package mounted in the same manner will have a θC-S value The value of 14.3˚C/W can be easily met using a TO-220 device. Calculating the value of θS-A required (assuming a value of θJ-C = 3˚C/W and θC-S = 1˚C/W): θS-A = θJ-A − (θJ-C + θC-S) θS-A = 14.3 − (3 + 1) = 10.3˚C/W Any heatsink may be used with a thermal resistance ≤ 10.3˚C/W @ 5.6W power dissipation (refer to manufacturer’s data sheet curves). Examples of suitable heatsinks are Thermalloy #6100B and IERC #LATO127B5CB. However, if the design must survive a sustained short on the output, the calculated θJ-A value of 3.2˚C/W eliminates the possibility of using a TO-220 package device. Assuming a TO-3 device is selected with a θJ-C value of 1.5˚C/W and θC-S = 0.4˚C/W, we can calculate the required value of θS-A: θS-A = θJ-A − (θJ-C + θC-S) θS-A = 3.2 − (1.5 + 0.4) = 1.3˚C/W A θS-A value ≤1.3˚C/W would require a relatively large heatsink, or possibly some kind of forced airflow for cooling. SHORT-CIRCUIT CURRENT LIMITING Short-circuit current limiting is easiliy implemented using a single external resistor (RSC). The value of RSC can be calculated from: RSC = VCL / ISC Where: ISC is the desired short circuit current. VCL is the current limit sense voltage. The value of VCL is 57 mV (typical), with guaranteed limits listed in the Electrical Characteristics section. When doing a worst-case calculation for power dissipation in the FET, it is important to consider both the tolerance of VCL and the tolerance (and temperature drift) of RSC. 11 www.national.com Application Hints For best results in most designs, the frequency of fz should fall between 5 kHz and 50 kHz. It must be noted that the values of COUT and ESR usually vary with temperature (severely in the case of aluminum electrolytics), and this must be taken into consideration. (Continued) For maximum accuracy, the INPUT and CURRENT LIMIT pins must be Kelvin connected to RSC, to avoid errors caused by voltage drops along the traces carrying the current from the input supply to the Source pin of the FET. For the design example (VOUT = 5V @ 1A), select a capacitor which meets the fz requirements. Solving the equation for ESR yields: EXTERNAL CAPACITORS The best capacitors for use in a specific design will depend on voltage and load current (examples of tested circuits for several different output voltages and currents are provided in a previous section.) Information in the next sections is provided to aid the designer in the selection of the external capacitors. INPUT CAPACITOR: Although not always required, an input capacitor is recommended. Good bypassing on the input assures that the regulator is working from a source with a low impedance, which improves stability. A good input capacitor can also improve transient response by providing a reservoir of stored energy that the regulator can utilize in cases where the load current demand suddenly increases. The value used for CIN may be increased without limit. Refer to the Reference Designs section for examples of input capacitors. OUTPUT CAPACITOR: The output capacitor is required for loop stability (compensation) as well as transient response. During sudden changes in load current demand, the output capacitor must source or sink current during the time it takes the control loop of the LP2975 to adjust the gate drive to the pass FET. As a general rule, a larger output capacitor will improve both transient response and phase margin (stability). The value of COUT may be increased without limit. OUTPUT CAPACITOR AND COMPENSATION: Loop compensation for the LP2975 is derived from COUT and, in some cases, the feed-forward capacitor CF (see next section). COUT forms a pole (referred to as fp) in conjuction with the load resistance which causes the loop gain to roll off (decrease) at an additional −20 dB/decade. The frequency of the pole is: fp = 0.16 / [ (RL + ESR) x COUT] ESR = 0.16 / (fz x COUT) Assuming fz = 5 kHz and 50 kHz, the limiting values of ESR for the 180 µF capacitor are found to be: 18 mΩ ≤ ESR ≤ 0.18Ω A good-quality, low-ESR capacitor type such as the Panasonic HFQ is a good choice. However, the 10V/180 µF capacitor (#ECA-1AFQ181) has an ESR of 0.3Ω which is not in the desired range. To assure a stable design, some of the options are: 1) Use a different type capacitor which has a lower ESR such as an organic-electrolyte OSCON. 2) Use a higher voltage capacitor. Since ESR is inversely proportional to the physical size of the capacitor, a higher voltage capacitor with the same C value will typically have a lower ESR (because of the larger case size). In this example, a Panasonic ECA-1EFQ181 (which is a 180 µF/25V part) has an ESR of 0.17Ω and would meet the desired ESR range. 3) Use a feed-forward capacitor (see next section). FEED-FORWARD CAPACITOR: Although not required in every application, the use of a feed-forward capacitor (CF) can yield improvements in both phase margin and transient response in most designs. The added phase margin provided by CF can prevent oscillations in cases where the required value of COUT and ESR can not be easily obtained (see previous section). CF can also reduce the phase shift due to the pole resulting from the Gate capacitance, stabilizing applications where this pole occurs at a low frequency (before cross-over) which would cause oscillations if left uncompensated (see later section GATE CAPACITANCE POLE FREQUENCY). Even in a stable design, adding CF will typically provide more optimal loop response (faster settling time). For these reasons, the use of a feed-forward capacitor is always recommended. Where: RL is the load resistance. COUT is the value of the output capacitor. ESR is the equivalent series resistance of COUT. As a general guideline, the frequency of fp should be ≤ 200 Hz. It should be noted that higher load currents correspond to lower values of RL, which requires that COUT be increased to keep fp at a given frequency. DESIGN EXAMPLE: Select the minimum required output capacitance for a design whose output specifications are 5V @ 1A: fp = 0.16 / [ (RL + ESR) x COUT] Re-written: COUT = 0.16 / [fp x (RL + ESR) ] Values used for the calculation: fp = 200 Hz, RL = 5Ω, ESR = 0.1Ω (assumed). Solving for COUT, we get 157 µF (nearest standard size would be 180 µF). The ESR of the output capacitor is very important for stability, as it creates a zero (fz) which cancels much of the phase shift resulting from one of the poles present in the loop. The frequency of the zero is calculated from: fz = 0.16 / (ESR x COUT) www.national.com CF is connected across the top resistor in the divider used to set the output voltage (see Typical Application Circuit). This forms a zero in the loop response (defined as fzf), whose frequency is: fzf = 6.6 x 10−6 / [CF x (VOUT / 1.24 − 1) ] When solved for CF, the fzf equation is: CF = 6.6 x 10−6 / [fzf x (VOUT / 1.24 − 1) ] For most applications, fzf should be set between 5 kHz and 50 kHz. ADJUSTING THE OUTPUT VOLTAGE If an output voltage is required which is not available as a standard voltage, the LP2975 can be used as an adjustable regulator (see Typical Application circuit). The external resistors R1 and R2 (along with the internal 24 kΩ resistor) set the output voltage. It is important to note that R2 is connected in parallel with the internal 24 kΩ resistor. If we define REQ as the total resistance between the COMP pin and ground, then its value will be the parallel combination of R2 and 24 kΩ: 12 Application Hints Where: RL is the load resistance. ESR is the equivalent series resistance of the output capacitor. (Continued) REQ = (R2 x 24k) / (R2 + 24k) It follows that the output voltage will be: VOUT = 1.24 [ (R1 / REQ) + 1] Some important considerations for an adjustable design: The tolerance of the internal 24 kΩ resistor is about ± 20%. Also, its temperature coefficient is almost certainly different than the TC of the external resistor that is used for R2. The term RL / / ESR is defined as: (RL x ESR) / (RL + ESR) It can be seen from these equations that CEFF varies with RL. To get the worst-case (maximum) value for CEFF, use the maximum value of load current, which also means the minimum value of load resistance RL. It should be noted that in most cases, the ESR is the dominant term which determines the value of RL / / ESR. For these reasons, it is recommended that R2 be set at a value that is not greater than 1.2k. In this way, the value of R2 will dominate REQ, and the tolerance and TC of the internal 24k resistor will have a negligible effect on output voltage accuracy. Gate Pin Output Impedance To determine the value for R1: R1 = REQ [ (VOUT / 1.24) − 1] External Capacitors (Adjustable Application) All information in the previous section EXTERNAL CAPACITORS applies to the adjustable application with the exception of how to select the value of the feed-forward capacitor. The feed-forward capacitor CC in the adjustable application (see Typical Application Circuit) performs exactly the same function as described in the previous section FEEDFORWARD CAPACITOR. However, because R1 is userselected, a different formula must be used to determine the value of CC: CC = 1 / (2 π x R1 x fzf) As stated previously, the optimal frequency at which to place the zero fzf is usually between 5 kHz and 50 kHz. DS100034-20 Gate Capacitance Pole Frequency (fpg) The pole frequency resulting from the Gate capacitance CEFF is defined as fpg and can be approximated from: fpg ≅ 0.16 / (RO x CEFF) Where: RO is the output impedance of the LP2975 Gate pin which drives the Gate of the FET. It is important to note that RO is a function of input supply voltage (see graph GATE PIN OUTPUT IMPEDANCE). As shown, the minimum value of RO is about 550Ω @ VIN = 24V, increasing to about 1.55 kΩ @ VIN = 3V. Using the equation for fpg, a family of curves are provided showing how fpg varies with CEFF for several values of RO (see graph fpg vs. CEFF): OPTIMIZING DESIGN STABILITY Because the LP2975 can be used with a variety of different applications, there is no single set of components that are best suited to every design. This section provides information which will enable the designer to select components that optimize stability (phase margin) for a specific application. Gate Capacitance An important consideration of a design is to identify the frequency of the pole which results from the capacitance of the Gate of the FET (this pole will be referred to as fpg). As fpg gets closer to the loop crossover frequency, the phase margin is reduced. Information will now be provided to allow the total Gate capacitance to be calculated so that fpg can be approximated. fpg vs. CEFF The first step in calculating fp is to determine how much effective Gate capacitance (CEFF) is present. The formula for calculating CEFF is: CEFF = CGS + CGD [1 + Gm (RL / / ESR) ] Where: CGS is the Gate-to-Source capacitance, which is found from the values (refer to FET data sheet for values of CISS and CRSS): CGS = CISS − CRSS GGD is the Gate-to-Drain capacitance, which is equal to: CGD = CRSS Gm is the transconductance of the FET. The FET data sheet specifies forward transconductance (Gfs) at some value of drain current (defined as ID). To find Gm at the desired value of load current (defined as IL), use the formula: Gm = Gfs x (IL / ID)1/2 DS100034-21 13 www.national.com Application Hints next section FEED-FORWARD COMPENSATION). This can improve phase margin by cancelling some of the excess phase shift. (Continued) As can be seen in the graph, values of CEFF in the 500 pF–2500 pF range produce values for fpg between 40 kHz and 700 kHz. To determine what effect fpg will have on stability, the bandwidth of the regulator loop must be calculated (see next section CROSSOVER FREQUENCY AND PHASE MARGIN). Feed-Forward Compensation Phase shift in the loop gain of the regulator results from fp (the pole from the output capacitor and load resistance), fpg (the pole from the FET gate capacitance), as well as the IC’s internal controller pole (see typical curve). If the total phase shift becomes excessive, instability can result. Crossover Frequency and Phase Margin The term fc will be used to define the crossover frequency of the regulator loop (which is the frequency where the gain curve crosses the 0 dB axis). The importance of this frequency is that it is the point where the loop gain goes below unity, which marks the usable bandwidth of the regulator loop. It is the phase margin (or lack of it) at fc that determines whether the regulator is stable. Phase margin is defined as the total phase shift subtracted from 180˚. In general, a stable loop requires at least 20˚-30˚ of phase margin at fc. fc can be approximated by the following equation (all terms have been previously defined): The total phase shift can be reduced using feed-forward compensation, which places a zero in the loop to reduce the effects of the poles. The feed-forward capacitor CF can accomplish this, provided it is selected to set the zero at the correct frequency. It is important to point out that the feed-forward capacitor produces both a zero and a pole. The frequency where the zero occurs will be defined as fzf, and the frequency of the pole will be defined as fpf. The equations to calculate the frequencies are: fzf = 6.6 x 10-6/ [CF x (VOUT/1.24 − 1) ] fpf = 6.6 x 10-6/ [CF x (1 − 1.24/VOUT)] In general, the feed-forward capacitor gives the greatest improvement in phase margin (provides the maximum reduction in phase shift) when the zero occurs at a frequency where the loop gain is > 1 (before the crossover frequency). The pole must occur at a higher frequency (the higher the better) where most of the phase shift added by the new pole occurs beyond the crossover frequency. For this reason, the pole-zero pair created by CF become more effective at improving loop stability as they get farther apart in frequency. In reviewing the equations for fzf and fpf, it can be seen that they get closer together in frequency as VOUT decreases. For this reason, the use of CF gives greatest benefit at higher output voltages, declining as VOUT approaches 1.24V (where CF has no effect at all). In selecting a value of feed-forward capacitor, the crossover frequency fc must first be calculated. In general, the frequency of the zero (fzf) set by this capacitor should be in the range: 0.2 fc ≤ fzf ≤ 1.0 fc DS100034-23 This equation assumes that no CF is used and fpg/fc > 1. If the frequency of the Gate capacitance pole fpg has been calculated (previous section), the amount of added phase shift may now be determined. As shown in the graph below (see graph PHASE SHIFT DUE TO fpg), the amount of added phase shift increases as fpg approaches fc. The amount of phase shift due to fpg that can occur before oscillation takes place depends on how much added phase shift is present as a result of the COUT pole (see previous section OUTPUT CAPACITOR). Phase Shift Due to fpg The equation to determine the value of the feed-forward capacitor in fixed-voltage applications is: CF = 6.6 x 10-6/ [fzf x (VOUT/1.24 − 1) ] In adjustable applications (using an external resistive divider) the capacitor is found using: CC = 1/(2 π x R1 x fzf) SUMMARY OF STABILITY INFORMATION This section will present an explanation of theory and terminology used to analyze loop stability, along with specific information related to stabilizing LP2975 applications. BODE PLOTS AND PHASE SHIFT Loop gain information is most often presented in the form of a Bode Plot, which plots Gain (in dB) versus Frequency (in Hertz). A Bode Plot also conveys phase shift information, which can be derived from the locations of the poles and zeroes. DS100034-22 Because of this, there is no exact number for fpg/fc that can be given as a fixed limit for stable operation. However, as a general guideline, it is recommended that fpg ≥ 3 fc. If this is not found to be true after inital calculations, the ratio of fpg/fc can be increased by either reducing CEFF (selecting a different FET) or using a larger value of COUT. POLE: A pole causes the slope of the gain curve to decrease by an additional −20 dB/decade, and it also causes phase lag (defined as negative phase shift) to occur. A single pole will cause a maximum −90˚ of phase lag (see graph EFFECTS OF A SINGLE POLE). It should be noted Along with these two methods, another technique for improving loop stability is the use of a feed-forward capacitor (see www.national.com 14 Application Hints STABILITY ANALYSIS OF TYPICAL APPLICATIONS (Continued) The first application to be analyzed is a fixed-output voltage regulator with no feed-forward capacitor (see graph STABLE PLOT WITHOUT FEED-FORWARD). that when the total phase shift at 0 dB reaches (or gets close to) −180˚, oscillations will result. Therefore, it can be seen that at least two poles in the gain curve are required to cause instability. ZERO: A zero has an effect that is exactly opposite to a pole. A zero will add a maximum +90˚ of phase lead (defined as positive phase shift). Also, a zero causes the slope of the gain curve to increase by an additional +20 dB/decade (see graph EFFECTS OF A SINGLE ZERO). Stable Plot without Feed-Forward Effects of a Single Pole DS100034-27 In this example, the value of COUT is selected so that the pole formed by COUT and RL (previously defined as fp) is set at 200 Hz. The ESR of COUT is selected so that zero formed by the ESR and COUT (defined as fz) is set at 5 kHz (these selections follow the general guidelines stated previously in this document). Note that the gate capacitance is assumed to be moderate, with the pole formed by the CGATE (defined as fpg) occurring at 100 kHz. To estimate the total phase margin, the individual phase shift contributions of each pole and zero will be calculated assuming fp = 200 Hz, fz = 5 kHz, fc = 10 kHz and fpg = 100 kHz: Controller pole shift = −90˚ fp shift = −arctan (10k/200) = −89˚ fz shift = arctan (10k/5k) = +63˚ fpg shift = −arctan (10k/100k) = −6˚ Summing the four numbers, the estimate for the total phase shift is −122˚, which corresponds to a phase margin of 58˚. This application is stable, but could be improved by using a feed-forward capacitor (see next section). EFFECT OF FEED-FORWARD: The example previously used will be continued with the addition of a feed-forward capacitor CF (see graph IMPROVED PHASE MARGIN WITH FEED-FORWARD). The zero formed by CF (previously defined as fzf) is set at 10 kHz and the pole formed by CF (previously defined as fpf) is set at 40 kHz (the 4X ratio of fpf/fzf corresponds to VOUT = 5V). DS100034-25 TOTAL PHASE SHIFT: The actual test of whether or not a regulator is stable is the amount of phase shift that is present when the gain curve crosses the 0 dB axis (the frequency where this occurs was previously defined as fc). The phase shift at fc can be estimated by looking at all of the poles and zeroes on the Bode plot and adding up the contributions of phase lag and lead from each one. As shown in the graphs, most of the phase lag (or lead) contributed by a pole (or zero) occurs within one decade of the frequency of the pole (or zero). In general, a phase margin (defined as the difference between the total phase shift and −180˚) of at least 20˚ to 30˚ is required for a stable loop. Effects of a Single Zero DS100034-26 15 www.national.com Application Hints (Continued) High ESR Unstable without Feed-Forward Improved Phase Margin with Feed-Forward DS100034-29 As shown, moving the location of fz lower in frequency extends the bandwidth, pushing the crossover frequency fc out to about 200 kHz. In viewing the plot, it can be seen that fp and fz essentially cancel out, leaving only the controller pole and fpg. However, since fpg now occurs well before fc, it will cause enough phase shift to leave very little phase margin. This application would either oscillate continuously or be marginally stable (meaning it would exhibit severe ringing on transient steps). This can be improved by adding a feed-forward capacitor CF, which adds a zero (fzf) and a pole (fpf) to the gain plot (see graph HIGH ESR CORRECTED WITH FEED-FORWARD). In this case, CF is selected to place fzf at about the same frequency as fpg (essentially cancelling out the phase shift due to fpg). Assuming the added pole fpf is near or beyond the fc frequency, it will add < 45˚ of phase lag, leaving a phase margin of > 45˚ (adequate for good stability). DS100034-28 To estimate the total phase margin, the individual phase shift contributions of each pole and zero will be calculated assuming fp = 200 Hz, fz = 5 kHz, fzf = 10 kHz, fpf = 40 kHz, fc = 50 kHz, and fpg = 100 kHz: Controller pole shift = −90˚ fp shift = −arctan (50k/200) = −90˚ fz shift = arctan (50k/5k) = +84˚ fzf shift = arctan (50k/100k) = +79˚ fpf shift = −arctan (50k/40k) = −51˚ fpg shift = −arctan (50k/100k) = −27˚ Summing the six numbers, the estimate for the total phase shift is −95˚, which corresponds to a phase margin of 85˚ (a 27˚ improvement over the same application without the feed-forward capacitor). For this reason, a feed-forward capacitor is recommended in all applications. Although not always required, the added phase margin typically gives faster settling times and provides some design guard band against COUT and ESR variations with temperature. High ESR Corrected with Feed-Forward CAUSES AND CURES OF OSCILLATIONS The most common cause of oscillations in an LDO application is the output capacitor ESR. If the ESR is too high or too low, the zero (fz) does not provide enough phase lead. HIGH ESR: To illustrate the effect of an output capacitor with high ESR, the previous example will be repeated except that the ESR will be increased by a factor of 20X. This will cause the frequency of the zero fz to decrease by 20X, which moves it from 5 kHz down to 250 Hz (see graph HIGH ESR UNSTABLE WITHOUT FEED-FORWARD). DS100034-31 LOW ESR: To illustrate how an output capacitor with low ESR can cause an LDO regulator to oscillate, the same example will be shown except that the ESR will be reduced sufficiently to increase the original fz from 5 kHz to 50 kHz. The plot now shows (see graph LOW ESR UNSTABLE WITHOUT FEED-FORWARD) that the crossover frequency fc has moved down to about 8 kHz. Since fz is 6X fc, it means that the zero fz can only provide about 9˚ of phase lead at fc, which is not sufficient for stability. www.national.com 16 Application Hints The use of a feed-forward capacitor CF will help reduce excess phase shift due to fpg, but its effectiveness depends on output voltage (see next section). (Continued) Low ESR Unstable without Feed-Forward LOW OUTPUT VOLTAGE AND CF The feed-forward capacitor CF will provide a positive phase shift (lead) which can be used to cancel some of the excess phase lag from any of the various poles present in the loop. However, it is important to note that the effectiveness of CF decreases with output voltage. This is due to the fact that the frequencies of the zero fzf and pole fpf get closer together as the output voltage is reduced (see equations in section FEED-FORWARD COMPENSATION). CF is more effective when the pole-zero pair are farther apart, because there is less self cancellation. The net benefit in phase shift provided by CF is the difference between the lead (positive phase shift) from fzf and the lag (negative phase shift) from fpf which is present at the crossover frequency fc. As the pole and zero frequency approach each other, that difference diminishes to nothing. The amount of phase lead at fC provided by CF depends both on the fzf/fpf ratio and the location of fz with respect to fc. To illustrate this more clearly, a graph is provided which shows how much phase lead can be obtained for VOUT = 12V, 5V, and 3.3V (see graph PHASE LEAD PROVIDED BY CF). DS100034-30 This application can also be improved by adding a feedforward capacitor. CF will add both a zero fzf and pole fpf to the gain plot (see graph LOW ESR CORRECTED WITH FEED-FORWARD). The crossover frequency fc is now about 10 kHz. If CF is selected so that fzf is about 5 kHz, and fpf is about 20 kHz (which means VOUT = 5V), the phase margin will be considerably improved. Calculating out all the poles and zeroes, the phase margin is increased from 9˚ to 43˚ (adequate for good stability). Phase Lead Provided by CF Low ESR Corrected with Feed-Forward DS100034-33 The most important information on the graph is the frequency range of fzf which will provide the maximum benefit (most positive phase shift): For VOUT = 12V: 0.1 fc < fz < 1.0 fc For VOUT = 5V: 0.2 fc < fz < 1.2 fc For VOUT = 3.3V: 0.2 fc < fz < 1.3 fc DS100034-32 EXCESSIVE GATE CAPACITANCE: Higher values of gate capacitance shift the pole fpg to lower frequencies, which can cause stability problems (see previous section GATE CAPACITANCE POLE FREQUENCY). As shown in the graph fpg vs. CEFF, the pole fpg will likely fall somewhere between 40 kHz and 500 kHz. How much phase shift this adds depends on the crossover frequency fc. The effect of gate capacitance becomes most important at high values of ESR for the output capacitor (see graph HIGH ESR UNSTABLE WITHOUT FEED-FORWARD). Higher values of ESR increase fc, which brings fpg more into the positive gain portion of the curve. As fpg moves to a lower frequency (corresponding to higher values of gate capacitance), this effect becomes even worse. This points out why FET’s should be selected with the lowest possible gate capacitance: it makes the design more tolerant of higher ESR values on the output capacitor. It’s also important to note how the maximum available phase shift that CF can provide drops off with VOUT. At 12V, more than 50˚ can be obtained, but at 3.3V less than 30˚ is possible. The lesson from this is that higher voltage designs are more tolerant of phase shifts from both fpg (the gate capacitance pole) and incorrect placement of fz (which means the output capacitor ESR is not at its nominal value). At lower values of VOUT, these parameters must be more precisely selected since CF can not provide as much correction. GENERAL DESIGN PROCEDURE Assuming that VIN, VOUT, and RL are defined: 17 www.national.com Application Hints maximum phase gain for the output voltage selected (see previous section LOW OUTPUT VOLTAGE AND CF). The formula for calculating CF is in the previous section FEEDFORWARD CAPACITOR. Lower ESR electrolytics are available which use organic electrolyte (OSCON types), but are more costly than typical aluminum electrolytics. If the calculated value of ESR is higher than what is found in the selected capacitor, an external resistor can be placed in series with COUT. LOW VOLTAGE DESIGNS: Designs which have a low output voltage (where the positive effects of CF are very small) may be marginally stable if the COUT and ESR values are not carefully selected. Also, if the FET gate capacitance is large (as in the case of a high-current FET), the pole fpg could possibly get low enough in frequency to cause a problem. (Continued) 1) Calculate the required value of capacitance for COUT so that the pole fp ≤ 200 Hz (see previous section OUTPUT CAPACITOR). For this calculation, an ESR of about 0.1Ω can be assumed for the purpose of determining COUT. IMPORTANT: If a smaller value of output capacitor is used (so that the value of fp > 200 Hz), the phase margin of the control loop will be reduced. This will result in increased ringing on the output voltage during a load transient. If the output capacitor is made extremely small, oscillations will result. To illustrate this effect, scope photos have been presented showing the output voltage of reference design #2 as the output capacitor is reduced to approximately 1/30 of the nominal value (the value which sets fp = 200 Hz). As shown, the effect of deviating from the nominal value is gradual and the regulator is quite robust in resisting going into oscillations. 2) Approximate the crossover frequency fc using the equation in the previous section CROSSOVER FREQUENCY AND PHASE MARGIN. 3) Calculate the required ESR of the output capacitor so that the frequency of the zero fz is set to 0.5 fc (see previous section OUTPUT CAPACITOR). 4) Calculate the value of the feed-forward capacitor CF so that the zero fzf occurs at the frequency which yields the www.national.com The solution in both cases is to increase the amount of output capacitance which will shift fp to a lower frequency (and reduce overall loop bandwidth). The ESR and CF calculations should be repeated, since this changes the crossover frequency fc. 18 LP2975 MOSFET LDO Driver/Controller Physical Dimensions inches (millimeters) unless otherwise noted Surface Mount Mini SO-8 Package NS Package Number MUA08A LIFE SUPPORT POLICY NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user. National Semiconductor Corporation Americas Tel: 1-800-272-9959 Fax: 1-800-737-7018 Email: [email protected] www.national.com National Semiconductor Europe Fax: +49 (0) 1 80-530 85 86 Email: [email protected] Deutsch Tel: +49 (0) 1 80-530 85 85 English Tel: +49 (0) 1 80-532 78 32 Français Tel: +49 (0) 1 80-532 93 58 Italiano Tel: +49 (0) 1 80-534 16 80 2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. National Semiconductor Asia Pacific Customer Response Group Tel: 65-2544466 Fax: 65-2504466 Email: [email protected] National Semiconductor Japan Ltd. Tel: 81-3-5639-7560 Fax: 81-3-5639-7507 National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.

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