NSC LM25037MT

LM25037/LM25037Q
Dual-Mode PWM Controller With Alternating Outputs
General Description
Features
The LM25037 PWM controller contains all the features necessary to implement balanced double-ended power converter
topologies, such as push-pull, half-bridge and full-bridge.
These double-ended topologies allow for higher efficiencies
and greater power densities compared to common singleended topologies such as the flyback and forward. The
LM25037 can be configured for either voltage mode or current
mode control with minimum external components. Two alternating gate drive outputs are provided, each capable of 1.2A
peak output current. The LM25037 can be configured to operate directly from the input voltage rail over an ultra-wide
range of 5.5V to 75V. Additional features include programmable maximum duty cycle limit, line under-voltage lockout, cycle-by-cycle current limit and a hiccup mode fault
protection with adjustable timeout delay, soft-start and a 2
MHz capable oscillator with synchronization capability, precision reference and thermal shutdown.
■ LM25037Q is an Automotive Grade product that is AEC■
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Q100 grade 1 qualified (-40°C to 125°C operating junction
temperature)
Ultra wide input operating range from 5.5V to 75V
Alternating outputs for double-ended topologies
Current-mode or feed-forward voltage-mode control
Programmable maximum duty cycle limit
2.0% feedback reference accuracy
High gain-bandwidth error amplifier
Programmable line under-voltage lockout (UVLO) with
adjustable hysteresis
Versatile dual mode over-current protection with hiccup
delay timer
Programmable soft-start
Precision 5V reference output
Current sense leading edge blanking
Resistor programmed 2 MHz capable oscillator
Oscillator synchronization capability with low frequency
lockout protection
Package
■ TSSOP-16
Simplified Push-Pull Power Converter
30065101
© 2009 National Semiconductor Corporation
300651
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LM25037/LM25037Q Dual-Mode PWM Controller With Alternating Outputs
June 30, 2009
LM25037/LM25037Q
Connection Diagram
30065102
Top View
16-Lead TSSOP Package
Ordering Information
Order Number
Package Type NSC Package Drawing
Supplied As
LM25037MT
92 Units per Rail
LM25037MTX
2500 Units on Tape and Reel
LM25037QMT
LM25037QMTX
TSSOP-16
MTC16
Feature
92 Units per Rail
AEC-Q100 Grade 1 qualified.
Automotive
Grade Production Flow*
2500 Units on Tape and Reel
*Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.
Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified
with the letter Q. For more information go to http://www.national.com/automotive.
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2
Pin
Name
Description
Application Information
1
RAMP
Pulse width modulator ramp
Modulation ramp for the PWM comparator. This ramp can be a representative
of the primary current (current mode) or proportional to input voltage (feedforward voltage mode). This pin is reset to ground at the conclusion of every
cycle by an internal FET.
2
UVLO
Line under-voltage lockout
An external voltage divider from the power source sets the shutdown and
standby comparator threshold levels. When UVLO exceeds the 0.45V
shutdown threshold, the VCC and REF regulators are enabled. When UVLO
exceeds the 1.25V standby threshold, the SS pin is released and the device
enters the active mode.
3
COMP
Input to the pulse width modulator
Output of the error amplifier and input to the PWM comparator.
4
FB
Feedback
Connected to inverting input of the error amplifier. An internal 1.25V reference
is connected to the non-inverting input of the error amplifier. In isolated
applications using an external error amplifier, this pin should be connected to
the AGND pin.
5
RT2
Oscillator dead-time control
The resistance connected between RT2 and AGND sets the forced dead-time
between switching periods of the alternating outputs.
6
AGND
Analog ground
Connect directly to Power Ground.
7
RT1
Oscillator maximum on-time control
The resistance connected between RT1 and AGND sets the oscillator
maximum on-time. The sum of this maximum on-time and the forced deadtime (set by RT2) sets the oscillator period.
8
CS
Current sense input
If CS exceeds 250 mV the output pulse will be terminated, entering cycle-bycycle current limit. An internal switch holds CS low for 65 nS after either output
switches high to blank leading edge transients.
9
RES
Restart timer
If cycle-by-cycle current limit is reached during any cycle, a 18 µA current is
sourced to the external RES pin capacitor. If the RES capacitor voltage
reaches 2.0V, the soft-start capacitor will be fully discharged and then
released with a pull-up current of 1 uA. After the first output pulse (when SS
= 1V), the SS pin charging current will increase to the normal level of 100 µA.
10
SS
Soft-start
An external capacitor and an internal 100uA current source set the soft-start
ramp. The SS current source is reduced to 1 µA following a restart event (RES
pin high).
11
PGND
Power ground
Connect directly to Analog Ground
12
OUTB
Output driver
Alternating gate drive output of the pulse width modulator. Capable of 1.2A
peak source and sink current.
13
OUTA
Output driver
Alternating gate drive output of the pulse width modulator. Capable of 1.2A
peak source and sink current.
14
VCC
Output of the high voltage start-up
regulator. The VCC voltage is
regulated to 7.7V.
If an auxiliary winding raises the voltage on this pin above the regulation set
point, the internal start-up regulator will shutdown thus reducing the IC power
dissipation. Locally decouple VCC with a 0.47 µF or greater capacitor.
15
REF
Output of a 5V reference
Locally decouple with a 0.1 µF or greater capacitor. Maximum output current
is 10 mA (typ).
16
VIN
Input voltage source
Input to the VCC Start-up regulator. Operating input range is 5.5V to 75V. For
power sources outside of this range, the LM25037 can be biased directly at
VCC by an external regulator.
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LM25037/LM25037Q
Pin Descriptions
LM25037/LM25037Q
Absolute Maximum Ratings (Note 1)
ESD Rating
Human Body Model
Storage Temperature Range
Junction Temperature
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN to GND
VCC, RAMP , OUTA, OUTB to GND
CS to GND
UVLO, FB, RT2, RT1, SS, REF to GND
COMP, RES(Note 2)
-0.3V to 76V
-0.3V to 16V
-0.3V to 1.0V
-0.3V to 7V
2kV
−65°C to + 150°C
150°C
Operating Ratings
VIN Voltage
External Voltage Applied to VCC
Operation Junction Temperature
5.5V to 75V
8V to 14V
−40°C to + 125°C
Electrical Characteristics
Specifications with standard typeface are for TJ = 25°C, and those with boldface type
apply over full operating junction temperature range of −40°C to + 125°C. VVIN = 12V, VVCC = 10V, RRT1 = 30.1 kΩ, RRT2 = 30.1
kΩ, VUVLO = 3V unless otherwise stated.
Symbol
Parameter
Conditions
Min
Typ
Max
7.7
8.1
Units
Startup Regulator (VCC Pin)
VCC voltage
IVCC = 10 mA
7.2
IVCC(Lim)
VVCC
VCC current limit
VVCC = 7V
20
VVCC(UV)
VCC Under-voltage threshold
4.6
Hysteresis
IVIN
Startup regulator current
Supply current into VCC from
external source
V
mA
5
5.4
V
µA
0.5
V
VVIN = 20V, VUVLO = 0V
35
58
VVIN = 75V, VUVLO = 0V
45
80
Outputs & COMP open, VVCC = 10V,
Outputs switching
4
µA
mA
Voltage Reference Regulator (REF Pin)
VREF
IREF(Lim)
REF Voltage
IREF = 0 mA
REF Voltage Regulation
IREF = 0 to 2.5 mA
REF Current Limit
VREF = 4.5V
VREF Under-Voltage Threshold
VREF(UV)
4.75
5
5.15
V
7
25
mV
5
10
3.7
4
Hysteresis
mA
4.3
0.4
V
V
Under-Voltage Lock Out and Shutdown (UVLO Pin)
VUVLO
Under-voltage threshold
IUVLO
Hysteresis current
UVLO pin sinking
Under-voltage Shutdown
Threshold
UVLO voltage rising
1.20
1.25
1.295
V
17
22
26
µA
0.35
0.45
0.6
V
Hysteresis
0.1
V
Current Sense Input (CS Pin)
VCS
Current Limit Threshold
CS delay to output
0.22
CS from zero to 1V. Time for OUTA and
OUTB to fall to 90% of VCC. Output load
= 0 pF.
Leading edge blanking time at CS
CS sink impedance (clocked)
0.255
0.29
V
27
ns
65
ns
21
45
Ω
1.9
2
2.2
V
Internal FET sink impedance
Current Limit Restart (RES Pin)
VRES
RES Threshold
Charge source current
VRES = 1.5V
14
18
21
µA
Discharge sink current
VRES = 1V
5
8
11
µA
Charging current in normal
operation
VSS = 0
70
100
130
µA
Charging current during a hiccup
mode restart
VSS = 0
0.6
1
1.4
µA
Soft-Start (SS Pin)
ISS
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4
Parameter
Soft-Stop Current Sink
Conditions
Min
Typ
Max
Units
VSS = 2.0V
70
100
130
µA
40
75
105
ns
Oscillator (RT1 and RT2 Pins)
DT1
Dead-Time 1
RRT2 = 15 kΩ
DT2
Dead-Time 2
RRT2 = 50 kΩ
FSW1
Frequency 1 (at OUTA, half
oscillator frequency)
RRT1 = 30.1 kΩ,
FSW2
Frequency 2 (at OUTA, half
oscillator frequency)
RRT1 = 11 kΩ,
250
ns
178
200
222
kHz
448
515
578
kHz
RRT2 = 30.1 kΩ,
RRT2 = 30.1 kΩ,
DC level
2
Input Sync threshold
2.5
3
V
3.4
V
PWM Controller (Comp Pin)
Delay to output
VPWM-OS
65
SS to RAMP offset
ns
0.7
1
1.2
V
Minimum duty cycle
VSS = 0V
0
%
COMP Open Circuit Voltage
VFB = 0V
4.5
4.75
5.0
V
COMP short circuit current
VFB = 0V, COMP = 0V
0.5
1
1.5
mA
RAMP sink impedance(Clocked)
5
20
Ω
Gain Bandwidth
4
Voltage Feed-Forward (RAMP Pin)
Error Amplifier
GBW
DC Gain
MHz
75
Input Voltage
VFB = COMP
COMP sink capability
VFB = 1.5V COMP=1V
dB
1.22
1.245
5
13
mA
10
nA
FB Bias Current
1.27
V
Main Output Drivers (OUTA and OUTB Pins)
Output high voltage
IOUT = 50 mA, (Source)
Vcc-0.5 Vcc-0.
25
V
Output low voltage
IOUT = 100 mA (Sink)
0.2
Rise time
CLOAD = 1 nF
17
ns
Fall time
CLOAD = 1 nF
18
ns
Peak source current
VVCC = 10V
1.2
A
Peak sink current
VVCC = 10V
1.2
A
Thermal Shutdown Threshold
165
°C
Thermal Shutdown Hysteresis
25
°C
0.5
V
Thermal Shutdown
TSD
Thermal Resistance
θJC
Junction to Case
TSSOP-16
29
°C/W
θJA
Junction to Ambient
TSSOP-16
125
°C/W
Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the
device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.
Note 2: COMP, RES are output pins. As such, it is not recommended that external power sources be connected to these pins.
Note 3: All limits are guaranteed. All electrical characteristics having room temperature limits are tested during production at TA = 25°C. All hot and cold limits
are guaranteed by correlating the electrical characteristics to process and temperature variations and applying statistical process control.
Note 4: Typical specifications represent the most likely parametric norm at 25°C operation.
Note 5: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin. 2kV for all pins.
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LM25037/LM25037Q
Symbol
LM25037/LM25037Q
Typical Performance Characteristics
Typical Application Circuit Efficiency
VVCC and VREF vs VVIN
30065150
30065103
Start-Up Regulator Current (UVLO = 0)
VVCC vs IVCC
30065104
30065105
VREF vs IREF
Feedback Amplifier Gain/Phase
30065107
30065106
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6
LM25037/LM25037Q
Oscillator Frequency vs RT1
Dead-Time vs RT2
30065108
30065110
VFB vs Temperature
Oscillator Frequency vs Temperature
30065112
30065111
Dead-Time vs Temperature
Soft-Start and Restart Current vs Temperature
30065113
30065114
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LM25037/LM25037Q
Block Diagram
30065118
FIGURE 1. Simplified Block Diagram
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8
The LM25037 PWM controller contains all the features necessary to implement double-ended power converter topologies such as push-pull, half-bridge and full-bridge. The unique
architecture allows the modulator to be configured for either
voltage-mode or current-mode control. The LM25037 provides two alternative gate driver outputs to drive the primary
side power MOSFETs with programmable forced dead-time.
The LM25037 can be configured to operate with bias voltages
ranging from 5.5V to 75V. Additional features include line under-voltage lockout, cycle-by-cycle current limit, voltage feedforward compensation, and hiccup mode fault protection with
adjustable delays, soft-start, and a 2MHz capable oscillator
with synchronization capability, precision reference and thermal shutdown. These rich set of features simplify the design
of double ended topologies. The functional block diagram is
shown in Figure 1.
REFERENCE
The REF pin is the output of a 5V linear regulator that can be
used to bias an opto-coupler transistor and external housekeeping circuits. The regulator output is internally current
limited to 10 mA (typical).
HIGH-VOLTAGE START-UP REGULATOR
The LM25037 contains an internal high voltage, low drop-out
start-up regulator that allows the input pin (VIN) to be connected directly to the supply voltage over a range of 5.5V to
a maximum of 75V. The regulator output at VCC (7.7V) is
internally current limited with a guaranteed minimum of 20
mA. When the UVLO pin potential is greater than 0.45V, the
VCC regulator is enabled to charge an external capacitor
connected to the VCC pin. The VCC regulator provides power
to the voltage reference (REF) and the gate drivers (OUTA
and OUTB). When the voltage on the VCC pin exceeds its
Under Voltage (VCC UV) threshold of 5.0V nominal, the internal voltage reference (REF) reaches its regulation set point
of 5V and the UVLO voltage is greater than 1.25V, the controller outputs are enabled. The value selected for the VCC
capacitor depends on the total system design, and its startup characteristics. The recommended range of values for the
VCC capacitor is 0.47 µF to 10 µF.The internal power dissipation of the LM25037 can be reduced by powering VCC from
an external supply. In typical applications, an auxiliary transformer winding is connected through a diode to the VCC pin.
This winding must raise the VCC voltage above 8.2V to shut
off the internal start-up regulator. Powering VCC from an auxiliary winding improves efficiency while reducing the
controller’s power dissipation. The VCC UV circuit will still
function in this mode, requiring that VCC never falls below
5.0V nominal during the start-up sequence. The VCC regulator series pass transistor includes a diode between VCC
and VIN that should not be forward biased in normal operation. Therefore the auxiliary VCC voltage should never exceed the VIN voltage.
An external DC bias voltage can be used instead of the internal regulator by connecting the external bias voltage to both
the VCC and the VIN pins. In this particular case, the external
bias must be greater than max VCC UV of 5.4V and less than
the VCC maximum operating voltage rating (14V).
ERROR AMPLIFIER
An internal high gain error amplifier is provided within the
LM25037. The amplifier’s non-inverting reference is tied to a
1.25V reference. In non-isolated applications the power converter output is connected to the FB pin via the voltage setting
resistors and loop compensation is connected between the
COMP and FB pins. A typical gain/phase plot is shown in
performance curves section.
For most isolated applications the error amplifier function is
implemented on the secondary side. Since the internal error
amplifier is configured as an open drain output, it can be disabled by connecting FB to ground. The internal 5K pull-up
resistor connected between the COMP pin and the 5V reference can be used as the pull-up for an opto-coupler or other
isolation device .
CYCLE-BY-CYCLE CURRENT LIMIT
The CS pin is to be driven by a signal representative of the
transformer primary current. The current sense signal can be
generated by using a sense resistor or a current sense transformer. If the voltage sensed at the CS pin exceeds 0.255V,
the current sense comparator terminates the output driver
pulse. If the high current condition persists, the controller operates in a cycle-by-cycle current limit mode with duty cycle
determined by the current sense comparator instead of the
PWM comparator. Cycle-by-cycle current limiting may eventually trigger the hiccup mode restart cycle; depending on the
configuration of the RES pin (see Overload Protection Timer
below). To suppress noise, a small R-C filter connected to the
CS pin and located near the controller is recommended. An
internal 21Ω MOSFET discharges the external current sense
filter capacitor at the conclusion of every cycle. The discharge
MOSFET remains on for an additional 65 ns after either OUTA
or OUTB driver switches high to blank leading edge transients
in the current sensing circuit. Discharging the CS pin filter
each cycle and blanking leading edge spikes reduces the filtering requirements and improves the current sense response
time. The current sense comparator is very fast and may respond to short duration noise pulses. Layout considerations
are critical for the current sense filter and sense resistor. The
capacitor associated with the CS filter must be placed very
close to the device and connected directly to the CS and AGND pins. If a sense resistor located in the source of the main
MOSFET switch is used for current sensing, a low inductance
LINE UNDER-VOLTAGE DETECTOR
The LM25037 contains a dual level line Under-Voltage Lock
Out (UVLO) circuit. When the UVLO pin voltage is below
0.45V, the controller is in a low current shutdown mode. When
the UVLO pin voltage is greater than 0.45V but less than
1.25V, the controller is in standby mode. In standby mode the
VCC and REF bias regulators are active while the controller
outputs are disabled. When the VCC and REF outputs exceed
their respective under-voltage thresholds and the UVLO pin
voltage is greater than 1.25V, the outputs are enabled and
normal operation begins. An external set-point voltage divider
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LM25037/LM25037Q
from VIN to GND can be used to set the minimum operating
voltage of the converter. The divider must be designed such
that the voltage at the UVLO pin will be greater than 1.25V
when VIN enters the desired operating range. UVLO hysteresis is accomplished with an internal 22 µA current source that
is switched on or off into the impedance of the set-point divider. When the UVLO pin voltage exceeds 1.25V threshold,
the current source is activated to quickly raise the voltage at
the UVLO pin. When the UVLO pin voltage falls below the
1.25V threshold, the current source is disabled causing the
voltage at the UVLO pin to quickly fall. The hysteresis of the
0.45V shutdown comparator is internally fixed at 100 mV.
The UVLO pin can also be used to implement various remote
enable/disable functions. Turning off the converter by forcing
the UVLO pin to standby condition provides a controlled softstop. See the Soft-Start section for more details.
Functional Description
LM25037/LM25037Q
•
type of resistor is required. When designing with a current
sense resistor, all the noise sensitive, low power ground connections should be connected together near the AGND pin,
and a single connection should be made to the power ground
(sense resistor ground point).
If the overload condition no longer exists after restart, the
RES pin will be held at ground by the 8 µA current sink and
normal operation resumes.
The overload timer function is very versatile and can be configured for the following modes of protection:
1. Cycle-by-cycle only: The hiccup mode can be
completely disabled by connecting a zero to 50 kΩ
resistor from the RES pin to AGND. In this configuration,
the cycle-by-cycle protection will limit the output current
indefinitely and no hiccup sequences will occur.
2. Hiccup only: The timer can be configured for immediate
activation of a hiccup sequence upon detection of an
overload by leaving the RES pin open circuit. In this
configuration, the first detection of current limit condition
by the CS pin comparator will initiate a hiccup cycle with
SS capacitor fully discharged and a delayed restart.
3. Delayed Hiccup: Connecting a capacitor to the RES pin
provides a programmed interval of cycle-by-cycle limiting
before initiating a hiccup mode restart, as previously
described. The dual advantages of this configuration are
that a short term overload will not cause a hiccup mode
restart but during extended overload conditions, the
average dissipation of the power converter will be very
low.
4. Externally Controlled Hiccup: The RES pin can also
be used as an input. By externally driving the pin to a level
greater than the 2.0V hiccup threshold, the controller will
be forced into the delayed restart sequence. For
example, the external trigger for a delayed restart
sequence could come from an over-temperature
protection circuit or an output over-voltage sensor
OVERLOAD PROTECTION TIMER
The LM25037provides a current limit restart timer to disable
the outputs and force a delayed restart (hiccup mode) if a
current limit condition is repeatedly sensed. The number of
cycle-by-cycle current limit events required to trigger the
restart is programmed by the external capacitor at the RES
pin. During each PWM cycle, the LM25037 either sources to
or sinks current from the RES pin capacitor. If no current limit
is detected during a cycle, a 8 µA discharge current sink is
enabled to pull the RES pin towards ground. If a current limit
is detected, the 8 µA sink current is disabled and an 18 µA
current source causes the voltage at the RES pin to gradually
increase. The LM25037 protects the converter with cycle-bycycle current limiting while the voltage at RES pin increases.
If the RES voltage reaches the 2.0V threshold, the following
restart sequence occurs (also see Figure 2):
• The RES capacitor and SS capacitors are fully discharged.
• The soft-start current source is reduced from 100 µA to 1
µA.
• The SS capacitor voltage slowly increases. When the SS
voltage reaches ≊1V, the PWM comparator will produce
the first narrow output pulse. After the first pulse occurs,
the SS source current reverts to the normal 100 µA level.
The SS voltage increases at its normal rate, gradually
increasing the duty cycle of the output drivers.
• If the overload condition persists after restart, cycle-bycycle current limiting will begin to increase the voltage on
the RES capacitor again, repeating the hiccup mode
sequence.
30065119
FIGURE 2. Current Limit Restart Circuit
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LM25037/LM25037Q
30065120
FIGURE 3. Current Limit Restart Timing
SOFT-START
The soft-start circuit allows the regulator to gradually reach a
steady state operating point, thereby reducing start-up stresses and current surges. When bias is supplied to the LM25037,
the SS pin capacitor is discharged by an internal MOSFET.
When the UVLO, VCC and REF pins reach their operating
thresholds, the SS capacitor is released and charged with a
100 µA current source. The PWM comparator control voltage
at the COMP pin is clamped to the SS pin voltage by an internal amplifier. When the PWM comparator input reaches
1V, output pulses commence with slowly increasing duty cycle. The voltage at the SS pin eventually increases to 5V,
while the voltage at the PWM comparator increases to the
value required for regulation as determined by the voltage
feedback loop.
One method to disable the regulator is to ground the SS pin.
This forces the internal PWM control signal to ground, reducing the output duty cycle quickly to zero. Releasing the SS pin
initiates a soft-start sequence and normal operation resumes.
A second shutdown method is discussed in the UVLO section.
RAMP PIN
The voltage at the RAMP pin provides the modulation ramp
for the PWM comparator. The PWM comparator compares
the modulation ramp signal at the RAMP pin to the loop error
signal to control the output duty cycle. The modulation ramp
can be implemented either as a ramp proportional to input
voltage, known as feed-forward voltage mode control, or as
a ramp proportional to the primary current, known as current
mode control. The RAMP pin is reset by an internal FET with
an RDS(ON) of 5Ω (typical) at the end of every cycle. The ability
to configure the RAMP pin for either voltage mode or current
mode allows the controller to be implemented for the optimum
control method for the selected power stage topology. Configuring RAMP pin is explained below and the differences
between voltage mode control and current mode control in
various double-ended topologies is explained in Applications
Information section.
FEED-FORWARD VOLTAGE MODE
An external resistor (RFF) and capacitor (CFF) connected to
VIN, AGND, and the RAMP pins is required to create the
PWM ramp signal as shown in Figure 4 below. It can be seen
that the slope of the signal at RAMP will vary in proportion to
the input line voltage. This varying slope provides line feedforward information necessary to improve line transient response with voltage mode control. The RAMP signal is
compared to the error signal by the pulse width modulator
comparator to control the duty cycle of the outputs. With a
constant error signal, the on-time (tON) varies inversely with
the input voltage (VIN) to stabilize the Volt • Second product
of the transformer primary. At the end of clock period, an internal FET will be enabled to reset the CFF capacitor. The
formulae for RFF and CFF and component selection criteria are
explained in Applications Information section. The amplitude
of the signal driving RAMP pin must not exceed the common
mode input voltage range of the PWM comparator (3.3V)
while in normal operation.
PWM COMPARATOR
The pulse width modulation (PWM) comparator compares the
voltage ramp signal at the RAMP pin to the loop error signal.
The loop error signal is derived from the internal error amplifier (COMP pin). The resulting control voltage passes through
a 1V level shift before being applied to the PWM comparator.
This comparator is optimized for speed in order to achieve
minimum controllable duty cycles. The common mode input
voltage range of the PWM comparator is from 0 to 4.3V.
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LM25037/LM25037Q
30065121
FIGURE 4. Feed-Forward Voltage Mode Configuration
capacitor and CS capacitor are reset through internal discharge FETs. The RDS(ON) of RAMP discharge FET is 5Ω
(typical); this ensures fast discharge of the RAMP reset capacitor. Any dc voltage source can be used in place of VREF
to generate the slope compensation ramp.
The timing diagram shown in Figure 6 depicts the current
mode waveforms and relative timing. When OUTA or OUTB
is enabled, the signal at the RAMP pin consists of the CS pin
signal (current ramp on a pedestal) plus the slope compensation ramp (dotted lines). When OUTA or OUTB is turned
off, the primary current component is absent but the voltage
at the RAMP pin continues to rise due to slope compensation
component until the end of the clock period, after which it is
reset by the RAMP discharge FET. A component selection
example is explained in detail in the Applications Information
section.
CURRENT MODE
The LM25037 can be configured for current mode control by
injecting a signal representative of primary current into the
RAMP pin. One way to achieve this is shown in Figure 5. Filter
components Rfilter and Cfilter are used to filter leading edge
noise spikes. The signal at the CS pin is thus a ramp on a
pedestal. The pedestal corresponds to the continuous conduction current in the transformer at the beginning of an
OUTA or OUTB conduction cycle. The R-C circuit (RSlope and
CSlope), shown in Figure 5, tied to VREF adds an additional
ramp to the current sense signal. This additional ramp signal,
known as slope compensation, is required to avoid instabilities at duty cycles above 50% (25% per phase). The compensated RAMP signal consists of two parts, the primary
current signal and the slope compensation. The compensated RAMP signal is compared to the error signal by the PWM
comparator to control the duty cycle of the outputs. The RAMP
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12
LM25037/LM25037Q
30065122
FIGURE 5. Current Mode Configuration with Slope Compensation
30065123
FIGURE 6. Timing Diagram for Current Mode Configuration
13
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LM25037/LM25037Q
It is recommended to set the dead-time range between 50 ns
and 250 ns. Beyond 250 ns, RT2 becomes excessively large,
and is prone to noise pickup. Fixed internal delays limit the
dead-time to greater than 50 ns. After the dead-time has been
programmed by RT2, the overall oscillator frequency can be
set by selecting resistor RT1 from :
OSCILLATOR
The LM25037 oscillator frequency and the maximum duty cycle are set by two external resistors connected between the
RT1 and RT2 pins to AGND. The minimum dead-time between OUTA and OUTB pulses is proportional to the RT2
resistor value and the overall oscillator frequency is inversely
proportional to RT1 and RT2 resistor values. Each output
switches at half the oscillator frequency. Initially, RT2 should
be selected for the desired dead-time or for the desired maximum duty cycle (Dmax).
For example, if the desired oscillator frequency is 400 kHz
(OUTA and OUTB each switching at 200 kHz) and desired
dead-time is 100 ns, the maximum duty cycle for each output
will be 96% and the values of RT1 and RT2 will be 15 kΩ and
20 kΩ respectively.
30065127
FIGURE 7. Timing Diagram of OUTA, OUTB and Dead-Time Set by RT2
As shown in Figure 7, the internal clock pulse width is the
same as the dead-time set by RT2. This dead-time pulse is
used to limit the maximum duty cycle for each of the outputs.
Also, the discharge FET connected to the RAMP pin is enabled during the dead-time every clock period. The voltages
at both the RT1 and RT2 pins are internally regulated to a
nominal 2V. Both the resistors RT1 and RT2 should be located as close as possible to the IC, and connected directly to
the pins. The tolerance of the external resistors and the frequency tolerance indicated in the Electrical Characteristics
table must be taken into account when determining the worst
case frequency range.
60% of the clock period under all conditions. When the synchronizing pulse transitions from low-to-high (rising edge), the
voltage at the RT1 pin must be driven to exceed 3.0V from its
nominal 2.0V volt dc level. During the synchronization clock
signal’s low time, the voltage at the RT1 pin will be clamped
at 2V volts by an internal regulator. The RT1 and RT2 resistors are always required, whether the oscillator is free running
or externally synchronized.
GATE DRIVER OUTPUTS (OUTA & OUTB)
The LM25037 provides two alternating gate driver outputs,
OUTA and OUTB. The internal gate drivers can each source
and sink 1.2A peak each. The maximum duty cycle is inherently limited to less than 50% and is based on the value of
RT2 resistor. As an example, if the COMP pin is in a high
state, RT1 = 15K and RT2 = 20K then the outputs will operate
at maximum duty cycle of 96%.
SYNC CAPABILITY
The LM25037 can be synchronized to an external clock by
applying a narrow ac pulse to the RT1 pin. The external clock
must be at least 10% higher than the free-running oscillator
frequency set by the RT1 and RT2 resistors. If the external
clock frequency is less than the programmed frequency, the
LM25037 will ignore the synchronizing pulses. The synchronization pulse width at the RT1 pin must be a minimum of 15
ns wide. The synchronization signal should be coupled into
the RT1 pin through a 100 pF capacitor or another value small
enough to ensure the sync pulse width at RT1 is less than
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THERMAL PROTECTION
Internal Thermal Shutdown circuitry is provided to protect the
integrated circuit in the event the maximum rated junction
temperature is exceeded. When activated, typically at 165°C,
the controller is forced into a low power standby state with the
output drivers (OUTA and OUTB) and the bias regulators
14
Using a line feed-forward ramp for PWM control requires very
little change in the voltage regulation loop to compensate for
changes in input voltage, as compared to a fixed slope oscillator ramp. Furthermore, voltage mode control is less susceptible to noise and does not require leading edge filtering,
and is therefore a good choice for wide input range power
converters. Voltage mode control requires a more complicated compensation network, due to the complex-conjugate
poles of the L-C output filter.
In push-pull and full-bridge topologies, any asymmetry in the
volt-second product applied to primary in one phase may not
be cancelled by subsequent phase, possibly resulting in a dc
current build-up in the transformer, which pushes the transformer core towards saturation. Special care in the transformer design, such as gapping the core, or adding ballasting
resistance in the primary is required to rectify this imbalance
when using voltage mode control with these topologies. Current mode control naturally corrects for any volt-second asymmetry in the primary.
The recommended capacitor value range for CFF is 100 pF to
1500 pF. Referring to Figure 8, it can be seen that value
CFF must be small enough such that the capacitor can be discharged within the clock (CLK) pulse width each cycle. The
CLK pulse width is same as the dead-time set by RT2. The
minimum possible dead-time for LM25037 is 50 ns and the
internal discharge FET RDS(ON) is 5Ω (typical),
The value of RFF required can be calculated from
Application Information
The following information is intended to provide guidelines for
the design process when applying the LM25037.
TOPOLOGY and CONTROL ALGORITHM CHOICE
The LM25037 has all the features required to implement double-ended power converter topologies such as push-pull, halfbridge and full-bridge with minimum external components.
One key feature is the flexibility in control algorithm selection,
i.e., the LM25037 can be used to implement either voltage
mode control or current mode control. Designers familiar with
these topologies recognize that conventionally, current mode
control is used for push-pull and full-bridge topologies while
voltage mode control is required for the half-bridge topology.
In limited applications, voltage mode control can be used for
push-pull and full-bridge topologies as well, with special care
to maintain flux balance, such as using a dc-blocking capacitor in the primary (full-bridge). The goal of this section is to
illustrate implementation of both current mode control and
voltage mode control using the LM25037 and aid the designer
in the design process.
VOLTAGE MODE CONTROL USING THE LM25037
An external resistor (RFF) and capacitor (CFF) connected to
VIN, AGND, and the RAMP pins is required to create a sawtooth modulation ramp signal shown in Figure 8. The slope of
the signal at RAMP will vary in proportion to the input line
voltage. The varying slope provides line feed-forward information necessary to improve line transient response with
voltage mode control. With a constant error signal, the ontime (tON) varies inversely with the input voltage (VIN) to
stabilize the Volt • Second product of the transformer primary.
For example, assuming a VRamp of 1 volt at VINmin (a good
compromise of signal range and noise immunity), oscillator
frequency, FOSC of 250 kHz, VINmin of 24 Volts, and CFF = 270
pF results in a value for RFF of 348 kΩ.
30065121
FIGURE 8. Feed-Forward Voltage Mode Configuration
15
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LM25037/LM25037Q
(VCC and REF) disabled. This helps to prevent catastrophic
failures from accidental device overheating. During thermal
shutdown, the soft-start capacitor is fully discharged and the
controller follows a normal start-up sequence after the junction temperature falls to the operating level (140°C).
LM25037/LM25037Q
tered and applied to the RAMP pin through capacitor Cslope,
for use as the modulation ramp. It can be seen that the signal
applied to the RAMP pin consists of the primary current information from the CS pin plus an additional ramp for slope
compensation, added by Rslope and Cslope.
CURRENT MODE CONTROL USING THE LM25037
The LM25037 can be configured in current mode control by
applying the primary current signal into the RAMP pin. One
way to achieve this is shown in Figure 9, which depicts a simplified push-pull converter. The primary current is sensed
using a sense resistor and the current information is then fil-
30065130
FIGURE 9. Current Mode Configuration
Current mode control inherently provides line voltage feedforward, cycle-by-cycle current limiting and ease of loop compensation as it removes the additional pole due to output
inductor. Also, in push-pull and full-bridge converters, current
mode control inherently balances volt-second product in both
the phases by varying the duty cycle as needed to terminate
the cycle at the same peak current for each output phase. For
duty cycles greater than 50% (25% for each phase), peak
current mode controlled circuits are subject to sub-harmonic
oscillation. Sub-harmonic oscillation is normally characterized by observing alternating wide and narrow duty cycles at
the controller output. Adding an artificial ramp (slope compensation) to the current sense signal will eliminate this potential oscillation. Current mode control is also susceptible to
noise and layout considerations. It is recommended that
CFilter and Cslope be placed as close to the IC as possible to
avoid any noise pickup and trace inductance. When the converter is operating at low duty cycles and light load, the
primary current amplitude is small and is susceptible to noise.
The artificial ramp, added to avoid sub-harmonic oscillations,
provides additional benefits by improving the noise immunity
of the converter.
Configuration and component selection for current mode control is recommended as follows: The current sense resistor is
selected such that during over current condition, the voltage
across the current sense resistor is above the minimum CS
threshold of 220 mV. It is recommended to set the
impedances of RFilter and CFilter as seen from Cslope at relawww.national.com
tively low values, so that the slope compensation is primarily
dictated by Rslope and Cslope components. For example, if the
filtering time (RFilter and CFilter) for leading edge noise is selected for 50 ns and if the value selected for RFilter = 25Ω, then
Resulting in a value of CFilter = 680 pF (approximated to a
standard value). In general, the amount of slope compensation required to avoid sub-harmonic oscillation is equal to at
least one-half the down-slope of the output inductor current,
transformed to the primary. To mitigate sub-harmonic oscillation after one switching period, the slope compensation has
to be equal to one times the down slope of the filter inductor
current transformed to primary. This is known as deadbeat
control. For circuits where primary current is sensed, the
amount of slope compensation for dead-beat control can be
calculated from:
Where, Turns-Ratio is referred with respect to the primary.
For example, for a 5V output converter with a turns ratio between secondary and primary of 1:2, an oscillator frequency
16
LM25037/LM25037Q
(FOSC) of 250 kHz, a filter inductance of 4 µH (LFilter) and a
current sense resistor (RCS) of 32 mΩ, slope compensation
of 80 mV will suffice. The slope compensation "volts" that results from the above expression is the maximum voltage of
the artificial ramp added linearly to the RAMP pin till the end
of maximum switching period. For circuits where a current
sense tramsformer is used for primary current sensing, the
turns-ratio of the current sense transformer has to be taken
into account.
Cslope should be selected such that it can be fully discharged
by the internal RAMP discharge FET. Capacitor values ranging from 100 pF to 1500 pF are recommended. The value
must be small enough such that the capacitor can be discharged within the clock (CLK) pulse width each cycle.
Rslope can be selected from the following formula:
30065134
FIGURE 10. Input Transient Protection
FOR APPLICATIONS WITH >75V INPUT
For applications where the system input voltage exceeds 75V
or the IC power dissipation is of concern, the LM25037 can
be powered from an external start-up regulator as shown in
Figure 11. In this configuration, the VIN and the VCC pins
should be connected together. The voltage at the VCC and
VIN pins must be at least 5.5V (> Max VCC UV voltage) yet
not exceed 14V. An auxiliary winding can be used to reduce
the power dissipation in the external regulator once the power
converter is active. The NPN base-emitter reverses breakdown voltage, which can be as low as 5V for some transistors,
should be considered when selecting the transistor.
For example, with a Cslope of 1500 pF, FOSC of 250 kHz, reference voltage of 5V (VREF), slope compensation of 80 mV
and Rfilter = 25Ω results in Rslope value of 165 kΩ.
VIN and VCC
The voltage applied to the VIN pin, which may be the same
as the system voltage applied to the power transformer’s primary (VPWR), can vary in the range from 5.5V to 75V. The
current into the VIN pin depends primarily on the gate charge
provided by the output drivers, the switching frequency, and
any external loads on the VCC and REF pins. It is recommended that the filter shown in Figure 10 be used to suppress
transients that may occur at the input supply. This is particularly important when VIN is operated close to the maximum
operating rating of the LM25037.
When power is applied to VIN and the UVLO pin voltage is
greater than 0.45V, the VCC regulator is enabled and supplies current into an external capacitor connected to the VCC
pin. When the voltage on the VCC pin reaches the regulation
point of 7.7V, the voltage reference (REF) is enabled. The
reference regulation set point is 5V. The outputs (OUTA and
OUTB) are enabled when the two bias regulators reach their
set point and the UVLO pin potential is greater than 1.25V. In
typical applications, an auxiliary transformer winding is connected through a diode to the VCC pin. This winding must
raise the VCC voltage above 8.1V to shut off the internal startup regulator.
After the outputs are enabled and the external VCC supply
voltage has begun supplying power to the IC, the current into
the VIN pin drops below 1 mA. VIN should remain at a voltage
equal to or above the VCC voltage to avoid reverse current
through protection diodes.
30065135
FIGURE 11. Start-up Regulator for VPWR >75V
CURRENT SENSE
The CS pin should receive an input signal representative of
the transformer’s primary current, either from a current sense
transformer or from a resistor in series with the source of the
OUTA and OUTB MOSFET switches. In both cases, the
sensed current creates a voltage ramp across R1, and the
RF/CF filter suppresses noise and transients as shown in Figure 12 and Figure 13. R1, RF and CF should be located as
close to the LM25037 as possible, and the ground connection
from the current sense transformer, or R1, should be a dedicated track to the AGND pin. The current sense components
must provide greater than 220 mV at the CS pin when an overcurrent condition exists.
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LM25037/LM25037Q
30065136
FIGURE 12. Current Sense Using Transformer
30065137
FIGURE 13. Current Sense Using Current Sense Resistor (R1)
UVLO DIVIDER SELECTION
A dedicated comparator connected to the UVLO pin detects
an input under-voltage condition. When the UVLO pin voltage
is below 0.45V, the LM25037 controller is in a low current
shutdown mode. For a UVLO pin voltage greater than 0.45V
but less than 1.25V, the controller is in standby mode with
VCC and REF regulators active but no switching. Once the
UVLO pin voltage is greater than 1.25V, the controller is fully
enabled. When the UVLO pin voltage rises above the 1.25V
threshold, an internal 22 µA current source as shown in Figure
14, is activated thus providing threshold hysteresis. The 22
µA current source is deactivated when the voltage at the UVLO pin falls below 1.25V. Resistance values for R1 and R2
can be determined from the following equations:
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Where VPWR is the desired turn-on voltage and VHYS is the
desired UVLO hysteresis at VPWR.
18
LM25037/LM25037Q
30065139
FIGURE 14. Basic UVLO Configuration
For example, if the LM25037 is to be enabled when VPWR
reaches 33V, and disabled when VPWR decreases to 30V, R1
should be 113 kΩ, and R2 should be 4.42 kΩ. The voltage at
the UVLO pin should not exceed 7V at any time. Be sure to
check both the power and voltage rating (0603 resistors can
be rated as low as 50V) for the selected R1 resistor. To main-
tain the UVLO threshold accuracy, a resistor tolerance of 1%
or better is recommended.
Remote control of the LM25037 operational modes can be
accomplished with open drain device(s) connected to the UVLO pin as shown in Figure 15.
30065140
FIGURE 15. Remote Standby and Disable Control
19
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LM25037/LM25037Q
initiation of a hiccup cycle is programmed by the selection of
the RES pin capacitor CRES as illustrated in Figure 16.
HICCUP MODE CURRENT LIMIT RESTART (RES)
The basic operation of the hiccup mode current limit is described in the functional description. The delay time to the
30065120
FIGURE 16. Hiccup Over-Load Restart Timing
In the case of continuous cycle-by-cycle current limit detection at the CS pin, the time required for CRES to reach the 2.0V
hiccup mode threshold is:
should be as physically close as possible to the IC, thereby
minimizing noise pickup on the PC board trace inductances.
Layout considerations are critical for the current sense filter.
If a current sense transformer is used, both leads of the transformer secondary should be routed to the sense filter components and to the IC pins. The ground side of the transformer
should be connected via a dedicated PC board trace to the
AGND pin, rather than through the ground plane.
If the current sense circuit employs a sense resistor in the
drive transistor source, low inductance resistors should be
used. In this case, all the noise sensitive, low-current ground
trace should be connected in common near the IC, and then
a single connection made to the power ground (sense resistor
ground point).
While employing current mode control, RAMP pin capacitor
and CS pin capacitor must be placed close to the IC. Also, a
short direct trace should be employed to connect RAMP capacitor to the CS pin.
The gate drive outputs of the LM25037 should have short,
direct paths to the power MOSFETs in order to minimize inductance in the PC board The two ground pins (AGND,
PGND) must be connected together with a short, direct connection, to avoid jitter due to relative ground bounce.
If the internal dissipation of the LM25037 produces high junction temperatures during normal operation, the use of multiple
vias under the IC to a ground plane can help conduct heat
away from the IC. Judicious positioning of the PC board within
the end product, along with use of any available air flow
(forced or natural convection) will help reduce the junction
temperatures. If using forced air cooling, avoid placing the
LM25037 in the airflow shadow of tall components, such as
input capacitors.
For example, if CRES = 0.01 µF the time t1 is approximately
2.0 ms. The cool down time, t2 is set by the soft-start capacitor
(CSS) and the internal 1 µA SS current source, and is equal
to:
If CSS = 0.01 µF, t2 is ≊10 ms.
The soft-start time t3 is set by the internal 100 µA current
source, and is equal to:
If CSS = 0.01 µF, t3 is ≊ 400 µs.
The time t2 provides a periodic cool-down time for the power
converter in the event of a sustained overload or short circuit.
This off time results in lower average input current and lower
power dissipation within the power components. It is recommended that the ratio of t2 / (t1 + t3) be in the range of 5 to
10 to take advantage of this feature.
If the application requires no delay from the first detection of
a current limit condition to the onset of the hiccup mode (t1 =
0), the RES pin can be left open (no external capacitor). If it
is desired to disable the hiccup mode entirely, the RES pin
should be connected to ground (AGND).
APPLICATION EXAMPLE
The following schematic shows an example of a 50W pushpull converter controlled by LM25037. The operating input
voltage range is 16V to 32V, and the output voltage is 5V. The
output current capability is 10 Amps. The converter is configured for current mode control with external slope compensation. An auxiliary winding of the output filter inductor L3 is used
to raise the VCC voltage to reduce the power dissipation.
PRINTED CIRCUIT BOARD LAYOUT
The LM25037 Current Sense and PWM comparators are very
fast, and respond to short duration noise pulses. The components at the CS, COMP, SS, UVLO, RT2 and the RT1 pins
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20
FIGURE 17. Schematic
30065144
LM25037/LM25037Q
21
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LM25037/LM25037Q
Physical Dimensions inches (millimeters) unless otherwise noted
16-Lead TSSOP Package
NS Package Number MTC16
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22
LM25037/LM25037Q
Notes
23
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LM25037/LM25037Q Dual-Mode PWM Controller With Alternating Outputs
Notes
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