LM3488 High Efficiency Low-Side N-Channel Controller for Switching Regulators General Description The LM3488 is a versatile Low-Side N-FET high performance controller for switching regulators. It is suitable for use in topologies requiring low side FET, such as boost, flyback, SEPIC, etc. Moreover, the LM3488 can be operated at extremely high switching frequency in order to reduce the overall solution size. The switching frequency of LM3488 can be adjusted to any value between 100kHz and 1MHz by using a single external resistor or by synchronizing it to an external clock. Current mode control provides superior bandwidth and transient response, besides cycle-by-cycle current limiting. Output current can be programmed with a single external resistor. The LM3488 has built in features such as thermal shutdown, short-circuit protection and over voltage protection. Power saving shutdown mode reduces the total supply current to 5µA and allows power supply sequencing. Internal soft-start limits the inrush current at start-up. Key Specifications n ± 1.5% (over temperature) internal reference n 5µA shutdown current (over temperature) Features 8-lead Mini-SO8 (MSOP-8) package Internal push-pull driver with 1A peak current capability Current limit and thermal shutdown Frequency compensation optimized with a capacitor and a resistor n Internal softstart n Current Mode Operation n Undervoltage Lockout with hysteresis n n n n Applications n Distributed Power Systems n Notebook, PDA, Digital Camera, and other Portable Applications n Offline Power Supplies n Set-Top Boxes n Wide supply voltage range of 2.97V to 40V n 100kHz to 1MHz Adjustable and Synchronizable clock frequency Typical Application Circuit 10138844 Typical SEPIC Converter © 2005 National Semiconductor Corporation DS101388 www.national.com LM3488 High Efficiency Low-Side N-Channel Controller for Switching Regulators May 2005 LM3488 Connection Diagram 10138802 8 Lead Mini SO8 Package (MSOP-8 Package) Package Marking and Ordering Information Order Number Package Type Package Marking Supplied As: LM3488MM MSOP-8 S21B 1000 units on Tape and Reel LM3488MMX MSOP-8 S21B 3500 units on Tape and Reel Pin Description www.national.com Pin Name Pin Number ISEN 1 Current sense input pin. Voltage generated across an external sense resistor is fed into this pin. Description COMP 2 Compensation pin. A resistor, capacitor combination connected to this pin provides compensation for the control loop. FB 3 Feedback pin. The output voltage should be adjusted using a resistor divider to provide 1.26V at this pin. AGND 4 Analog ground pin. PGND 5 Power ground pin. DR 6 Drive pin of the IC. The gate of the external MOSFET should be connected to this pin. FA/SYNC/SD 7 Frequency adjust, synchronization, and Shutdown pin. A resistor connected to this pin sets the oscillator frequency. An external clock signal at this pin will synchronize the controller to the frequency of the clock. A high level on this pin for ≥ 30µs will turn the device off. The device will then draw less than 10µA from the supply. VIN 8 Power supply input pin. 2 LM3488 Absolute Maximum Ratings (Note 1) Lead Temperature MM Package Vapor Phase (60 sec.) Infared (15 sec.) If Military/Aerospace specified devices are required, please contact the National Semiconductor Sales Office/ Distributors for availability and specifications. 215˚C 220˚C 45V DR Pin Voltage −0.4V ≤ VDR ≤ 8V FB Pin Voltage -0.4V < VFB < 7V ILIM Pin Voltage 600mV FA/SYNC/SD Pin Voltage -0.4V < VFA/SYNC/SD < 7V Operating Ratings (Note 1) Input Voltage Peak Driver Output Current ( < 10µs) 1.0A Power Dissipation Internally Limited Storage Temperature Range −65˚C to +150˚C Junction Temperature Junction Temperature Range +150˚C ESD Susceptibilty Human Body Model (Note 2) 2.97V ≤ VIN ≤ 40V Supply Voltage −40˚C ≤ TJ ≤ +125˚C 100kHz ≤ FSW ≤ 1MHz Switching Frequency 2kV Electrical Characteristics Specifications in Standard type face are for TJ = 25˚C, and in bold type face apply over the full Operating Temperature Range. Unless otherwise specified, VIN = 12V, RFA = 40kΩ Symbol VFB Parameter Feedback Voltage Conditions Typical VCOMP = 1.4V, 2.97 ≤ VIN ≤ 40V 1.26 ∆VLINE Feedback Voltage Line Regulation 2.97 ≤ VIN ≤ 40V 0.001 ∆VLOAD Output Voltage Load Regulation IEAO Source/Sink ± 0.5 VUVLO Input Undervoltage Lock-out 2.85 Input Undervoltage Lock-out Hysteresis 170 VUV(HYS) Limit Units 1.2507/1.24 1.2753/1.28 V V(min) V(max) %/V %/V (max) 2.97 V V(max) 130 210 mV mV (min) mV (max) 370 420 kHz kHz(min) kHz(max) Nominal Switching Frequency RFA = 40KΩ RDS1 (ON) Driver Switch On Resistance (top) IDR = 0.2A, VIN = 5V 16 Ω RDS2 (ON) Driver Switch On Resistance (bottom) IDR = 0.2A 4.5 Ω VDR (max) Maximum Drive Voltage Swing(Note 6) VIN < 7.2V VIN V Dmax Maximum Duty Cycle(Note 7) 100 Tmin (on) Minimum On Time 325 Fnom ISUPPLY IQ VSENSE 400 VIN ≥ 7.2V 7.2 Supply Current (switching) (Note 9) Quiescent Current in Shutdown Mode VFA/SYNC/SD = 5V(Note 10), VIN = 5V Current Sense Threshold Voltage VIN = 5V 2.0 % 230 550 nsec nsec(min) nsec(max) 2.6 mA mA (max) 7 µA µA (max) 140/ 135 195/ 200 mV mV (min) mV (max) 5 165 3 www.national.com LM3488 Electrical Characteristics (Continued) Specifications in Standard type face are for TJ = 25˚C, and in bold type face apply over the full Operating Temperature Range. Unless otherwise specified, VIN = 12V, RFA = 40kΩ Symbol VSC VSL VOVP VOVP(HYS) Gm AVOL IEAO VEAO Parameter Conditions Typical Short-Circuit Current Limit Sense Voltage VIN = 5V Internal Compensation Ramp Voltage VIN = 5V Output Over-voltage Protection (with respect to feedback voltage) (Note 8) VCOMP = 1.4V Output Over-Voltage Protection Hysteresis(Note 8) VCOMP = 1.4V Error Ampifier Transconductance VCOMP = 1.4V IEAO = 100µA (Source/Sink) 800 Error Amplifier Voltage Gain VCOMP = 1.4V IEAO = 100µA (Source/Sink) 38 Error Amplifier Output Current (Source/ Sink) Source, VCOMP = 1.4V, VFB = 0V 110 Sink, VCOMP = 1.4V, VFB = 1.4V −140 Error Amplifier Output Voltage Swing Limit Units 235 395 mV mV (min) mV (max) 52 132 mV mV(min) mV(max) 32/ 25 78/ 85 mV mV(min) mV(max) 20 110 mV mV(min) mV(max) 600/ 365 1000/ 1265 µmho µmho (min) µmho (max) 26 44 V/V V/V (min) V/V (max) 80/ 50 140/ 180 µA µA (min) µA (max) −100/ −85 −180/ −185 µA µA (min) µA (max) 1.8 2.4 V V(min) V(max) 0.2 1.0 V V(min) V(max) 325 92 50 60 Upper Limit VFB = 0V COMP Pin = Floating 2.2 Lower Limit VFB = 1.4V 0.56 TSS Internal Soft-Start Delay VFB = 1.2V, VCOMP = Floating 4 msec Tr Drive Pin Rise Time Cgs = 3000pf, VDR = 0 to 3V 25 ns Tf Drive Pin Fall Time Cgs = 3000pf, VDR = 0 to 3V 25 ns VSD Shutdown and Synchronization signal threshold (Note 5) Output = High ISD Shutdown Pin Current 1.27 Output = Low 1.35 V V (max) 0.35 V V (min) 0.65 VSD = 5V −1 VSD = 0V +1 µA TSD Thermal Shutdown 165 ˚C Tsh Thermal Shutdown Hysteresis 10 ˚C θJA Thermal Resistance 200 ˚C/W www.national.com MM Package 4 (Continued) Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics. Note 2: The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin. Note 3: All limits are guaranteed at room temperature (standard type face) and at temperature extremes (bold type face). All room temperature limits are 100% tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL). Note 4: Typical numbers are at 25˚C and represent the most likely norm. Note 5: The FA/SYNC/SD pin should be pulled to VIN through a resistor to turn the regulator off. Note 6: The voltage on the drive pin, VDR is equal to the input voltage when input voltage is less than 7.2V. VDR is equal to 7.2V when the input voltage is greater than or equal to 7.2V. Note 7: The limits for the maximum duty cycle can not be specified since the part does not permit less than 100% maximum duty cycle operation. Note 8: The over-voltage protection is specified with respect to the feedback voltage. This is because the over-voltage protection tracks the feedback voltage. The over-voltage thresold can be calculated by adding the feedback voltage, VFB to the over-voltage protection specification. Note 9: For this test, the FA/SYNC/SD Pin is pulled to ground using a 40K resistor . Note 10: For this test, the FA/SYNC/SD Pin is pulled to 5V using a 40K resistor. 5 www.national.com LM3488 Electrical Characteristics LM3488 Typical Performance Characteristics Unless otherwise specified, VIN = 12V, TJ = 25˚C. IQ vs Temperature & Input Voltage ISupply vs Input Voltage (Non-Switching) 10138834 10138803 ISupply vs VIN Switching Frequency vs RFA 10138835 10138804 Frequency vs Temperature Drive Voltage vs Input Voltage 10138854 www.national.com 10138805 6 Current Sense Threshold vs Input Voltage (Continued) COMP Pin Voltage vs Load Current 10138862 10138845 Efficiency vs Load Current (3.3V In and 12V Out) Efficiency vs Load Current (5V In and 12V Out) 10138859 10138858 Efficiency vs Load Current (9V In and 12V Out) Efficiency vs Load Current (3.3V In and 5V Out) 10138860 10138853 7 www.national.com LM3488 Typical Performance Characteristics Unless otherwise specified, VIN = 12V, TJ = 25˚C. LM3488 Typical Performance Characteristics Unless otherwise specified, VIN = 12V, TJ = 25˚C. Error Amplifier Gain (Continued) Error Amplifier Phase 10138855 10138856 COMP Pin Source Current vs Temperature Short Circuit Protection vs Input Voltage 10138857 10138836 Compensation Ramp vs Compensation Resistor Shutdown Threshold Hysteresis vs Temperature 10138846 10138851 www.national.com 8 (Continued) Current Sense Voltage vs Duty Cycle 10138852 9 www.national.com LM3488 Typical Performance Characteristics Unless otherwise specified, VIN = 12V, TJ = 25˚C. LM3488 Functional Block Diagram 10138806 The voltage sensed across the sense resistor generally contains spurious noise spikes, as shown in Figure 1. These spikes can force the PWM comparator to reset the RS latch prematurely. To prevent these spikes from resetting the latch, a blank-out circuit inside the IC prevents the PWM comparator from resetting the latch for a short duration after the latch is set. This duration is about 150ns and is called the blank-out time. Under extremely light load or no-load conditions, the energy delivered to the output capacitor when the external MOSFET is on during the blank-out time is more than what is delivered to the load. An over-voltage comparator inside the LM3488 prevents the output voltage from rising under these conditions. The over-voltage comparator senses the feedback (FB pin) voltage and resets the RS latch under these conditions. The latch remains in reset state till the output decays to the nominal value. Functional Description The LM3488 uses a fixed frequency, Pulse Width Modulated (PWM), current mode control architecture. In a typical application circuit, the peak current through the external MOSFET is sensed through an external sense resistor. The voltage across this resistor is fed into the ISEN pin. This voltage is then level shifted and fed into the positive input of the PWM comparator. The output voltage is also sensed through an external feedback resistor divider network and fed into the error amplifier negative input (feedback pin, FB). The output of the error amplifier (COMP pin) is added to the slope compensation ramp and fed into the negative input of the PWM comparator. At the start of any switching cycle, the oscillator sets the RS latch using the SET/Blank-out and switch logic blocks. This forces a high signal on the DR pin (gate of the external MOSFET) and the external MOSFET turns on. When the voltage on the positive input of the PWM comparator exceeds the negative input, the RS latch is reset and the external MOSFET turns off. www.national.com 10 LM3488 Functional Description (Continued) 10138807 FIGURE 1. Basic Operation of the PWM comparator From the above equation, when D > 0.5, ∆I1 will be greater than ∆IO. In other words, the disturbance is divergent. So a very small perturbation in the load will cause the disturbance to increase. To prevent the sub-harmonic oscillations, a compensation ramp is added to the control signal, as shown in Figure 3. With the compensation ramp, SLOPE COMPENSATION RAMP The LM3488 uses a current mode control scheme. The main advantages of current mode control are inherent cycle-bycycle current limit for the switch, and simpler control loop characteristics. It is also easy to parallel power stages using current mode control since as current sharing is automatic. Current mode control has an inherent instability for duty cycles greater than 50%, as shown in Figure 2. In Figure 2, a small increase in the load current causes the switch current to increase by ∆IO. The effect of this load change, ∆I1, is : 10138809 FIGURE 2. Sub-Harmonic Oscillation for D > 0.5 11 www.national.com LM3488 Functional Description (Continued) 10138811 FIGURE 3. Compensation Ramp Avoids Sub-Harmonic Oscillation The compensation ramp has been added internally in LM3488. The slope of this compensation ramp has been selected to satisfy most of the applications. The slope of the internal compensation ramp depends on the frequency. This slope can be calculated using the formula: MC = VSL.FS Volts/second In the above equation, VSL is the amplitude of the internal compensation ramp. Limits for VSL have been specified in the electrical characteristics. In this equation, ∆VSL is equal to 40.10-6RSL. Hence, In order to provide the user additional flexibility, a patented scheme has been implemented inside the IC to increase the slope of the compensation ramp externally, if the need arises. Adding a single external resistor, RSL(as shown in Figure 4) increases the slope of the compensation ramp, MC by : www.national.com ∆VSL versus RSL has been plotted in Figure 5 for different frequencies. 12 LM3488 Functional Description (Continued) 10138813 FIGURE 4. Increasing the Slope of the Compensation Ramp 10138851 FIGURE 5. ∆VSL vs RSL this resistor is dependent on the off time of the synchronization pulse, TOFF(SYNC). Table 1 shows the range of resistors to be used for a given TOFF(SYNC). FREQUENCY ADJUST/SYNCHRONIZATION/SHUTDOWN The switching frequency of LM3488 can be adjusted between 100kHz and 1MHz using a single external resistor. This resistor must be connected between FA/SYNC/SD pin and ground, as shown in Figure 6. Please refer to the typical performance characteristics to determine the value of the resistor required for a desired switching frequency. The LM3488 can be synchronized to an external clock. The external clock must be connected to the FA/SYNC/SD pin through a resistor, RSYNC as shown in Figure 7. The value of TABLE 1. 13 TOFF(SYNC) (µsec) RSYNC range (kΩ) 1 5 to 13 2 20 to 40 3 40 to 65 www.national.com LM3488 Functional Description The FA/SYNC/SD pin also functions as a shutdown pin. If a high signal (refer to the electrical characteristics for definition of high signal) appears on the FA/SYNC/SD pin, the LM3488 stops switching and goes into a low current mode. The total supply current of the IC reduces to less than 10µA under these conditions. Figure 8 and Figure 9 show implementation of shutdown function when operating in Frequency adjust mode and synchronization mode respectively. In frequency adjust mode, connecting the FA/SYNC/SD pin to ground forces the clock to run at a certain frequency. Pulling this pin high shuts down the IC. In frequency adjust or synchronization mode, a high signal for more than 30ms shuts down the IC. (Continued) TABLE 1. (Continued) TOFF(SYNC) (µsec) RSYNC range (kΩ) 4 55 to 90 5 70 to 110 6 85 to 140 7 100 to 160 8 120 to 190 9 135 to 215 10 150 to 240 It is also necessary to have the width of the synchronization pulse narrower than the duty cycle of the converter. It is also necessary to have the synchronization pulse width ≥ 300nsecs. 10138816 FIGURE 6. Frequency Adjust 10138815 FIGURE 7. Frequency Synchronization 10138816 FIGURE 8. Shutdown Operation in Frequency Adjust Mode www.national.com 14 LM3488 Functional Description (Continued) 10138817 FIGURE 9. Shutdown Operation in Synchronization Mode SHORT-CIRCUIT PROTECTION When the voltage across the sense resistor (measured on ISEN Pin) exceeds 350mV, short-circuit current limit gets activated. A comparator inside LM3488 reduces the switching frequency by a factor of 5 and maintains this condition till the short is removed. Typical Applications The LM3488 may be operated in either continuous or discontinuous conduction mode. The following applications are designed for continuous conduction operation. This mode of operation has higher efficiency and lower EMI characteristics than the discontinuous mode. ferred to the load and output capacitor. The ratio of these two cycles determines the output voltage. The output voltage is defined as: BOOST CONVERTER The most common topology for LM3488 is the boost or step-up topology. The boost converter converts a low input voltage into a higher output voltage. The basic configuration for a boost regulator is shown in Figure 10. In continuous conduction mode (when the inductor current never reaches zero at steady state), the boost regulator operates in two cycles. In the first cycle of operation, MOSFET Q is turned on and energy is stored in the inductor. During this cycle, diode D is reverse biased and load current is supplied by the output capacitor, COUT. In the second cycle, MOSFET Q is off and the diode is forward biased. The energy stored in the inductor is trans- (ignoring the drop across the MOSFET and the diode), or where D is the duty cycle of the switch, VD is the forward voltage drop of the diode, and VQ is the drop across the MOSFET when it is on. The following sections describe selection of components for a boost converter. 15 www.national.com LM3488 Typical Applications (Continued) 10138822 FIGURE 10. Simplified Boost Converter Diagram (a) First cycle of operation. (b) Second cycle of operation POWER INDUCTOR SELECTION The inductor is one of the two energy storage elements in a boost converter. Figure 11 shows how the inductor current varies during a switching cycle. The current through an inductor is quantified as: 10138824 FIGURE 11. A. Inductor current B. Diode current If VL(t) is constant, diL(t)/dt must be constant. Hence, for a given input voltage and output voltage, the current in the inductor changes at a constant rate. www.national.com 16 PROGRAMMING THE OUTPUT VOLTAGE AND OUTPUT CURRENT (Continued) The important quantities in determining a proper inductance value are IL (the average inductor current) and ∆iL (the inductor current ripple). If ∆iL is larger than IL, the inductor current will drop to zero for a portion of the cycle and the converter will operate in discontinuous conduction mode. If ∆iL is smaller than IL, the inductor current will stay above zero and the converter will operate in continuous conduction mode. All the analysis in this datasheet assumes operation in continuous conduction mode. To operate in continuous conduction mode, the following conditions must be met: IL > ∆iL The output voltage can be programmed using a resistor divider between the output and the feedback pins, as shown in Figure 12. The resistors are selected such that the voltage at the feedback pin is 1.26V. RF1 and RF2 can be selected using the equation, A 100pF capacitor may be connected between the feedback and ground pins to reduce noise. The maximum amount of current that can be delivered at the output can be controlled by the sense resistor, RSEN. Current limit occurs when the voltage that is generated across the sense resistor equals the current sense threshold voltage, VSENSE. Limits for VSENSE have been specified in the electrical characteristics. This can be expressed as: Isw(peak) * RSEN = VSENSE VSENSE represents the maximum value of the control signal as shown in Figure 2. This control signal, however, is not a constant value and changes over the course of a period as a result of the internal compensation ramp (see Figure 3). Therefore the current limit will also change as a result of the internal compensation ramp. The actual command signal, VCS, can be better expressed as a function of the sense voltage and the internal compensation ramp: VCS = VSENSE − (D * VSL) VSL is defined as the internal compensation ramp voltage, limits are specified in the electrical characteristics. Choose the minimum IOUT to determine the minimum L. A common choice is to set ∆iL to 30% of IL. Choosing an appropriate core size for the inductor involves calculating the average and peak currents expected through the inductor. In a boost converter, The peak current through the switch is equal to the peak inductor current. Isw(peak) = IL + ∆iL Therefore for a boost converter and IL_peak = IL(max) + ∆iL(max), where A core size with ratings higher than these values should be chosen. If the core is not properly rated, saturation will dramatically reduce overall efficiency. The LM3488 can be set to switch at very high frequencies. When the switching frequency is high, the converter can be operated with very small inductor values. With a small inductor value, the peak inductor current can be extremely higher than the output currents, especially under light load conditions. The LM3488 senses the peak current through the switch. The peak current through the switch is the same as the peak current calculated above. Combining the three equation yields an expression for RSEN 17 www.national.com LM3488 Typical Applications LM3488 Typical Applications (Continued) 10138820 FIGURE 12. Adjusting the Output Voltage In the above equation, IOUT is the output current and ∆IL has been defined in Figure 11 CURRENT LIMIT WITH ADDITIONAL SLOPE COMPENSATION If an external slope compensation resistor is used (see Figure 4) the internal control signal will be modified and this will have an effect on the current limit. The control signal is given by: VCS = VSENSE − (D * VSL) Where VSENSE and VSL are defined parameters in the electrical characteristics section. If RSL is used, then this will add to the existing slope compensation. The command voltage will then be given by: VCS = VSENSE − (D * ( VSL + ∆VSL) ) Where ∆VSL is the additional slope compensation generated and can be calculated by use of Figure 5 or is equal to 40 x 10−6 * RSL. This changes the equation for RSEN to: The peak reverse voltage for boost converter is equal to the regulator output voltage. The diode must be capable of handling this voltage. To improve efficiency, a low forward drop schottky diode is recommended. POWER MOSFET SELECTION The drive pin of LM3488 must be connected to the gate of an external MOSFET. In a boost topology, the drain of the external N-Channel MOSFET is connected to the inductor and the source is connected to the ground. The drive pin (DR) voltage depends on the input voltage (see typical performance characteristics). In most applications, a logic level MOSFET can be used. For very low input voltages, a sublogic level MOSFET should be used. The selected MOSFET directly controls the efficiency. The critical parameters for selection of a MOSFET are: 1. Minimum threshold voltage, VTH(MIN) 2. On-resistance, RDS(ON) 3. Total gate charge, Qg 4. Reverse transfer capacitance, CRSS 5. Maximum drain to source voltage, VDS(MAX) Therefore RSL can be used to provide an additional method for setting the current limit. The off-state voltage of the MOSFET is approximately equal to the output voltage. VDS(MAX) of the MOSFET must be greater than the output voltage. The power losses in the MOSFET can be categorized into conduction losses and ac switching or transition losses. RDS(ON) is needed to estimate the conduction losses. The conduction loss, PCOND, is the I2R loss across the MOSFET. The maximum conduction loss is given by: POWER DIODE SELECTION Observation of the boost converter circuit shows that the average current through the diode is the average load current, and the peak current through the diode is the peak current through the inductor. The diode should be rated to handle more than its peak current. The peak diode current can be calculated using the formula: ID(Peak) = IOUT/ (1−D) + ∆IL www.national.com 18 (Continued) OUTPUT CAPACITOR SELECTION The output capacitor in a boost converter provides all the output current when the inductor is charging. As a result it sees very large ripple currents. The output capacitor should be capable of handling the maximum rms current. The rms current in the output capacitor is: where DMAX is the maximum duty cycle. The turn-on and turn-off transitions of a MOSFET require times of tens of nano-seconds. CRSS and Qg are needed to estimate the large instantaneous power loss that occurs during these transitions. Where The amount of gate current required to turn the MOSFET on can be calculated using the formula: IG = Qg.FS The required gate drive power to turn the MOSFET on is equal to the switching frequency times the energy required to deliver the charge to bring the gate charge voltage to VDR (see electrical characteristics and typical performance characteristics for the drive voltage specification). PDrive = FS.Qg.VDR and D, the duty cycle is equal to (VOUT − VIN)/VOUT. The ESR and ESL of the output capacitor directly control the output ripple. Use capacitors with low ESR and ESL at the output for high efficiency and low ripple voltage. Surface Mount tantalums, surface mount polymer electrolytic and polymer tantalum, Sanyo- OSCON, or multi-layer ceramic capacitors are recommended at the output. Designing SEPIC Using LM3488 INPUT CAPACITOR SELECTION Due to the presence of an inductor at the input of a boost converter, the input current waveform is continuous and triangular, as shown in Figure 11. The inductor ensures that the input capacitor sees fairly low ripple currents. However, as the input capacitor gets smaller, the input ripple goes up. The rms current in the input capacitor is given by: Since the LM3488 controls a low-side N-Channel MOSFET, it can also be used in SEPIC (Single Ended Primary Inductance Converter) applications. An example of SEPIC using LM3488 is shown in Figure 14. As shown in Figure 14, the output voltage can be higher or lower than the input voltage. The SEPIC uses two inductors to step-up or step-down the input voltage. The inductors L1 and L2 can be two discrete inductors or two windings of a coupled transformer since equal voltages are applied across the inductor throughout the switching cycle. Using two discrete inductors allows use of catalog magnetics, as opposed to a custom transformer. The input ripple can be reduced along with size by using the coupled windings of transformer for L1 and L2. Due to the presence of the inductor L1 at the input, the SEPIC inherits all the benefits of a boost converter. One main advantage of SEPIC over boost converter is the inherent input to output isolation. The capacitor CS isolates the input from the output and provides protection against shorted or malfunctioning load. Hence, the A SEPIC is useful for replacing boost circuits when true shutdown is required. This means that the output voltage falls to 0V when the switch is turned off. In a boost converter, the output can only fall to the input voltage minus a diode drop. The input capacitor should be capable of handling the rms current. Although the input capacitor is not as critical in a boost application, low values can cause impedance interactions. Therefore a good quality capacitor should be chosen in the range of 100µF to 200µF. If a value lower than 100µF is used, then problems with impedance interactions or switching noise can affect the LM3478. To improve performance, especially with VIN below 8 volts, it is recommended to use a 20Ω resistor at the input to provide a RC filter. The resistor is placed in series with the VIN pin with only a bypass capacitor attached to the VIN pin directly (see Figure 13). A 0.1µF or 1µF ceramic capacitor is necessary in this configuration. The bulk input capacitor and inductor will connect on the other side of the resistor with the input power supply. 10138893 FIGURE 13. Reducing IC Input Noise 19 www.national.com LM3488 Typical Applications LM3488 Designing SEPIC Using LM3488 (Continued) The duty cycle of a SEPIC is given by: In the above equation, VQ is the on-state voltage of the MOSFET, Q, and VDIODE is the forward voltage drop of the diode. 10138844 FIGURE 14. Typical SEPIC Converter SELECTION OF INDUCTORS L1 AND L2 Proper selection of the inductors L1 and L2 to maintain constant current mode requires calculations of the following parameters. Average current in the inductors: POWER MOSFET SELECTION As in boost converter, the parameters governing the selection of the MOSFET are the minimum threshold voltage, VTH(MIN), the on-resistance, RDS(ON), the total gate charge, Qg, the reverse transfer capacitance, CRSS, and the maximum drain to source voltage, VDS(MAX). The peak switch voltage in a SEPIC is given by: VSW(PEAK) = VIN + VOUT + VDIODE The selected MOSFET should satisfy the condition: VDS(MAX) > VSW(PEAK) The peak switch current is given by: IL2AVE = IOUT Peak to peak ripple current, to calculate core loss if necessary: The rms current through the switch is given by: maintains the condition IL > ∆iL/2 to ensure constant current mode. POWER DIODE SELECTION The Power diode must be selected to handle the peak current and the peak reverse voltage. In a SEPIC, the diode peak current is the same as the switch peak current. The off-state voltage or peak reverse voltage of the diode is VIN + VOUT. Similar to the boost converter, the average diode current is equal to the output current. Schottky diodes are recommended. www.national.com 20 having high rms current ratings relative to size. Ceramic capacitors could be used, but the low C values will tend to cause larger changes in voltage across the capacitor due to the large currents. High C value ceramics are expensive. Electrolytics work well for through hole applications where the size required to meet the rms current rating can be accommodated. There is an energy balance between CS and L1, which can be used to determine the value of the capacitor. The basic energy balance equation is: (Continued) Peak current in the inductor, to ensure the inductor does not saturate: Where IL1PK must be lower than the maximum current rating set by the current sense resistor. is the ripple voltage across the SEPIC capacitor, and The value of L1 can be increased above the minimum recommended to reduce input ripple and output ripple. However, once DIL1 is less than 20% of IL1AVE, the benefit to output ripple is minimal. By increasing the value of L2 above the minimum recommended, ∆IL2 can be reduced, which in turn will reduce the output ripple voltage: is the ripple current through the inductor L1. The energy balance equation can be solved to provide a minimum value for CS: where ESR is the effective series resistance of the output capacitor. If L1 and L2 are wound on the same core, then L1 = L2 = L. All the equations above will hold true if the inductance is replaced by 2L. A good choice for transformer with equal turns is Coiltronics CTX series Octopack. Input Capacitor Selection Similar to a boost converter, the SEPIC has an inductor at the input. Hence, the input current waveform is continuous and triangular. The inductor ensures that the input capacitor sees fairly low ripple currents. However, as the input capacitor gets smaller, the input ripple goes up. The rms current in the input capacitor is given by: SENSE RESISTOR SELECTION The peak current through the switch, ISW(PEAK) can be adjusted using the current sense resistor, RSEN, to provide a certain output current. Resistor RSEN can be selected using the formula: The input capacitor should be capable of handling the rms current. Although the input capacitor is not as critical in a boost application, low values can cause impedance interactions. Therefore a good quality capacitor should be chosen in the range of 100µF to 200µF. If a value lower than 100µF is used, then problems with impedance interactions or switching noise can affect the LM3478. To improve performance, especially with VIN below 8 volts, it is recommended to use a 20Ω resistor at the input to provide a RC filter. The resistor is placed in series with the VIN pin with only a bypass capacitor attached to the VIN pin directly (see Figure 13). A 0.1µF or 1µF ceramic capacitor is necessary in this configuration. The bulk input capacitor and inductor will connect on the other side of the resistor with the input power supply. Sepic Capacitor Selection The selection of SEPIC capacitor, CS, depends on the rms current. The rms current of the SEPIC capacitor is given by: The SEPIC capacitor must be rated for a large ACrms current relative to the output power. This property makes the SEPIC much better suited to lower power applications where the rms current through the capacitor is relatively small (relative to capacitor technology). The voltage rating of the SEPIC capacitor must be greater than the maximum input voltage. Tantalum capacitors are the best choice for SMT, Output Capacitor Selection The ESR and ESL of the output capacitor directly control the output ripple. Use low capacitors with low ESR and ESL at 21 www.national.com LM3488 Designing SEPIC Using LM3488 LM3488 Output Capacitor Selection mount tantalums, surface mount polymer electrolytic and polymer tantalum, Sanyo- OSCON, or multi-layer ceramic capacitors are recommended at the output for low ripple. (Continued) the output for high efficiency and low ripple voltage. Surface mount tantalums, surface mount polymer electrolytic and polymer tantalum, Sanyo- OSCON, or multi-layer ceramic capacitors are recommended at the output. The output capacitor of the SEPIC sees very large ripple currents (similar to the output capacitor of a boost converter. The rms current through the output capacitor is given by: The ESR and ESL of the output capacitor directly control the output ripple. Use low capacitors with low ESR and ESL at the output for high efficiency and low ripple voltage. Surface www.national.com 22 LM3488 Other Application Circuits 10138843 FIGURE 15. Typical High Efficiency Step-Up (Boost) Converter 23 www.national.com LM3488 High Efficiency Low-Side N-Channel Controller for Switching Regulators Physical Dimensions inches (millimeters) unless otherwise noted National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications. For the most current product information visit us at www.national.com. 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