OBSOLETE
July 9, 2009
170W Class D Audio Power Amplifier Solution
General Description
Key Specifications
The IC combination of the LM4651 driver and the LM4652
power MOSFET provides a high efficiency, Class D subwoofer amplifier solution.
The LM4651 is a fully integrated conventional pulse width
modulator driver IC. The IC contains short circuit, under voltage, over modulation, and thermal shut down protection circuitry. The LM4651also contains a standby function which
shuts down the pulse width modulation minimizing supply
current. The LM4652 is a fully integrated H-bridge power
MOSFET IC in a TO-220 power package. The LM4652 has a
temperature sensor built in to alert the LM4651 when the die
temperature of the LM4652 exceeds the threshold. Together,
these two IC's form a simple, compact high power audio amplifier solution complete with protection normally seen only in
Class AB amplifiers. Few external components and minimal
traces between the IC's keep the PCB area small and aids in
EMI control.
The near rail-to-rail switching amplifier substantially increases
the efficiency compared to Class AB amplifiers. This high efficiency solution significantly reduces the heat sink size compared to a Class AB IC of the same power level. This two-chip
solution is optimum for powered subwoofers and self powered
speakers.
■
■
■
■
Output power into 4Ω with < 10% THD.
170W (Typ)
THD at 10W, 4Ω, 10 − 500Hz.
< 0.3% THD (Typ)
Maximum efficiency at 125W
85% (Typ)
Standby attenuation.
>100dB (Min)
Features
■
■
■
■
■
■
■
Conventional pulse width modulation.
Externally controllable switching frequency.
50kHz to 200kHz switching frequency range.
Integrated error amp and feedback amp.
Turn−on soft start and under voltage lockout.
Over modulation protection (soft clipping).
Externally controllable output current limiting and thermal
shutdown protection.
■ Self checking protection diagnostic.
Applications
■ Powered subwoofers for home theater and PC's
■ Car booster amplifier
■ Self-powered speakers
Connection Diagrams
LM4651 Plastic Package
LM4652 Plastic Package (Note 8)
10127773
10127772
Top View
Order Number LM4651N
See NS Package Number N28B
Isolated TO-220 Package
Order Number LM4652TF
See NS Package Number TF15B
or
Non-Isolated TO-220 Package
Order Number LM4652TA
See NS Package Number TA15A
Overture® is a registered trademark of National Semiconductor.
© 2009 National Semiconductor Corporation
101277
101277 Version 9 Revision 2
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Print Date/Time: 2009/07/09 16:41:57
LM4651 & LM4652 Overture® 170W Class D Audio Power Amplifier Solution
LM4651 & LM4652
Overture™ Audio Power
Amplifier
LM4651 & LM4652
Absolute Maximum Ratings (Notes 1, 2)
Thermal Resistance
LM4651 N Package
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Supply Voltage
Output Current (LM4652)
Power Dissipation (LM4651) (Note 3)
Power Dissipation (LM4652) (Note 3)
ESD Susceptibility (LM4651) (Note 4)
LM4652 (pins 2,6,10,11)
ESD Susceptibility (LM4651) (Note 5)
LM4652 (pins 2,6,10,11)
Junction Temperature (Note 6)
Soldering Information
N, TA and TF Package (10 seconds)
Storage Temperature
Operating Ratings
± 22V
10A
1.5W
32W
2000V
500V
200V
100V
150°C
θJA
52°C/W
θJC
22°C/W
LM4652 TF, TO−220 Package
θJA
43°C/W
θJC
2.0°C/W
LM4652 T, TO−220 Package
θJA
37°C/W
θJC
1.0°C/W
260°C
−40°C to + 150°C
(Notes 1, 2)
−40°C ≤ TA ≤ +85°
C
22V to 44V
Temperature Range
Supply Voltage |V+| + |V−|
System Electrical Characteristics for LM4651 and LM4652
(Notes 1, 2)
The following specifications apply for +VCC = +20V, −VEE = −20V, f SW = 125kHz, fIN = 100Hz, RL = 4Ω, unless otherwise specified.
Typicals apply for TA = 25°C. For specific circuit values, refer to Figure 1 (Typical Audio Application Circuit).
LM4651 & LM4652
Symbol
Parameter
Conditions
Typical
Limit
(Note 7)
ICQ
ISTBY
AM
PO
Total Quiescent Power Supply
Current
Units
(Limits)
VIN = 0V, IO = 0mA
RDLY = 0Ω
RDLY = 10kΩ
237
124
mA
mA
Standby Current
VPIN13 = 5V, Stby: On
17
mA
Standby Attenuation
VPIN13 = 5V, Stby: On
>115
dB
RL = 4Ω, 1% THD
125
W
RL = 4Ω, 10% THD
155
W
RL = 8Ω, 1% THD
75
W
RL = 8Ω, 10% THD
90
W
fSW = 75kHz, RL = 4Ω, 1% THD
135
W
fSW = 75kHz, RL = 4Ω, 10% THD
170
W
Output Power (Continuous Average)
η
Efficiency at PO = 5W
PO = 5W, RDLY = 5kΩ
55
%
η
Efficiency
(LM4651 & LM4652)
PO = 125W, THD = 1%
85
%
PO = 125W, THD = 1% (max)
22
W
Pd
Power Dissipation
(LM4651 + LM4652)
fSW = 75kHz, PO = 135W,
THD = 1% (max)
22
W
0.3
%
A Weighted, no signal, RL = 4Ω
550
µV
A-Wtg, Pout = 125W, RL = 4Ω
92
dB
22kHz BW, Pout = 125W, RL = 4Ω
89
dB
THD+N
10W, 10Hz ≤ fIN ≤ 500Hz,
Total Harmonic Distortion Plus Noise AV = 18dB
10Hz ≤ BW ≤ 80kHz
εOUT
SNR
Output Noise
Signal-to-Noise Ratio
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101277 Version 9 Revision 2
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Symbol
Parameter
Conditions
Typical
Limit
(Note 7)
VOS
Output Offset Voltage
VIN = 0V, IO = 0mA, ROFFSET = 0Ω
PSRR
Power Supply Rejection Ratio
RL = 4Ω, 10Hz ≤ BW ≤ 30kHz
+VCCAC = −VEEAC = 1VRMS,
Units
(Limits)
0.07
V
37
dB
fAC = 120Hz
Electrical Characteristics for LM4651
(Notes 1, 2, 7)
The following specifications apply for +VCC = +20V, −VEE = −20V, fSW = 125kHz, unless otherwise specified. Limits apply for TA =
25°C. For specific circuit values, refer to Figure 1 (Typical Audio Application Circuit).
LM4651 & LM4652
Symbol
Parameter
Conditions
Typical
Limit
(Note 7)
LM4652 not connected, IO = 0mA,
Units
(Limits)
15
45
mA (min)
mA (max)
0.8
V (max)
2.5
V (min)
ICQ
Total Quiescent Current
VIL
Standby Low Input Voltage
Not in Standby Mode
VIH
Standby High Input Voltage
In Standby Mode
fSW
Switching Frequency Range
fSWerror
50% Duty Cycle Error
ROSC = 4kΩ, fSW = 125kHz
1
Tdead
Dead Time
RDLY = 0Ω
27
ns
TOverMod
Over Modulation Protection Time
Pulse Width Measured at 50%
310
ns
|VCC+| + |VEE-|, RDLY = 0Ω
36
2.0
ROSC = 15kΩ
65
kHz
ROSC = 0Ω
200
kHz
3
% (max)
Electrical Characteristics for LM4652
(Notes 1, 2, 7)
The following specifications apply for +VCC = +20V, −VEE = −20V, unless otherwise specified. Limits apply for TA = 25°C. For
specific circuit values, refer to Figure 1 (Typical Audio Application Circuit).
LM4651 & LM4652
Symbol
Parameter
Conditions
Typical
Limit
(Note 7)
V(BR)DSS
Drain−to−Source Breakdown
Voltage
VGS = 0
IDSS
Drain−to−Source Leakage Current
VGSth
Gate Threshold Voltage
RDS(ON)
Static Drain−to−Source On
Resistance
tr
Rise Time
RGATE = 0Ω
tf
Fall Time
ID
Maximum Saturation Drain Current
Units
(Limits)
55
V
VDS = 44VDC, VGS = 0V
1.0
mA
VDS = VGS, ID = 1mADC
0.85
V
VGS = 6VDC, ID = 6ADC
200
VGD = 6VDC, VDS = 40VDC,
300
mΩ (max)
25
ns
RGATE = 0Ω
26
ns
VGS = 6VDC, VDS = 10VDC
10
VGD = 6VDC, VDS = 40VDC,
8
ADC (min)
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions
which guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters
where no limit is given, however, the typical value is a good indication of device performance.
Note 2: All voltages are measured with respect to the GND pin unless otherwise specified.
Note 3: For operating at case temperatures above 25°C, the LM4651 must be de−rated based on a 150°C maximum junction temperature and a thermal resistance
of θJA = 62 °C/W (junction to ambient), while the LM4652 must be de−rated based on a 150°C maximum junction temperature and a thermal resistance of θJC =
2.0 °C/W (junction to case) for the isolated package (TF) or a thermal resistance of θJC = 1.0°C/W (junction to case) for the non-isolated package (T).
Note 4: Human body model, 100 pF discharged through a 1.5 kΩ resistor.
Note 5: Machine Model, 220pF-240pF discharge through all pins.
Note 6: The operating junction temperature maximum, Tjmax is 150°C.
Note 7: Limits are guaranteed to National's AOQL (Average Outgoing Quality Level).
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LM4651 & LM4652
LM4651 & LM4652
LM4651 & LM4652
Note 8: The LM4652TA package TA15A is a non-isolated package, setting the tab of the device and the heat sink at −V potential when the LM4652 is directly
mounted to the heat sink using only thermal compound. If a mica washer is used in addition to thermal compound, θCS (case to sink) is increased, but the heat
sink will be isolated from −V.
10127768
FIGURE 1. Typical Application Circuit and Test Circuit
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101277 Version 9 Revision 2
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LM4651 & LM4652
LM4651 Pin Descriptions
Pin No.
Symbol
1
OUT1
Description
The reference pin of the power MOSFET output to the gate drive circuitry.
2,27
BS1,BS2
The bootstrap pin provides extra bias to drive the upper gates, HG1,HG2.
3
HG1
High−Gate #1 is the gate drive to a top side MOSFET in the H-Bridge.
4
HG2
High−Gate #2 is the gate drive to a top side MOSFET in the H-Bridge.
5,15
GND
The ground pin for all analog circuitry.
6
+6VBYP
7
+VCC
8
−6VBYP
The internally regulated negative voltage output for analog circuitry. This pin is available for
internal regulator bypassing only.
9
FBKVO
The feedback instrumentation amplifier output pin.
10
ERRIN
The error amplifier inverting input pin. The input audio signal and the feedback signal are
summed at this input pin.
11
ERRVO
The error amplifier output pin.
The internally regulated positive voltage output for analog circuitry. This pin is available for
internal regulator bypassing only.
The positive supply input for the IC.
12
TSD
13
STBY
The thermal shut down input pin for the thermal shut down output of the LM4652.
Standby function input pin. This pin is CMOS compatible.
14
FBK1
The feedback instrumentation amplifier pin. This must be connected to the feedback filter from
VO1 (pin 15 on the LM4652 ).
16
OSC
The switching frequency oscillation pin. Adjusting the resistor from 15.5kΩ to 0Ω changes the
switching frequency from 75kHz to 225kHz.
17
Delay
The dead time setting pin.
18
SCKT
Short circuit setting pin. Minimum setting is 10A.
19
FBK2
The feedback instrumentation amplifier pin. This must be connected to the feedback filter from
VO2 (pin 7 on the LM4652 ).
20,21
−VDDBYP
22,23
−VEE
24
START
25
LG1
Low−Gate #1 is the gate drive to a bottom side MOSFET in the H-Bridge.
26
LG2
Low−Gate #2 is the gate drive to a bottom side MOSFET in the H-Bridge.
28
OUT2
The reference pin of the power MOSFET output to the gate drive circuitry.
The regulator output for digital blocks. This pin is for bypassing only.
The negative voltage supply pin for the IC.
The start up capacitor input pin. This capacitor adjusts the start up time of the diagnostic
sequence for the modulator. Refer to Start up Sequence and Timing in the Application
Information section.
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LM4651 & LM4652
LM4652 Pin Descriptions
Pin No.
Symbol
1
GND
A ground reference for the thermal shut down circuitry.
Description
2
LG1
Low−Gate #1 is the gate input to a bottom side MOSFET in the H-Bridge.
3
−VEE
The negative voltage supply input for the power MOSFET H-Bridge.
4
TSD
The thermal shut down flag pin. This pin transitions to 6V when the die temperature exceeds
150°C.
5
NC
No connection
6
LG2
Low−Gate #2 is the gate input to a bottom side MOSFET in the H-Bridge.
7
VO2
The switching output pin for one side of the H-Bridge.
8
NC
No connection.
9
NC
No connection.
10
HG2
High−Gate #2 is the gate input to a top side MOSFET in the H-Bridge.
11
NC
No connection.
12
NC
No connection.
13
+VCC
The positive voltage supply input for the power MOSFET H-Bridge.
14
HG1
High−Gate #1 is the gate input to a top side MOSFET in the H-Bridge.
15
VO2
The switching output pin for one side of the H-Bridge.
Note: NC, no connect pins are floating pins. It is best to connect the pins to GND to minimize any noise from being
coupled into the pins.
External Components Description
(Refer to Figure 1)
Components
Functional Description
1.
R1
Works with R2, Rfl1 and Rfl2 to set the gain of the system. Gain = {[R2/(R1 + 100)] x [(Rfl1 +
Rfl2)/Rfl2] − [R2/(R1 + 100)] + 0.5} + [(VCC - 20) * 0.0175] (V/V).
2.
R2
See description above for R1.
3.
Rf
4.
Cf
5.
RfI1
Provides a reduction in the feedback with RfI2. RfI1should be 10 X RfI2 minimum to reduce
effects on the pole created by RfI2 and CfI1. See also note for R1, R2 for effect on System Gain.
6.
RfI2
RfI2 and CfI1 creates a low pass filter with a pole at fC = 1/(2πRfI2CfI1) (Hz). See also note for
R1, R2 for effect on System Gain.
7.
CfI1
See description above for RfI2.
8.
RfI3
Establish the second pole for the low pass filter in the feedback path at fC = 1/(2πRfI3CfI2)
(Hz).
9.
CfI2
See description above for RfI3.
10.
L1
11.
C1
12.
Cbyp
13.
CB1−CB4
14.
CBT
Sets the gain and bandwidth of the system by creating a low pass filter for the Error Amplifier's
feedback with Cf. 3dB pole is at fC = 1/(2πRfCf) (Hz).
See description above for Rf.
Combined with CBYP creates a 2−pole, low pass output filter that has a −3dB pole at fC = 1/
{2π[L1(2CBYP + C1)]1/2} (Hz).
Filters the commom mode high frequency noise from the amplifier's outputs to GND.
Recommended value is 0.1µF to 1µF.
See description for L1.
Bypass capacitors for VCC, VEE, analog and digital voltages (VDD, +6V, −6V). See Supply
Bypassing and High Frequency PCB Design in the Application Information section for
more information.
Provides the bootstrap capacitance for the boot strap pin.
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101277 Version 9 Revision 2
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RDLY
16.
CSTART
Controls the startup time with TSTART = (8.5x104) CSTART (seconds) or CSTART = TSTART /
(8.5x104) (F).
17.
RSCKT
Sets the output current limit with ISCKT = (1x105)/(10kΩ ‖ RSCKT) (A) or RSCKT = [(1x109)/
RDLY = [TDLY/(1.7x10−12)] - 500 (Ω).
ISCKT] / [10k - (1x105/ISCKT)] (Ω).
18.
ROSC
Controls the switching frequency with fSW = 1x109 / (4000 + ROSC) (Hz) or ROSC = (1x109/
fSW) - 4000 (Ω).
19.
D1
20.
CSBY1, CSBY2, CSBY3
Schottky diode to protect the output MOSFETs from fly back voltages.
Supply de-coupling capacitors. See Supply Bypassing in the Application Information
section.
21.
ROFFSET
Provides a small DC voltage at the input to minimize the output DC offset seen by the load.
This also minimize power on pops and clicks.
22.
CIN
Blocks DC voltages from being coupled into the input and blocks the DC voltage created by
ROFFSET from the source.
23.
Rgate
Slows the rise and fall time of the gate drive voltages that drive the output FET's.
Typical Performance Characteristics
Output Power vs. Supply Voltage
Output Power vs. Supply Voltage
10127704
10127705
THD+N vs. Output Power
RL = 4Ω
THD+N vs. Output Power
RL = 8Ω
10127706
10127707
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LM4651 & LM4652
Sets the dead time or break before make time to TDLY = (1.7x10−12)(500 + RDLY) (seconds) or
15.
LM4651 & LM4652
THD+N vs. Output Power
RL = 4Ω
THD+N vs. Output Power
RL = 8Ω
10127708
10127709
THD+N vs. Frequency vs. Bandwidth
RL = 4Ω
THD+N vs. Frequency vs. Bandwidth
RL = 8Ω
10127710
10127711
THD+N vs. Frequency vs. Bandwidth
RL = 4Ω
THD+N vs. Frequency vs. Bandwidth
RL = 8Ω
10127712
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10127713
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LM4651 & LM4652
Power Dissipation & Efficiency
vs. Output Power
Clipping Power Point & Efficiency
vs. Switching Frequency (fSW)
10127716
10127717
Frequency Response
RL = 4Ω
Supply Current vs. Switching Frequency
(LM4651 & LM4652)
10127720
10127718
Supply Current vs. Supply Voltage
(LM4651 & LM4652)
RDS(ON) vs. Temperature
10127723
10127721
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LM4651 & LM4652
The value of CSTART sets the time it takes for the IC to go
though the start-up sequence and the frequency that the diagnostic circuitry checks to see if an error condition has been
corrected. An Error condition occurs if current limit, thermal
shut down, under voltage detection, or standby are sensed.
The self-diagnostic circuit checks to see if any one of these
error flags has been removed at a frequency set by the
CSTART capacitor. For example, if the value of CSTART is 10µF
then the diagnostic circuitry will check approximately every
second to see if an error condition has been corrected. If the
error condition is no longer present, the LM4651/52 will return
to normal operation.
Application Information
GENERAL FEATURES
System Functional Information
The LM4651 is a conventional pulse width modulator/driver.
As Figure 2 shows the incoming audio signal is compared with
a triangle waveform with a much higher frequency than the
audio signal (not drawn to scale). The comparator creates a
variable duty cycle squarewave. The squarewave has a duty
cycle proportional to the audio signal level. The squarewave
is then properly conditioned to drive the gates of power MOSFETs in an H-bridge configuration, such as the LM4652. The
pulse train of the power MOSFETs are then fed into a low
pass filter (usually a LC) which removes the high frequency
and delivers an amplified replica of the audio input signal to
the load.
10127770
10127701
FIGURE 3. Startup Timing Diagram
FIGURE 2. Conventional Pulse Width Modulation
Current Limiting and Short Circuit Protection
The resistor value connected between the SCKT pin and
GND determines the maximum output current. Once the output current is higher than the set limit, the short circuit protection turns all power MOSFETs off. The current limit is set
to a minimum of 10A internally but can be increased by adjusting the value of the RSCKT resistor. Equation (3) shows
how to find RSCKT.
Standby Function
The standby function of the LM4651 is CMOS compatible, allowing the user to perform a muting of the music by shutting
down the pulse width waveform. Standby has the added advantage of minimizing the quiescent current. Because standby shuts down the pulse width waveform, the attenuation of
the music is complete (>120dB), EMI is minimized, and any
output noise is eliminated since there is no modulation waveform. When in Standby mode, the outputs of the LM4652 will
both be at VCC. By placing a logic "1" or 5V at pin 13, the
standby function will be enabled. A logic "0" or 0V at pin 13
will disable the standby function allowing modulation by the
input signal.
ISCKT = 1X105/(10kΩ‖ RSCKT)
Thermal Protection
The LM4651 has internal circuitry (pin 12) that is activated by
the thermal shutdown output signal from the LM4652 (pin 4).
The LM4652 has thermal shut down circuitry that monitors the
temperature of the die. The voltage on the TSD pin (pin 4 of
the LM4652) goes high (6V) once the temperature of the
LM4652 die reaches 150°C. This pin should be connected
directly to the TSD pin of the LM4651 (pin 12). The LM4651
disables the pulse width waveform when the LM4652 transmits the thermal shutdown flag. The pulse width waveform
remains disabled until the TSD flag from the LM4652 goes
low, signaling the junction temperature has cooled to a safe
level.
Start Up Sequence and Self-Diagnostic Timing
The LM4651 has an internal soft start feature (see Figure 3)
that ensures reliable and consistent start-up while minimizing
turn-on thumps or pops. During the start-up cycle the system
is in standby mode. This start-up time is controlled externally
by adjusting the capacitance (CSTART) value connected to the
START pin. The start-up time can be controlled by the capacitor value connected to the START pin given by Equation
(1) or (2):
CSTART =
TSTART/(8.5x104)
(seconds)
(1)
(Farads)
(2)
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Dead Time Setting
The DELAY pin on the LM4651 allows the user to set the
amount of dead time or break before make of the system. This
is the amount of time one pair of FETs are off before another
pair is switched on. Increased dead time will reduce the shoot
through current but has the disadvantage of increasing THD.
The dead time should be reduced as the desired bandwidth
of operation increases. The dead time can be adjusted with
the RDLY resistor by Equation (4):
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101277 Version 9 Revision 2
(3)
This feature is designed to protect the MOSFETs by setting
the maximum output current limit under short circuit conditions. It is designed to be a fail-safe protection when the output
terminals are shorted or a speaker fails and causes a short
circuit condition.
Under Voltage Protection
The under voltage protection disables the output driver section of the LM4651 while the supply voltage is below ± 10.5V.
This condition can occur as power is first applied or when low
line, changes in load resistance or power supply sag occurs.
The under voltage protection ensures that all power MOSFETs are off, eliminating any shoot-through current and minimizing pops or clicks during turn-on and turn-off. The under
voltage protection gives the digital logic time to stabilize into
known states providing a popless turn on.
tSTART = (8.4x104)CSTART
(Amps)
Print Date/Time: 2009/07/09 16:41:57
(Seconds)
(4)
instrumentation amplifier has an internally fixed gain of 1. The
use of an instrumentation amplifier serves two purposes.
First, it's input are high impedance so it doesn't load down the
output stage. Secondly, an IA has excellent common-mode
rejection when its gain setting resistors are properly matched.
This feature allows the IA to derive the true feedback signal
from the differential output, which aids in improving the system performance.
Currently, the recommended value is 5kΩ.
Oscillator Control
The modulation frequency is set by an external resistor,
ROSC, connected between pin 16 and GND. The modulation
frequency can be set within the range of 50kHz to 225kHz
according to the design requirements. The values of ROSC and
fOSC can be found by Equation (5) and (6):
fOSC = 1x109/ (4000 + ROSC)
(Hz)
(5)
ROSC = (1x109/ fOSC) − 4000 (Ω) (6)
Equations (5) and (6) are for RDLY = 0. Using a value of
RDLY greater than zero will increase the value needed for
ROSC. For RDLY = 5kΩ, ROSC will need to be increased by
about 2kΩ. As the graphs show, increasing the switching frequency will reduce the THD but also decreases the efficiency
and maximum output power level before clipping. Increasing
the switching frequency increases the amount of loss because switching currents lower the efficiency across the output power range. A higher switching frequency also lowers
the maximum output power before clipping or the 1% THD
point occur.
10127703
FIGURE 5. Feedback instrumentation Amplifier
Schematic
Over-Modulation Protection
The over-modulation protection is an internally generated
fixed pulse width signal that prevents any side of the H-bridge
power MOSFETs from remaining active for an extended period of time. This condition can result when the input signal
amplitude is higher than the internal triangle waveform. Lack
of an over modulation signal can increase distortion when the
amplifier's output is clipping. Figure 4 shows how the over
modulation protection works.
Error Amplifier
The purpose of the error amplifier is to sum the input audio
signal with the feedback signal derived from the output. This
inverting amplifier's gain is externally configurable by resistors Rf and R1. The parallel feedback capacitor and resistor
form a low pass filter that limits the frequency content of the
input audio signal and the feedback signal. The pole of the
filter is set by Equation (7).
fIP = 1/(2πRfCf)
(7)
On-Board Regulators
The LM4651 has its own internal supply regulators for both
analog and digital circuits. Separate ±6V regulators exist
solely for the analog amplifiers, oscillator and PWM comparators. A separate voltage regulator powers the digital logic that
controls the protection, level shifting, and high−/low−side driver circuits. System performance is enhanced by bypassing
each regulator's output. The ±6V regulator outputs, labeled
+6VBYP (pin 6) and −6VBYP (pin 8) should be bypassed to
ground. The digital regulator output, −VDDBYP (pins 20 & 21)
should be bypassed to −VEE (pins 22 & 23). The voltage level
of −VDDBYP should be always be 6V closer to ground than the
negative rail, −VEE. As an example, if −VEE = −20V, then
−VDDBYP should equal −14V. Recommended capacitor values
and type can be found in Figure 1, Typical audio Application
Circuit.
10127702
FIGURE 4. Over Modulation Protection
The over modulation protection also provides a "soft clip" type
response on the top of a sine wave. This minimum pulse time
is internally set and cannot be adjusted. As the switching frequency increases this minimum time becomes a higher percentage of the period (TPERIOD = 1/fSW). Because the over
modulation protection time is a higher percentage of the period, the peak output voltage is lower and, therefore, the
output power at clipping is lower for the same given supply
rails and load.
APPLICATIONS HINTS
Introduction
National Semiconductor (NSC) is committed to providing application information that assists our customers in obtaining
the best performance possible from our products. The following information is provided in order to support this commitment. The reader should be aware that the optimization of
performance was done using a reference PCB designed by
NSC and shown in Figure 7 through Figure 11. Variations in
performance can occur because of physical changes in the
printed circuit board and the application. Therefore, the designer should know that component value changes may be
required in order to optimize performance in a given applica-
Feedback Amplifier and Filter
The purpose of the feedback amplifier is to differentially sample the output and provide a single-ended feedback signal to
the error amplifier to close the feedback loop. The feedback
is taken directly from the switching output before the demodulating LC filter to avoid the phase shift caused by the output
filter. The signal fed back is first low pass filtered with a single
pole or dual pole RC filter to remove the switching frequency
and its harmonics. The differential signal, derived from the
bridge output, goes into the high input impedance instrumentation amplifier that is used as the feedback amplifier. The
11
101277 Version 9 Revision 2
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Print Date/Time: 2009/07/09 16:41:57
www.national.com
LM4651 & LM4652
TDLY = 1.7x10−12 (500 + RDLY)
LM4651 & LM4652
where εo = 0.22479pF/in and εr = 4.1
tion. The values shown in this data sheet can be used as a
starting point for evaluation purposes. When working with
high frequency circuits, good layout practices are also critical
to achieving maximum performance.
A is the common PCB area and d is the distance between the
planes. The designer should target a value of 100pF or
greater for both the positive supply to ground capacitance and
negative supply to ground capacitance. Signal traces that
cross over each other should be laid out at 90° to minimized
any coupling.
Input Pre-Amplifier with Subwoofer Filter
The LM4651 and LM4652 Class D solution is designed for low
frequency audio applications where low gain is required. This
necessitates a pre−amplifier stage with gain and a low pass
audio filter. An inexpensive input stage can be designed using
National's LM833 audio operational amplifier and a minimum
number of external components. A gain of 10 (20dB) is recommended for the pre−amplifier stage. For a subwoofer application, the pole of the low pass filter is normally set within
the range of 60Hz − 180Hz. For a clean sounding subwoofer
the filter should be at least a second-order filter to sharply roll
off the high frequency audio signals. A higher order filter is
recommended for stand-alone self-powered subwoofer applications. Figure 6 shows a simple input stage with a gain of
10 and a second-order low pass filter.
Output Offset Voltage Minimization
The amount of DC offset voltage seen at the output with no
input signal present is already quite good with the LM4651/52.
With no input signal present the system should be at 50% duty
cycle. Any deviation from 50% duty cycle creates a DC offset
voltage seen by the load. To completely eliminate the DC offset, a DC voltage divider can be used at the input to set the
DC offset to near zero. This is accomplished by a simple resistor divider that applies a small DC voltage to the input. This
forces the duty cycle to 50% when there is no input signal.
The result is a LM4651 and LM4652 system with near zero
DC offset. The divider should be a 1.8MΩ from the +6V output
(pin 6) to the input (other side of 25k, R1). R1 acts like the
second resistor in the divider. Also use a 1µF input capacitor
before R1 to block the DC voltage from the source. R1 and the
1µF capacitor create a high pass filter with a 3dB point at
6.35Hz. The value of ROFFSET is set according to the application. Variations in switching frequency and supply voltage will
change the amount of offset voltage requiring a different value
than stated above. The value above (1.8MΩ) is for ±20V and
a switching frequency of 125kHz.
Output Stage Filtering
As common with Class D amplifier design, there are many
trade-offs associated with different circuit values. The output
stage is not an exception. National has found good results
with a 50µF inductor and a 5µF Mylar capacitor (see Figure
1, Typical Audio Application Circuit) used as the output LC
filter. The two-pole filter contains three components; L1 and
CBYP because the LM4651 and LM4652 have a bridged output. The design formula for a bridge output filter is fC = 1/{2π
[L1(2CBYP + C1)]½} (Hz).
A common mistake is to connect a large capacitor between
ground and each output. This applies only to single-ended
applications. In bridge operation, each output sees CBYP. This
causes the extra factor of 2 in the formula. The alternative to
CBYP is a capacitor connected between each output, VO, and
VO2, and ground. This alternative is, however, not size or cost
efficient because each capacitor must be twice CBYP's value
to achieve the same filter cutoff frequency. The additional
small value capacitors connected between each output and
ground (C1) help filter the high frequency from the output to
ground . The recommended value for C1 is 0.1µF to 1µF or
2% to 20% of CBYP."
10127777
FIGURE 6. Pre−amplifier Stage with Low Pass Filter
Supply Bypassing
Correct supply bypassing has two important goals. The first
is to ensure that noise on the supply lines does not enter the
circuit and become audible in the output. The second is to
help stabilize an unregulated power supply and provide current under heavy current conditions. Because of the two
different goals multiple capacitors of various types and values
are recommended for supply bypassing. For noise de-coupling, generally small ceramic capacitors (.001µF to .1µF)
along with slightly larger tantalum or electrolytic capacitors
(1µF to 10µF) in parallel will do an adequate job of removing
most noise from the supply rails. These capacitors should be
placed as close as possible to each IC's supply pin(s) using
leads as short as possible. For supply stabilizing, large electrolytic capacitors (3,300µF to 15,000µF) are needed. The
value used is design and cost dependent.
High Frequency PCB Design
A double-sided PCB is recommended when designing a class
D amplifier system. One side should contain a ground plane
with the power traces on the other side directly over the
ground plane. The advantage is the parasitic capacitance
created between the ground plane and the power planes. This
parasitic capacitance is very small (pF) but is the value needed for coupling high frequency noise to ground. At high frequencies, capacitors begin to act more like inductors because
of lead and parasitic inductance in the capacitor. For this reason, bypassing capacitors should be surface mount because
of their low parasitic inductance. Equation (8) shows how to
determine the amount of power to ground plane capacitance.
C = εoεrA/d
(Farads)
Modulation Frequency Optimization
Setting the modulation frequency depends largely on the application requirements. To maximize efficiency and output
power a lower modulation frequency should be used. The
lower modulation frequency will lower the amount of loss
caused by switching the output MOSFETs increasing the efficiency a few percent. A lower switching frequency will also
increase the peak output power before clipping because the
over modulation protection time is a smaller percentage of the
total period. Unfortunately, the lower modulation frequency
has worse THD+N performance when the output power is
below 10 watts. The recommended switching frequency to
balance the THD+N performance, efficiency and output power is 125kHz to 145kHz.
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101277 Version 9 Revision 2
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Determining Maximum Power Dissipation
Power dissipation within the integrated circuit package is a
very important parameter. An incorrect maximum power dissipation (PD) calculation may result in inadequate heat sinking, causing thermal shutdown circuitry to operate intermittently. There are two components of power dissipation in a
class D amplifier. One component of power dissipation in the
LM4652 is the RDS(ON) of the FET times the RMS output current
when operating at maximum output power. The other component of power dissipation in the LM4652 is the switching
loss. If the output power is high enough and the DC resistance
of the filter coils is not minimized then significant loss can occur in the output filter. This will not affect the power dissipation
in the LM4652 but should be checked to be sure that the filter
coils with not over heat.
The first step in determining the maximum power dissipation
is finding the maximum output power with a given voltage and
load. Refer to the graph Output Power verses Supply Voltage to determine the output power for the given load and
supply voltage. From this power, the RMS output current can
be calculated as IOUTRMS = SQRT(POUT/RL). The power dissipation caused by the output current is PDOUT = (IOUTRMS)2 *
(2 * RDS(ON)). The value for RDS(ON) can be found from the Electrical Characteristics for the LM4652 table above. The
percentage of loss due to the switching is calculated by Equation (9):
THERMAL CONSIDERATIONS
%LOSSSWITCH = (tr+ tf + TOVERMOD) * fSW
PDSWITCH = (%LOSSSWITCH * POUTMAX) /
(1−%LOSSSWITCH)
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101277 Version 9 Revision 2
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tr, tf and TOVERMOD can be found in the Electrical Characteristic for the LM4651 and Electrical Characteristic for the
LM4652 sections above. The system designer determines
the value for fSW (switching frequency). Power dissipation
caused by switching loss is found by Equation (10). POUTMAX is the 1% output power for the given supply voltage and
the load impedance being used in the application. POUTMAX
can be determined from the graph Output Power vs. Supply
Voltage in the Typical Performance Characteristics section above.
Heat Sinking
The choice of a heat sink for the output FETs in a Class D
audio amplifier is made such that the die temperature does
not exceed TJMAX and activate the thermal protection circuitry
under normal operating conditions. The heat sink should be
chosen to dissipate the maximum IC power which occurs at
maximum output power for a given load. Knowing the maximum output power, the ambient temperature surrounding the
device, the load and the switching frequency, the maximum
power dissipation can be calculated. The additional parameters needed are the maximum junction temperature and the
thermal resistance of the IC package (θJC, junction to case),
Print Date/Time: 2009/07/09 16:41:57
(Watts)
(10)
www.national.com
LM4651 & LM4652
both of which are provided in the Absolute Maximum Ratings and Operating Ratings sections above.
It should be noted that the idea behind dissipating the power
within the IC is to provide the device with a low resistance to
convection heat transfer such as a heat sink. Convection
cooling heat sinks are available commercially and their manufacturers should be consulted for ratings. It is always safer
to be conservative in thermal design.
Proper IC mounting is required to minimize the thermal drop
between the package and the heat sink. The heat sink must
also have enough metal under the package to conduct heat
from the center of the package bottom to the fins without excessive temperature drop. A thermal grease such as Wakefield type 120 or Thermalloy Thermacote should be used
when mounting the package to the heat sink. Without some
thermal grease, the thermal resistance θCS (case to sink) will
be no better than 0.5°C/W, and probably much worse. With
the thermal grease, the thermal resistance will be 0.2°C/W or
less. It is important to properly torque the mounting screw.
Over tightening the mounting screw will cause the package to
warp and reduce the contact area with the heat sink. It can
also crack the die and cause failure of the IC. The recommended maximum torque applied to the mounting screw is 40
inch-lbs. or 3.3 foot-lbs.
THD+N Measurements and Out of Audio Band Noise
THD+N (Total Harmonic Distortion plus Noise) is a very important parameter by which all audio amplifiers are measured.
Often it is shown as a graph where either the output power or
frequency is changed over the operating range. A very important variable in the measurement of THD+N is the bandwidth limiting filter at the input of the test equipment.
Class D amplifiers, by design, switch their output power devices at a much higher frequency than the accepted audio
range (20Hz - 20kHz). Switching the outputs makes the amplifier much more efficient than a traditional Class A/B amplifier. Switching the outputs at high frequency also increases
the out-of-band noise. Under normal circumstances this outof-band noise is significantly reduced by the output low pass
filter. If the low pass filter is not optimized for a given switching
frequency, there can be significant increase in out-of-band
noise.
THD+N measurements can be significantly affected by outof-band noise, resulting in a higher than expected THD+N
measurement. To achieve a more accurate measurement of
THD, the bandwidth at the input of the test equipment must
be limited. Some common upper filter points are 22kHz,
30kHz, and 80kHz. The input filter limits the noise component
of the THD+N measurement to a smaller bandwidth resulting
in a more real-world THD+N value.
The output low pass filter does not remove all of the switching
fundamental and harmonics. If the switching frequency fundamental is in the measurement range of the test equipment,
the THD+N measurement will include switching frequency
energy not removed by the output filter. Whereas the switching frequency energy is not audible, it's presence degrades
the THD+N measurement. Reducing the bandwidth to 30kHz
and 22kHz reveals the true THD performance of the Class D
amplifier. Increasing the switching frequency or reducing the
cutoff frequency of the output filter will also reduce the level
of the switching frequency fundamental and it's harmonics
present at the output. This is caused by a switching frequency
that is higher than the output filter cutoff frequency and, therefore, more attenuation of the switching frequency.
In-band noise is higher in switching amplifiers than in linear
amplifiers because of increased noise from the switching
waveform. The majority of noise is out of band (as discussed
above), but there is also an increase of audible noise. The
output filter design (order and location of poles) has a large
effect on the audible noise level. Power supply voltage also
has an effect on noise level. The output filter removes a certain amount of the switching noise. As the supply increases,
the attenuation by the output fiter is constant. However, the
switching waveform is now much larger resulting in higher
noise levels.
LM4651 & LM4652
PDMAX for the LM4652 is found by adding the two components
(PDSWITCH + PDOUT) of power dissipation together.
sired, RDLY should be a lower value resistor. If a zero value
for RDLY is desired, connect the LM4651's pin 17 to GND.
Determining the Correct Heat Sink
Once the LM4652's power dissipation known, the maximum
thermal resistance (in °C/W) of a heat sink can be calculated.
This calculation is made using Equation (11) and is based on
the fact that thermal heat flow parameters are analogous to
electrical current flow properties.
Determine the Value of L1, CBYP, C1, Rfl1 Rfl2, Cfl1 Cfl2, Rf,
Cf (the Output and Feedback Filters)
All component values show in Figure 1 Typical Audio Application Circuit, are optimized for a subwoofer application.
Use the following guidelines when changing any component
values from those shown. The frequency response of the output filter is controlled by L1 and CBYP. Refer to the Application
Information section titled Output Stage Filtering for a detailed explanation on calculating the correct values for L1 and
CBYP.
C1 should be in the range of 0.1µF to 1µF or 2 - 20% of
CBYP.
Rfl1 and Rfl2 are found by the ratio Rfl1 = 10Rfl2.
A lower ratio can be used if the application is for lower output
voltages than the 125Watt, 4Ω solution show here.
The feedback RC filter's pole location should be higher than
the output filter pole. The reason for two capacitors in parallel
instead of one larger capacitor is to reduce the possible EMI
from the feedback traces. Cfl1 is placed close as possible to
the output of the LM4652 so that an audio signal is present
on the feedback trace instead of a high frequency square
wave. Cfl2 is then placed as close as possible to the feedback
inputs (pins 14, 19) of the LM4651 to filter off any noise picked
up by the feedback traces. The combination lowers EMI and
provides a cleaner audio feedback signal to the LM4651. Rf
should be in range of 100kΩ to1MΩ. Cf controls the bandwidth
of the error signal and should be in the range of 100pF to
470pF.
PDMAX = (TJMAX − TAMBIENTMAX) / θJA (Watts)
(11)
Where θJA = θJC + θCS + θSA
Since we know θJC, θCS, and TJMAX from the Absolute Maximum Ratings and Operating Ratings sections above
(taking care to use the correct θJC for the LM4652 depending
on which package type is being used in the application) and
have calculated PDMAX and TAMBIENTMAX, we only need θSA,
the heat sink's thermal resistance. The following equation is
derived from Equation (11):
θSA = [(TJMAX − TAMBIENTMAX) / PDMAX] − θJC − θCS
Again, it must be noted that the value of θSA is dependent
upon the system designer's application and its corresponding
parameters as described previously. If the ambient temperature surrounding the audio amplifier is higher than
TAMBIENTMAX, then the thermal resistance for the heat sink,
given all other parameters are equal, will need to be lower.
Example Design of a Class D Amplifier
The following is an example of how to design a class D amplifier system for a power subwoofer application utilizing the
LM4651 and LM4652 to meet the design requirements listed
below:
• Output Power, 1% THD
125W
• Load Impedance
4Ω
• Input Signal level
3V RMS (max)
• Input Signal Bandwidth
10Hz − 150Hz
• Ambient Temperature
50°C (max)
Determine the Value for CSTART (Start Up Delay)
The start-up delay is chosen to be 1 second to ensure minimum pops or clicks when the amplifier is powered up. Using
Equation (2), the value of CSTART is 11.7µF. A standard value
of 10µF is used.
Determine the Value of Gain, R1, and R2
The gain is set to produce a 125W output at no more than 1%
distortion with a 3VRMS input. A dissipation of 125W in a 4Ω
load requires a 22.4VRMS signal. To produce this output signal, the LM4651/LM4652 amplifier needs an overall closedloop gain of 22.4VRMS/3VRMS, or 7.5V/V (17.5db). Equation
(12) shows all the variables that affect the system gain.
Determine the Supply Voltage
From the graph Output Power verses Supply voltage at 1%
THD the supply voltage needed for a 125 watt, 4Ω application
is found to be ±20V.
Determine the Value for ROSC(Modulation Frequency)
The oscillation frequency is chosen to obtain a satisfactory
efficiency level while also maintaining a reasonable THD performance. The modulation frequency can be chosen using the
Clipping Power Point and Efficiency verses Switching
Frequency graph. A modulation frequency of 125kHz is
found to be a good middle ground for THD performance and
efficiency. The value of the resistor for ROSC is found from
Equation (6) to be 3.9 kΩ.
Gain = {[R2/(R1 + 100)] x [(Rfl1 + Rfl2)/Rfl2] − [R2/(R1 +
100)] + 0.5} + [(VCC - 20) * 0.0175] (V/V)
The values for RfI1, RfI2, and Rf were found in the Determine
the Value of the Filters section above and shown in Figure
1. Therefore, RfI1 = 620kΩ, RfI2 = 62kΩ and Rf = 390kΩ. The
value of VCC was also found as the first step in this example
to be ±20V. Inserting these values into equation (12) and reducing gives the equation below:
Determine the Value for RSCKT (Circuit Limit)
The current limit is internally set as a failsafe to 10 amps. The
inductor ripple current and the peak output current must be
lower than 10 amps or current limit protection will turn on. A
typical 4Ω load driven by a filter using 50µH inductors does
not require more than 10A. The current limit will have to be
increased when loads less than 4Ω are used to acheive higher
output power. With RSCKT equal to 100kΩ, the current limit is
10A.
R2 = 0.7(R1 + 100)
The input resistance is desired to be 20kΩ so R1 is set to
20kΩ. R2 is then found to require a value of 14.1kΩ. Standard
resistor values are 14.0kΩ giving a gain of 7.43V/V or
14.3kΩ giving a gain of 7.58V/V.
Lowering R2 direcly affects the noise of the system. Changing
R1 to increase gain with the lower value for R2 has very little
affect on the noise level. The percent change in noise is about
what whould be expected with a higher gain. The drawback
to a lower R1 value is a larger CIN value, necessary to properly
couple the lowest desired signal frequencies. If a 20kΩ input
Determine the Value for RDLY (Dead Time Control)
The delay time or dead time is set to the recommended value
so RDLY equals 5kΩ. If a higher bandwidth of operation is dewww.national.com
14
101277 Version 9 Revision 2
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Print Date/Time: 2009/07/09 16:41:57
The total power dissipation in the LM4652 is the sum of these
two power losses giving:
Determine the Needed Heat Sink
The only remaining design requirement is a thermal design
that prevents activating the thermal protection circuitry. Use
Equations (9) - (11) to calculate the amount of power dissipation for the LM4652. The appropriate heat sink size, or
thermal resistance in °C/W, will then be determined.
Equation (9) determines the percentage of loss caused by the
switching. Use the typical values given in the Electrical Characteristics for the LM4651 and Electrical Characteristics
for the LM4652 tables for the rise time, fall time and over
modulation time:
PDTOTAL = 6.6W + 14.4W = 21W
The value for Maximum Power Dissipation given in the System Electrical Characteristics for the LM4651 and
LM4652 is 22 watts. The difference is due to approximately 1
watt of power loss in the LM4651. The above calculations are
for the power loss in the LM4652.
Lastly, use Equation (11) to determine the thermal resistance
of the LM4652's heat sink. The values for θJC and TJMAX are
found in the Operating Ratings and the Absolute Maximum
Ratings section above for the LM4652. The value of θJC is 2°
C/W for the isolated (TF) package or 1°C/W for the non-isolated (T) package. The value for TJMAX is 150°C. The value
for θCS is set to 0.2°C/W since this is a reasonable value when
thermal grease is used. The maximum ambient temperature
from the design requirements is 50°. The value of θSA for the
isolated (TF) package is:
%Loss = (25ns+26ns+350ns) * 125kHz
%Loss = 5.0%
This switching loss causes a maximum power dissipation,
using Equation (10), of:
θSA = [(150°C − 50°C)/21W] − 2°C/W − 0.2°C/W
PDSWITCH = (5.0% * 125W) / (1−5.0%)
θSA = 2.5°C/W
PDSWITCH = 6.6W
and for the non-isolated (T) package without a mica washer
to isolate the heat sink from the package:
Next the power dissipation caused by the RDS(ON) of the output FETs is found by multiplying the output current times the
RDS(ON). Again, the value for RDS(ON) is found from the Electrical Characteristics for the LM4652 table above. The
value for RDS(ON) at 100°C is used since we are calculating
the maximum power dissipation.
θSA = [(150°C − 50°C)/21W] − 1°C/W − 0.2°C/W
θSA = 3.5°C/W
To account for the use of a mica washer simply subtract the
thermal resistance of the mica washer from θSA calculated
above.
IOUTRMS = SQRT(125watts/4Ω) = 5.59 amps
PRDS(ON) = (5.59A)2 * (0.230Ω*2)
PRDS(ON) = 14.4W
RECOMMENDATIONS FOR CRITICAL EXTERNAL COMPONENTS
Circuit Symbol
Suggested
Value
Suggested Type
CfI1
330pF
Ceramic Disc
CfI2
100pF
Ceramic Disc
Supplier/Contact Information
Supplier Part #
Cf
470pF
Ceramic Disc
CB2
1.0µF - 10µF
Resin Dipped Solid Tantalum
CB1 & CBT
0.1µF
Monolithic Ceramic
CB3
0.001µF - 0.1µF
Monolithic Ceramic
C2
0.1µF - 1.0µF
Metallized Polypropylene or
Polyester Film
CBYP
1.0µF - 10µF
Metallized Polypropylene or
Polyester Film
Bishop Electronics Corp.
(562) 695 - 0446
http://www.bishopelectronics.com/
BEC-9950
A11A-50V
CBYP
1.0µF - 10µF
Metallized Polypropylene or
Polyester Film
Nichicon Corp.
(847) 843-7500
http://www.nichicon-us.com/
QAF2Exx
or
QAS2Exx
D1
1A, 50V
Fast Schottky Diode
15
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
www.national.com
LM4651 & LM4652
impedance is not required, then the recommended values
shown in Figure 1, Typical Audio Application Circuit should
be used: with R1's value set to 4.7kΩ and then using a value
of 3.4kΩ for R2 for a gain of 7.5V/V.
LM4651 & LM4652
L1
25µH, 5A
High Current Toroid Inductor
(with header)
J.W. Miller
(310) 515-1720
http://www.jwmiller.com/
6702
L1
47µH, 5A
High Saturation Open Core
(Vertical Mount Power Chokes)
CoilCraft
(847) 639-6400
http://www.coilcraft.com/
PCV-0473-05
L1
50µH, 5.6A
High Saturation Flux Density
Ferrite Rod
J.W. Miller
(310) 515-1720
http://www.jwmiller.com/
5504
L1
68µH, 7.3A
High Saturation Flux Density
Ferrite Rod
J.W. Miller
(310) 515-1720
http://www.jwmiller.com/
5512
10127764
FIGURE 7. Reference PCB Schematic
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16
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
LM4651 & LM4652
10127781
FIGURE 8. Reference PCB Silk Screen Layer
10127780
FIGURE 9. Reference PCB Silk Screen and Solder Mask Layers
17
101277 Version 9 Revision 2
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LM4651 & LM4652
10127782
FIGURE 10. Reference PCB Top Layer
10127779
FIGURE 11. Reference PCB Bottom Layer
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18
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
Symbol
Value
Tolerance
Type
# per
Board
RFL1
620kΩ
1%
1/8 - 1/4 watt
2
RFL2
62kΩ
1%
1/8 - 1/4 watt
2
RFL3
0Ω
1%
1/8 - 1/4 watt
2
RF
1MΩ
1%
1/8 - 1/4 watt
1
R1
4.7kΩ
1%
1/8 - 1/4 watt
1
R2
4.7kΩ
1%
1/8 - 1/4 watt
1
RLP
2.2kΩ
1%
1/8 - 1/4 watt
1
ROFFSET
0
1%
1/8 - 1/4 watt
0
RDLY
5.1kΩ
10%
1/8 - 1/4 watt
1
RSCKT
39kΩ
10%
1/8 - 1/4 watt
1
ROSC
6.8kΩ
10%
1/8 - 1/4 watt
1
Supplier/Comment
Part #
Shorting Jumper
** NOT USED **
Can also use as a 5.6kΩ resistor
All Caps. are Radial lead except CBYP, C1.
Symbol
Value
Tolerance
Type
Voltage
# per
Board
Supplier/Comment
Part #
CIN
1µF
10%
Metal Polyester
100V
1
Digi-Key (800) 344-4539
EF1105-ND
CLP
0.47µF
10%
Metal Polyester
25V
1
Digi-Key (800) 344-4539
EF1474-ND
CF
470pF
5%
Ceramic Disc
25V
1
Digi-Key (800) 344-4539
1321PH-ND
CFL1
0
5%
Ceramic Disc
25V
0
** NOT USED **
1319PH-ND
CFL2
100pF
5%
Ceramic Disc
25V
2
Digi-Key (800) 344-4539
1313PH-ND
CBT
0.1µF
10% - 20%
Monolithic
Ceramic
100V
2
Digi-Key (800) 344-4539
P4924-ND
CB1
0.1µF
10% - 20%
Monolithic
Ceramic
100V
6
Digi-Key (800) 344-4539
P4924-ND
CB2
1µF
10%
Tantalum
Radial lead
35V
6
Digi-Key (800) 344-4539
P2059-ND
CB3
0.001µF
10% - 20%
Monolithic
Ceramic
100V
3
Digi-Key (800) 344-4539
P4898-ND
CB4
47µF
10% - 20%
Electrolytic
Radial
16V
1
Digi-Key (800) 344-4539
P914-ND
CStart
1.5µF
10%
Tantalum
Radial lead
25V
1
Digi-Key (800) 344-4539
P2044-ND
C1
0
10%
Metal Polyester
25V
0
** NOT USED **
C2
1µF
10%
Metal Polyester
25V
2
Digi-Key (800) 344-4539
EF1105-ND
CBYP
4.7µF
10% - 20%
Metal Polyester
50V
1
Digi-Key (800) 344-4539
EF1475-ND
CSBY1
4,700µF
20%
Electrolytic
Radial
25V
2
Digi-Key (800) 344-4539
P5637A-ND
CSBY2
0.1µF
20%
Ceramic Disc
25V
2
Digi-Key (800) 344-4539
P4201-ND
CSBY3
0
50V
0
** NOT USED **
10% - 20% Mylar Axial lead
19
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
www.national.com
LM4651 & LM4652
BILL OF MATERIALS FOR REFERENCE PCB
LM4651 & LM4652
One or more pairs of coils from the list below is included with the reference PCB.
Symbol
Value
Tolerance
Type
Voltage
# per
Board
Supplier/Comment
Part #
L1
25µH
15%
High Current
Toroid with
Header
5.5 amp
2
J.W. Miller (310) 515-1720
6702
L1
47µH
10%
Ferrite Bobbin
Core
5.0 amp
2
CoilCraft (847) 639-6400
http://www.coilcraft.com
PVC-2-473-05
L1
50µH
10%
Ferrite Core
5.6 amp
2
J.W. Miller (310) 515-1720
5504
# per
Board
Supplier/Comment
Part #
(SPDT) on-on, switch for STBY
1
Mouser (800) 346-6873
1055-TA2130
Standoffs
Plastic Round, 0.875", 4-40
4
Newark (800) 463-9275
92N4905
RCA Input
PCB Mount
1
Mouser (800) 346-6873
16PJ097
Banana jack BLACK
5
Mouser (800) 346-6873
164-6218
Wakefield 603K, 2” high X 2” wide, ~ 7°C/W
1
Newark (800) 463-9275
58F537 (603K)
1A, 50Volt Schottky (40A surge current, 8.3mS)
4
Digi-Key (800) 344-4539
SR105CT-ND
Symbol
S1
Banana
Jack
Heat sink
D1
Description
Additional Formulas for Reference PCB:
Pole due to CIN:
f3dB = 1/[2π(R1 + RLP)CIN] or CIN = 1/[2π(R1 + RLP)f3dB]
Pole due to RLP and CLP:
f3dB = 1/[2π(R1 // RLP)CLP] or CLP = 1/[2π(R1 // RLP)f3dB]
where:
(R1 // RLP) = 1/[1/R1 + 1/RLP]
Gain for Reference PCB:
Gain = {[R2/(R1 + 100 + Rlp)] x [(Rfl1 + Rfl2)/Rfl2] − [R2/(R1 + 100 + Rlp)] + 0.5} + [(VCC - 20) * 0.0175]
www.national.com
20
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
PRE-AMPLIFIER AND INPUT FILTER
For a complete solution and best performance a pre-amplifier
is required. With the addition of a pre-amplifier the gain of the
class D stage can be greatly reduced to improve performance. The pre-amplifier gain is set to 10V/V allowing for low
gain on the class D stage with total system gain high enough
to be a complete solution from line level (1VRMS) sources.
Without the pre-amplifier stage the class D stage must have
much higher gain and will result in decreased performance in
the form of much higher THD.
With an extra op. amp. available on the other side of the
LM833N the passive RC input filter is changed to an active
two pole filter. The input filter does not noticeable increase
THD performance but will help maintain a flat frequency response as the Q of the output filter changes with load
impedance. A real speaker load impedance varies with frequency changing the Q of the output filter. The input filter is
recommended to maintain flat response. For the pre-amplifier
and input filter stage the circuit in Figure 6 was used with the
complete input stage shown in Figure 12.
FILTERS
To achieve full bandwidth operation there are several filter
points that must be modified. They are the output filter, the
feedback filters, the error amplifier filter and the input filter. If
any of the filter points are too low there will be large phase
shifts in the upper audio frequencies reducing the resolution
and clarity of the highs. For this reason the frequency response of the system should be flat out to 20kHz. The mistake
is often made to set the –3dB point near 20kHz resulting in
good bench performance but poor quality in listening test.
The output filter is made up of L1, L2, CBYP, CF1, CF2 (see
Figure 13). The output filter design is determined by the load
impedance along with the frequency response. The filter must
have a 3dB point beyond 20kHz and a Q factor close to 0.707
for best performance. The output filter is the only filter that
changes with the load impedance (See Bill of Materials for
Full Audio Bandwidth Reference PCB for values). Standard
inductor values were used for both 4Ω and 8Ω filters.
The feedback filters and error amplifier filters will interact with
the output filter if the individual pole locations of each are too
close together. The feedback filter point is moved by reducing
the value of CFL1, CFL3 to 50pF putting the feedback filter
points approximately 5kHz higher than the output filter point.
The error amplifier filter point is determined by Equation 7.
Reducing the value of CF to 390pF gave the best results.
The input filter in the typical application is a simple passive,
single pole RC filter. For improved performance an active two
pole filter was added as discussed below.
SWITCHING FREQUENCY
A switching frequency from 75kHz to 125kHz is adequate for
subwoofer applications. A lower switching frequency has
higher efficiency and higher output power at the start of clipping. For a full audio bandwidth application a higher switching
frequency is needed. The switching frequency must be increased not only for waveform resolution for the higher audio
frequencies but also to decrease the noise floor. A switching
frequency of 175kHz was used for the performance graphs
shown below. The Audio Precision AUX-0025 Switching Amplifier Measurement Filter was placed before the input to the
Audio Precision unit for the THD+N graphs below.
21
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
www.national.com
LM4651 & LM4652
FULL AUDIO BANDWIDTH OPERATION
There is nothing in the design of the LM4651/52 class D
chipset that prevents full audio bandwidth (20 – 20kHz) operation. For full bandwidth operation there are several external circuit changes required. Additional external circuitry is
helpful to achieve a complete solution with the best performance possible with the LM4651/52 class D chipset. The
additional sections and figures below detail the changes
needed for either a 60W / 8Ω or 100W / 4Ω (10% THD+N)
complete solution using a +/-17V supply.
LM4651 & LM4652
TYPICAL PERFORMANCE FOR FULL RANGE APPLICATION
Frequency Response
±17V, fSW = 175kHz, POUT = 5W = 0dB
RL = 8Ω, No Filters
THD+N vs Frequency
±17V, fSW = 175kHz, POUT = 1W & 25W
RL = 8Ω, 30kHz BW
10127785
10127787
THD+N vs Output Power
±17V, fSW = 175kHz
RL = 8Ω, 30kHz BW
Frequency Response
±17V, fSW = 175kHz, POUT = 5W = 0dB
RL = 4Ω, No Filters
10127789
10127784
THD+N vs Frequency
±17V, fSW = 175kHz, POUT = 1W & 50W
RL = 4Ω, 30kHz BW
THD+N vs Output Power
±17V, fSW = 175kHz
RL = 4Ω, 30kHz BW
10127786
www.national.com
10127788
22
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
LM4651 & LM4652
10127797
FIGURE 12. Input Pre-Amplifier And Filter Schematic
23
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
www.national.com
LM4651 & LM4652
10127796
FIGURE 13. Full Audio Bandwidth Schematic
www.national.com
24
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
LM4651 & LM4652
FULL AUDIO BANDWIDTH REFERENCE BOARD ARTWORK
10127792
FIGURE 14. Composite Top View
10127791
FIGURE 15. Composite Bottom View
25
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
www.national.com
LM4651 & LM4652
10127793
FIGURE 16. Silk Screen Layer
10127794
FIGURE 17. Top Layer
www.national.com
26
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
LM4651 & LM4652
10127790
FIGURE 18. Bottom Layer
27
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
www.national.com
LM4651 & LM4652
BILL OF MATERIALS FOR FULL AUDIO BANDWIDTH REFERENCE PCB
Symbol
Value
Tolerance
Type
Supplier/ Comment
RFL1, RFL2
620kΩ
1%
1/8 – 1/4 Watt
RFL3, RFL4
62kΩ
1%
1/8 – 1/4 Watt
RF
1MΩ
1%
1/8 – 1/4 Watt
R1
10kΩ
1%
1/8 – 1/4 Watt
R2
3.3kΩ
1%
1/8 – 1/4 Watt
ROFFSET
RDLY
Part #
** NOT USED **
5.1kΩ
5%
1/8 – 1/4 Watt
RSCKT
39kΩ
5%
1/8 – 1/4 Watt
ROSC
20kΩ
20%
Trim Potentiometer
RG1, RG2, RG3,
RG4
3.3Ω
5%
1/8 – 1/4 Watt
RTSD
100kΩ
5%
1/8 – 1/4 Watt
RPA
10kΩ
1%
1/8 – 1/4 Watt
Ri
1kΩ
1%
1/8 – 1/4 Watt
RLP1, RLP2
2.7kΩ
1%
1/8 – 1/4 Watt
RIN
47kΩ
5%
1/8 – 1/4 Watt
RV1, RV2
750kΩ
5%
Mouser (800) 346–6873 323–409H-20K
1/4 Watt
All Capacitors are
Radial lead
Symbol
Value
Tolerance
Type
Voltage
Supplier/Comment
Part #
CIN
1µF
10%
Metal Polyester
100V
Digi-Key (800) 344–
4539
EF1105–ND
CLP1
0.0022µF
10%
Ceramic Disc
25V
Digi-Key (800) 344–
4539
P4053A-ND
CLP2
0.001µF
10%
Ceramic Disc
25V
Digi-Key (800) 344–
4539
P4049A-ND
CBT1, CBT2
0.1µF
20%
Monolithic Ceramic
100V
Digi-Key (800) 344–
4539
P4924–ND
CF
390pF
10%
Metal Polyester
25V
Digi-Key (800) 344–
4539
P4932–ND
CFL1, CFL3
47pF
10%
Metal Polyester
50V
Digi-Key (800) 344–
4539
P4845–ND
CFL2
** NOT USED **
CSTART
1.5μF
10%
Tantalum Radial lead
25V
Digi-Key (800) 344–
4539
P2044–ND
CB1, CB2
0.001μF
20%
Monoilthic Ceramic
100V
Digi-Key (800) 344–
4539
P4898–ND
CB3 – CB12
0.1μF
20%
Monolithic Ceramic
100V
Digi-Key (800) 344–
4539
P4924–ND
CS1 – CS5
1μF
10%
Tantalum Radial lead
35V
Digi-Key (800) 344–
4539
P2059–ND
CVD1
0.001μF
20%
Monolithic Ceramic
100V
Digi-Key (800) 344–
4539
P4898–ND
CVD2
47μF
20%
Electrolytic Radial
16V
Digi-Key (800) 344–
4539
P914–ND
CF1, CF2
0.1μF
10%
Metal Polyester
25V
Digi-Key (800) 344–
4539
EF1104–ND
CBYP (4Ω)
0.47μF
10%
Metal Polyester
50V
Digi-Key (800) 344–
4539
EF1474–ND
www.national.com
28
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
0.22μF
10%
Metal Polyester
50V
Digi-Key (800) 344–
4539
EF1224–ND
CSBY1 , CSBY2
4,700μF
20%
Electrolytic Radial
25V
Digi-Key (800) 344–
4539
P10289-ND
CSBY3 , CSBY4
1,000μF
20%
Electrolytic Radial
25V
Digi-Key (800) 344–
4539
P10279–ND
CSBY5 , CSBY6
0.1μF
Ceramic Disc
25V
Digi-Key (800) 344–
4539
P4201–ND
CSBY7, CSBY8
47μF
20%
Electrolytic Radial
16V
Digi-Key (800) 344–
4539
P914–ND
Symbol
Value
Tolerance
Type
Rating
Supplier/Comment
Part #
20%
L1, L2 (4Ω)
10μH
10%
Ferrite Bobbin Core
5.0 amp
CoilCraft (847) 639–
6400
PVC–2–103–05
http://www.coilcraft.com
L1, L2 (8Ω)
22μH
10%
Ferrite Bobbin Core
5.0 amp
CoilCraft (847) 639–
6400
PVC–2–223–05
http://www.coilcraft.com
Symbol
Description
Supplier/Comment
Part #
S1
(SPDT) on-on,
switch for STBY
Mouser (800) 346–6873
1055–TA2130
D1 – D4
1A, 50V Schottky
(40A surge
current, 8.3ms)
Digi-Key (800) 344–
4539
SR105CT-ND
ZDV1, ZDV2
12V, 500mW
Zener diode
Digi-Key (800) 344–
4539
1N5242
Plastic Round,
0.875”, 4–40
Newark (800) 463–9275 92N4905
Banana jack RED
Mouser (800) 346–6873
164–6219
J5
Banana jack
BLACK
Mouser (800) 346–6873
164–6218
J6
RCA jack, PCB
mount
Mouser (800) 346–6873
16PJ097
U1
Dual audio Op.
Amp.
National Semiconductor
LM833N
U2
Integrated Class
D controller and
amplifier
National Semiconductor
LM4651N
U3
H-Bridge Power
MOSFET
National Semiconductor
LM4652
Heat sink
Wakefield 603K,
2” high x 2” wide,
Standoffs
J1, J2, J3, J4
Newark (800) 463–9275 58F537 (603K)
∼7°C/W
29
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
www.national.com
LM4651 & LM4652
CBYP (8Ω)
LM4651 & LM4652
Physical Dimensions inches (millimeters) unless otherwise noted
Order Number LM4651N
NS Package Number N28B
Isolated TO-220 15-Lead Package
Order Number LM4652TF
NS Package Number TF15B
www.national.com
30
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
LM4651 & LM4652
Non-Isolated TO-220 15-Lead Package
Order Number LM4652TA
NS Package Number TA15A
31
101277 Version 9 Revision 2
Print Date/Time: 2009/07/09 16:41:57
www.national.com
LM4651 & LM4652 Overture® 170W Class D Audio Power Amplifier Solution
Notes
For more National Semiconductor product information and proven design tools, visit the following Web sites at:
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www.national.com/amplifiers
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www.national.com/tempsensors SolarMagic™
www.national.com/solarmagic
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PowerWise® Design
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