AOZ1013 EZBuck™ 3A Simple Regulator General Description Features The AOZ1013 is a high efficiency, simple to use, 3A buck regulator. The AOZ1013 works from a 4.5V to 16V input voltage range, and provides up to 3A of continuous output current with an output voltage adjustable down to 0.8V. ● 4.5V to 16V operating input voltage range ● 50mΩ internal PFET switch for high efficiency: up to 95% ● Internal Schottky diode ● Internal soft start ● Output voltage adjustable to 0.8V ● 3A continuous output current ● Fixed 500kHz PWM operation ● Cycle-by-cycle current limit ● Short-circuit protection ● Thermal shutdown ● Small size SO-8 package The AOZ1013 comes in an SO-8 package and is rated over a -40°C to +85°C ambient temperature range. N m o c e ot r d n e m r o f ed d w ne . s n esig Applications ● Point of load DC/DC conversion ● PCIe graphics cards ● Set top boxes ● DVD drives and HDD ● LCD panels ● Cable modems ● Telecom/networking/datacom equipment Typical Application VIN C1 22μF C6 NC VIN From μPC L1 U1 EN AOZ1013 VOUT LX C4 NU COMP RC CC R1 C2 10μF FB C5 1000pF AGND Rs NU GND D1 C3 100μF R2 10kΩ Cs NU Figure 1. 3.3V/3A Buck Regulator Rev. 1.2 October 2009 www.aosmd.com Page 1 of 14 AOZ1013 Ordering Information Part Number Ambient Temperature Range Package Environmental AOZ1013AI* -40°C to +85°C SO-8 RoHS * Not recommended for new designs. Replacement part is AOZ1017. All AOS Products are offering in packaging with Pb-free plating and compliant to RoHS standards. Please visit www.aosmd.com/web/quality/rohs_compliant.jsp for additional information. Pin Configuration VIN 1 8 LX PGND 2 7 LX AGND 3 6 EN FB 4 5 COMP SO-8 (Top View) Pin Description Pin Number Pin Name Pin Function 1 VIN 2 PGND Power ground. Electrically needs to be connected to AGND. 3 AGND Reference connection for controller section. Also used as thermal connection for controller section. Electrically needs to be connected to PGND. 4 FB The FB pin is used to determine the output voltage via a resistor divider between the output and GND. 5 COMP 6 EN The enable pin is active high. Connect EN pin to VIN if not used. Do not leave the EN pin floating. 7, 8 LX PWM output connection to inductor. Thermal connection for output stage. Supply voltage input. When VIN rises above the UVLO threshold the device starts up. External loop compensation pin. Block Diagram VIN UVLO & POR EN Internal +5V 5V LDO Regulator OTP + ISen – Reference & Bias Softstart Q1 ILimit + 0.8V + EAmp FB – – PWM Comp PWM Control Logic + Level Shifter + FET Driver LX LX COMP 500kHz Oscillator AGND Rev. 1.2 October 2009 www.aosmd.com PGND Page 2 of 14 AOZ1013 Absolute Maximum Ratings Recommend Operating Ratings Exceeding the Absolute Maximum ratings may damage the device. The device is not guaranteed to operate beyond the Maximum Operating Ratings. Parameter Rating Supply Voltage (VIN) Parameter 18V Rating Supply Voltage (VIN) 4.5V to 16V LX to AGND -0.7V to VIN+0.3V Output Voltage Range 0.8V to VIN EN to AGND -0.3V to VIN+0.3V Ambient Temperature (TA) -40°C to +85°C FB to AGND -0.3V to 6V 82°C/W COMP to AGND -0.3V to 6V Package Thermal Resistance SO-8 (ΘJA)(2) PGND to AGND -0.3V to +0.3V Junction Temperature (TJ) +150°C Storage Temperature (TS) -65°C to +150°C ESD Rating (1) Note: 2. The value of ΘJA is measured with the device mounted on 1-in2 FR-4 board with 2oz. Copper, in a still air environment with TA = 25°C. The value in any given application depends on the user's specific board design. 2kV Note: 1. Devices are inherently ESD sensitive, handling precautions are required. Human body model rating: 1.5kΩ in series with 100pF. Electrical Characteristics TA = 25°C, VIN = VEN = 12V, VOUT = 3.3V unless otherwise specified(3) Symbol VIN VUVLO Parameter Conditions Supply Voltage Min. Typ. 4.5 Input Under-Voltage Lockout Threshold VIN Rising VIN Falling Max. Units 16 V 4.00 3.70 V Supply Current (Quiescent) IOUT = 0, VFB = 1.2V, VEN > 1.2V 2 3 mA IOFF Shutdown Supply Current VEN = 0V 3 20 mA VFB Feedback Voltage 0.8 0.818 IIN 0.782 V Load Regulation 0.5 % Line Regulation 1 % IFB Feedback Voltage Input Current VEN EN Input Threshold VHYS EN Input Hysteresis 200 Off Threshold On Threshold 0.6 2.0 100 nA V mV MODULATOR fO Frequency 350 DMAX Maximum Duty Cycle 100 DMIN Minimum Duty Cycle 500 600 kHz % 6 % Error Amplifier Voltage Gain 500 V/ V Error Amplifier Transconductance 200 µA / V PROTECTION ILIM Current Limit Over-Temperature Shutdown Limit tSS 4 TJ Rising TJ Falling Soft Start Interval 5 A 145 100 °C 4 ms OUTPUT STAGE High-Side Switch On-Resistance VIN = 12V VIN = 5V 40 65 50 85 mΩ Note: 3. Specification in BOLD indicate an ambient temperature range of -40°C to +85°C. These specifications are guaranteed by design. Rev. 1.2 October 2009 www.aosmd.com Page 3 of 14 AOZ1013 Typical Performance Characteristics Circuit of Figure 1. TA = 25°C, VIN = VEN = 12V, VOUT = 3.3V unless otherwise specified. Light Load (DCM) Operation Full Load (CCM) Operation Vin ripple 50mV/div Vin ripple 0.1V/div Vo ripple 50mV/div Vo ripple 50mV/div IL 2A/div IL 2A/div VLX 10V/div VLX 10V/div 1μs/div 1μs/div Startup to Full Load Full Load to Turnoff Vin 5V/div Vin 5V/div Vo 1V/div Vo 1V/div lin 1A/div lin 1A/div 1ms/div 1ms/div 50% to 100% Load Transient No Load to Turnoff Vo ripple 0.1V/div Vin 5V/div Vo 1V/div lo 2A/div 100μs/div Rev. 1.2 October 2009 lin 1A/div 1s/div www.aosmd.com Page 4 of 14 AOZ1013 Typical Performance Characteristics (Continued) Circuit of Figure 1. TA = 25°C, VIN = VEN = 12V, VOUT = 3.3V unless otherwise specified. Short Circuit Protection Short Circuit Recovery Vo 2V/div Vo 2V/div IL 2A/div IL 2A/div 100μs/div 1ms/div AOZ1013AI Efficiency Efficiency (VIN = 12V) vs. Load Current 8.0V OUTPUT 95 5.0V OUTPUT Efficieny (%) 90 3.3V OUTPUT 85 80 75 0 0.5 1.0 1.5 2.0 Load Current (A) 2.5 3.0 Thermal de-rating curves for SO-8 package part under typical input and output condition based on the evaluation board. 25°C ambient temperature and natural convection (air speed < 50LFM) unless otherwise specified. Derating Curve at 5V Input Derating Curve at 12V Input 3.5 3.5 1.8V, 3.3V, 5.0V OUTPUT 2.5 2.0 1.5 1.0 0 25 1.8V, 5.0V, 8.0V OUTPUT 3.0 Output Current (IO) Output Current (IO) 3.0 3.3V OUTPUT 2.5 2.0 1.5 1.0 35 45 55 65 75 85 0 25 Ambient Temperature (TA) Rev. 1.2 October 2009 35 45 55 65 75 85 Ambient Temperature (TA) www.aosmd.com Page 5 of 14 AOZ1013 Detailed Description The AOZ1013 is a current-mode step down regulator with integrated high side PMOS switch. It operates from a 4.5V to 16V input voltage range and supplies up to 3A of load current. The duty cycle can be adjusted from 6% to 100% allowing a wide range of output voltage. Features include enable control, Power-On Reset, input under voltage lockout, fixed internal soft-start and thermal shut down. V O_MAX = V IN – I O × ( R DS ( ON ) + R inductor ) The AOZ1013 is available in SO-8 package. Enable and Soft Start The AOZ1013 has internal soft start feature to limit in-rush current and ensure the output voltage ramps up smoothly to regulation voltage. A soft start process begins when the input voltage rises to 4.0V and voltage on EN pin is HIGH. In soft start process, the output voltage is ramped to regulation voltage in typically 4ms. The 4ms soft start time is set internally. The EN pin of the AOZ1013 is active high. Connect the EN pin to VIN if enable function is not used. Pulling EN to ground will disable the AOZ1013. Do not leave it open. The voltage on EN pin must be above 2.0V to enable the AOZ1013. When voltage on EN pin falls below 0.6V, the AOZ1013 is disabled. If an application circuit requires the AOZ1013 to be disabled, an open drain or open collector circuit should be used to interface to EN pin. Steady-State Operation Under steady-state conditions, the converter operates in fixed frequency and Continuous-Conduction Mode (CCM). The AOZ1013 integrates an internal P-MOSFET as the high-side switch. Inductor current is sensed by amplifying the voltage drop across the drain to source of the high side power MOSFET. Output voltage is divided down by the external voltage divider at the FB pin. The difference of the FB pin voltage and reference is amplified by the internal transconductance error amplifier. The error voltage, which shows on the COMP pin, is compared against the current signal, which is sum of inductor current signal and ramp compensation signal, at PWM comparator input. If the current signal is less than the error voltage, the internal high-side switch is on. The inductor current flows from the input through the inductor to the output. When the current signal exceeds the error voltage, the high-side switch is off. The inductor current is freewheeling through the external Schottky diode to output. Rev. 1.2 October 2009 The AOZ1013 uses a P-Channel MOSFET as the high side switch. It saves the bootstrap capacitor normally seen in a circuit which is using an NMOS switch. It allows 100% turn-on of the upper switch to achieve linear regulation mode of operation. The minimum voltage drop from VIN to VO is the load current times DC resistance of MOSFET plus DC resistance of buck inductor. It can be calculated by equation below: where; VO_MAX is the maximum output voltage, VIN is the input voltage from 4.5V to 16V, IO is the output current from 0A to 3A, RDS(ON) is the on resistance of internal MOSFET, the value is between 40mΩ and 70mΩ depending on input voltage and junction temperature, and Rinductor is the inductor DC resistance. Switching Frequency The AOZ1013 switching frequency is fixed and set by an internal oscillator. The practical switching frequency could range from 350kHz to 600kHz due to device variation. Output Voltage Programming Output voltage can be set by feeding back the output to the FB pin with a resistor divider network. In the application circuit shown in Figure 1. The resistor divider network includes R1 and R2. Usually, a design is started by picking a fixed R2 value and calculating the required R1 with equation below: R 1⎞ ⎛ V O = 0.8 × ⎜ 1 + -------⎟ R 2⎠ ⎝ Some standard values of R1 and R2 for the most commonly used output voltage values are listed in Table 1. Table 1. R1 (kΩ) VO (V) R2 (kΩ) 0.8 1.0 Open 1.2 4.99 10 1.5 10 11.5 1.8 12.7 10.2 2.5 21.5 10 3.3 31.6 10 5.0 52.3 10 www.aosmd.com Page 6 of 14 AOZ1013 The combination of R1 and R2 should be large enough to avoid drawing excessive current from the output, which will cause power loss. Thermal Protection Since the switch duty cycle can be as high as 100%, the maximum output voltage can be set as high as the input voltage minus the voltage drop on upper PMOS and inductor. An internal temperature sensor monitors the junction temperature. It shuts down the internal control circuit and high side PMOS if the junction temperature exceeds 145°C. The regulator will restart automatically under the control of soft-start circuit when the junction temperature decreases to 100°C. Protection Features Application Information The AOZ1013 has multiple protection features to prevent system circuit damage under abnormal conditions. The basic AOZ1013 application circuit is shown in Figure 1. Component selection is explained below. Input Capacitor Over Current Protection (OCP) The sensed inductor current signal is also used for over current protection. Since the AOZ1013 employs peak current mode control, the COMP pin voltage is proportional to the peak inductor current. The COMP pin voltage is limited to be between 0.4V and 2.5V internally. The peak inductor current is automatically limited cycle by cycle. The cycle by cycle current limit threshold is set between 4A and 5A. When the load current reaches the current limit threshold, the cycle by cycle current limit circuit turns off the high side switch immediately to terminate the current duty cycle. The inductor current stops rising. The cycle by cycle current limit protection directly limits inductor peak current. The average inductor current is also limited due to the limitation on peak inductor current. When the cycle by cycle current limit circuit is triggered, the output voltage drops as the duty cycle is decreasing. The AOZ1013 has internal short circuit protection to protect itself from catastrophic failure under output short circuit conditions. The FB pin voltage is proportional to the output voltage. Whenever FB pin voltage is below 0.2V, the short circuit protection circuit is triggered. As a result, the converter is shut down and hiccups at a frequency equal to 1/8 of normal switching frequency. The converter will start up via a soft start once the short circuit condition disappears. In short circuit protection mode, the inductor average current is greatly reduced because of the low hiccup frequency. Power-On Reset (POR) The input capacitor must be connected to the VIN pin and PGND pin of the AOZ1013 to maintain steady input voltage and filter out the pulsing input current. The voltage rating of input capacitor must be greater than maximum input voltage plus ripple voltage. The input ripple voltage can be approximated by the equation below: VO ⎞ VO IO ⎛ ΔV IN = ----------------- × ⎜ 1 – ---------⎟ × --------f × C IN ⎝ V IN⎠ V IN Since the input current is discontinuous in a buck converter, the current stress on the input capacitor is another concern when selecting the capacitor. For a buck circuit, the RMS value of input capacitor current can be calculated by: VO ⎛ VO ⎞ - ⎜ 1 – --------⎟ I CIN_RMS = I O × -------V IN ⎝ V IN⎠ If let m equal the conversion ratio: VO -------- = m V IN The relationship between the input capacitor RMS current and voltage conversion ratio is calculated and shown in Figure 2 on the next page. It can be seen that when VO is half of VIN, CIN is under the worst current stress. The worst current stress on CIN is 0.5 x IO . A power-on reset circuit monitors the input voltage. When the input voltage exceeds 4V, the converter starts operation. When input voltage falls below 3.7V, the converter shuts down. Rev. 1.2 October 2009 www.aosmd.com Page 7 of 14 AOZ1013 The peak inductor current is: 0.5 ΔI L I Lpeak = I O + -------2 0.4 High inductance gives low inductor ripple current but requires a larger size inductor to avoid saturation. Low ripple current reduces inductor core losses. Low ripple current also reduces RMS current through the inductor and switches, which results in less conduction loss. ICIN_RMS(m) 0.3 IO 0.2 0.1 0 0 0.5 m 1 Figure 2. ICIN vs. Voltage Conversion Ratio For reliable operation and best performance, the input capacitors must have current rating higher than ICIN_RMS at the worst operating conditions. Ceramic capacitors are preferred for input capacitors because of their low ESR and high ripple current rating. Depending on the application circuits, other low ESR tantalum capacitors or aluminum electrolytic capacitors may also be used. When selecting ceramic capacitors, X5R or X7R type dielectric ceramic capacitors are preferred for their better temperature and voltage characteristics. Note that the ripple current rating from capacitor manufactures is based on certain amount of life time. Further de-rating may be necessary for practical design requirement. Inductor The inductor is used to supply constant current to the output when it is driven by a switching voltage. For a given input and output voltage, inductance and switching frequency together decide the inductor ripple current, which is: VO ⎛ VO ⎞ -⎟ ΔI L = ----------- × ⎜ 1 – -------f×L ⎝ V IN⎠ When selecting the inductor, make sure it is able to handle the peak current at the highest operating temperature without saturation. The inductor takes the highest current in a buck circuit. The conduction loss on the inductor needs to be checked for thermal and efficiency requirements. Surface mount inductors in different shape and styles are available from Coilcraft, Elytone and Murata. Shielded inductors are small and radiate less EMI noise, but they cost more than unshielded inductors. The choice depends on EMI requirement, price and size. Table 2 lists some inductors for typical output voltage design. Output Capacitor The output capacitor is selected based on the DC output voltage rating, output ripple voltage specification, and ripple current rating. The selected output capacitor must have a higher rated voltage specification than the maximum desired output voltage including ripple. De-rating needs to be considered for long term reliability. Output ripple voltage specification is another important factor for selecting the output capacitor. In a buck converter circuit, output ripple voltage is determined by inductor value, switching frequency, output capacitor Table 2. Typical Inductors VOUT 5.0V 3.3V 1.8V Rev. 1.2 October 2009 L1 Manufacture Shielded, 6.8µH, MSS1278-682MLD Coilcraft Shielded, 6.8µH, MSS1260-682MLD Coilcraft Un-shielded, 4.7µH, DO3316P-472MLD Coilcraft Shielded, 4.7µH, DO1260-472NXD Coilcraft Shielded, 3.3µH, ET553-3R3 ELYTONE Shield, 2.2µH, ET553-2R2 ELYTONE Un-shielded, 3.3µH, DO3316P-222MLD Coilcraft Shielded, 2.2µH, MSS1260-222NXD Coilcraft www.aosmd.com Page 8 of 14 AOZ1013 value and ESR. It can be calculated by the equation below: 1 ΔV O = ΔI L × ⎛ ESR CO + -------------------------⎞ ⎝ 8×f×C ⎠ O With peak current mode control, the buck power stage can be simplified to be a one-pole and one-zero system in frequency domain. The pole is dominant pole and can be calculated by: 1 f P1 = ----------------------------------2π × C O × R L where, CO is output capacitor value, and The zero is a ESR zero due to output capacitor and its ESR. It is can be calculated by: ESRCO is the equivalent series resistance of the output capacitor. When low ESR ceramic capacitor is used as output capacitor, the impedance of the capacitor at the switching frequency dominates. Output ripple is mainly caused by capacitor value and inductor ripple current. The output ripple voltage calculation can be simplified to: 1 ΔV O = ΔI L × ------------------------8×f×C 1 f Z1 = -----------------------------------------------2π × C O × ESR CO where; CO is the output filter capacitor, RL is load resistor value, and ESRCO is the equivalent series resistance of output capacitor. O If the impedance of ESR at switching frequency dominates, the output ripple voltage is mainly decided by capacitor ESR and inductor ripple current. The output ripple voltage calculation can be further simplified to: ΔV O = ΔI L × ESR CO The compensation design is actually to shape the converter close loop transfer function to get desired gain and phase. Several different types of compensation networks can be used for AOZ1013. For most cases, a series capacitor and resistor network connected to the COMP pin sets the pole-zero and is adequate for a stable high-bandwidth control loop. For lower output ripple voltage across the entire operating temperature range, X5R or X7R dielectric type of ceramic, or other low ESR tantalum capacitor or aluminum electrolytic capacitor may also be used as output capacitors. In the AOZ1013, FB pin and COMP pin are the inverting input and the output of internal transconductance error amplifier. A series R and C compensation network connected to COMP provides one pole and one zero. The pole is: In a buck converter, output capacitor current is continuous. The RMS current of output capacitor is decided by the peak to peak inductor ripple current. It can be calculated by: G EA f P2 = ------------------------------------------2π × C C × G VEA ΔI L I CO_RMS = ---------12 GEA is the error amplifier transconductance, which is 200 x 10-6 A/V, GVEA is the error amplifier voltage gain, which is 500 V/V, and Usually, the ripple current rating of the output capacitor is a smaller issue because of the low current stress. When the buck inductor is selected to be very small and inductor ripple current is high, output capacitor could be overstressed. Loop Compensation The AOZ1013 employs peak current mode control for easy use and fast transient response. Peak current mode control eliminates the double pole effect of the output L&C filter. It greatly simplifies the compensation loop design. Rev. 1.2 October 2009 where; CC is compensation capacitor. The zero given by the external compensation network, capacitor CC and resistor RC ,is located at: 1 f Z2 = ----------------------------------2π × C C × R C To design the compensation circuit, a target crossover frequency fC for close loop must be selected. The system crossover frequency is where control loop has unity gain. The crossover frequency is also called the converter bandwidth. Generally a higher bandwidth means faster response to load transient. However, the bandwidth should not be too high because of system stability www.aosmd.com Page 9 of 14 AOZ1013 concerns. When designing the compensation loop, converter stability under all line and load condition must be considered. Usually, it is recommended to set the bandwidth to be less than 1/10 of switching frequency. The AOZ1013 operates at a fixed switching frequency range from 350kHz to 600kHz. The recommended crossover frequency is less than 30kHz. Table 3. VOUT L1 RC CC 1.8V 2.2µH 49.9kΩ 1.5nF 3.3V 4.7µH 20kΩ 2.2nF 5V 6.8µH 49.9kΩ 1.2nF 8V 10µH 49.9kΩ 1.2nF Thermal Management and Layout Consideration f C = 30kHz The strategy for choosing RC and CC is to set the cross over frequency with RC and set the compensator zero with CC. Using selected crossover frequency, fC, to calculate RC: VO 2π × C O R C = f C × ---------- × ----------------------------V G ×G FB EA CS where; fC is the desired crossover frequency, VFB is 0.8V, GEA is the error amplifier transconductance, which is 200 x 10-6 A/V, and GCS is the current sense circuit transconductance, which is 6.68 A/V. The compensation capacitor CC and resistor RC together make a zero. This zero is put somewhere close to the dominate pole, fP1, but lower than 1/5 of the selected crossover frequency. CC can is selected by: 1.5 C C = ----------------------------------2π × R C × f P1 In the AOZ1013 buck regulator circuit, high pulsing current flows through two circuit loops. The first loop starts from the input capacitors, to the VIN pin, to the LX pins, to the filter inductor, to the output capacitor and load, and then return to the input capacitor through ground. Current flows in the first loop when the high side switch is on. The second loop starts from inductor, to the output capacitors and load, to the anode of Schottky diode, to the cathode of Schottky diode. Current flows in the second loop when the low side diode is on. In PCB layout, minimizing the two loops area reduces the noise of this circuit and improves efficiency. A ground plane is strongly recommended to connect input capacitor, output capacitor, and PGND pin of the AOZ1013. In the AOZ1013 buck regulator circuit, the two major power dissipating components are the AOZ1013, the Schottky diode, and output inductor. The total power dissipation of converter circuit can be measured by input power minus output power. P total_loss = V IN × I IN – V O × I O The power dissipation in Schottky can be approximately calculated as: The previous equation can also be simplified to: P diode_loss = IO × ( 1 – D ) × V FW_Schottky CO × RL C C = --------------------RC where; VFW_Schottky is the Schottky diode forward voltage drop. An easy-to-use application software which helps to design and simulate the compensation loop can be found at www.aosmd.com. Table 3 lists the values for a typical output voltage design when output is 44µF ceramics capacitor. The power dissipation of inductor can be approximately calculated by output current and DCR of inductor. P inductor_loss = IO2 × R inductor × 1.1 The actual junction temperature can be calculated with power dissipation in the AOZ1013 and thermal impedance from junction to ambient is: T junction = ( P total_loss – P inductor_loss ) × Θ JA Rev. 1.2 October 2009 www.aosmd.com Page 10 of 14 AOZ1013 The maximum junction temperature of AOZ1013 is 145°C, which limits the maximum load current capability. Please see the thermal de-rating curves for maximum load current of the AOZ1013 under different ambient temperatures. The thermal performance of the AOZ1013 is strongly affected by the PCB layout. Extra care should be taken by users during the design process to ensure that the IC will operate under the recommended environmental conditions. Several layout tips are listed below for the best electric and thermal performance. Figure 3 below illustrates a PCB layout example as a reference. 1. Do not use thermal relief connection to the VIN and the PGND pin. Pour a maximized copper area to the PGND pin and the VIN pin to help thermal dissipation. 3. A ground plane is preferred. If a ground plane is not used, separate PGND from AGND and connect them only at one point to avoid the PGND pin noise coupling to the AGND pin. 4. Make the current trace from LX pins to L to Co to the PGND as short as possible. 5. Pour copper plane on all unused board area and connect it to stable DC nodes, like VIN, GND, or VOUT. 6. The two LX pins are connected to the internal PFET drain. They are low resistance thermal conduction path and most noisy switching node. Connect a copper plane to the LX pin to help thermal dissipation. This copper plane should not be too large otherwise switching noise may be coupled to other parts of the circuit. 7. Keep sensitive signal traces away from the LX pins. 2. Input capacitor should be connected to the VIN pin and the PGND pin as close as possible. L Cin VIN 1 PGND 2 AGND FB 8 LX 7 LX 3 6 EN 4 5 COMP SO-8 Cc Cout Rc Figure 3. AOZ1013 PCB Layout Rev. 1.2 October 2009 www.aosmd.com Page 11 of 14 AOZ1013 Package Dimensions, SO-8L D Gauge Plane Seating Plane e 0.25 8 L E E1 h x 45° 1 C θ 7° (4x) A2 A 0.1 b A1 Dimensions in millimeters 2.20 5.74 1.27 Min. 1.35 0.10 1.25 0.31 Nom. Max. 1.65 — 1.50 — 1.75 0.25 1.65 0.51 c D E1 0.17 4.80 3.80 — 4.90 0.25 5.00 e E 0.80 Unit: mm Symbols A A1 A2 b h L θ 3.90 4.00 1.27 BSC 5.80 6.00 6.20 0.25 — 0.50 0.40 — 1.27 0° — 8° Dimensions in inches Symbols A A1 A2 b Min. 0.053 0.004 0.049 0.012 Nom. 0.065 — 0.059 — Max. 0.069 0.010 0.065 0.020 c D E1 0.007 0.189 0.150 — 0.193 0.154 0.010 0.197 0.157 e E 0.228 h L θ 0.010 0.016 0° 0.050 BSC 0.236 0.244 — — — 0.020 0.050 8° Notes: 1. All dimensions are in millimeters. 2. Dimensions are inclusive of plating 3. Package body sizes exclude mold flash and gate burrs. Mold flash at the non-lead sides should be less than 6 mils. 4. Dimension L is measured in gauge plane. 5. Controlling dimension is millimeter, converted inch dimensions are not necessarily exact. Rev. 1.2 October 2009 www.aosmd.com Page 12 of 14 AOZ1013 Tape and Reel Dimensions SO-8 Carrier Tape P1 D1 See Note 3 P2 T See Note 5 E1 E2 E See Note 3 B0 K0 A0 D0 P0 Feeding Direction Unit: mm Package SO-8 (12mm) A0 6.40 ±0.10 B0 5.20 ±0.10 K0 2.10 ±0.10 D0 1.60 ±0.10 D1 1.50 ±0.10 E 12.00 ±0.10 SO-8 Reel E1 1.75 ±0.10 E2 5.50 ±0.10 P0 8.00 ±0.10 P1 4.00 ±0.10 P2 2.00 ±0.10 T 0.25 ±0.10 W1 S G N M K V R H W N Tape Size Reel Size M W 12mm ø330 ø330.00 ø97.00 13.00 ±0.10 ±0.30 ±0.50 W1 17.40 ±1.00 H K ø13.00 10.60 +0.50/-0.20 S 2.00 ±0.50 G — R — V — SO-8 Tape Leader/Trailer & Orientation Trailer Tape 300mm min. or 75 empty pockets Rev. 1.2 October 2009 Components Tape Orientation in Pocket www.aosmd.com Leader Tape 500mm min. or 125 empty pockets Page 13 of 14 AOZ1013 AOZ1013 Package Marking Z1013AI FAYWLT Part Number Code Assembly Lot Code Fab & Assembly Location Year & Week Code This data sheet contains preliminary data; supplementary data may be published at a later date. Alpha & Omega Semiconductor reserves the right to make changes at any time without notice. LIFE SUPPORT POLICY ALPHA & OMEGA SEMICONDUCTOR PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body or (b) support or sustain life, and (c) whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury of the user. Rev. 1.2 October 2009 2. A critical component in any component of a life support, device, or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. www.aosmd.com Page 14 of 14