® ® TOPSwitch Flyback Design Methodology Application Note AN-16 Designing an off-line switching power supply involves many aspects of electrical engineering: analog and digital circuits, bipolar and MOS power device characteristics, magnetics, thermal considerations, safety requirements, control loop stability, etc. This presents an enormous challenge involving complex trade-offs with a large number of design variables. However, with TOPSwitch’s high level of integration, this design task has been greatly simplified. Because of the significantly reduced number of design variables and built-in loop stability, it is possible to develop a simple step-by-step design method that is easy to follow and quickly leads to satisfactory results. simplified version of the design method which, at implementation level, guides the reader from a set of given system requirements/ specifications all the way to the completion of the desired TOPSwitch flyback power supply using rules of thumb, look up tables and a simple spread sheet program. The information section, at optimization level, makes available the key background information for the design method, such as equations and guidelines. Cross references are provided among the three which allow the reader to switch among conceptual, implementation and optimization levels at any given stage for an in-depth understanding and/or further optimization. Basic circuit configuration Introduction The design of a switching power supply, by nature, is an iterative process with many variables that have to be adjusted to optimize the design. The design method described below consists of three parts: a complete design flow chart, a simplified step-by-step design procedure and an in-depth information section. The flow chart, at conceptual level, serves as a map providing an overall picture and guideline for the complete design methodology. The step-by-step design procedure is a Because of the high level integration of TOPSwitch, many power supply design issues are resolved in the chip. Far fewer issues are left to be addressed externally, resulting in one basic circuit configuration remaining unchanged from application to application. Different output power levels may require the use of different values for some circuit components, but the circuit configuration itself stays valid. Application specific issues outside of the basic flyback converter requirements (such as constant current, constant power outputs, etc.) are beyond the Output Capacitor Clamp Zener Output Post Filter L, C +VD- V+ + VO - FUSE VAC CX +VDB- CM CHOKE + VB - Blocking Diode CIN TOPSwitch Bias Capacitor DRAIN SOURCE CONTROL Feedback Circuit V- Control Pin Capacitor and Series Resistor PI-1849-050696 Figure 1. Typical TOPSwitch Flyback Power Supply. June 1996 AN-16 1. System Requirements VACMIN, VACMAX, fL, VO, PO, η, Z Step 1-2 Determine System Level Requirements and Choose Feedback Circuit 2. Choose Feedback Circuit & VB 3. Determine CIN, VMIN 4. Determine VOR, VCLO 5. Determine DMAX 6. Set KRP 7. Determine IAVG, IP, IR, IRMS Step 3-11 Choose The Smallest TOPSwitch For The Required Power 8. Choose TOPSwitch 9. Calculate TOPSwitch loss PD Y 10. PD Too High N N 11. IP= 0.9 x ILIMIT or KRP= 1 Y To Step 12 PI-1868-052896 Figure 2A. TOPSwitch Flyback Design Flow Chart, Step 1 to 11. scope of this application note. However, such requirements are usually implemented by adding additional circuitry to the basic converter configuration. The only part of the circuit configuration that may change is the feedback circuitry. Depending on the power supply output requirement, one of four possible circuits, shown in Figures 3-6, will be chosen for the application. The basic circuit configuration used in most TOPSwitch flyback power supplies is shown in Figure 1 which also serves as the reference circuit for component identifications used in the descriptions throughout this application note. 2 A 6/96 Design Flow Figure 2A, B and C present a design flow chart showing the complete design procedure in 35 steps. With the basic circuit configuration shown in Figure 1 as its foundation, the logic behind this design approach can be summarized as following: 1. Determine system requirements and decide on feedback circuit accordingly. 2. Find the smallest TOPSwitch capable for the application. 3. Design the smallest transformer for the TOPSwitch chosen. 4. Select all other components in Figure 1 to complete the design. AN-16 From Step 11 12. Determine LP 13. Choose Core & Bobbin Determine A , Le,AL, BW e Y 22. NS, L Iterated N 14. Set NS, L 15. Calculate NP, NB 16. Calculate BM N 17. 2000 ≤ BM≤ 3000 Y Step 12-24 Design The Smallest Transformer To Work with The TOPSwitch Chosen 18. Calculate Lg N 19. Lg > 0.051 mm Y 20. Calculate OD, DIA, CMA N 21. 200 ≤ CMA ≤ 500 Y 23. Calculate ISP, ISRMS, IRIPPLE DIAS, ODS 24. Calculate PIVS, PIVB To Step 25 PI-1869-052896 Figure 2B. TOPSwitch Flyback Design Flow Chart, Step 12 to 24. The overriding objective of this procedure is “design for cost effectiveness”. Using smaller components will usually lead to a less expensive power supply. However, for applications with stringent size or weight limitations, the designer may need to strike a compromise between cost and specific design requirements in order to achieve the ultimate cost effectiveness at the end product. A 6/96 3 AN-16 From Step 24 25. Select Clamp Zener & Blocking Diode 26. Select Output Rectifier 27. Select Output Capacitor 28. Switching Ripple Too High N Y 29. Select Output Post Filter L, C 30. Select Bias Rectifier Step 25-35 Select Other Components 31. Select Bias Capacitor 32. Select Control Pin Capacitor & Series Resistor 33. Select Feedback Circuit Compenents According to Reference Designs: RD1, ST202A, ST204A 34. Select Bridge Rectifier 35. Design Complete PI-1870-052896 Figure 2C. TOPSwitch Flyback Design Flow Chart, Step 25 to 35. 4 A 6/96 AN-16 Step by Step Design Procedure This design procedure uses the AN-17 spreadsheet (available from Power Integrations), which contains all the important equations required for a TOPSwitch flyback power supply design, and automates most calculations. Designers therefore are relieved from the tedious calculations involved in the complicated and highly iterative design process. Anytime a parameter is involved in a calculation, whether it is an input or an output, a cell location for the parameter will be shown in parenthesis at the right side of the page. For example (A1) denotes column A and row 1. Note that all user provided inputs are in column B and all spreadsheet calculated results are in column D. Column C is reserved for intermediate variables needed in some complicated calculations. Look up tables and rules of thumb are also provided wherever appropriate, to facilitate the design task. For questions regarding any particular step of this procedure, please refer to the corresponding step in the information section, where in-depth explanation is provided. Step 1. Determine system requirements: VACMAX, VACMIN, fL, fS, VO, PO, η, Z • Set minimum AC input voltage, VACMIN, per Table 1 • Set maximum AC input voltage, VACMAX, per Table 1 Input (VAC) 100/115 Universal 230 VACMIN (VAC) 85 85 195 (B3) (B4) VACMAX (VAC) 132 265 265 Table 1 • • • • • • Line frequency, fL: 50Hz or 60Hz TOPSwitch switching frequency, fS: 100KHz Output voltage, VO: in Volts Output power, PO: in Watts Power supply efficiency, η: 0.8 if no better reference data available Loss allocation factor, Z: 0.5 if no better reference data available (B5) (B6) (B7) (B8) (B9) (B10) Step 2. Choose feedback circuit and bias voltage VB based on output requirements: • Select a feedback circuit (Figures 3-6) based on output specification: Feedback Circuit Output VB (V) Accuracy Primary/basic 5.7 Primary/enhanced 27.7 Opto/Zener 12 Opto/TL431 12 ±10% ±5% ±5% ±1% Load Regulation Line Regulation Reference Design ±5% ±2.5% ±1% ±0.2% ±1.5% ±1.5% ±0.5% ±0.2% RD1 RD1 ST202A ST204A Table 2 • Choose required bias voltage, VB , per Table 2 (B11) A 6/96 5 AN-16 Step 3. Determine input storage capacitor CIN and minimum DC input voltage VMIN based on input voltage and P O: • Set bridge rectifier conduction time, tC = 3 mS • Choose input storage capacitor, CIN , per Table 3 • Derive minimum DC input voltage, VMIN Input (VAC) 100/115 Universal 230 CIN (µF/Watt of P O) 2~3 2~3 1 (B12) (B13) (D33) VMIN (V) ≥90 ≥90 ≥240 Table 3 Step 4. Determine reflected output voltage VOR and clamp Zener voltage VCLO based on input voltage: • Set VOR based on input voltages per Table 4 Input (VAC) 100/115 Universal 230 VOR (V) 60 135 135 (B16) VCLO (V) 90 200 200 Table 4 Note : VCLO is to be used in Step 25 for clamp Zener selection Step 5. Determine DMAX based on VMIN and VOR: • Set TOPSwitch Drain to Source voltage, VDS = 10 V • Determine maximum duty cycle at low line, DMAX (B17) (D37) Step 6. Set value for primary ripple current IR to primary peak current IP ratio, KRP: • Starting with: KRP = 0.4 for 100/115 VAC or universal input 0.6 for 230 VAC • KRP must be kept within the range specified in Table 5 throughout iteration KRP Input (VAC) 100/115 Universal 230 Table 5 6 A 6/96 Minimum (Most Continuous) 0.4 0.4 0.6 Maximum (Discontinuous) 1.0 1.0 1.0 (B20) AN-16 Step 7. Determine primary waveform parameters IAVG, IP, IR, IRMS: • • • • Calculate average input current, IAVG : in Amps Calculate primary Peak current, IP : in Amps Calculate primary ripple current, IR : in Amps Calculate primary RMS current, IRMS : in Amps (D38) (D39) (D40) (D41) Step 8 to Step 10. Choose the smallest possible TOPSwitch for the job under practical thermal limitation • Start with the smallest TOPSwitch based on minimum current limit spec such that 0.9 x ILIMIT(minimum) ≥ IP • Refer to AN-14 Table 2 for thermal considerations. Select larger TOPSwitch if necessary. Step 11. Check minimum ILIMIT of the selected TOPSwitch against required peak current IP. Increase KRP until KRP = 1.0 or IP = 0.9 x ILIMIT(minimum) • Enter new value of KRP • Monitor IP • Iterate until KRP = 1.0 or IP = 0.9 x ILIMIT (minimum) Step 12. Calculate primary inductance LP (B20) (D39) (D44) Step 13. Choose core and bobbin based on PO using AN-18 Appendix A, Table 2 and determine Ae, L e, AL and BW from core and bobbin catalog: • • • • Core effective cross sectional area, Ae: in cm2 Core effective path length, Le : in cm Core ungapped effective inductance, AL: in nH/turn2 Bobbin width, BW : in mm (B24) (B25) (B26) (B27) Step 14. Set value for number of primary layers L and number of secondary turns NS (may need iteration): • Starting with L = 2 (Keep 1.0 ≤ L ≤ 2.0 throughout iteration) • Starting with NS= 1 turn/volt for 100/115 VAC 0.6 turn/volt for 230 VAC and universal inputs • Both L and NS may need iteration (B29) (B30) Step 15. Calculate number of primary turns NP and number of bias turns NB : • • • • • Diode voltages: use 0.7V for P/N diode and 0.4V for schottky diode Set output rectifier forward voltage, VD Set bias rectifier forward voltage, VDB Calculate number of primary turns NP Calculate number of bias turns NB (B18) (B19) (D45) (D46) A 6/96 7 AN-16 Step 16 to Step 22. Check BM, CMA and Lg. Iterate if necessary by changing L, NS or core/bobbin until within specified range: • Set safety margin, M. Use 3 mm (118 mils) for margin wound with 230 VAC or universal input and 1.5 mm (59 mils) for 110/115 VAC input. Set to zero for triple insulated secondary. • Maximum flux density, BM: 3000 ≥ B M ≥ 2000; in Gauss • Gap length, Lg: Lg ≥ 0.051 mm • Primary winding current capacity, CMA: 500 ≥ CMA ≥ 200; in circular mils per Amp • Iterate by changing L, NS, core/bobbin according to Table 6 • Primary minimum conductor diameter, DIA: in mm • Primary maximum wire outside diameter, OD: in mm BM Lg (B28) (D48) (D51) (D58) (D55) (D53) CMA L ↑ - - ↑ (B29) NS Core ↑ ↑ ↓ ↓ ↑ ↑ ↓ ↑ (B30) (B24/25/26/27) Table 6 Step 23. Determine secondary parameters ISP, ISRMS, IRIPPLE, DIAS, ODS: • • • • • Secondary Peak current, ISP : in Amps Secondary RMS current, ISRMS : in Amps Output capacitor ripple current, IRIPPLE : in Amps Secondary minimum conductor diameter, DIAS : in mm Secondary maximum wire outside diameter, ODS : in mm (D61) (D62) (D64) (D68) (D69) Step 24. Determine maximum peak inverse voltages PIVS, PIVB for secondary and bias windings: • Secondary winding maximum peak inverse voltage PIVS: in Volts • Bias winding maximum peak inverse voltage PIVB: in Volts (D74) (D75) Step 25. Select clamp Zener and blocking diode for primary clamping per Table 7 based on input voltage and VCLO (from Step 4): Input (VAC) VCLO (V) Zener Diode 100/115 Universal 230 90 200 200 P6KE91 P6KE200 P6KE200 BYV26B BYV26C BYV26C Table 7 Notes: 1. P6KE91: 91V/5W; Motorola P6KE200: 200V/5W; Motorola BYV26B: 400V/1A, UFR; Philips BYV26C: 600V/1A, UFR; Philips 2. Ishizuka 180V Zener may be used for lower power TOP210, TOP200, TOP201 applications 8 A 6/96 AN-16 Step 26. Select output rectifier per Table 8 such that: • VR ≥ 1.25 x PIVS; where PIVS is from Step 24 and VR is the rated reverse voltage of the rectifier diode • ID ≥ 3 x IO; where ID is the diode rated DC current and IO = PO/VO Rectifier Diode VR (V) ID (A) Manufacturer Schottky 1N5819 1N5822 MBR745 MBR1045 MBR1645 UFR UF4002 MUR110 MUR120 UF4003 BYV27-200 UF5401 UF5402 MUR410 MUR420 MUR810 MUR820 BYW29-200 BYV32-200 40 40 45 45 45 100 100 200 200 200 100 200 100 200 100 200 200 200 Motorola Motorola Motorola Motorola Motorola GI Motorola Motorola GI Philips, GI GI GI Motorola Motorola Motorola Motorola Philips, GI Philips 1.0 3.0 7.5 10 16 1.0 1.0 1.0 1.0 2.0 3.0 3.0 4.0 4.0 8.0 8.0 8.0 20 Table 8 Step 27. Select output capacitor based on IRIPPLE (from Step 23): • • • • Capacitor ripple current specified @ 105 OC, 100KHz must be equal or larger than IRIPPLE, where I RIPPLE is from Step 23. Use low ESR, electrolytic capacitor. Output switching ripple voltage is ISP x ESR , where ISP is from Step 23. Use parallel capacitors to increase ripple current capacity for high current outputs. Examples: Output Output Capacitor 5V to 24V, 1A 330uF, 35V, low ESR, electrolytic United Chemicon LXF35VB331M10X20LL Nichicon UPL1V331MRH Panasonic ECA-1VFQ331L 5V to 24V, 2A 1000uF, 35V, low ESR, electrolytic UnitedChemicon LXF35VB102M12.5X30LL Nichicon UPL1V102MRH Panasonic ECA-1VFQ102L A 6/96 9 AN-16 Step 28. to Step 29. Add output LC post filter if and only if output switching ripple voltage is not within specification: • Inductor L: 2.2 to 4.7µH. Use ferrite bead for low current (≤ 1A) output and standard off the shelf choke for higher current output. Increase choke current rating or wire size if necessary to avoid significant DC voltage drop. • Capacitor, C: 120uF, 35V, low ESR electrolytic United Chemicon LXF35VB121M8X12LL Nichicon UPL1V121MRH Panasonic ECA-1VFQ121L Step 30. Select bias rectifier per Table 9 such that VR ≥ 1.25 x PIVB ; where PIVB is from Step 24 and V R is the rated reverse voltage of the rectifier diode. Rectifier V R (V) Manufacturer 1N4148 BAV21 UF4003 75 200 200 Motorola Philips GI Table 9 Step 31. Select bias capacitor: • Use 0.1uF, 50V, ceramic Step 32. Select Control pin capacitor and series resistor: • Control pin capacitor: use 47uF, 10V, low cost electrolytic (Do not use low ESR capacitor). • Series resistor: use 6.2 Ω, 1/4 Watt (Not needed if KRP = 1, e.g. discontinuous mode). Step 33. Select feedback circuit components according to applicable Reference Design: RD1, ST202A, ST204A. • Applicable reference design: identified in Step 2. • Detailed component information: refer to appropriate reference design board documentation. Step 34. Select input bridge rectifier such that: • VR ≥ 1.25 x (1.414 x VACMAX); where VACMAX is from Step 1. • IACRMS ≥ 2 x ID; where ID is the bridge rectifier rated RMS current and IACRMS is the input RMS current. Note: I ACRMS = PO η × VACMIN × PF ; where VACMIN is from step 1 and PF is the power factor of the power supply which is typically between 0.5 and 0.7. If no better reference data is available, use 0.5. Step 35. TOPSwitch flyback power supply design complete. 10 A 6/96 AN-16 In-depth Information Step 1. Determine system requirements: VACMAX, VACMIN, fL, fS, VO, PO, η, Z The step-by-step procedure uses predetermined parameter values such as VACMAX, VACMIN, VMIN, VOR and VCLO for most commonly encountered input voltage ranges: 85 to 132 VAC for 100/115 VAC, 195 to 265 VAC for 230 VAC and 85 to 265 VAC for universal input. A ±15% line voltage variation is assumed in all cases. Applications with a different input voltage range can be handled by following the information and methods provided in Step 3, 4 and 5 of this in-depth information section to derive appropriate values for CIN, VOR , VCLO and VMIN. Efficiency η is the ratio of output power to input power. Since efficiency can vary significantly with output voltage due to secondary diode loss, it is best to use a number that is representative of similar power supplies. Switching power supply efficiencies typically range from 75% for supplies delivering most of their power at low voltage outputs (5 or 3.3V) to 85% for those supplying most of their power through higher voltage outputs (12V and above). If this data is not available, 80% is a reasonable choice. For a power supply with an output power PO and an efficiency η, PO x (1-η)/η watts of power is lost somewhere in the system: part in the secondary circuits, and the balance in the primary circuits. It is important to know the loss distribution between primary and secondary because only the secondary losses represent power that must be processed by the transformer and considered in the transformer design. Note that the power dissipated at the primary clamp is considered as secondary loss because this power is processed by the transformer before being delivered to the clamp circuit. The ratio of the secondary loss to the total loss is defined as the loss allocation factor, Z, which should be set based on experience. A value of 0.5 should be used if no reference data is available. Step 2. Decide on a Feedback/sense circuit and bias voltage VB Four types of feedback/sense circuits are recommended. The primary feedback circuit, shown in Figure 3, is the least expensive but has lower absolute accuracy and regulation and is suitable only for low power and higher output voltage (VO >5V). Output accuracy can be improved for the primary feedback circuit by adding a 22V Zener and a capacitor as shown in Figure 4. The D2 1N5822 1 L1 3.3 µH +5V 8 + C2 330 µF 10 V VR1 C3 100 µF 10 V 5 D1 D3 1N4148 2 DC INPUT 4 C1 10 nF 400 V 3 T1 C TOP 210 S R1 15 Ω TRD1 U1 D RTN S C5 47 µF 10 V - PI-1850-050696 Figure 3. RD1 Reference Design Board. A 6/96 11 AN-16 D2 1N5822 1 L1 3.3 µH +5V 8 + C2 330 µF 10 V VR1 C3 100 µF 10 V 5 D1 D3 BAV21 2 DC INPUT RTN 4 C1 10nF 400 V T1 TRD1-1 R1 15 Ω 3 U1 D C TOP 210 S C5 47 µF 10 V S VR 2 1N5251D 22 V 1% C4 100 nF 50 V - PI-1851-050696 Figure 4. RD1 Reference Design Board (Enhanced). L1 3.3 µH D2 UG8BT 7.5 V R1 39 Ω C2 680 µF 25 V VR1 C3 120 µF 25 V U2 NEC2501 D1 BR1 400 V L2 22 mH C1 33 µF 400 V C6 0.1 µF F1 3.15 A L R2 68 Ω VR2 1N5234B 6.2 V RTN D3 1N4148 C5 47µF C7 1 nF Y1 T1 DRAIN SOURCE CONTROL U1 TOP202YAI C4 0.1 µF N J1 PI-1852-050696 Figure 5. ST202A Reference Design Board. 12 A 6/96 AN-16 D2 MUR610CT L1 3.3 µH 15 V C1 47 µF 400 V L2 33 mH R2 200 Ω 1/2 W C2 1000 µF 35 V VR1 RTN D1 D3 1N4148 BR1 400 V U2 NEC2501 C4 0.1 µF R1 510 Ω T1 L C5 R3 47 µF 6.2 Ω U3 TL431 DRAIN R5 10 kΩ SOURCE CONTROL N J1 R4 49.9 kΩ C9 0.1 µF C6 0.1 µF F1 3.15 A C3 120 µF 25 V C7 1 nF Y1 U1 TOP204YAI PI-1853-050696 Figure 6. ST204A Reference Design Board. opto-coupler feedback using an accurate reference/comparator (Figure 6) such as the TL431 for sensing provides high accuracy and regulation at a slightly added cost and is applicable to all power and voltage ranges. An intermediate solution is to use an opto-coupler with a Zener sense circuit (Figure 5). This technique is suitable for medium power levels (up to 30W) and is reasonably accurate, especially at output voltages higher than 5V. Step 3. Determine input capacitor CIN and minimum DC input voltage VMIN When the full wave rectified AC line is filtered with an input capacitance CIN (Figure 1), the resulting High Voltage DC bus (V+) has a ripple voltage as shown in Figure 7. The minimum DC voltage VMIN occurring at the lowest line voltage VACMIN is an important parameter for the design of the power supply. A rule of thumb on choosing the CIN value is to use 2 to 3 µF per watt of output power for 100/115 VAC or universal input, and 1 µF/Watt for 230VAC. This results in a VMIN of 90VDC for 100/115VAC or universal input and 240VDC for 230VAC, respectively. The CIN value obtained by using this rule represents a nearly optimum design in terms of system cost in most applications. Higher values of CIN increase capacitor cost without a corresponding pay back in terms of higher VMIN or lower ripple, whereas lower values of CIN result in significantly lower VMIN increasing TOPSwitch cost due to increased peak operating current demand. Lower values of CIN also increase input ripple voltage, which could increase output ripple voltage if the control loop gain is a limiting factor. The accurate calculation of VMIN for a given CIN (or vice versa) is a very complicated task which involves the solving of an equation with no closed form solution. The equation shown below represents a good first order approximation which is accurate enough for most situations. VMIN 1 2 × PO × − tC 2 × fL 2 = (2 × VACMIN )− η × CIN tC is typically 3 ms, and can be verified by direct measurement. Step 4. Determine reflected output voltage VOR and clamp Zener voltage VCLO: A 6/96 13 AN-16 VACMIN × 2 V+ VMIN tC PO = Output Power fL = Line Frequency (50 or 60Hz) 2 VMIN = (2 × VACMIN )−( 1 − tC ) 2 × fL ) η × CIN 2 × PO × ( tC = Conduction Angle Use 3ms if unknown η = Efficiency - Assume 0.8 if unknown PI-1854-050696 Figure 7. Input Voltage Waveform. A typical flyback circuit using TOPSwitch is shown in Figure 1. When the TOPSwitch is off and the secondary is conducting, the voltage on the secondary is reflected to the primary side of the transformer by the turns ratio. This reflected voltage VOR adds to the input DC voltage at the TOPSwitch drain node. Worst case voltage at the drain occurs at high line when the DC input voltage is at its maximum value. The maximum DC input voltage can be calculated as: VMAX = 2 × VACMAX In addition to VMAX+VOR the drain also sees a large voltage spike at turn off that is caused by the energy stored in the leakage inductance of the primary winding (see Figures 8 and 9). To keep this voltage spike from exceeding the rated minimum drain breakdown voltage BVDSS, a clamp circuit is needed across the primary winding. A Zener clamp as shown in Figure 1 is highly recommended over the usual RC clamp as it is much more effective in clamping the leakage energy during start up transients. The nominal value of Zener clamp voltage VCLO needs to be 50% (determined empirically) greater than the reflected voltage so that the Zener clamps only the leakage energy and does not impede the switch-over of current from the primary to the secondary. Experimental measurements show that this voltage margin is needed for the secondary current to be quickly established through the leakage inductance. Lower 14 A 6/96 clamp voltage should not be used, because part of the stored energy in the core would be delivered to the Zener, dramatically increasing Zener dissipation. The nominal clamp Zener voltage VCLO is usually specified at low current values and at room temperature. High voltage Zeners have a strong positive temperature coefficient and are quite resistive. Consequently, the clamp voltage at high current and high temperature VCLM can be much higher. Experimental data has shown that the VCLM can be as high as 40% above the specified VCLO VCLM = 1.4 × VCLO This needs to be taken into consideration when choosing a clamp Zener. In addition, it is important to allow an additional 20V for the spike due to the forward recovery time of the blocking diode in series with the clamp Zener. With all those factors considered, the maximum voltage that the TOPSwitch drain may experience is: VDRAIN = VMAX + (1.4 × 1.5 × VOR ) + 20V To minimize power supply cost, it is important to maximize the VOR consistent with the TOPSwitch breakdown voltage rating after taking into account all of the above effects. As will be seen AN-16 BVDSS MARGIN = 17 V BLOCKING DIODE FORWARD RECOVERY = 20 V 350 V 333 V 313 V 277 V VCLM VCLO VMAX 247 V VOR = 60 V 187 V VCLO = 1.5 x VOR = 90V VCLM = 1.4 x VCLO =126V D ≅ 24% 0V 0V For 100/115 VAC Input Using 350 V TOPSwitch Use VOR = 60 V and 90 V Zener Clamp PI-1855-050696 Figure 8. Reflected Voltage (VOR) and Clamp Zener Voltage (VCLO ) - 100/115 VAC Input. BVDSS MARGIN = 25 V BLOCKING DIODE FORWARD RECOVERY = 20 V 700 V 675 V 655V 575 V VCLM VCLO VMAX 510 V VOR = 135 V 375 V VCLO = 1.5 x VOR = 200V VCLM = 1.4 x VCLO = 280V 0V D ≅ 26% 0V For Universal/230 VAC Input Using 700 V TOPSwitch Use VOR = 135 V and 200 V Zener Clamp PI-1856-050696 Figure 9. Reflected Voltage (VOR) and Clamp Zener Voltage (VCLO ) - Universal/230 VAC Input. A 6/96 15 AN-16 later, a higher VOR will result in a larger DMAX which reduces TOPSwitch operating current for the same output power. If DMAX comes close to the maximum allowable duty cycle of the TOPSwitch (64%) then VOR should not be increased any further. For a 100/115 VAC power supply the VACMAX based on 115 VAC would be 132 VAC which corresponds to: Step 5. Determine maximum duty cycle at low line DMAX using VOR and VMIN Once the VOR and VMIN are known, it is easy to calculate the DMAX: DMAX = VMAX = 2 × 132 = 187V As can be seen in Figure 8, going through the above exercise for a VMAX of 187V using a 350V TOPSwitch results in a standard clamp Zener voltage of 90V and VOR of 60V and a margin of 17V. Likewise in 230VAC or Universal application, a VACMAX of 265VAC corresponds to a VMAX of 375V. At this value of VMAX, a 700V TOPSwitch will allow for a standard Zener value of 200V with corresponding VOR of 135V leaving a margin of 25V (see Figure 9). If these margins seem too small, it is important to remember that this analysis uses all worst case values added together and typical margins will be much greater. Also, TOPSwitch breakdown voltage increases at high temperature, providing additional margin. VOR VOR + (VMIN − VDS ) VDS is the average Drain to Source voltage during TOPSwitch ON time. As shown in Figures 10 and 11, with VDS set to zero, the value of DMAX ranges from 36%/40% for single input voltage applications to 60% for the universal input application. In reality, VDS should be set to approximately 10V which results in a slight increase in DMAX. Higher VMIN directly increases the output power capability of a given TOPSwitch, while lower VMAX allows larger VOR and consequently larger DMAX, also increasing the output power of a given TOPSwitch. Therefore, a narrower input voltage range always leads to either a higher output power or a lower power supply cost. BVDSS 350 V (700 V) 236 V (540 V) 216 V (520 V) BLOCKING DIODE FORWARD RECOVERY = 20 V 180 V (440 V) VCLM VCLO VMIN 0V DMAX ≅ 40% (36%) VOR = 60 V (135 V) 150 V (375 V) 90 V (240 V) VCLM = 1.4 x VCLO =126V (200 V) 0V PI-1857-050696 Figure 10. Determine DMAX - 100/115 VAC (230 VAC) Input 16 A 6/96 AN-16 BVDSS 700 V 390 V 370 V BLOCKING DIODE FORWARD RECOVERY = 20 V 290 V VCLM VCLO 225 V VOR = 135 V VMIN 90 V DMAX ≅ 60% 0V 0V PI-1858-050696 Figure 11. Determine DMAX - Universal Input Step 6. Set ripple current IR to peak current IP ratio KRP (see Figure 12) K RP = IR IP • Starting with KRP = 0.4 for 100/115 VAC or universal input 0.6 for 230 VAC for most continuous operation ` • KRP may be increased to higher values for less continuous operation • KRP , by definition, can not be larger than 1.0 and may not be set smaller than above values Many power supply design engineers prefer to use discontinuous mode (KRP =1) design as the control loop is easier to stabilize. With TOPSwitch, because of the built-in loop compensation, it is possible to use one simple external RC network to stabilize the loop independent of operating mode. Setting KRP to the values recommended above allows continuous mode operation at low input line voltage, minimizing the peak primary current for a given output power, and allowing the use of the smallest possible TOPSwitch for the application. A KRP of 0.6 is recommended for 230VAC (compared to 0.4 for 100/115 VAC and universal input) to accommodate a significantly taller and wider leading edge current spike caused by the discharge of the drain node capacitance at the higher voltage levels. Step 7. Determine primary waveform parameters IAVG, IP, IR and IRMS The average DC current IAVG at low line is simply the input power divided by VMIN, where the input power is equal to the output power divided by the efficiency. I AVG = PO η × VMIN With KRP and DMAX already determined, the shape of the current waveform is known. Due to the simple geometry of the waveform, the Primary peak current IP, ripple current IR and RMS current IRMS can be easily derived as a function of IAVG: A 6/96 17 AN-16 2 PIR = I RMS × RDS ( ON ) (100°C ) DRAIN CURRENT WAVEFORM SHAPES • Calculate TOPSwitch switching loss at low line: KRP = IP IR < 1.0 IP PCXT ≅ } IR where CXT is the external capacitance at the drain node. Continuous Mode (a) IP 1 × CXT × (VMAX + VOR )2 × fS 2 • Calculate junction temperature Tj of TOPSwitch as a function of total loss KRP = 1.0 } IR TJ = 25°C + ( PIR + PCXT ) × θ JA • If Tj > 100oC, choose bigger TOPSwitch. Discontinuous Mode (b) PI-1902-61096 • For non-critical applications, refer to AN-14 Table 2 for TOPSwitch recommendations with practical heatsinking. Figure 12. Primary Current Waveform. IP = I AVG 1 − K RP × D MAX 2 I R = IP × K RP K2 I RMS = IP × DMAX × RP − K RP + 1 3 Step 8. Select TOPSwitch based on TOPSwitch data sheet minimum ILIMIT specification and required IP (from Step 7) such that: 0.9 × minimum I LIMIT ≥ IP The minimum value of current limit ILIMIT in TOPSwitch data sheet is specified at room temperature. To accommodate the slight reduction of this parameter at high temperature, the room temperature limit should be derated by 10%. This can be accomplished by dividing the IP by 0.9 and comparing this value to the minimum ILIMIT in the datasheet. The smallest TOPSwitch that has an ILIMIT higher than this value should be selected as the first choice for the lowest cost. Step 9 to Step 10. Check thermal limitation - Use bigger TOPSwitch if necessary to reduce power loss • Calculate TOPSwitch conduction loss at low line: 18 A 6/96 TOPSwitch thermal environment can vary significantly from application to application. Fully enclosed lap top adapters with no ventilation pose significant limitations on the power that can be dissipated inside the box without exceeding acceptable surface temperatures on the outside of the box. Heat sinks in this application only help to distribute the heat across the surface of the box. The actual power capability at a given surface temperature is determined largely by the surface area of the box. In contrast, a PC power supply has a fan which provides forced air cooling. Here a larger heat sink could be the answer to higher power dissapation. It is therefore important to first estimate the losses in the TOPSwitch to see whether it is acceptable in a given application. The conduction losses (PIR) at low line tend to be the dominant loss factor and can be calculated using the IRMS and the RDS(ON) at 100°C from the output characteristic curve in the TOPSwitch data sheet. If the losses are unacceptable, a larger TOPSwitch with a lower RDS(ON) could be chosen to lower the power dissipation. Switching losses at low line due to internal drain capacitance are negligible and can be ignored. If significant external capacitance CXT is present, the switching losses (PCXT ) should also be estimated. Even though low line is usually the worst case for TOPSwitch losses, it is prudent to verify this by calculating the conduction and switching losses at high line, especially if there is significant external capacitance on the drain. Once the worst case loss in the TOPSwitch is known, the maximum die temperature at worst case ambient (internal ambient should be used for enclosed supplies) can be estimated using the thermal impedance from die to tab/heat sink of the AN-16 package, ØJC (specified in the TOPSwitch datasheet), and from heat sink to ambient, ØCA (usually specified in the heat sink data sheet). If a package without a heatsink tab is used, such as an 8 pin DIP, then a typical die to ambient thermal impedance, ØJA, for a board mounted part can be found in the data sheet for these calculations. It is recommended that the die temperature be kept below 100°C under all conditions. Step 11. Check minimum I LIMIT of the selected TOPSwitch against required IP. Increase KRP, if possible, for least continuous operation. Using continuous mode operation at low line decreases the peak current required for a given output power, allowing the use of a smaller TOPSwitch. However, if so desired, a trade-off between TOPSwitch and core size can be accomplished by increasing the KRP value. Larger KRP allows the use of a smaller core at the price of a larger TOPSwitch, as larger KRP implies less continuous operation and lower inductance LP, but higher peak current IP. This is very important when the best suited (smallest possible) TOPSwitch that can be chosen for a design still ends up with significant extra current capability. It is then best to trade this extra current capability for a reduced core size by using a higher KRP. In addition to affecting the size of the transformer core, KRP also influences supply efficiency. Larger KRP results in higher primary RMS current IRMS and higher TOPSwitch conduction loss while lower KRP results in lower IRMS and lower TOPSwitch loss. For applications with tight physical size/weight limitation and/or efficiency requirements, an intermediate KRP value can offer the optimum solution between cost and performance. Although this design method is designed to use the highest possible KRP once TOPSwitch is first chosen, the flexibility is certainly available for other design options. Experienced engineers should make their own judgment on KRP value based on the specific requirements of their application. Step 13. Chose core and bobbin as a function of PO based on AN-18, Appendix A, Table 2 and determine Ae, Le, AL and BW from core and bobbin catalog AN-18 Appendix A provides a table of recommended core types for various power ranges. Notice that there are two transformer construction types shown in the table. For single output designs, a triple insulated secondary simplifies transformer construction and allows the use of the smallest size core and bobbin for a given output power. Margin winding, which is suitable for both single and multiple output secondaries, will require wider bobbins and therefore, longer/taller cores. If there is no specific form factor requirement, it is best to start with the smallest EE type core for the power level. EE cores are usually the least expensive type. The two digit number following the core type indicates the core size in mm. For 100KHz operation, the selection of core material is not very critical. TDK PC40 material is a good first choice. Other ferrite materials with similar characteristics are available from many manufacturers. Lower frequency core materials such as Philips 3C85 and its equivalents will also work at 100 KHz, and could be used if there is a cost advantage. Once a core has been selected from the catalog, a suitable bobbin can be easily identified. Manufacturer specified core parameters Ae, L e, AL and bobbin parameter BW are usually found in the same catalog. Step 14. Set number of primary layers L and number of secondary turns NS (see Step 16 to 22) Step 15. Calculate number of primary turns NP and number of bias turns NB (see Step 16 to 22) Step 12. Determine primary inductance LP Because the energy transferred from primary to secondary each switching cycle is simply the difference between 1/2 x LP x IP2 and 1/2 x LP x (IP - IR)2. The primary inductance LP can be expressed as a function of IP, KRP , fS, PO, η and Z: LP = 10 6 × PO Z × (1 − η) + η × K η IP2 × K RP × 1 − RP × fS 2 η is the efficiency and Z is the loss allocation factor. If Z=1, all losses are on the secondary side. If Z = 0, all losses are on the primary side. Z is simply the ratio of secondary loss to total loss. If no better reference information is available, Z should be set to 0.5. Step 16 to Step 22. Check BM, CMA and Lg. Iterate if necessary by changing L, NS, core/bobbin until within specified range In addition to the selection of core and bobbin, a total of nine parameters must be specified in the construction of a transformer: primary inductance Lp, core gap length Lg, number of turns for primary NP, secondary NS and bias NB, wire outside diameter for primary OD and secondary ODS, bare conductor diameter for primary DIA and secondary DIAS. Because the bias winding carries very little current (typically less than 10 mA), the wire size of the bias winding is never a problem. Except for LP, the above parameters are all interdependent. A good starting point is to pick a number for the secondary turns. Using 1 turn/volt for 100/115 VAC and 0.6 turn/volt for 230 VAC A 6/96 19 AN-16 or universal inputs is a good assumption. As an example, for a 115VAC input and an output voltage VO of 15V plus the rectifier forward drop VD of 0.7V, a 16 turn secondary would be used as the initial value. The primary number of turns NP is related to the secondary number of turns N S by the ratio between V OR and V O + V D N P = NS × VOR VO + VD where VOR is the reflected output voltage, VO is the output voltage and VD is the output rectifier forward voltage drop. Similarly, the number of bias winding turns NB can be derived from N B = NS × VB + VDB VO + VD where VB is the bias voltage and VDB is the bias rectifier forward voltage drop. From the core/bobbin size, it is possible to determine the outside diameter of the primary wire OD in mm that is required to accommodate the primary turns in one or two full layers allowing for margins as appropriate. OD = BWE NP BWE is the effective bobbin width, which takes into account physical bobbin width BW, margins M (all in mm), and the number of winding layers L: BWE = L × [ BW − (2 × M )] The closest standard magnet wire gauge that is less than or equal to this diameter can be selected. Determine the bare conductor diameter DIA of this wire gauge using information from a wire table. The next step is to find out if this conductor size is sufficient for the maximum IRMS. The current capacity for magnet wire is specified in terms of “Circular mils per Amp” or CMA, which is the inverse of current density: CMA = 1.27 × DIA2 × I RMS π 2 4 × 1000 25.4 If the CMA is less than 200, a larger gauge wire is needed to handle the current. This could be accommodated by adding a 20 A 6/96 second layer if there is only one existing layer and/or by using a larger core/bobbin and/or a smaller NP. On the other hand, a CMA greater than 500 would indicate that a smaller core/ bobbin and/or a larger NP could be used. Note that in the AN-17 spreadsheet, DIA is actually derived from OD using an empirical equation. A practical wire size, AWG (American Wire Gauge), is determined according to DIA (see AN-18 Appendix A, Table 2 for wire size information). CMA is then calculated from AWG. Another critical parameter that must be checked is the maximum flux density in the core (BM ). BM = 100 × IP × LP N P × Ae Ae is the effective cross sectional area of the core. If BM is greater than 3000 Gauss, either the core cross sectional area (core size) or NP must be increased to bring it within the 2000 to 3000 range. On the other hand, if B M is less than 2000 Gauss, a smaller core or fewer turns on the primary can be used. In addition to BM , the core gap length Lg required to generate inductance LP with number of primary turns NP must also be checked: N P2 1 Lg = 40 × π × Ae × − 1000 × LP AL The core cross sectional area Ae and ungapped effective inductance AL can be found from the data sheets for the core. Lg is usually incorporated as an air gap ground into the center leg of the core and needs to be at least 51 µm or (2 mils) for manufacturability. If Lg is less than 51 µm, once again the core size or NP must be increased. One other parameter always required by transformer manufacturer is the gapped core effective inductance, ALG, which can be determined only after NP is fixed: ALG = 1000 × LP N P2 As can be seen, the transformer design is a highly iterative process in itself. When NP is changed, NS and NB will change according to ratios already established. Similarly, any change in core size requires a recalculation of CMA, BM and Lg to make sure that they are within the specified limits. AN-16 Step 23. Determine secondary parameters ISP, ISRMS, IRIPPLE, DIAS, ODS winding consisting of NS turns of two parallel strands of 26 AWG will be a good choice. The secondary peak current ISP can be derived from the primary peak current IP and the turns ratio between primary and secondary NP/NS Note that if triple insulated wire is to be used for secondary, the insulated wire diameter is actually larger than DIAS by twice the thickness of the insulator. Therefore, the maximum outside diameter ODS (in mm) must be calculated: ISP = IP × NP NS The KRP of the secondary is always identical to that of the primary, since it is only a reflected version of the primary current with duty cycle (1-D). Therefore, the secondary RMS current ISRMS can be expressed in a manner similar to the primary RMS current, only with DMAX replaced by (1-DMAX). K2 ISRMS = ISP × (1 − DMAX ) × RP − K RP + 1 3 ODS = A triple insulated wire should be specified with a conductor diameter equal to or greater than DIAS and an insulated outside diameter equal to or less than ODS. Step 24. Determine maximum peak inverse voltages PIVS, PIVB for secondary and bias windings. The peak inverse voltage across the secondary rectifier diode is given by: N PIVS = VO + VMAX × S NP IRIPPLE is the RMS ripple current of the output capacitor. Because of current conservation, it is found that: I RIPPLE = I 2SRMS − I O2 Similarly, the peak inverse voltage across the bias rectifier diode is given by: IO is the power supply output current which can be calculated, if not already specified, as IO = PO VO With the secondary RMS current ISRMS available, the minimum secondary wire diameter DIAS (in mm), can be calculated as follows: BW − (2 × M ) NS N PIVB = VB + VMAX × B NP Step 25. Select clamp Zener and blocking diode for primary clamping based on input voltage and VCLO (see Step 4) Step 26. Select output rectifier DIAS = 4 × CMA × ISRMS 25.4 × 1.27 × π 1000 Note that in the AN-17 spreadsheet, a practical wire size, AWGS, is derived from primary current capacity CMA and secondary RMS current ISRMS using an empirical equation. DIAS is then determined from AWGS. If the required secondary wire diameter turns out to be larger than that of the 26 AWG wire which corresponds to twice the skin depth at 100 KHz, a parallel configuration of windings using a gauge equal to or smaller than 26 AWG should be used to provide the same effective cross sectional area. The parallel windings must have identical number of turns equal to NS. For example, if the equation above indicates a 23 AWG wire, a The peak inverse voltage across the secondary diode PIVS is calculated in Step 24. The diodes should be chosen with a reverse voltage rating VR equal to or greater than 1.25 X PIVS to keep the PIVS at no more than 80% of the diode VR rating. The rule of thumb on the diode current rating is to choose one with rated DC current of at least three times the maximum output DC current. Schottky diodes are recommended for VR less than 45V which would correspond to low output voltages such as 5V or 3.3V. For VR requirements that are higher than 45V, ultra fast recovery PN diodes should be used for the lowest cost. (See Table 8 for recommended diodes.) A 6/96 21 AN-16 Step 27. Select Output capacitor ESR is the most important parameter for output filter capacitor selection. Capacitor ESR directly determines the output ripple voltage of the power supply and the ripple current rating of the capacitor while the actual capacitance value only affects control loop bandwidth. Below 35V, ESR is mainly determined by capacitor case size. Consider two Nichicon PL series capacitors: 1500µF/6.3V and 390µF/35V. Both capacitors have a case size of 10 mm diameter and 25 mm length, and both have the same ESR of 55 mΩ. To keep control loop bandwidth high, the smaller capacitance, higher voltage rating capacitor is preferred. Ripple current is typically specified at 105oC ambient which is much higher than the ambient temperature required in most applications. Therefore, it is possible to operate the capacitor at higher ripple currents determined by a multiplier factor from the capacitor data sheet. Actual ripple current of the output capacitor can be calculated as follows: 2 I RIPPLE = ISRMS − IO2 where ISRMS is the secondary winding RMS current and IO is the DC output current. Step 32. Select Control pin capacitor and series resistor A 47 µF, 10V low cost standard grade electrolytic capacitor across the Control pin and Source pin of the TOPSwitch takes care of loop compensation for all types of feedback configurations. Low ESR capacitors should not be used for this purpose, as the ESR resistance of the standard grade capacitor (2 Ω typical) improves the loop stability by introducing a zero. In fact, a 6.2Ω resistor in series with this capacitor is recommended to improve phase margin in designs that either have excessive gain in the secondary (such as the TL431 circuit shown in Figure 6), or a KRP value of less than one (continuous mode). Step 33. Select feedback circuit components • Primary feedback: Refer to RD1 • Opto/Zener feedback: Refer to ST202A • Opto/TL431: Refer to ST204A • Select opto-coupler with CTR between 50% and 200% (Refer to AN-14, Table 3) Step 34. Select bridge rectifier based on input voltage VACMAX and input RMS current IACRMS Maximum operating current for the input bridge rectifier occurs at low line: Step 28 to Step 29. Select Output post filter L, C If the measured switching ripple voltage at the output capacitor is higher than the required specification, an LC post filter consisting of a 2.2 to 4.7µH inductor or ferrite bead (only for power levels below 5W) with a 120 uF/35V, low ESR electrolytic capacitor is recommended. This will provide a lower cost solution compared to increasing the capacitance value and/or lowering the ESR of the main output filter capacitor. I ACRMS = PO η × VACMIN × PF PF is the power factor of the power supply. Typically, for a power supply with a capacitor input filter, PF is between 0.5 and 0.7. Use 0.5 if there is no better reference data available. Select the bridge rectifier such that: The output post filter, to a first order, is independent of output power except that the DC voltage drop across the inductor may be a concern at high currents. Inductors with larger gauge wire and higher current rating solve this problem. • ID ≥ 2 x IACRMS, where ID is the rated RMS current of the bridge rectifier • VR ≥ 1.25 x 1.414 x VACMAX; where VR is the rated reverse voltage of the rectifier diode Step 30. Select bias rectifier Step 35. Design complete Bias rectifier selection is similar to output rectifier selection with the exception that since the bias winding carries very little current (typically less than 10 mA), the considerations for current capability and very fast recovery no longer apply. Step 31. Select bias capacitor Because of the low voltage and the minimal power required at the bias output, a 0.1 uF, 50V ceramic capacitor always meets the requirement. 22 A 6/96 Following the step-by-step procedure completes the design of a basic TOPSwitch flyback converter. Once built, the power supply should be fully functional and capable of delivering maximum rated output power at minimum input line voltage, while meeting all specifications. Minor adjustments may be necessary to center the output voltage. AN-16 Issues Beyond the Design Method Issues outside the basic converter requirements are beyond the scope of this application note. However, design guidelines for various issues are available in the following documentation: • Constant current/power output • PC board layout • Transformer design • Transformer construction • Efficiency • EMI and safety • Transient : DN-14 : AN-14 : AN-17 : AN-18 : AN-19 : AN-15 : AN-20 Application specific requirements such as constant current and/or constant power outputs (DN-14), input under voltage protection, soft start etc. (refer to AN-14) are usually implemented by adding minimal circuitry to the basic converter. General design guidelines for EMI, safety and input transient are provided in AN-15 and AN-20 respectively. However, the optimum solution for any particular design can only be found through experimentation. Transformer construction techniques are very critical in the successful development of a TOPSwitch flyback. AN-18 provides practical guidelines that should be followed carefully to minimize parasitics such as leakage inductance, inter-winding capacitance etc. A 6/96 23 AN-16 Appendix A TOPSwitch Flyback Fundamentals This appendix explains the operation of a flyback power supply using the TOPSwitch power integrated circuit. TOPSwitch is a monolithic device combining a high voltage power MOSFET switch with all the analog and digital control circuitry required to implement isolated, regulated, and protected switching power supplies. Designing the power supply is greatly simplified because few external components are required. The high switching frequency of 100 KHz reduces the size of the power supply by allowing the use of smaller energy storage components. The TOPSwitch was designed for use in isolated power supplies or DC to DC converters. Power levels up to 50 Watts can be delivered from AC voltages of 85 to 265 VAC, or 100W with a 195 to 265 VAC input range. Operation from lower input voltages is also possible with reduced levels of output power. The flyback power supply is described in detail. Ideal and nonideal circuit operation is explained. The difference between the discontinuous and continuous mode of operation is discussed. The benefits of high frequency operation are presented. Other types of power supplies using both linear and switching techniques are examined and compared with the flyback topology. The Flyback Power Supply The flyback topology, shown in Figure 1, is recommended for off-line, isolated, power supply applications. The flyback supply has a low parts count, wide input voltage range, inherent feedback voltage sensing, single or multiple output voltage capability, output voltages that can be higher or lower than the input voltage, and ability to provide both positive and negative voltages. Almost all off-line switching power supplies require isolation D2 NS VIN C1 Another important advantage of the flyback topology is that a feedback voltage proportional to the output voltage can be obtained directly by adding a “feedback” winding to the power transformer. This means that secondary side regulation can be accomplished on the primary side of the power supply without using an optocoupler or similar isolation device between the primary and secondary circuitry. Single or multiple, higher or lower, positive or negative output voltages are primarily a function of the construction of the power transformer. Comparison to Other Techniques Alternatives to flyback power supplies for low power applications include linear supplies and other switching topologies such as the buck converter and the forward converter. These are briefly examined below. Additional information can be found in some of the references listed at the end of this appendix. Linear Power Supplies VO T1 between primary and secondary components to satisfy the requirements of domestic and international safety regulations. This isolation, along with any necessary voltage transformation, requires a power transformer. Most switching power supplies also need an inductor as the energy storage component and also as part of the low pass filter required to transform the pulse width modulated switching waveform into a DC output. The flyback topology is attractive for low power isolated switching power supplies because the transformer is combined with the inductor in a single magnetic component providing energy storage, isolation, and voltage transformation. As compared to other topologies such as the forward converter, the flyback has the fewest magnetic components and the lowest parts count, resulting in the lowest cost. The flyback topology retains these advantages at power levels up to 100 watts, or output currents up to 10 amperes. Component stress levels above 100 watts or 10 amperes output current require the use of more expensive components, allowing other topologies to become more cost effective. RL NP The linear power supply is characterized by the use of an AC line frequency (50-60 Hz) transformer, rectifier, filter, and linear regulator as shown in Figure 2. This type of power supply is inexpensive and reliable but suffers from the following disadvantages: TOPSwitch • • • • DRAIN SOURCE CONTROL FEEDBACK PI-1615-021296 Figure 1. Basic Flyback Converter Circuit. 24 A 6/96 Largest size Highest weight Poorest efficiency Narrow input voltage range AN-16 Switching Power Supplies 60 Hz TRANSFORMER LINEAR REGULATOR AC IN VO PI-1735-021296 There are many different switching power supply topologies available. The buck, boost, and forward converters are described below. Multiswitch and resonant converters are also briefly discussed. Buck Converter - The buck converter, shown in Figure 3, is useful for stepping down from a higher voltage to a lower voltage. The key points are: Figure 2. Linear Regulator Circuit. VIN • Not isolated • High side switch requires level shift or bootstrap circuit to drive • Limited to approximate 10:1 conversion range by duty cycle requirements • Provides only down converted, positive output voltages VO RL CONTROL FEEDBACK PI-1788-021296 Figure 3. Buck Converter Circuit. VIN VO RL CONTROL Boost Converter - The boost converter, shown in Figure 4, is useful for stepping up from a lower voltage to a higher voltage. The key points are: • Not isolated • Limited to approximate 10:1 conversion range by duty cycle requirements • Provides only up converted, positive output voltages Forward Converter - The forward converter, shown in Figure 5, is an isolated version of the Buck. Single or multiple, positive or negative, higher or lower output voltages are available by transformer design. This topology can be useful for output power of 100 W to 300 W. The key points are: • Inductor required for each output voltage • Extra diode required for each output voltage • Additional isolated feedback circuit required FEEDBACK PI-1789-021296 Figure 4. Boost Converter Circuit. VIN RL CONTROL FEEDBACK WITH ISOLATION PI-1720-120595 Figure 5. Forward Converter Circuit. Multiple Switch Converters - Multiple switch converter topologies include the push-pull, half bridge, full bridge, two transistor flyback, and two transistor forward converters. All these circuits require at least one additional power switch and are much more complex and costly. They are used to implement power supplies ranging from 200 watts to several kilowatts and are inappropriate for low power, low cost designs. Resonant and Quasi-Resonant Converters - Resonant converters are switching power supplies that use resonant tank circuits to process power with sinusoidal waveforms rather than the pulse width modulated quasi-square waves employed by conventional switching power supplies. Quasi resonant power supplies are switching power supplies that use resonant circuits to smooth the turn on and turn off edges in the switching waveform. In general, resonant and quasi-resonant converters are used at frequencies considerably higher than 100 KHz, and require more components than the traditional quasi-square wave A 6/96 25 AN-16 switching power supply. Peak voltage or current stress levels are higher than quasi square wave power converters, depending on whether a zero-voltage switching or a zero-current switching topology is used. The most effective resonant converters use both high side and low side switches, adding to circuit complexity. Resonant converters are not cost effective at low output power levels. into the load circuit, supplying current to the load RL and replenishing the charge depleted from C1 during the on time. The initial value of the secondary current at the instant the TOPSwitch turns off will be equal to IP x NS/NP, where IP is the peak value of IPRI at the end of TOPSwitch on time and NP is the number of primary turns and NS is the number of secondary turns. The secondary current decays from its initial value according to Equation (2). Flyback Theory Basic Flyback Operation ISEC A basic flyback power supply circuit utilizing TOPSwitch is shown in Figure 1. Transformer T1 is used both for energy storage, output isolation, and output voltage transformation. When the TOPSwitch is on, secondary diode D2 is reverse biased, and current ramps up in the transformer primary winding according to the equation IPRI (V − VDS ( ON ) ) × tON = I I + IN LP IP × N P (VO + VD 2 ) × tOFF × N P2 = − NS NS2 × LP (1) IPRI is the primary current in amperes, II is the initial value of the primary current in amperes, VIN is the DC input voltage after the bridge, VDS(ON) is the drain to source voltage drop across the TOPSwitch output MOSFET, tON is the on time of the TOPSwitch, and LP is the transformer primary inductance in Henries. Since the transformer is isolated from the output load circuit by the reverse biased D2, energy is supplied to RL from the output capacitor C1 during the TOPSwitch on time. When the TOPSwitch turns off, the magnetic flux in the transformer core starts to decay, and the polarity of the secondary winding is reversed. D2 turns on, and the energy stored in the transformer during the on time of the TOPSwitch is discharged ( ISEC ≥ 0) (2) VO is the output voltage of the supply, VD2 is the forward voltage drop of D2, and tOFF is the TOPSwitch off time. If the secondary current decays to zero during the off time of the primary switch, the output current is then supplied by the output capacitor C1. There are two distinct modes of flyback supply operation, depending on the value of ISEC at the end of the TOPSwitch off time. If ISEC decays to zero at or before the end of the TOPSwitch off time, the supply is running in the discontinuous mode. If ISEC is greater than zero at the end of the off time, the supply is running in the continuous mode of operation. Ideal Model (Discontinuous Mode) There are three distinct intervals of circuit operation for flyback power supplies operating in the discontinuous mode as shown in Figure 6. The first interval (1) of operation occurs when the TOPSwitch is on. Current IPRI ramps up linearly in the transformer primary winding, causing a magnetic field to build in the transformer VIN +VOR VOR D2 + VIN VDRAIN IPRI + ISEC VIN Interval 1 2 3 0 C1 VO RL - IP IPRI - + DRAIN SOURCE CONTROL VDS (ON) ISEC PI-1616-021496 Figure 6. Ideal Flyback Converter Waveforms - Discontinuous Mode. 26 A 6/96 AN-16 core. The drain to source voltage VDS(ON) across TOPSwitch is nearly zero during this interval. The output diode prevents current flow in the secondary due to the transformer dot polarity. Since the transformer secondary is isolated from the output by the reverse biased diode D2, current is supplied to the output from C1. flows in the primary or secondary of the transformer (which defines the discontinuous mode of operation). Note that the drain to source voltage across the TOPSwitch has decayed to the level of the input voltage. Since the stored energy of the transformer has decayed to zero, the output load current is again supplied by output capacitor C1. The second interval (2) of operation starts when TOPSwitch turns off. The energy stored in the magnetic field of the transformer causes the voltage across both the primary and secondary windings to reverse polarity. In an ideal circuit the primary current IPRI instantly stops flowing while the secondary current ISEC instantly starts flowing (it will be shown later how important it is to consider non-ideal behavior). The voltage across the secondary winding is equal to the sum of the output voltage and diode forward voltage. The secondary voltage is “reflected” back through the transformer turns ratio to the primary winding. Note that the drain to source voltage across the TOPSwitch during this interval of operation is equal to the sum of the reflected output voltage VOR and the input voltage VIN . This reflected voltage must be taken into account when selecting the transformer turns ratio to avoid excessive voltage stress on TOPSwitch. The reflected voltage can also be used to indirectly sense the output voltage of the supply from the primary side of the transformer through a bias or control winding referenced to the primary return, making primary side control of the supply possible. The energy delivered to the load each cycle by the transformer is given by The energy stored in the primary inductance of the transformer during the first interval of operation supplies current to the load circuit during the second interval of operation and replenishes the charge depleted from output capacitor C1 during the first and third intervals. The third interval (3) of operation occurs when the magnetic field within the core has decayed to zero (ISEC = 0). No current VIN +VOR VIN VDRAIN Interval 1 2 0 IPRI ISEC PI-1736-021496 Figure 7. Ideal Flyback Converter Waveforms - Continuous Mode. E= 1 × LP × IP2 × η 2 thus the output power is defined by PO = 1 × LP × IP2 × η × fS 2 where fS is the operating frequency of the power supply, and η is the efficiency. Substituting the expression of Equation (1) for IP (with II = 0 and VDS(ON) = 0), and defining t ON as D/fS, where D is the duty cycle, and fS is the TOPSwitch operating frequency. One obtains the expression PO = VIN2 × D2 × η 2 × LP × fS (3) In a power supply operating in the discontinuous mode, the controller will adjust the duty cycle of the primary switch to deliver enough power to the load to maintain the desired output voltage. The duty cycle is a function of both the input voltage and the output load. Ideal Model (Continuous Mode) Refer to Figure 7 for the characteristic waveforms for the continuous mode of operation. The reference circuit is the same as in Figure 6. The secondary current ISEC does not decay completely to zero as in the discontinuous mode, so that the third interval of operation (3) does not exist. The primary current IPRI starts with a current step equal to the final value of the secondary current ISEC reflected back through the transformer turns ratio. The drain to source voltage across TOPSwitch at the instant of turn on is also different since the third interval has been eliminated as previously discussed. The reflected output voltage state persists for the balance of the off cycle until TOPSwitch turns on again. In order to maintain a constant output voltage, the amount of current ramped up in the primary inductance during the on time must be balanced by the current ramped down during the off A 6/96 27 AN-16 Leakage Spike Voltage VP VIN +VOR LKP LKS VIN VDRAIN IPRI 2 1 Interval 0 3 ISEC VIN + D2 + C1 VO RL - IPRI + Slope = di/dt DRAIN SOURCE CONTROL ISEC VDS - Crossover Interval CDRAIN = COSS+CXT PI-1617-021496 Figure 8. Non-ideal Flyback Converter Waveforms - Discontinuous Mode. time. This means that continuous and discontinuous operation. This equation is derived by assuming that the integral of the output current of the power supply over the entire switching cycle is exactly equal to the integral of the transformer secondary output current over the off time period. This means that during the off time, the transformer delivers exactly enough energy to balance the energy delivered to the load over the entire switching cycle, with no energy left over, and runs out exactly at the end of the off time. (VIN − VDS ( ON ) ) × D (VO + VD 2 ) × (1 − D) (4) = NS LP × fS × LP × fS NP Solving for VO, one obtains the expression D N VO = (VIN − VDS ( ON ) ) × × S − VD 2 (5) 1 − D NP As long as the power supply is running in the continuous mode, it can be seen from the above expression that there is no direct dependence of the output voltage on the output loading. To a first order, the duty cycle of the supply will remain constant as the load is changed, and the initial value of the primary current waveform will change instead. The primary inductance of the power transformer, output loading, and the TOPSwitch off time determine continuous or discontinuous operation. This dependence is shown in Equation (2). The boundary of continuous versus discontinuous operation is defined by the equation IOB = V × VO 2 IN N 2 × fS × LP × S × VIN + VO N P 2 (6) Where IOB is the output current at the boundary between 28 A 6/96 If the output current is greater than the right hand side of Equation (6) , the supply is operating in the continuous mode. If the output current is less than or equal to the right hand side of the equation, the supply is operating in the discontinuous mode. A smaller transformer primary inductance will give up the energy stored in the magnetic field at a faster rate and result in discontinuous conduction mode. Conversely, a larger primary inductance will not give up all the energy stored in the core each cycle and operate in continuous mode. If the load current is reduced below IOB, the supply will run in the discontinuous mode. Also, if the input voltage is increased for a given load, the supply can transition to the discontinuous mode, as IOB increases with increasing input voltage. Non-ideal Model (Discontinuous and Continuous Mode) The circuit for the non-ideal flyback power supply and the associated waveforms for the discontinuous and continuous operating modes are shown in Figures 8 and 9. The non-ideal flyback has three additional parasitic circuit elements: two inductors and one capacitor. The inductor LKP is the leakage inductance of the primary winding on the power transformer. The inductor LKS is the leakage inductance of the secondary AN-16 winding on the power transformer. The capacitor CDRAIN is the sum of COSS and CXT which are the TOPSwitch output capacitance and the transformer winding capacitance, respectively. These parasitic circuit elements are present in any real-life flyback power supply circuit, and greatly affect supply performance. As previously shown, the discontinuous mode circuit has three intervals of operation per switching cycle (see Figure 8). The impact on circuit operation of the parasitic circuit elements in each of three intervals of operation is discussed below. In the first interval (1) the TOPSwitch turns on, discharging COSS and CXT. The energy stored by these capacitances at the end of the previous cycle is dissipated in the TOPSwitch at the beginning of the turn on interval. This dissipated energy is proportional to the square of the voltage on the parasitic capacitances. Because of this effect, large values of parasitic capacitance can dramatically lower the power supply efficiency, especially at high input voltage. Leakage inductance has little effect during the turn on interval, since the transformer has no stored energy, and the initial value of the secondary output current is zero. In interval (2) of operation, the TOPSwitch turns off. The energy stored in the transformer magnetic field during the previous interval is now transferred to the secondary circuit. A problem that arises during this transfer is that leakage inductances LKP and LKS are both trying to oppose changes in current flow. LKP is trying to maintain primary current flow, and LKS is trying to block secondary current flow. There is a “crossover region” during which the primary current ramps down and the secondary current ramps up. The primary current ramps down to zero with a slope determined by the value of leakage inductance and circuit voltage levels. The secondary current ramps up to the final value with a slope determined by the value of leakage inductance and circuit voltage levels. The big problem is that the primary current must continue to flow during this crossover Leakage Spike Voltage VP VIN +VOR V IN VDRAIN 0 Interval IPRI 2 1 Slope = di/dt ISEC Crossover Interval D1 Reverse Recovery Current Spike PI-1618-021496 Figure 9. Non-ideal Flyback Converter Waveforms - Continuous Mode. interval. The decaying primary current ends up flowing into COSS and CXT which charge up to a peak voltage VP. This peak voltage, caused by leakage inductance, will be referred to as the “leakage spike”. In a practical TOPSwitch flyback supply, the leakage spike should be clamped to a value below the TOPSwitch breakdown voltage rating. During interval (3) of operation, the reflected output voltage goes to zero. The transformer magnetic field has given up all the energy stored during the first interval. The TOPSwitch drain to source voltage makes a transition from the level equal to the sum of the reflected output voltage VOR and input voltage VIN down to a level equal to the input voltage VIN alone. This transition excites the resonant tank circuit formed by the stray capacitance and the primary inductance to create a decaying oscillatory waveform, which persists until the TOPSwitch turns on again. This waveform “modulates” the voltage on (and the amount of energy stored in) COSS and CXT, determining the power loss when TOPSwitch turns on at the beginning of the next cycle. In the continuous mode of operation, the same parasitic elements are present as in the discontinuous mode. In addition, the nonideal aspects of the output rectifier characteristic become important. An ideal rectifier has no forward voltage drop, and switches infinitely fast. An actual diode has a finite forward voltage drop, and takes a finite time to switch off. A PN junction diode has a finite reverse recovery time ( trr) due to the fact that the minority charge carriers must be swept from the junction by the applied reverse voltage before the diode junction can reverse bias and switch to the off state. In the case of a Schottky diode, this finite recovery time is caused by junction capacitance. This recovery time ( trr) is associated with a reverse recovery current spike that persists until the diode switches off. This current spike causes reverse power dissipation in the output rectifier, and loads down the TOPSwitch during its turn on transition. The amplitude and duration of this current spike is dependent on the speed of the diode. For 100 KHz power supplies, ultrafast diodes (trr < 50 nsec) are recommended. Use of slower diodes will cause a loss in efficiency due to excessive reverse recovery power dissipation, and can result in thermal runaway of the output rectifier diode. Non-ideal operating waveforms of a continuous mode flyback converter are shown in Figure 9. During the interval (1) of operation, TOPSwitch turns on while current is still flowing in the transformer secondary. This means that the drain voltage at the instant of turn on is equal to the sum of the input voltage and the secondary voltage reflected back through the transformer turns ratio. This results in higher TOPSwitch turn-on power dissipation than in the discontinuous mode, due to the extra energy stored in the parasitic capacitances of the primary circuit. In addition, the current in the secondary leakage inductance must be discharged before the secondary output can be turned off. This results in a turn on current crossover while A 6/96 29 AN-16 the secondary current ramps down and the primary current ramps up. Once the secondary leakage inductance is discharged, the output rectifier D2 is reverse biased, and the charge carriers in the diode junction are withdrawn, resulting in a reverse recovery current spike that is reflected to the primary and appears at the leading edge of the primary current waveform. Depending on the diode characteristics, this initial current spike can be comparable in amplitude or higher than the final value of the primary current. This can result in spurious operation of a current limit protection circuit. The TOPSwitch provides built-in leading edge current limit blanking to prevent the initial current spike from spuriously triggering the current limit protection circuitry. References When TOPSwitch turns off, operation in the continuous mode is similar to that of the discontinuous mode. The primary and secondary current experience a crossover region due to the effects of the transformer leakage inductance. This gives rise to a primary leakage spike, as in the discontinuous operating mode. The TOPSwitch drain to source voltage rises to the sum of the input supply voltage and the output voltage reflected back through the transformer turns ratio. Unlike the discontinuous mode model, this reflected voltage persists until TOPSwitch turns on again, so that there is no interval (3) where the reflected secondary voltage decays to zero. 5. Col. William McLyman, Transformer and Inductor Design Handbook, New York, Marcel Dekker, Inc., 1978 1. Power Integrations, Power Integrated Circuit Data Book 2. Ralph E. Tarter, Solid State Power Conversion Handbook, New York, John Wiley & Sons, Inc., 1993 3. Abraham I. Pressman, Switching Power Supply Design (2nd ed.), New York, McGraw-Hill, Inc., 1991 4. Application Information 472, C. van Velthooven, Properties of DC-to-DC converters for switched-mode power supplies, Philips Components, 1975 (Ordering Code 9399 324 47201) 6. Col. William McLyman, Magnetic Core Selection for Transformers and Inductors, New York, Marcel Dekker, Inc., 1982 7. Philips Components, Ferroxcube Magnetic Design Manual, Bulletin 550, 1971 8. Ferdinand C. Geerlings, “SMPS Power Inductor and Transformer Design, Part 1”, Powerconversion International, November/December 1979, pp. 45-52 9. Ferdinand C. Geerlings, “SMPS Power Inductor Design and Transformer Design, Part 2”, Powerconversion International, January/February 1980, pp. 33-40 10. Philips Semiconductors, Power Semiconductor Applications, 1991, (Ordering Code 9398 651 40011) 11. Technical Information 042, Using very fast recovery diodes on SMPS, Philips Components, 1978 (Ordering Code 9399 450 34201) 12. Brian Huffman, “Build Reliable Power Supplies by Limiting Capacitor Dissipation”, EDN, March 31, 1993, pp. 93-98 13. Jon Schleisner, “Selecting the Optimum Voltage Transient Suppressor”, General Instrument Data Book, 11th Edition, pp. 629-634 30 A 6/96 AN-16 A 6/96 31 AN-16 Power Integrations reserves the right to make changes to its products at any time to improve reliability or manufacturability. Power Integrations does not assume any liability arising from the use of any device or circuit described herein, nor does it convey any license under its patent rights or the rights of others. PI Logo and TOPSwitch are registered trademarks of Power Integrations, Inc. ©Copyright 1994, Power Integrations, Inc. 477 N. Mathilda Avenue, Sunnyvale, CA 94086 WORLD HEADQUARTERS Power Integrations, Inc. 477 N. Mathilda Avenue Sunnyvale, CA 94086 USA Main: 408•523•9200 Customer Service: Phone: 408•523•9265 Fax: 408•523•9365 AMERICAS For Your Nearest Sales/Rep Office Please Contact Customer Service Phone: 408•523•9265 Fax: 408•523•9365 JAPAN Power Integrations, K.K. Keihin-Tatemono 1st Bldg. 12-20 Shin-Yokohama 2-Chome, Kohoku-ku, Yokohama-shi, Kanagawa 222 Japan Phone: 81•(0)•45•471•1021 Fax: 81•(0)•45•471•3717 ASIA & OCEANIA For Your Nearest Sales/Rep Office Please Contact Customer Service Phone: 408•523•9265 Fax: 408•523•9365 32 A 6/96 EUROPE & AFRICA Power Integrations (Europe) Ltd. Mountbatten House Fairacres Windsor SL4 4LE United Kingdom Phone: 44•(0)•1753•622•208 Fax: 44•(0)•1753•622•209 APPLICATIONS HOTLINE World Wide 408•523•9260 APPLICATIONS FAX Americas 408•523•9361 Europe/Africa 44•(0)•1753•622•209 Japan 81•(0)•45•471•3717 Asia/Oceania 408•523•9364

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