ETC AN-37

®
LinkSwitch-TN
Design Guide
Application Note AN-37
Introduction
• Universal input – the same power supply/product can be
used worldwide
• High power density – smaller size, no µF’s of X class
capacitance needed
• High efficiency – Full load efficiencies >75% typical for
12 V output
• Excellent line and load regulation
• High efficiency at light load – ON/OFF control maintains
high efficiency even at light load
• Extremely energy efficient – input power <100 mW at no
load
• Entirely manufacturable in SMD
• More robust to drop test mechanical shock
• Fully fault protected (overload, short circuit and thermal
faults)
• Scalable – LinkSwitch-TN family allows the same basic
design to be used from <50 mA to 360 mA
LinkSwitch-TN combines a high voltage power MOSFET switch
with an ON/OFF controller in one device. It is completely selfpowered from the DRAIN pin, has a jittered switching frequency
for low EMI and is fully fault protected. Auto-restart limits
device and circuit dissipation during overload and output short
circuit while over temperature protection disables the internal
MOSFET during thermal faults. The high thermal shutdown
threshold is ideal for applications where the ambient temperature
is high while the large hysteresis protects the PCB and
surrounding components from high average temperatures.
LinkSwitch-TN is designed for any application where a nonisolated supply is required such as appliances (coffee machines,
rice cookers, dishwashers, microwave ovens etc.), nightlights,
emergency exit signs and LED drivers. LinkSwitch-TN can be
configured in all common topologies to give a line or neutral
referenced output and an inverted or non-inverted output voltage
- ideal for applications using triacs for AC load control. Using
a switching power supply rather than a passive dropper
(capacitive or resistive) gives a number of advantages, some of
which are listed below.
Scope
This application note is for engineers designing a non-isolated
power supply using the LinkSwitch-TN family of devices. This
DFB
RFB
CBP
DIN2
LIN
FB
RBIAS
S
D
L
LinkSwitch-TN
AC
Input
DIN2
CIN1
CFB
BP
DFW
CIN2
+
RF
CO
RPL
VO
PI-3764-121003
1 (a)
DFB
RFB
CBP
RBIAS
DIN1
LIN
FB
S
D
AC
Input
LinkSwitch-TN
DIN2
CIN1
CFB
BP
CIN2
1 (b)
DFW
L
CO
RPL
VO
+
RF
PI-3765-121003
Figure 1 (a). Basic Configuration using LinkSwitch-TN in a Buck Converter. Figure 1 (b) Basic Configuration using LinkSwitch-TN in a
Buck-Boost Converter.
January 2004
AN-37
Quick Start
document describes the design procedure for buck and buckboost converters using the LinkSwitch-TN family of integrated
off-line switchers. The objective of this document is to provide
power supply engineers with guidelines in order to enable them
to quickly build efficient and low cost buck or buck-boost
converter based power supplies using low cost off-the-shelf
inductors. Complete design equations are provided for the
selection of the converter’s key components. Since the power
MOSFET and controller are integrated into a single IC the
design process is greatly simplified, the circuit configuration
has few parts and no transformer is required. Therefore a quick
start section is provided that allows off-the-shelf components to
be selected for common output voltages and currents.
Readers wanting to start immediately can use the following
information to quickly select the components for a new design,
using Figure 1 and Tables 1 and 2 as references.
1) For AC input designs select the input stage (Table 9).
2) Select the topology (Tables 1 and 2).
- If better than ±10% output regulation is required,
then use opto coupler feedback with suitable reference.
3) Select the LinkSwitch-TN device, L, RFB or VZ, RBIAS, CFB,
RZ and the reverse recovery time for DFW
(Table 3: Buck, table 4:Buck-Boost).
4) Select freewheeling diode to meet trr determined in step 3
(Table 5).
5) For direct feedback designs, if the minimum load < 3 mA
then calculate RPL = VO / 3 mA.
6) Select CO as 100 µF, 1.25 • VO, low ESR type.
7) Construct prototype and verify design.
In addition to this application note a design spreadsheet is
available within the PIXls tool in the PI Expert design software
suite. The reader may also find the LinkSwitch-TN DAK
engineering prototype board useful as an example of a working
supply. Further details of support tools and updates to this
document can be found at www.powerint.com.
TOPOLOGY
High-Side
Buck –
Direct
Feedback
BASIC CIRCUIT SCHEMATIC
FB
+
BP
S
D
LinkSwitch-TN
VIN
KEY FEATURES
+
1)
2)
3)
4)
Output referenced to input
Positive output (VO) with respect to -VIN
Step down – VO < VIN
Low cost direct feedback (±10% typ.)
VO
PI-3751-121003
High-Side
Buck Boost –
Direct
Feedback
FB
+
VIN
D
1)
2)
3)
4)
5)
BP
S
LinkSwitch-TN
VO
+
PI-3794-121503
Output referenced to input
Negative output (VO) with respect to -VIN
Step down – VO > VIN or VO < VIN
Low cost direct feedback (± 10% typ.)
Fail-safe – output is not subjected to input
voltage if the internal MOSFET fails
6) Ideal for driving LEDs – better accuracy and
temperature stability than low-side Buck
constant current LED driver
Notes
1. Low cost, directly sensed feedback typically achieves overall regulation tolerance of ± 10%.
2. To ensure output regulation a pre-load may be required to maintain a minimum load current of 3 mA (Buck and Buck-Boost only).
3. Boost topology (step up) also possible but not shown.
Table 1. LinkSwitch-TN Circuit Configurations Using Directly Sensed Feedback.
2
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AN-37
TOPOLOGY
High-Side
Buck –
Optocoupler
Feedback
BASIC CIRCUIT SCHEMATIC
BP
FB
+
KEY FEATURES
D
S
+
RZ
LinkSwitch-TN
VO
VIN
VZ
PI-3796-121903
Low-Side
Buck –
Optocoupler
Feedback
+
+
RZ
LinkSwitch-TN
VO
VIN
1)
2)
3)
4)
Output referenced to input
Positive output (VO) with respect to -VIN
Step down – VO < VIN
Optocoupler feedback
- Accuracy only limited by reference choice
- Low cost non-safety rated optocoupler
- No pre-load required
5) Minimum no-load consumption
1)
2)
3)
4)
VZ
BP
S
Low-Side
Buck Boost –
Optocoupler
Feedback
FB
D
PI-3797-121903
+
VZ
LinkSwitch-TN
VO
VIN
RZ
BP
S
FB
D
+
PI-3798-121903
Output referenced to input
Negative output (VO) with respect to +VIN
Step down – VO < VIN
Optocoupler feedback
- Accuracy only limited by reference choice
- Low cost non-safety rated optocoupler
- No pre-load required
1)
2)
3)
4)
Output referenced to input
Positive output (VO) with respect to +VIN
Step up/down – VO > VIN or VO < VIN
Optocoupler feedback
- Accuracy only limited by reference choice
- Low cost non-safety rated optocoupler
- No pre-load required
5) Fail-safe – output is not subjected to input
voltage if the internal MOSFET fails
6) Minimum no-load consumption
Notes
1. Performance of opto feedback only limited by accuracy of reference (Zener or IC).
2. Optocoupler does not need to be safety approved.
3. Reference bias current provides minimum load. The value of RZ is determined by Zener test current or reference IC bias current.
Typically 470 Ω to 2 kΩ, 1/8 W, 5%
4. Boost topology (step-up) is also possible but not shown.
5. Optocoupler feedback provides lowest no-load consumption.
Table 2. LinkSwitch-TN Circuit Configurations Using Optocoupler Feedback.
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3
AN-37
INDUCTOR
TOKIN
COILCRAFT
220
230
320
340
440
430
SBC2-681-211
SBC2-681-211
SBC3-681-211
SBC4-681-211
SBC4-681-211
SBC4-681-211
RFB0807-681
RFB0807-681
RFB0810-681
RFB0810-681
RFB0810-681
RFB0810-681
680
1000
1500
680
1000
680
1500
180
230
320
340
440
430
400
SBC2-681-211
SBC3-102-281
SBC3-152-251
SBC3-681-361
SBC4-102-291
SBC4-681-431
SBC6-152-451
RFB0807-681
RFB0807-102 LNK304
RFB0810-152
RFB0810-681 LNK305
RFB0810-102
RFB0810-681
LNK306
RFB1010-152
≤70
120
160
175
225
280
360
680
1200
1800
820
1200
820
1500
160
210
210
310
310
390
390
SBC2-681-211
SBC6-152-451
RFB0807-681
RFB0807-122
RFB0810-182
RFB0810-821
RFB1010-122
RFB1010-821
RFB1010-152
≤50
120
160
175
225
280
360
680
1500
2200
1200
1500
1200
2200
130
190
180
280
280
350
360
SBC2-681-211
SBC4-152-221
SBC4-222-211
SBC6-152-451
SBC6-222-351
RFB0807-681
RFB0810-152
RFB0810-222
RFB0810-122
RFB1010-152
RFB1010-122
-
VOUT
IOUT(MAX)
5
≤120
160
175
225
280
360
680
680
680
680
680
680
12
≤85
120
160
175
225
280
360
15
24
µH IRMS (mA)
Other Standard Components
RBIAS: 2 kΩ, 1%, 1/8 W
CBP: 0.1 µF, 50 V Ceramic
CFB: 10 µF, 1.25 • VO
DFB: 1N4005GP
RZ:
470 Ω to 2 kΩ, 1/8 W, 5%
Table 3. Components Quick Select for Buck Converters.
4
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1/04
LNK30X
LNK304
LNK305
LNK306
LNK304
LNK305
LNK306
LNK304
LNK305
LNK306
MODE DIODE trr
RFB
VZ
MDCM
CCM
MDCM
CCM
MDCM
CCM
MDCM
MDCM
CCM
MDCM
CCM
MDCM
CCM
≤75 ns
≤35 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
3.84 kΩ 3.9 V
≤75 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
11.86 kΩ 11 V
MDCM
MDCM
CCM
MDCM
CCM
MDCM
CCM
MDCM
MDCM
CCM
MDCM
CCM
MDCM
CCM
≤75 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
15.29 kΩ 13 V
≤75 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
25.6 kΩ
22 V
AN-37
INDUCTOR
TOKIN
COILCRAFT
220
230
340
320
440
430
SBC2-681-211
SBC2-681-211
SBC3-681-361
SBC4-681-431
SBC4-681-431
SBC4-681-431
RFB0807-681
RFB0807-681
RFB0810-681
RFB0810-681
RFB0810-681
RFB0810-681
680
1200
1800
820
1200
820
1800
180
220
210
320
310
410
410
SBC2-681-211
-
RFB0807-681
RFB1010-122
RFB0807-182
RFB0807-821
RFB0810-122
RFB0810-821
RFB1010-182
≤50
120
160
175
225
280
360
680
1500
2200
1000
1500
1200
2200
180
220
220
320
320
400
410
SBC2-681-211
SBC3-152-251
SBC4-222-211
SBC4-102-291
SBC4-152-251
SBC6-222-351
RFB0807-681
RFB0807-152
RFB0810-222
RFB0810-102
RFB0810-152
RFB0810-122
RFB1010-222
≤35
120
160
175
225
280
360
680
2200
3300
1800
2200
1800
3300
180
210
210
300
290
370
410
SBC2-681-211
SBC3-222-191
SBC4-332-161
SBC4-222-211
-
RFB0807-681
RFB0810-222 LNK304
RFB0810-332
RFB0810-182
LNK305
RFB1010-222
RFB1010-182
LNK306
-
VOUT
IOUT(MAX)
5
≤120
160
175
225
280
360
680
680
680
680
680
680
≤70
120
160
175
225
280
360
12
15
24
µH IRMS (mA)
LNK30X
LNK304
LNK305
LNK306
LNK304
LNK305
LNK306
LNK304
LNK305
LNK306
MODE DIODE trr
RFB
VZ
MDCM
CCM
MDCM
CCM
MDCM
CCM
≤75 ns
≤35 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
3.84 kΩ 3.9 V
MDCM
MDCM
CCM
MDCM
CCM
MDCM
CCM
≤75 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
11.86 kΩ 11 V
MDCM
MDCM
CCM
MDCM
CCM
MDCM
CCM
≤75 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
MDCM
MDCM
CCM
MDCM
CCM
MDCM
CCM
≤75 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
≤75 ns
≤35 ns
15.29 kΩ 13 V
25.6 kΩ 22 V
Other Standard Components
RBIAS: 2 kΩ, 1%, 1/8 W
CBP: 0.1 µF, 50 V Ceramic
CFB: 10 µF, 1.25 • VO
DFB: 1N4005GP
RZ:
470 Ω to 2 kΩ, 1/8 W, 5%
Table 4. Components Quick Select for Buck-Boost Converters.
PART NO.
VRRM
(V)
IF
(A)
t rr
(ns)
MUR160
600
1
50
Leaded
Vishay
UF4005
600
1
75
Leaded
Vishay
BYV26C
600
1
30
Leaded
Vishay/Philips
FE1A
600
1
35
Leaded
Vishay
STTA10 6
600
1
20
Leaded
ST Microelectronics
STTA10 6U
600
1
20
SMD
ST Microelectronics
US1J
600
1
75
SMD
Vishay
PACKAGE MANUFACTURER
Table 5. List of Ultra-Fast Diodes Suitable for use as the Freewheeling Diode.
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5
AN-37
LinkSwitch-TN Circuit Design
To regulate the output, an ON/OFF control scheme is used as
illustrated in Table 6. As the decision to switch is made on a
cycle by cycle basis, the resultant power supply has extremely
good transient response and removes the need for control loop
compensation components. If no feedback is received for 50 ms
then the supply enters auto restart.
LinkSwitch-TN Operation
The basic circuit configuration for a Buck converter using
LinkSwitch-TN is shown in Figure 1a.
Reference
Schematic
and Key
FB
S
D
+
= MOSFET
Enabled
BP
+
LinkSwitch-TN
VIN
VO
= MOSFET
Disabled Cycle Skipped
PI-3784-121603
ID
Is IFB
>49 µA? No
At beginning of each cycle the FEEDBACK
(FB) pin is sampled.
• If IFB < 49 µA then next cycle occurs
• If IFB > 49 µA then next switching cycle
is skipped
No
Yes
No
No
Yes
Yes
No
Normal
Operation
High load – few cycles skipped
Low load – many cycles skipped
PI-3767-121903
IFB < 49 µA, > 50 ms
= Auto Restart
Auto Restart
50 ms
800 ms
Auto Restart = 50 ms ON / 800 ms OFF
PI-3768-121603
Table 6. LinkSwitch-TN Operation.
6
A
1/04
If no feedback (IFB < 49 µA) for > 50 ms
then output switching is disabled for
approximately 800 ms.
AN-37
To allow direct sensing of the output voltage, without the need
for a reference (Zener diode or reference IC), the FB pin voltage
is tightly toleranced over the entire operating temperature
range. For example, this allows a 12 V design with an overall
output tolerance of ±10%. For higher performance, an optocoupler can be used with a reference as shown in table 2. Since
the optocoupler just provides level shifting, it does not need to
be safety rated or approved. The use of an optocoupler also
allows flexibility in the location of the device, for example it
allows a buck converter configuration with the LinkSwitch-TN
in the low side return rail, reducing EMI as the SOURCE pins
and connected components are no longer part of the switching
node.
freewheeling diode, and the average current through the output
inductor are slightly lower in the Buck topology as compared to
the Buck-Boost topology.
Selecting the Operating Mode – MDCM and CCM
Operation
Selecting the Topology
At the start of a design, select between mostly discontinuous
conduction mode (MDCM) and continuous conduction mode
(CCM) as this decides the selection of the LinkSwitch-TN
device, freewheeling diode and inductor. For maximum output
current select CCM, for all other cases MDCM is recommended.
Overall, select the operating mode and components to give the
lowest overall solution cost. Table 7 summarizes the trade-offs
between the two operating modes.
If possible use the Buck topology, the Buck topology maximizes
the available output power from a given LinkSwitch-TN and
inductor value. Also, the voltage stress on the power switch and
Additional differences between CCM and MDCM include
better transient response for DCM and lower output ripple (for
same capacitor ESR) for CCM. However these differences, at
COMPARISON OF CCM AND MDCM OPERATING MODES
OPERATING MODE
MDCM
CCM
IL
IL
IO
Operating
Description
IO
t
tON
tOFF
tIDLE
PI-3769-121803
t
tON
tOFF
PI-3770-121503
Inductor current falls to zero during tOFF,
Borderline between MDCM and CCM when
tIDLE = 0.
Current flows continuously in the inductor for
the entire duration of a switching cycle.
Lower Cost
Lower value, smaller size.
Higher Cost
Higher value, larger size.
Freewheeling
Diode
Lower Cost
75 ns ultra-fast reverse recovery type
(≤35 ns for ambient >70 °C).
Higher Cost
35 ns ultra-fast recovery type required.
LinkSwitch-TN
Potentially Higher Cost
May require larger device to deliver required
output current–depends on required output
current.
Potentially Lowest Cost
May allow smaller device to deliver required
output current–depends on required output
current.
Efficiency
Higher Efficiency
Lower switching losses.
Lower Efficiency
Higher switching losses.
Overall
Typically Lower Cost
Typically Higher Cost
Inductor
Table 7. Comparison of Mostly Discontinuous Conduction (MDCM) and Continuous Conduction (CCM) Modes of Operation.
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7
AN-37
the low output currents of LinkSwitch-TN applications, are
normally not significant.
Output Power, PO: in Watts.
Power supply efficiency, η: 0.7 for a 12 V output, 0.55 for a
5 V output if no better reference data available.
The conduction mode CCM or MDCM of a Buck or BuckBoost converter primarily depends on input voltage, output
voltage, output current and device current limit. The input
voltage, output voltage and output current are fixed design
parameters therefore the LinkSwitch-TN (current limit) is the
only design parameter that sets the conduction mode.
Total Capacitance CIN(TOTAL)
µF/POUT (CIN1 + CIN2)
AC Input
Voltage (VAC)
100/115
230
Universal
The phrase “mostly discontinuous” is used as with on-off
control, since a few switching cycles may exhibit continuous
inductor current, the majority of the switching cycles will be in
the discontinuous conduction mode. A design can be made
fully discontinuous but that will limit the available output
current, making the design less cost effective.
Half Wave
Rectification
6-8
1-2
6-8
Full Wave
Rectification
3-4
1
3-4
Table 10. Suggested Total Input Capacitance Values for Different
Input Voltage Ranges.
Step 2. Determine AC Input Stage
Step-by-Step Design Procedure
The input stage comprises fusible resistor(s) input rectification
diodes and line filter network. The fusible resistor should be
chosen as flame proof and depending on the differential line
input surge requirements, a wire wound type may be required.
The fusible resistor(s) provides fuse safety, inrush current
limiting and differential mode noise attenuation.
Step 1. Determine System Requirements VACMIN,
VACMAX, PO, VO, fL, η
Determine the input voltage range from Table 8.
Input (VAC)
100/115
230
Universal
VACMIN
85
195
85
VACMAX
132
265
265
For designs ≤1 W it is lower cost to use half-wave rectification,
>1 W full wave rectification (smaller input capacitors). The
EMI performance of half wave rectified designs is improved by
adding a second diode in the lower return rail. This provides
EMI gating (EMI currents only flow when the diode is
conducting) and also doubles differential surge withstand as the
surge voltage is shared across two diodes. Table 9 shows the
recommended input stage based on output power for a universal
input design while Table 10 shows how to adjust the input
capacitance for other input voltage ranges.
Table 8. Standard Worldwide Input Line Voltage Ranges.
Line Frequency, fL: 50 or 60 Hz, for half-wave rectification
use fL/2.
Output Voltage, VO: in Volts.
≤ 0.25 W
POUT
0.25-1 W
>1W
DIN1-4
RF1
DIN1
*
DIN2*
+
**
CIN
AC
IN
RF2
85-265 VAC
Input Stage
Comments
RF1
DIN1
AC
IN
**
CIN1
A
1/04
+
CIN2
DIN2*
RF1, RF2: 100-470 Ω,
0.5 W, Fusible
CIN: ≥2.2 µF, 400 V
DIN1, DIN2: 1N4007, 1 A,
1000 V
RF1
DIN1
AC
IN
LIN
**
CIN1
+
RF1
CIN2
AC IN
LIN
**
CIN1
+
CIN2
DIN2*
PI-3772-121603
PI-3771-121603
RF1: 8.2 Ω, 1 W Fusible
RF2: 100 Ω, 0.5 W,
Flame proof
CIN1, CIN2: ≥3.3 µF,
400 V each
DIN1, DIN2: 1N4007, 1 A,
1000 V
PI-3773-121603
RF1: 8.2 Ω, 1 W Fusible
LIN: 470 µH-2.2 mH,
0.05 A-0.3 A
CIN1, CIN2: ≥4 µF/WOUT,
400 V each
DIN1, DIN2: 1N4007, 1 A,
1000 V
PI-3774-121603
RF1: 8.2 Ω, 1 W Fusible
LIN: 470 µH-2.2 mH,
0.05 A-0.3 A
CIN1, CIN2: ≥2 µF/WOUT,
400 V each
DIN1, DIN2: 1N4005, 1 A,
600 V
*Optional for improved EMI and line surge performance. Remove for designs requiring no impedance in return rail.
**Increase value to meet required differential line surge performance.
Table 9. Recommended AC Input Stages For Universal Input.
8
RF2
AN-37
Step 3. Determine Minimum and Maximum DC Input
Voltages VMIN and VMAX Based on AC Input Voltage
Step 5. Select the Output Inductor
Tables 3 and 4 provide inductor values and RMS current ratings
for common output voltages and currents based on the
calculations in the design spreadsheet. Select the next nearest
higher voltage and/or current above the required output
specification. Alternatively the PIXls spreadsheet tool in the
PI Expert software design suite or Appendix A can be used to
calculate the exact inductor value (Eq. A7) and RMS current
rating (Eq. A20).
Calculate VMAX as
(1)
VMAX = 2 ⋅ VACMAX
Assuming that the value of input fusible resistor is small, the
voltage drop across it can be ignored.
Assume bridge diode conduction time of tc = 3 ms if no other
data available.
It is recommended that the value of inductor chosen should be
closer to LTYP rather than 1.5 • LTYP due to lower DC resistance
and higher RMS rating. The lower limit of 680 µH limits the
maximum di/dt to prevent very high peak current values.
Tables 3 and 4 provide reference part numbers for standard
inductors from two suppliers.
Derive minimum input voltage VMIN
VMIN =
(2 ⋅ VACMIN 2 )
 1

2 ⋅ PO 
− tC 
 2 ⋅ fL

η ⋅ CIN ( TOTAL )
(2)
680 µH < LTYP < L < 1.5 ⋅ LTYP
If VMIN is ≤70 V then increase value of CIN(TOTAL).
For LinkSwitch-TN designs the mode of operation is not
dependent on the inductor value. The mode of operation is a
function of load current and current limit of the chosen device,
the inductor value merely sets the average switching frequency.
Step 4. Select LinkSwitch-TN Device Based on
Output Current and Current Limit
Decide on operating mode - refer to Table 7.
Figure 2 shows a typical standard inductor manufacturer’s data
sheet. The value of off-the-shelf “drum core / dog bone / I core”
inductors will drop up to 20% in value as the current increases.
The constant KL_TOL in equation (A7) and the design spreadsheet
adjusts for both this drop and the initial inductance value
tolerance.
For MDCM operation, the output current (IO) should be less
than or equal to half the value of the minimum current limit of
the chosen device from the data sheet.
(3)
I LIMIT _ MIN > 2 ⋅ IO
For example if a 680 µH, 360 mA inductor is required, referring
to Figure 2, the tolerance is 10% and an estimated 9.5% for the
reduction in inductance at the operating current (approximately
[0.36/0.38] • 10). Therefore the value of KL_TOL = 1.195 (19.5%).
For CCM operation, the device should be chosen such that the
output current IO, is more than 50%, but less than 80% of the
minimum current limit ILIMIT_MIN.
If no data is available assume a KL_TOL of 1.15 (15%).
0.5 ⋅ I LIMIT _ MIN < IO < 0.8 ⋅ I LIMIT _ MIN
(4)
Not all the energy stored in the inductor is delivered to the load,
due to losses in the inductor itself. To compensate for this a loss
Please see data sheet for LinkSwitch-TN current limit values.
Inductance and
Tolerance
SBC3 Series (SBC3-
-
Current Rating
for 20 °C Rise
Current Rating
for Value -10%
Current Rating
for 40 °C Rise
)
Inductance
L(mH/ at 10 kHz
Rdc
(W)
max.
Rated Current
(A)
∆T = 20 °C
680±10%
1000±10%
1500±10%
2200±10%
3300±10%
1.62
2.37
3.64
5.62
7.66
0.36
0.28
0.25
0.19
0.15
Model
681-361
102-281
152-251
222-191
332-151
(5)
Figure 2. Example of Standard Inductor Data Sheet.
Current (Reference Value)
(A)
∆T = 40 °C
L change rate -10%
0.50
0.39
0.35
0.26
0.21
0.38
0.31
0.26
0.21
0.17
PI-3783-121003
A
1/04
9
AN-37
factor KLOSS is used. This has a recommended value of between
50% and 66% of the total supply losses as given by equation (5).
For example, a design with an overall efficiency (η) of 0.75
would have a KLOSS value of between 0.875 and 0.833.
 (1 − η) 
 2(1 − η) 
K LOSS = 1 − 
 to 1 − 

 2 
 3 
(6)
Let the value of RBIAS = 2 kΩ; this biases the feedback network
at a current of ∼0.8 mA. Hence the value of RFB is given by
RFB =
(VO − VFB ) ⋅ RBIAS = (VO − 1.65 V) ⋅ 2 kΩ
VO − VFB
=
VFB
1.748 V
+ I FB VFB + ( I FB ⋅ RBIAS )
RBIAS
(10)
Step 9. Select the Feedback Diode and Capacitor
Step 6. Select Freewheeling Diode
For MDCM operation at tAMB ≤70 °C, select an ultra-fast diode
with trr ≤75 ns. At tAMB >70 °C, trr ≤ 35 ns.
For the feedback capacitor, use a 10 µF general purpose
electrolytic capacitor with a voltage rating ≥1.25 • VO.
For CCM operation, select an ultra-fast diode with trr ≤35 ns.
For the feedback diode, use a glass passivated 1N4005GP or
1N4937GP device with a voltage rating of ≥1.25 • VMAX.
Allowing 25% design margin for the freewheeling diode,
Step 10. Select Bypass Capacitor
VPIV > 1.25 ⋅ VMAX
(7)
The diode must be able to conduct the full load current. Thus
I F > 1.25 ⋅ IO
(8)
Table 5 lists common freewheeling diode choices.
Use 0.1 µF, 50 V ceramic capacitor.
Step 11. Select Pre-load Resistor
For direct feedback designs if the minimum load <3 mA then
calculate RPL = VO / 3 mA.
Other information
Step 7. Select Output Capacitor
Startup Into Non-Resistive Loads
The output capacitor should be chosen based on the output
voltage ripple requirement. Typically the output voltage ripple
is dominated by the capacitor ESR and can be estimated as:
ESRMAX =
VRIPPLE
I LIMIT
(9)
where VRIPPLE is the maximum output ripple specification and
ILIMIT is the LinkSwitch-TN current limit. The capacitor ESR
value should be specified approximately at the switching
frequency of 66 kHz.
Capacitor values above 100 µF are not recommended as they
can prevent the output voltage from reaching regulation during
the 50 ms period prior to auto-restart. If more capacitance is
required, then a soft-start capacitor should be added (see Other
Information section).
If the total system capacitance is >100 µF or the output voltage
is >12 V, then the output may fail to reach regulation during
start-up. This may also be true when the load is not resistive, for
example the output is supplying a motor or fan.
To increase the startup time, a soft-start capacitor can be added
across the feedback resistor, as shown in Figure 3. The value of
this soft-start capacitor is typically in the range of 0.47 µF to
47 µF with a voltage rating of 1.25 • VO. Figure 4 shows the
effect of CSS used on a 12 V, 150 mA design driving a motor
load.
CSS
RFB
Step 8. Select the Feedback Resistors
FB
The values of RFB and RBIAS are selected such that at the
regulated output voltage, the voltage on the FEEDBACK pin
(VFB) is 1.65 V. This voltage is specified for a FEEDBACK pin
current (IFB) of 49 µA.
+
VIN
D
BP
S
LinkSwitch-TN
+
VO
PI-3775-121003
Figure 3. Example Schematic Showing Placement of Soft-Start
Capacitor.
10
A
1/04
AN-37
12
No soft-start capacitor. Output
never reaches regulation (in
auto-restart).
10
Voltage (V)
PI-3785-010504
14
FB
+
+7 V
LinkSwitch-TN
VIN
8
BP
S
D
6V8
RTN
6
5V1
4
-5 V
PI-3776-121003
2
0
Figure 5. Example Circuit – Generating Dual Output Voltages.
-2
0
2.5
5
Generating Negative and Positive Outputs
Time (s)
PI-3786-010504
14
12
Voltage (V)
10
8
6
4
Soft-start capacitor value too
small – output still fails to reach
regulation before auto-restart.
2
In appliance applications there is often a requirement to generate
both an AC line referenced positive and negative output. This
can be accomplished using the circuit in Figure 5. The two
zener diodes have a voltage rating close to the required output
voltage for each rail and ensure that regulation is maintained
when one rail is lightly and the other heavily loaded. The
LinkSwitch-TN circuit is designed as if it were a single output
voltage with an output current equal to the sum of both outputs.
The magnitude sum of the output voltages in this example being
12 V.
Constant Current Circuit Configuration (LED Driver)
0
-2
0
2.5
5
Time (s)
PI-3787-010503
14
The circuit shown in Figure 5 is ideal for driving constant
current loads such as LEDs. It uses the tight tolerance and
temperature stable FEEDBACK pin of LinkSwitch-TN as the
reference to provide an accurate output current.
12
RFB
300 Ω
Voltage (V)
10
Correct value of soft-start
capacitor – output reaches
regulation before auto-restart.
8
FB
6
+
D
BP
RBIAS
2 kΩ
VRFB DFB
RSENSE
S
VIN
IO
DFW
LinkSwitch-TN
4
Optional
See Text
L
CSENSE
CO
2
0
PI-3795-122403
-2
0
2.5
5
Time (s)
Figure 4. Example of Using Soft-Start Capacitor to Enable Driving
a 12 V, 0.15 A Motor Load. All Measurements were made
at 85 VAC (worst case condition).
Figure 6. High-Side Buck-Boost Constant Current Output
Configuration.
To generate a constant current output, the average output
current is converted to a voltage by resistor RSENSE and capacitor
CSENSE and fed into the FEEDBACK pin via RFB and RBIAS.
With the values of RBIAS and RFB as shown, the value of RSENSE
should be chosen to generate a voltage drop of 2 V at the
A
1/04
11
AN-37
required output current. Capacitor CSENSE filters the voltage
across RSENSE, which is modulated by inductor ripple current.
The value of CSENSE should be large enough to minimize the
ripple voltage, especially in MDCM designs. A value of CSENSE
is selected such that the time constant (t) of RSENSE and CSENSE is
greater than 20 times that of the switching period (15 µs). The
peak voltage seen by CSENSE is equal to RSENSE • ILIMIT(MAX).
The output capacitor is optional; however with no output
capacitor the load will see the full peak current (ILIMIT) of the
selected LinkSwitch-TN. Increase the value of CO (typically in
the range of 100 nF to 10 uF) to reduce the peak current to an
acceptable level for the load.
If the load is disconnected, feedback is lost and the large output
voltage which results may cause circuit failure. To prevent this,
a second voltage control loop, DFB and VRFB, can be added as
shown if Figure 6. This also requires that CO is fitted. The
voltage of the Zener is selected as the next standard value above
the maximum voltage across the LED string when it is in
constant current operation.
Recommended Layout Considerations
Traces carrying high currents should be as short in length and
thick in width, as possible. These are the traces which connect
the input capacitor, LinkSwitch-TN, inductor, freewheeling
diode and the output capacitor.
Most off-the-shelf inductors are drum core inductors or dogbone inductors. These inductors do not have a good closed
magnetic path, and are a source of significant magnetic coupling.
They are a source of differential mode noise and for this reason,
they should be placed as far away as possible from the AC input
lines.
Appendix A
Calculations for Inductor Value for Buck and BuckBoost Topologies
There is a minimum value of inductance that is required to
deliver the specified output power, regardless of line voltage
and operating mode.
The same design equations / design spreadsheet can be used as
for a standard buck-boost design, with the following additional
considerations.
1. VO = LED VF • Number of LEDs per string
2. IO = LED IF • Number of strings
3. Lower efficiency estimate due to RSENSE losses (enter
RSENSE into design spreadsheet as inductor resistance)
4. Set RBIAS = 2 kΩ and RFB = 300 Ω
5. RSENSE = 2/IO
6. CSENSE = 20 • (15 µs/RSENSE)
7. Select CO based on acceptable output ripple current
through the load
8. If the load can be disconnected or for additional fault
protection, add voltage feedback components DFB and
VRFB, in addition to CO.
VIN-VO
VL
t
VO
ILimit
IL
IO
t
tON
tOFF
tIDLE
PI-3778-121803
Figure 7. Inductor Voltage and Inductor Current of a Buck
Converter in DCM.
Thermal Environment
To ensure good thermal performance, the SOURCE pin
temperature should be maintained below 100 °C, by providing
adequate heatsinking.
For applications with high ambient temperature (>50 °C), it is
recommended to build and test the power supply at the maximum
operating ambient temperature, and ensure that there is adequate
thermal margin. The figures for maximum output current
provided in the data sheet correspond to an ambient temperature
of 50 °C, and may need to be thermally derated. Also, it is
recommended to use ultra fast (≤35 ns) low reverse recovery
diodes at higher operating temperatures (>70 °C).
12
A
1/04
As a general case, Figure 7 shows the inductor current in
discontinuous conduction mode (DCM). The following
expressions are valid for both CCM as well as DCM operation.
There are three unique intervals in DCM as can be seen from
Figure 7. Interval tON is when the LinkSwitch-TN is ON and the
freewheeling diode is OFF. Current ramps up in the inductor
from an initial value of zero. The peak current is the current limit
ILIMIT of the device. Interval tOFF is when the LinkSwitch-TN is
OFF and the freewheeling diode is ON. Current ramps down to
zero during this interval. Interval tIDLE is when both the
LinkSwitch-TN and freewheeling diode are OFF, and the inductor
current is zero.
AN-37
In CCM this idle state does not exist and thus tIDLE = 0.
Neglecting the forward voltage drop of the freewheeling diode,
we can express the current swing at the end of interval tON in a
buck converter as
∆I (tON ) = I RIPPLE =
VMIN − VDS − VO
⋅ tON
LMIN
(
I RIPPLE = 2 ⋅ I LIMIT _ MIN − IO
)
I RIPPLE = I LIMIT _ MIN ,
VO
⋅ tOFF
LMIN
(A2)
The initial current through the inductor at the beginning of each
switching cycle can be expressed as
I INITIAL = I LIMIT _ MIN − I RIPPLE
1
1⋅ I
⋅ 
LIMIT _ MIN + I INITIAL ⋅ tON +
2

2


 I LIMIT _ MIN + I INITIAL ⋅ tOFF + 0 ⋅ t IDLE 
(
(
(
)
2 ⋅ (VO ⋅ IO ) ⋅ (VMIN − VDS − VO )
( I LIMIT _ MIN 2 − I INITIAL 2 ) ⋅ FSMIN ⋅ (VMIN − VDS )
(A6)
This however does not account for the losses within the inductor
(resistance of winding and core losses) and the freewheeling
diode, which will limit the maximum power delivering capability
and thus reduce the maximum output current. The minimum
inductance must compensate for these losses in order to deliver
specified full load power. An estimate of these losses can be
made by estimating the total losses in the power supply, and
then allocating part of these losses to the inductor and diode.
This is done by the loss factor KLOSS which increases the size of
the inductor accordingly.
Furthermore, typical inductors for this type of application are
bobbin core or dog bone chokes. The specified current rating
refer to a temperature rise of 20 °C or 40 °C and to an inductance
drop of 10%. We must incorporate an Inductance Tolerance
Factor KL_TOL within the expression for minimum inductance, to
account for this manufacturing tolerance. The typical inductance
value thus can be expressed as
(A3)
The average current through the inductor over one switching
cycle is equal to the output current IO. This current can be
expressed as
TSW _ MAX
)
t IDLE > 0 ( for DCM )
Similarly, we can express the current swing at the end of
interval tOFF as
IO =
TSW _ MAX
t IDLE = 0 ( for CCM )
where
IRIPPLE = Inductor Ripple Current
ILIMIT_MIN = Minimum current limit
VMIN = Minimum DC Bus Voltage
VDS = On state Drain to Source Voltage drop
VO = Output Voltage
LMIN = Minimum Inductance
1
(
1
LMIN =
(A1)
∆I (tOFF ) = I RIPPLE =
IO =
I RIPPLE ⋅ LMIN 
1⋅ I
LIMIT _ MIN + I INITIAL
2
VMIN − VDS − VO 


I RIPPLE ⋅ LMIN 
+ 1 ⋅ I
I
+
INITIAL
 2 LIMIT _ MIN

VO


(A5)
)
)
LTYP =
V ⋅I 
2 ⋅ K L _ TOL ⋅  O O  ⋅ (VMIN − VDS − VO )
 K LOSS 
( I LIMIT _ MIN 2 − I INITIAL 2 ) ⋅ FSMIN ⋅ (VMIN − VDS )
(A7)
where
KLOSS is a loss factor, which accounts for the off-state total losses
of the inductor.
(A4)
where
IO = Output Current.
TSW_MAX = the switching interval corresponding to minimum
switching frequency FSMIN.
KL_TOL is the Inductor Tolerance Factor and can be between 1.1
and 1.2. A typical value is 1.15.
With this typical inductance we can express maximum output
power as
1
⋅ LTYP ⋅ I LIMIT _ MIN 2 − I INITIAL 2 ⋅
2
K
VMIN − VDS
(A8)
FSMIN ⋅
⋅ LOSS
VMIN − VDS − VO K L _ TOL
PO _ MAX =
Substituting for tON and tOFF from equations (A1) and (A2) we
have
(
)
A
1/04
13
AN-37
Similarly for Buck-Boost topology the expressions for LTYP and
PO_MAX are
LTYP =
V ⋅I 
2 ⋅ K L _ TOL ⋅  O O 
 K LOSS 
( I LIMIT _ MIN 2 − I INITIAL 2 ) ⋅ FSMIN
PO _ MAX =
The current through the LinkSwitch-TN as a function of time is
given by
iSW (t ) = I INITIAL +
(A9)
1
⋅ LTYP ⋅ ( I LIMIT _ MIN 2 − I INITIAL 2 ) (A10)
2
VMIN − VDS − VO
⋅ t , 0 < t ≤ tON
L
iSW (t ) = 0 , tON < t ≤ tON
The current through the Freewheeling diode as a function of
time is given by
iD (t ) = 0,
Average Switching Frequency
Since LinkSwitch-TN uses an on-off type of control, the frequency
of switching is non-uniform due to cycle skipping. We can
average this switching frequency by substituting the maximum
power as the output power in equation (A8). Simplifying, we have
FSAVG =
2 ⋅ VO ⋅ IO ⋅ K L _ TOL
(
2
2
)
L ⋅ I LIMIT − I INITIAL K LOSS
⋅
Similarly for Buck-Boost converter, simplifying equation (A9)
we have
FSAVG
K
2 ⋅ VO ⋅ IO
=
⋅ L _ TOL
2
2
L ⋅ I LIMIT − I INITIAL K LOSS K LOSS
(
0 < t ≤ tON
VO
, tON < t ≤ t SW
L
V
I LIMIT _ MIN − O ⋅ t < 0
L
iD (t ) = I LIMIT _ MIN −
(A15)
iD (t ) = 0,
(A16)
And the current through the inductor as a function of time is
given by
VMIN − VDS − VO
VMIN − VDS
(A11)
(A14)
iL (t ) = iSW ( t ) + iD (t )
(A17)
From the definition of RMS currents we can express the RMS
currents through the switch, freewheeling diode and inductor as
follows
t ON
1
iSW _ RMS =
TAVG
)
∫ iSW (t )2 ⋅ dt
(A18)
0
(A12)
iD _ RMS =
Calculation of RMS Currents
The RMS current value through the inductor is mainly required
to ensure that the inductor is appropriately sized and will not
over heat. Also, RMS currents through the LinkSwitch-TN and
freewheeling diode are required to estimate losses in the power
supply.
Assuming CCM operation, the initial current in the inductor in
steady state is given by
I INITIAL = I LIMIT _ MIN −
VO
⋅ tOFF
L
For DCM operation this initial current will be zero.
14
A
1/04
(A13)
iL _ RMS =
t ON + t OFF
1
TAVG
1
TAVG
∫
t ON
iD (t )2 ⋅ dt
TAVG
2
∫ (iSW (t ) + iD (t )) ⋅ dt
0
(A19)
(A20)
Since the switch and freewheeling diode currents fall to zero
during the turn off and turn on intervals respectively, the RMS
inductor current is simplified to
iL _ RMS = iSW _ RMS 2 + iD _ RMS 2
(A21)
AN-37
Table A1 lists the design equations for important parameters
using the Buck and Buck-Boost topologies.
PARAMETER
BUCK
BUCK-BOOST
LTYP
LTYP =
FAVG
iSW(t)
LinkSwitch-TN
Current
id(t)
Diode
Forward
Current
iL(t) Inductor
Current
Max Drain
Voltage
FSTYP =
 V ⋅I 
2 ⋅ K L ⋅  O O  ⋅ (VMIN − VDS − VO )
 K L _ LOSS 
( I LIMIT _ MIN 2 − I INITIAL 2 ) ⋅ FSMIN ⋅ (VMIN − VDS )
V
2 ⋅ VO ⋅ IO ⋅ K L
− VDS − VO
⋅ MIN
2
V
L ⋅ I LIMIT − I INITIAL ⋅ K L _ LOSS
MIN − VDS
(
)
VMIN − VDS − VO
⋅ t , t ≤ tON
L
iSW (t ) = 0 , t > tON
iSW (t ) = iINIT +
VO
⋅ t , t > tON
L
V
iD (t ) = 0 , I LIMIT _ MIN − O ⋅ t < 0
L
iD (t ) = 0 , t ≤ tON
iD (t ) = I LIMIT _ MIN −
LTYP =
FSAVG =
 V ⋅I 
2 ⋅ KL ⋅  O O 
 K L _ LOSS 
( I LIMIT _ MIN 2 − I INITIAL 2 ) ⋅ FSMIN
2 ⋅ VO ⋅ IO
KL
⋅
2
2
K
L ⋅ I LIMIT − I INITIAL
L _ LOSS
(
)
VMIN − VDS
⋅ t , t ≤ tON
L
iSW (t ) = 0 , t > tON
iSW (t ) = iINIT +
VO
⋅ t , t > tON
L
V
iD (t ) = 0 , I LIMIT _ MIN − O ⋅ t < 0
L
iD (t ) = 0 , t ≤ tON
iD (t ) = I LIMIT _ MIN −
iL (t ) = iSW (t ) + iD (t )
iL (t ) = iSW (t ) + iD (t )
VMAX
VMAX + VO
Table A1. Circuit Characteristics for Buck and Buck-Boost Topologies.
A
1/04
15
AN-37
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Power Integrations reserves the right to make changes to its products at any time to improve reliability or manufacturability. Power Integrations does not
assume any liability arising from the use of any device or circuit described herein, nor does it convey any license under its patent rights or the rights of others.
The products and applications illustrated herein (including circuits external to the products and transformer construction) may be covered by one or more U.S.
and foreign patents or potentially by pending U.S. and foreign patent applications assigned to Power Integrations. A complete list of Power Integrations’ patents
may be found at www.powerint.com.
LIFE SUPPORT POLICY
POWER INTEGRATIONS' PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR
SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF POWER INTEGRATIONS. As used herein:
1. Life support devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform, when
properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in a significant injury to the user.
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support device or system, or to affect its safety or effectiveness.
The PI logo, TOPSwitch, TinySwitch, LinkSwitch and EcoSmart are registered trademarks of Power Integrations.
PI Expert and DPA-Switch are trademarks of Power Integrations. ©Copyright 2004, Power Integrations
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16
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