ETC UCC38510

 SLUS517− DECEMBER 2002
FEATURES
D Provides Control of PFC and PWM Power
D
D
D
D
DESCRIPTION
Stages In One Device
Leading-Edge PFC, Trailing-Edge PWM
Modulation for Reduced Ripple
Built-In Sequencing of PFC and PWM
Turn-On
2-A Source and 3-A Sink Gate Drive for Both
PFC and PWM Stages
Typical 16-ns Rise Time and 7-ns Fall Time
into 1-nF Loads
PFC Features
− Average-Current-Mode Control for
Continuous Conduction Mode Operation
− Highly-Linear Multiplier for Near-Unity
Power Factor
− Input Voltage Feedforward Implementation
− Improved Load Transient Response
− Accurate Power Limiting
− Zero Power Detect
PWM Features
−
−
−
−
−
Peak-Current-Mode Control Operation
1:1 or 1:2 PFC:PWM Frequency Options
Programmable maximum duty cycle
Programmable Soft-Start
Two Hysteresis Options for Differing
Hold-Up Time Requirements
10.2
16
10.2
PWM
HYSTERESIS
(V)
1.45
3.20
PFC:PWM
FREQUENCY RATIO
1:1
1:2
UCC28510
UCC28514
UCC28511
UCC28515
UCC28512
UCC28516
UCC28513
UCC28517
Based on the average current mode control
architecture with input voltage feedforward of prior
PFC/PWM combination controllers, these devices
offer performance advantages. Two new key
PWM features are programmable maximum duty
cycle and the 2x PWM frequency options to the
base PFC frequency. For the PFC stage, the
devices feature an improved multiplier and the
use of a transconductance amplifier for enhanced
transient response.
The core of the PFC section is in a three-input
multiplier that generates the reference signal for
the line current. The UCC28510 series features a
highly linearized multiplier circuit capable of
producing a low distortion reference for the line
current over the full range of line and load
conditions. A low-offset, high-bandwidth current
error amplifier ensures that the actual inductor
current (sensed through a resistor in the return
path) follows the multiplier output command
signal. The output voltage error is processed
through a transconductance voltage amplifier.
AVAILABLE OPTIONS
PFC UVLO
TURN−ON
THRESHOLD
(V)
16
The UCC28510 series of combination PFC/PWM
controllers provide complete control functionality
for any off-line power system requiring
compliance with the IEC1000−3−2 harmonic
reduction requirements. By combining the control
and drive signals for the PFC and the PWM stages
into a single device, significant performance and
cost benefits are gained. By managing the
modulation mechanisms of the two stages
(leading-edge modulation for PFC and
trailing-edge modulation for PWM), the ripple
current in the boost capacitor is minimized.
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Copyright  2002, Texas Instruments Incorporated
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1
SLUS517− DECEMBER 2002
DESCRIPTION (CONTINUED)
The transient response of the circuit is enhanced by allowing a much faster charge/discharge of the voltage
amplifier output capacitance when the output voltage falls outside a certain regulation window. A number of
additional features such as UVLO circuit with selectable hysteresis levels, an accurate reference voltage for the
voltage amplifier, zero power detect, OVP/enable, peak current limit, power limiting, high-current output gate
driver characterize the PFC section.
The PWM section features peak current mode control (with a ramp signal available to add slope compensation),
programmable soft-start, accurate maximum duty cycle clamp, peak current limit and high-current output gate
driver. The oscillator for the combination controller is available in two versions. In UCC28510, UCC28511,
UCC28512, and UCC28513, the PWM and the PFC circuits are switched at the same frequency. In the
UCC28514, UCC28515, UCC28516, and UCC28517, the PWM stage frequency is twice that of the PFC
frequency. The PWM stage is suppressed until the PFC output has reached 90% of its programmed value during
startup. During line dropout and turn off, the device allows the PWM stage to operate until the PFC output has
dropped to 47% (UCC28512, UCC28513, UCC28516, and UCC28517) or 71% (UCC28510, UCC28511,
UCC28514, and UCC28515) of its nominal value. See available options table on page 1 for a summary of
options.
The UCC28510 family also features leading-edge modulation for the PFC stage and trailing-edge modulation
for the PWM stage in order to reduce the ripple current in the boost output capacitor. The current amplifier
implementation associated with this scheme also results in better noise immunity.
Available in 20-pin N and DW packages.
SIMPLIFIED APPLICATION DIAGRAM
PRIMARY
SECONDARY
+
+
RECT
D1
−
−
VAC
+
BIAS
UCC2851X
11
PWRGND
GT2
10
12
GT1
VCC
9
13
SS2
ISENSE2
8
14
PKLMT
VERR
7
15
CAOUT
16
ISENSE1
17
MOUT
18
−
REF
Z
6
5
D_MAX
4
IAC
VSENSE
3
19
VFF
RT
2
20
VREF
VAOUT
1
+
GND
CT_BUFF
Z
REF
2
VOUT
Z
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PWM
V−LOOP
SLUS517− DECEMBER 2002
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature (unless otherwise noted)†}
Supply voltage VCC
Idle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 V
Operating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 V
Gate drive current (GT1, GT2)
Continuous . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.4 A
Pulsed
Sourcing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −2.5 A
Sinking . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.5 A
Maximum GT1, GT2 voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.5 V to VCC+0.3 V
Input voltage
VSENSE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0 V to 11 V
D_MAX, SS2, CAOUT, ISENSE1, MOUT, VFF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.5 V to VREF+0.3 V
VAOUT, CT_BUFF, ISENSE2, PKLMT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.5 V to 6 V
Pin Current
RT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.5 mA
VFF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −5 mA
CT_BUFF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 mA
VAOUT, VERR, ISENSE2, SS2, CAOUT, IAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10 mA
Maximum pin capacitance
ISENSE2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220 pF
Operating junction temperature range, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −55 0C to 150 0C
Storage Temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65 0C to 150 0C
Lead temperature 1.6mm (1/16 inch from case for 10 seconds) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300 0C
Power dissipation
PDIP (N) package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 W
SOIC (DW) package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.7 W
† Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied.
Exposure to absolute−maximum−rated conditions for extended periods may affect reliability.
‡ Currents are positive into, negative out of the specified terminal. All voltages are referenced to GND.
ELECTROSTATIC DISCHARGE (ESD) PROTECTION
PARAMETER
MAX
Human body model
2.5
CDM
0.5
UNITS
kV
AVAILABLE OPTIONS{}
OPTIONS
PFC:PWM
FREQUENCY
RATIO
PACKAGED DEVICES
PFC UVLO
TURN-ON (V)
PFC UVLO
HYSTERESIS (V)
PWM UVLO2
TURN-OFF (V)
PWM UVLO2
HYSTERESIS (V)
PDIP−20
(N)
SOIC W−20
(DW)
1:1
16
6.3
5.30
1.45
UCC28510N
UCC28510DW
1:1
10.2
0.5
5.30
1.45
UCC28511N
UCC28511DW
1:1
16
6.3
3.55
3.2
UCC28512N
UCC28512DW
1:1
10.2
0.5
3.55
3.2
UCC28513N
UCC28513DW
1:2
16
6.3
5.30
1.45
UCC28514N
UCC28514DW
1:2
10.2
0.5
5.30
1.45
UCC28515N
UCC28515DW
1:2
16
6.3
3.55
3.2
UCC28516N
UCC28516DW
1:2
10.2
0.5
3.55
3.2
UCC28517N
UCC28517DW
† The DW package is available taped and reeled. Add R suffix to device type (e.g. UCC28510DWR) to order quantities of 2000 devices per reel.
‡ All devices are rated from −40°C to +105°C.
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3
SLUS517− DECEMBER 2002
ELECTRICAL CHARACTERISTICS
TA = –40°C to 105°C for the UCC2851x, TA = TJ, VCC = 12 V, RT = 156 kΩ, RCT_BUFF = 10 kΩ
(unless otherwise noted)
supply current
PARAMETER
TEST CONDITIONS
Supply current, off
VCC turn-on threshold −300 mV
Supply current, on
no load on GT1 or GT2
MIN
TYP
MAX
UNITS
100
150
µA
4
6
mA
PFC stage undervoltage lockout (UVLO)
PARAMETER
VCC turn-on threshold
VCC turn-off threshold
UVLO hysteresis
TEST CONDITIONS
MIN
TYP
MAX
UCC28510
UCC28512
UCC28514
UCC28516
15.4
16
16.6
UCC28511
UCC28513
UCC28515
UCC28517
9.7
10.2
10.8
UCC2851X
9.1
9.7
10.6
UCC28510
UCC28512
UCC28514
UCC28516
5.8
6.3
6.8
UCC28511
UCC28513
UCC28515
UCC28517
0.3
0.5
0.8
UNITS
V
voltage amplifier
PARAMETER
Input voltage
TEST CONDITIONS
MIN
TYP
MAX
25°C
7.39
7.50
7.61
Over temperature
7.35
7.50
7.65
100
300
VSENSE bias current
Open loop gain
VSENSE = VREF
2 V ≤ VAOUT ≤ 4 V
High-level output voltage
ILOAD = –150 µA
ILOAD = 150 µA
UNITS
V
nA
50
60
5.3
5.5
5.6
0.00
0.05
0.15
70
100
130
−1
−3.5
1
3.5
MIN
TYP
MAX
UNITS
VREF
+ 0.440
VREF
+ 0.490
VREF
+ 0.540
V
Hysteresis
300
500
600
Enable threshold
1.7
1.9
2.1
Enable hysteresis
0.08
0.2
0.3
Low-level output voltage
gM conductance
Maximum source current
IVAOUT = −20 µA to 20 µA
Maximum sink current
dB
V
µS
mA
overvoltage protection and enable
PARAMETER
TEST CONDITIONS
Overvoltage reference window
4
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mV
V
SLUS517− DECEMBER 2002
ELECTRICAL CHARACTERISTICS
TA = –40°C to 105°C for the UCC2851x, TA = TJ, VCC = 12 V, RT = 156 kΩ, RCT_BUFF = 10 kΩ
(unless otherwise noted)
current amplifier
PARAMETER
Input offset voltage
Input bias current
Input offset current
Open loop gain
TEST CONDITIONS
VCM = 0 V,
VCM = 0 V,
VCAOUT = 3 V
VCAOUT = 3 V
VCM = 0 V,
VCM = 0 V,
VCAOUT = 3 V
2 V ≤ VCAOUT ≤ 5 V
High-level output voltage
0 V ≤ VCM ≤ 1.5 V, VCAOUT = 3 V
ILOAD = –500 µA
Low-level output voltage
Gain bandwidth product(1)
ILOAD = 500 µA
See Note 1
Common−mode rejection ratio
MIN
TYP
–5
MAX
UNITS
0
5
−50
−100
25
100
mV
nA
90
dB
80
5.6
6.3
7.0
0
0.2
0.5
2.0
V
MHz
oscillator
PARAMETER
TEST CONDITIONS
fPWM, PWM frequency, initial accuracy
Frequency, voltage stability
TA = 25°C
10.8 V ≤ VCC ≤ 15 V
Frequency, total variation
Line, Temp
MIN
170
dc-to-dc ramp peak voltage
TYP
MAX
200
230
−1%
1%
160
240
4.5
dc-to-dc ramp amplitude voltage(1)
(peak-to-peak)
5.0
UNITS
kHz
kHz
5.5
4.0
V
PFC ramp peak voltage
4.5
5.0
5.5
PFC ramp amplitude voltage (peak-to-peak)
3.5
4.0
4.5
voltage reference
PARAMETER
Input voltage
Load regulation
TEST CONDITIONS
MIN
TYP
MAX
UNITS
25°C
7.39
7.50
7.61
V
Over temperature
7.35
7.50
7.65
V
5
15
1
10
–25
−50
Line regulation
IREF = −1 mA to −6 mA
10.8 V ≤ VCC ≤ 15 V
Short circuit current
VREF = 0V
−20
mV
mA
peak current limit
PARAMETER
TEST CONDITIONS
PKLMT reference voltage
PKLMT propagation delay
PKLMT to GT1
MIN
TYP
MAX
UNITS
–20
0
20
mV
150
300
500
ns
multiplier
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
IMOUT, high-line low-power output current
IMOUT, high-line high-power output current
IAC = 500 µA, VFF = 4.7 V,
IAC = 500 µA, VFF = 4.7 V,
VAOUT = 1.25 V
−3
–6
−9
VAOUT = 5 V
−75
–90
−110
IMOUT, low-line low-power output current
IMOUT, low-line high-power output current
IAC = 150 µA, VFF = 1.4 V,
IAC = 150 µA, VFF = 1.4 V,
VAOUT = 1.25 V
−10
–15
−50
VAOUT = 5 V
−245
–290
−330
IMOUT, IAC-limited output current
Gain constant (k)
IAC = 150 µA, VFF = 1.3 V,
IAC = 300 µA, VFF = 2.8 V,
VAOUT = 5 V
−245
–290
−330
0.8
1
1.2
1/V
IAC = 150 µA, VFF = 1.4 V,
IAC = 500 µA, VFF = 4.7 V,
VAOUT = 0.25 V
0
–0.2
µA
IMOUT, zero current
VAOUT = 0.25 V
0
–0.2
µA
IAC = 500 µA, VFF = 4.7 V,
IAC = 150 µA, VFF = 1.4 V,
VAOUT = 0.5 V
Power limit (IMOUT × VFF)
VAOUT = 2.5 V
VAOUT = 5 V
−343
µA
0
–0.2
µA
–406
−462
µW
1. Ensured by design. Not 100% tested in production.
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5
SLUS517− DECEMBER 2002
ELECTRICAL CHARACTERISTICS
TA = –40°C to 105°C for the UCC2851x, TA = TJ, VCC = 12 V, RT = 156 kΩ, RCT_BUFF = 10 kΩ
(unless otherwise noted)
zero power
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
Zero power comparator threshold
Measured on VAOUT,
falling edge
0.20
0.33
0.50
V
Zero power comparator hysteresis
Measured on VAOUT,
rising edge
40
90
140
mV
PFC gate driver
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
GT1 pull-up resistance
−100 mA ≤ ∆IOUT ≤ −200 mA
5
12
GT1 pull-down resistance
IOUT = 100 mA
2
10
16
25
7
15
93%
95%
100%
120
150
200
GT1 output rise time
RLOAD = 10 Ω
CLOAD = 1 nF,
GT1 output fall time
Maximum duty cycle
Minimum controllable pulse width
UNITS
Ω
ns
ns
PWM stage undervoltage lockout (UVLO2)
PARAMETER
PWM turn-on reference
PWM turn-off threshold
Hysteresis
TEST CONDITIONS
UCC2851X
MIN
TYP
6.30
6.75
UCC28510
UCC28511
UCC28514
UCC28515
5.3
UCC28512
UCC28513
UCC28516
UCC28517
3.55
MAX
UNITS
7.30
V
UCC28510
UCC28511
UCC28514
UCC28515
1.16
1.45
1.74
UCC28512
UCC28513
UCC28516
UCC28517
2.56
3.20
3.84
PWM stage soft-start
PARAMETER
SS2 charge current
SS2 discharge current
Input voltage (VERR)
TEST CONDITIONS
VSENSE = 7.5 V,
SS2 = 0 V
VSENSE = 2.5 V,
SS2 = 2.5 V,
(UVLO2 = Low, ENABLE = High)
MIN
TYP
MAX
UNITS
–7.0
–10.5
–14.0
µA
6
10
14
mA
300
mV
IVERR = 2 mA,UVLO2 = Low
PWM stage duty cycle clamp
PARAMETER
Maximum duty cycle
TEST CONDITIONS
D_MAX = 4.15 V
MIN
70%
TYP
75%
MAX
UNITS
80%
PWM stage pulse-by-pulse current sense
PARAMETER
Current sense comparator offset voltage
TEST CONDITIONS
ISENSE2 = 0 V,
measured on VERR
1. Ensured by design. Not 100% tested in production.
6
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MIN
1.35
TYP
1.50
MAX
1.65
UNITS
V
SLUS517− DECEMBER 2002
ELECTRICAL CHARACTERISTICS
TA = –40°C to 105°C for the UCC2851x, TA = TJ, VCC = 12 V, RT = 156 kΩ, RCT_BUFF = 10 kΩ
(unless otherwise noted)
PWM stage overcurrent limit
PARAMETER
TEST CONDITIONS
MIN
Peak current comparator threshold voltage
Input bias current(1)
1.15
TYP
MAX
1.30
1.45
50
UNITS
V
nA
PWM stage gate driver
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
GT2 pull-up resistance
−100 mA ≤ ∆IOUT ≤ −200 mA
5
12
GT2 pull-down resistance
IOUT = 100 mA
2
10
Ω
16
25
ns
7
15
ns
GT2 output rise time
RLOAD = 10 Ω
CLOAD = 1 nF,
GT2 output fall time
Ω
1. Ensured by design. Not 100% tested in production.
TERMINAL FUNCTIONS
TERMINAL
I/O
DESCRIPTION
PFC
O
Output of the current control amplifier of the PFC stage. CAOUT is internally connected
to the PWM comparator input in the PFC stage
5
PWM
O
Internally buffered PWM stage oscillator ramp output, typically used to program slope
compensation with a single resistor
D_MAX
4
PWM
I
Positive input to set the maximum duty cycle clamp level of the PWM stage
GND
6
−
−
Analog ground
GT1
12
PFC
O
PFC stage gate drive output
GT2
10
PWM
O
PWM stage gate drive output
IAC
18
PFC
I
Multiplier current input that is proportional to the instantaneous rectified line voltage
ISENSE1
16
PFC
I
Non-inverting input to the PFC stage current amplifier
ISENSE2
8
PWM
I
Input for PWM stage current sense and peak current limit
MOUT
17
PFC
I/O
PKLMT
14
PFC
I
Voltage input to the PFC peak current limit comparator
PWRGND
11
−
−
Power ground for GT1, GT2 and high current return paths
RT
2
−
I
Oscillator programming pin that is set with a single resistor to GND
SS2
13
PWM
I
Soft start for the PWM stage
VAOUT
1
PFC
I/O
VCC
9
−
I
Positive supply voltage pin
VERR
7
PWM
I
Feedback error voltage input for the PWM stage, typically connected to an optocoupler
output
VFF
19
PFC
I
Voltage feedforward pin for the PFC stage, sources an IAC/2 current that should be
externally filtered
VREF
20
−
O
Precision 7.5-V reference output
VSENSE
3
PFC
I
Inverting input to the PFC transconductance voltage amplifier, and input to the OVP,
ENABLE and UVLO2 comparators
NAME
NO.
Stage
CAOUT
15
CT_BUFF
PFC multiplier high−impedance current output, internally connected to the current amplifier inverting input
Output of the PFC transconductance voltage amplifier and it is internally connected to
the Zero Power Detect comparator input and the multiplier input
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7
SLUS517− DECEMBER 2002
BLOCK DIAGRAM
OSC
CLK1
1x:2x Option Only
CLK2
6.75 V
CT_BUFF
D_MAX
VERR
ISENSE2
5
4
7
8
VCC
SS2
13
9
UVLO2
+
PWM STAGE
SOFT START
ENABLE
1.9 V
7.5 V
REFERENCE
20 VREF
+
RT
2
3V
PFC:PWM
Frequency
1:1 = IRT
1:2 = 0.5IRT
UVLO
16 V, 9.7 V
10.2 V, 9.7 V
+
VCC
IRT
D_MAX
COMP
ILIMIT
PWM
10
1.3 V
1.5 V
R
R
S
+
+
CLK1
GT2
Q
PWM
+
CLK2
CLK2
PWM
PWM
8.0 V
PFC
VAOUT
3
PFC
ZERO
POWER
1
gM VOLTAGE
ERROR AMP
VSENSE
PFCOVP
+
0.33 V
+
VCC
X
÷ MULT
X
+
7.5 V
VFF
19
(VFF)2
CURRENT
AMP
MIRROR
2:1
IAC
18
MOUT
17
12
GT1
11
PWRGND
14
PKLMT
+
CLK1
+
PWM
R
R
S
Q
PFC
ILIMIT
+
16
15
6
ISENSE1
CAOUT
GND
DETAILED PIN DESCRIPTIONS
CAOUT (Pin 15): This is the output of a wide-bandwidth operational amplifier that senses line current and
commands the PFC stage PWM comparator to force the correct duty cycle. This output can swing close to GND
to command maximum duty cycle, and above the PFC ramp peak voltage to force zero duty cycle when
necessary. Connect current loop compensation components between CAOUT and MOUT.
CT_BUFF (Pin 5): The 4-V amplitude oscillator ramp is internally buffered at this pin to allow a resistor to be
connected directly from this pin to ISENSE2 for slope compensation. The internal buffer can drive a typical
500-µA resistive load at this pin.
D_MAX (Pin 4): Program the maximum duty cycle at GT2 by applying a dc voltage to this pin. Between 0.09
and 0.90, the maximum duty ratio is linearly related to D_MAX. Usually, this voltage is set with a precision
resistor divider powered by VREF. A first order approximation, with the CT_BUFF frequency near 200 kHz, is
estimated by:
D MAX ^
V DX * 1
4
where, DMAX is a dimensionless ratio
VDX is the voltage at D_MAX in volts
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DETAILED PIN DESCRIPTIONS (CONTINUED)
The maximum duty ratio is modestly dependent on the switching frequency. A more accurate estimate of the
maximum duty cycle that is valid over the full range of switching frequencies (65 kHz to 600 kHz) is given by:
ǒ
D MAX + 0.26 * ǒ4.4
10 *8
Ǔ
f SǓ V DX ) 6.9
10 *8
f S * 0.31
where, fS is the oscillator frequency measured at CT_BUFF in Hz
This pin can also be used to set DMAX to 0 by setting VDX less than 0.7 V.
GND (Pin 6): Signal ground for the integrated circuit. All voltages measured with respect to ground are
referenced to this pin. The bypass capacitors for VCC and VREF should connect to this pin with as little lead
length as possible. PWRGND must be externally connected to this pin. For best results, use a single small circuit
trace to electrically connect between the circuits that use the GND return path and the circuits that use the
PWRGND return path.
GT1 (Pin 12): A 2-A peak source and 3-A peak sink current totem pole MOSFET gate driver for the PFC stage.
Some overshoot at GT1 can be expected when driving a capacitive load, but adding a minimal series resistor
of about 2 Ω between GT1 and the external MOSFET gate can reduce this overshoot. GT1 is disabled unless
VCC is outside the UVLO region and VREF is on.
GT2 (Pin 10): A 2-A peak source and 3-A peak sink current totem pole MOSFET gate driver for the PWM stage,
identical to the driver at GT1.
IAC (Pin 18): This multiplier input senses the rectified ac line voltage. A resistor between IAC and the line
voltage converts the instantaneous line voltage waveform into a current input for the analog multiplier. The
recommended maximum IAC current is 500 µA.
ISENSE1 (Pin 16): This pin is the non-inverting input terminal of the current amplifier. Connect a resistor
between this pin and the grounded side of the PFC stage current sensing resistor. The resistor connected to
this pin should have a value that equals the value of the resistor that is connected between the MOUT pin and
the ungrounded side of the PFC current sense resistor.
ISENSE2 (Pin 8): A voltage across the PWM stage external current sense resistor generates the input signal
to this pin, with the peak limit threshold set to 1.3 V for peak current mode control. An internal 1.5-V level shift
between ISENSE2 and the input to the PWM comparator provides greater noise immunity. The oscillator ramp
can also be summed into this pin for slope compensation.
MOUT (Pin 17): The output of the multiplier and the input to the current amplifier in the PFC stage are internally
connected at this pin. Set the power range of the PFC stage with a resistor tied between the MOUT pin and the
non-grounded end of the PFC current sense resistor. Connect impedance between the MOUT pin and the
CAOUT pin to compensate the PFC current control loop. The multiplier output is a current and the current
amplifier input is high impedance. The multiplier output current is given by:
I MOUT +
ǒVVAOUT * 1.0Ǔ
K
ǒVVFFǓ
I IAC
2
where, K is the multiplier gain constant, in volts−1.
PKLMT (Pin 14): Program the peak current limit of the PFC stage using this pin. The threshold for peak limit
is 0 V. Use a resistor divider between VREF and the non-grounded side of the PFC current sense resistor in
order to shift the level of this signal to a voltage that corresponds to the desired overcurrent threshold voltage,
measured across the PFC current sense resistor.
PWRGND (Pin 11): Ground for the output drivers at GT1 and GT2. This ground should be tied to GND externally
via a single Kelvin connection.
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DETAILED PIN DESCRIPTIONS (CONTINUED)
RT (Pin 2): A resistor between RT and GND programs the oscillator frequency, measured at CT_BUFF. In all
options, the PWM stage operates at the frequency that is measured at CT_BUFF. In the UCC28510, UCC28511,
UCC28512 and UCC28513, the PFC stage operates at the same frequency as the PWM stage. In the
UCC28514, UCC28515, UCC28516 and UCC28517, the PFC stage operates at half the frequency of the PWM
stage. The voltage is dc (nominally 3 V); do not connect a capacitor to this pin in an attempt to stabilize the
voltage. Instead, connect the GND side of the oscillator-programming resistor closer to the GND pin. The
recommended range of resistors is 45 kΩ to 500 kΩ for a frequency range of 600 kHz to 65 kHz, respectively.
Resistor RT programs the oscillator frequency fS, as measured at CT_BUFF, according to the following
equation:
RT +
31
ǒ
1
1 * 2.0
10 *12 f S
Ǔ
10 *7
where, RT is in Ω
fS is in Hz
SS2 (pin 13): A capacitor between SS2 and GND programs the softstart duration of the PWM stage gate drive.
When the UVLO2 comparator enables the PWM stage, an internal 10.5-µA current source charges the external
capacitor at SS2 to 3 V to ramp the voltage at VERR during startup. This allows the GT2 duty cycle to increase
from 0% to the maximum clamped by the duty cycle comparator over a controlled time delay tSS given by:
C SS2 +
t SS
10.5
3
10 *6
,
Farads
In the event of a disable command or a UVLO2 dropout, SS2 quickly discharges to ground to disable the PWM
stage gate drive.
VAOUT (Pin 1): This transconductance voltage amplifier output regulates the PFC stage output voltage and
operates between GND and 5.5 V maximum to prevent overshoot. Connect the voltage compensation
components between VAOUT and GND. When this output goes below 1 V, the multiplier output current goes
to zero. When this output falls below 0.33 V, the zero power detect comparator ensures the PFC stage gate drive
is turned off. In the linear range, this pin sources or sinks up to 30 µA. A slew rate enhancement feature enables
VAOUT to sink or source up to 3.3 mA, when operating outside the linear range.
VCC (Pin 9): Chip positive supply voltage that should be connected to a stable source of at least 20 mA between
12 V and 17 V for normal operation. Bypass VCC directly to GND with a 0.1 µF or larger ceramic capacitor to
absorb supply current spikes caused by the fast charging of the external MOSFET gate capacitances.
VERR (Pin 7): The voltage at this pin controls the GT2 duty cycle and is connected to the feedback error signal
from an external amplifier in the PWM stage. This pin is clamped to a maximum of 3 V and can demand 100%
duty cycle at GT2. The typical pull-up current flowing out of this pin is 10 µA.
VFF (Pin 19): The output current from this pin comes from an internal current mirror that divides the IAC input
current by 2. The input voltage feedforward signal for the multiplier is then generated across an external
single-pole R/C filter connected between VFF and GND. At low line, the VFF voltage should be set to 1.4 V.
VREF (Pin 20): This is the output of an accurate 7.5-V reference that powers most of the internal circuitry and
can deliver over 10 mA, with a typical load regulation of 5 mV ensured for an external load of up to 6 mA. The
internal reference is current limited to 25 mA, which protects the part if VREF is short-circuited to ground. VREF
should be bypassed directly to GND with a ceramic capacitor between 0.1 µF and 10 µF for stability. VREF is
disabled and remains at 0 V when VCC is below the 9.7-V UVLO threshold.
VSENSE (Pin 3): Inverting input to the PFC transconductance voltage amplifier, which serves as the PFC
feedback connection point. When VSENSE operates within +/− 0.35 V of its steady-state value, the current at
VAOUT is proportional to the difference between the VREF and VSENSE voltages by a factor of gM. Outside
this range, the magnitude of the current of VAOUT is increased in order to enhance the slew rate for rapid voltage
control recovery in the PFC stage. Decisive activation and deactivation of the voltage control recovery is
internally implemented with about 120 mV of hysteresis at VSENSE. VSENSE is internally connected to the
OVP, Enable and UVLO2 comparators as well.
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APPLICATION INFORMATION
D2
L1
D3
PRIMARY
SECONDARY
+
R3
R6
T1
D4
C2
+
D1
VOUT
C1
R1
–
Q1
−
Q2
R4
R2
R5
GND2
VAC
PGND
R9
R8
PGND
C5
AGND
C6
R13
C7
R14
12 GT1
VCC
9
13 SS2
ISENSE2
8
14 PKLMT
VERR
7
15 CAOUT
GND
6
16 ISENSE1 CT_BUFF
5
D_MAX
4
18 IAC
VSENSE
3
19 VFF
RT
2
VAOUT
1
17 MOUT
20 VREF
R15
D5
C3
AGND
UCC2851X
GT2 10
11 PWRGND
R7
R12
R11
R10
U1
C8
REF
C9
REF
R16
C4
R17
R22
D6
R18
GND2
C14
AGND PGND
R19
R20
C10
U2
R23
C13
R25
R26
C12
C11
U3
TL431
R21
AGND
R24
GND2
Figure 1. Typical Application Circuit: Boost PFC and Flyback PWM Power System
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APPLICATION INFORMATION
The UCC28510 series of combination controllers include a power factor correction (PFC) controller that is
synchronized with a pulse width modulator (PWM) controller integrated into one chip. The PFC controller has
all of the features for an average current mode controlled PFC. The PWM controller has all of the features for
an isolated peak current program mode controlled converter. The two controllers are synchronized at a fixed
frequency so that the PFC controller is leading edge modulated (LEM) and the PWM controller is trailing edge
modulated (TEM). The LEM/TEM combination reduces the ripple current in the energy storage capacitor of the
PFC stage. A comparison between the ripple current in the energy storage capacitor with traditional TEM/TEM
modulation versus LEM/TEM modulation is shown in Figure 2.
PFC BOOST CONVERTER
i IN
i D1
L1
i Q2
D1
+
i ES
D AC
BUCK DERIVED CONVERTER
iL
Q2
L2
C ES
Q1
D2
LOAD
C OUT
–
VAC
T
T
ON
OFF
Q1
OFF
ON
i D1
ON
OFF
Q2
ON
OFF
i Q2
i ES
i ES = i D1 − iQ2
TEM/TEM
LEM/TEM
Figure 2. Equivalent PFC+PWM power supply system and the comparison of the energy storage
capacitor current for traditional TEM/TEM with LEM/TEM controllers.
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APPLICATION INFORMATION
selection of controller options
The UCC2851x is optimized for the most common combination of PFC/PWM stages, which is a boost PFC stage
cascaded by a buck-derived PWM stage. Other topology combinations can be used with this controller, as well.
The programmable PWM duty ratio limit feature is especially useful when using two-transistor forward and
flyback topologies for the PWM stage. The PFC boost stage is typically designed for continuous conduction
mode (CCM) of operation at full rated load in order to minimize line filter requirements. The PWM stage can be
designed for either continuous or discontinuous mode operation, as necessary.
Eight different options are available for the UCC2851x. This device is available in two PFC under voltage lock
out (UVLO) turn-on thresholds, two PWM UVLO hysteresis levels and, two combinations of PFC/PWM
switching frequencies as shown in Table 1.
Table 1. Available Options
PFC UVLO TURN−
ON THRESHOLD
(V)
16
10.2
16
10.2
PFC:PWM
FREQUENCY RATIO
PWM
HYSTERESIS
(V)
1.45
3.20
1:1
1:2
UCC28510
UCC28514
UCC28511
UCC28515
UCC28512
UCC28516
UCC28513
UCC28517
Select the PFC UVLO option first, based on biasing topology. Then, select the PFC versus PWM switching
frequency based on the allowable switching loss of the intended PWM stage. Last, select the PWM UVLO option
based on bulk ripple voltage and load transients.
The PFC UVLO turn-on threshold is selected based on line range, bias supply topology and gate drive voltage
requirements. The 16-V turn-on options are intended for applications where the bias voltage is self-generated
from an auxiliary winding, with little or no regulation. The 10.2-V turn-on / 0.5-V hysteresis options are intended
for applications where the bias voltage is derived from an auxiliary supply source and is regulated.
The PWM UVLO hysteresis level option is selected based on the desired operational range of the energy
storage capacitor voltage. A narrow range permits a highly optimized PWM stage. However, the wider range
permits larger energy storage capacitor voltage ripple and load transients.
Two options are available for the PFC:PWM switching frequency, 1:1 and 1:2. Both versions are synchronized
as LEM/TEM oscillators. The best minimization of the energy storage capacitor ripple current occurs with the
1:1 option. However, the diode in the PFC stage often has high reverse recovery currents that restrict the
switching frequency of the PFC stage. Situations where the switching losses of the PWM stage permit higher
switching frequencies can benefit from the 1:2 option. For example, the 1:2 option would be a good choice for
PWM stages that have Schottky diode output rectifiers. The energy storage capacitor ripple current for a system
that is controlled by the 1:2 option will be larger than if it were controlled by a 1:1 option. However the capacitor
current of the 1:2 option is less than a system that is TEM/TEM modulated.
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APPLICATION INFORMATION
design procedure
The following discussion steps through the typical design process of a PFC/PWM converter system that is
controlled by one of the UCC28510 options. The design process begins with the power stage elements, then
the control elements for the PFC stage, then the control elements for the PWM stage. Keep in mind that a
general design process is often iterative. Iteration typically begins after either simulating and/or testing the
completed PFC/PWM system. This design procedure refers to the typical application in Figure 1.
A design begins with a list of requirements for output voltage, output power and ac line voltage range. Other
details, such as efficiency and permissible current harmonics could be given at the onset, or developed
throughout the product design cycle. The need for power factor correction arises from either an agency
requirement, such as IEC−61000, or if the available line power is nearly equal to the output power of the power
system. Hold-up time requirements are also necessary at the early stages of design. Typically, the hold-up time,
tHU, is at least the period of 1.5 line cycles.
The general structure of the PFC/PWM stage power system is two switched-mode converters connected in
cascade. Each stage has an associated efficiency and each stage has its own set of fault limiting controls that
must be properly set in order to achieve the desired line harmonic and load regulation performance,
simultaneously. The PFC stage must always be designed to supply sufficient average power to the PWM stage.
The cycle-by-cycle current limit of the PFC stage should be programmed to activate at a slightly larger power
level at low ac line voltage than the average power clamp in order to allow for PFC current sense tolerances.
This will allow power factor correction for the full range of maximum rated load. If the instantaneous load nearly
equals the average load, then the fault clamps for the PWM stage can be programmed to limit power at a level
that is slightly less than or equal to the average power clamp of the PFC stage. The margin for the clamping
action should allow for measurement tolerances and efficiency. Conversely, if the instantaneous load has high
peaks that are much shorter than the hold-up time, the current limit and duty ratio limits of the PWM stage can
clamp at a higher level than the average power clamp in the PFC stage. In order to simplify the design procedure,
the average and the peak loads of the PWM stage are assumed to be equal. Thus, all of the current limits and
duty cycle limits are programmed to clamp power at a slightly lower level (10%) than the average power clamp
on the PFC stage.
developing the internal parameters
Select the energy storage voltage VC1. Since the PFC stage is a boost converter, the voltage across C1 must
be larger than the peak ac line voltage by enough to permit controllability in the event of load transients. Typically,
this will be around 5% which is about 400 V for a universal ac line application of 85 VAC to 265 VAC.
Once the energy storage voltage, VC1, is determined, the range of the PFC stage duty ratio, D1, is set. For CCM
operation of the PFC stage, the minimum PFC duty ratio is given by:
D 1(min) + 1 *
14
Ǹ2
VAC MIN
V C1
(1)
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APPLICATION INFORMATION
Select the regulation constant, k1R, of the energy storage voltage, as described by equation 2.
k 1R +
^
V REF * (PWM turn−on reference) ) (PWM hysteresis)
V REF
(2)
for UCC28510, UCC28511, UCC28514, UCC28515
ƪ0.29
ƫ
0.53 for UCC28512, UCC28513, UCC28516, UCC28517
where,
the nominal PWM turn-on reference = 6.75 V
the nominal PWM hysteresis is = 1.45 V or 3.2 V
There are effectively two options for k1R that directly relate to the two PWM hysteresis options, k1R = 0.29 and
k1R = 0.53. Select the large PWM hysteresis option if the system load has large, sudden step changes during
steady state operation. Select the small PWM hysteresis option if the system load has moderate step changes
or slow load changes during steady state operation. The PWM stage can be optimized best with the small PWM
hysteresis range because the maximum primary current of transformer T1 (which occurs at minimum VC1) is
smallest with the small PWM hysteresis range.
Select an approximate switching frequency for the PFC stage. A good starting frequency for a MOSFET based
PFC stage is in the range of 100 kHz to 200 kHz, depending on maximum line voltage and maximum line current.
Adjustments in switching frequency may result from meeting switching loss requirements in Q1 and D3, or in
order to optimize the design of L1.
Select an appropriate topology for the PWM stage using the information about the power requirements and the
magnitude of VC1. For simplicity, the typical application in Figure 1 shows a flyback converter in the PWM stage.
In most cases, the PWM stage topology must have transformer isolation and the topology must require only one
pulse-width signal. Topologies that have these features include:
•
•
•
•
single-transistor forward converter
single-transistor flyback converter
two-transistor forward converter
two-transistor flyback converter
Estimate the nominal and the maximum duty ratios of the PWM stage (D2(nom) , D2(max) and the associated peak
Q2 drain current, iQ2(peak)), based on the topology, PWM hysteresis option and output voltage requirements of
the PWM stage. Also estimate whether or not it is appropriate to operate the PWM stage at the same switching
frequency as the PFC stage or if the PWM stage can operate at twice the switching frequency of the PFC stage.
Base the estimation for the switching frequency of the PWM stage on the maximum voltages and currents of
the power MOSFETs and power diodes. Program the oscillator frequency of the PWM stage with the value of
R20.
R20 +
31
ǒ
1
1 * 2.0
10 *12 f S(pwm)
Ǔ
10 *7 ,
W
(3)
Most applications require that the PWM stage regulates at the minimum energy storage capacitance voltage.
Maximum duty ratio D2(max) and iQ2(peak) should be calculated for the minimum energy storage voltage in order
to estimate the peak current stresses for transformer T1 and any other inductive element in the PWM stage.
V C1(min) + ǒ1 * k 1RǓ
V C1(nom)
(4)
At this point, enough information is available to estimate which member of the UCC28510 family should be
selected.
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APPLICATION INFORMATION
power stage elements
The power stage elements include the following elements: C1−3, D1−5, L1, R2, R5, Q1, Q2, T1. Details
concerning the PWM stage elements C2, D4, D5, Q2 and T1 will not be discussed in detail here, due to their
dependence on the choice PWM stage topology. The PWM stage is an isolated dc-to-dc topology with the same
stresses and loss mechanisms that are typical for the selected topology. An estimation of the average steady
state duty ratio of the PWM stage and the Q2 switch current will be needed for stress estimations in the PFC
stage. Also, the natural step response of the PWM stage is required to estimate the soft start capacitor, C5, and
the bias supply capacitor, C3.
The selection process of the PFC stage elements C1, C3, D1−3 and Q1 are discussed in detail here. In general,
the selection process for the PFC stage elements is the same as for a typical fixed switching frequency PFC
design, except for capacitor C1 due to PFC/PWM stage synchronization.
Diode bridge D1 is selected to withstand the rms line current and the peak ac line voltage. Diode D2 allows
capacitor C1 to charge during initial power up without saturating L1 and it is selected to withstand the peak inrush
current and peak of the maximum ac line voltage. Additional inrush current limiting circuitry in series with the
ac line could be required, depending on agencies or situations.
The PFC stage inductor, L1, is selected to have a maximum current ripple at the minimum ac line voltage.
Typically a ripple factor, kRF, is chosen to be about 0.2. If the line current has excessive crossover distortion,
a larger ripple factor (perhaps 0.3) will reduce the distortion but the line current will have more switching ripple.
Initially, the inductance can be estimated by approximating the input power equal to the output power.
L1 +
V AC(min)
where, k RF +
2
D 1(min)
k RF
T S(pfc)
P IN
(5)
Di L1(p−p)
i L1(max)
Inductor L1 must be designed to withstand the maximum ac rms line current without saturation at the peak ac
line current.
Select power MOSFET Q1 and diode D3 with the same criteria that is normally used for fixed switching
frequency PFC design. They must have sufficient voltage rating to withstand the energy storage voltage, VC1
and they must have sufficient current ratings. Gate drive resistor R9 is necessary to limit the source and sink
current from the GT1 pin. Some circumstances require additional gate drive components for improved
protection and performance.[10] A similar gate drive resistor, R10, is required between the GT2 pin and the gate
of Q2 for the same reason.
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APPLICATION INFORMATION
The current sense resistor for the PFC stage, R2, is selected to operate over a 1-V dynamic range (VDYNAMIC).
The sense resistor must also have a large enough power rating to permit safe operation with the maximum RMS
line current.
R2 +
V DYNAMIC
i L1(max) ) 0.5 Di L1(p−p)
where, i L1(max) +
(6)
Ǹ2 P
IN
V AC(min)
The PFC ILIMIT comparator threshold is at the ground reference for the controller device. So, the PFC current
sense voltage, measured at PKLMT must be biased with a positive voltage to cross 0.0 V when the
instantaneous PFC current is at its maximum. The bias voltage is established with R14 and R7, as shown in
equation 7, and resistor R14 is arbitrarily chosen around 10 kΩ.
R7 +
R14
1
VREF
*1
iL1(max) R2
(7)
The capacitance value of the energy storage capacitor, C1 is selected to meet hold-up time requirements (tHU)
by the equation:
C1 +
2
V C1
P OUT
2
t HU
k R1ǒ2 * k R1Ǔ
(8)
Capacitor C1 must be rated for the selected energy storage voltage and it must be able to withstand the rms
ripple current, IC1(rms), that is produced by the combined action of the PFC stage and the PWM stage. The
average Q2 drain current during the interval that GT2 activates MOSFET Q2 is used to find IC1(rms). An initial
estimate can be made using the inequality in equation 9, then consult Figure 3 or Figure 4 for better accuracy.
I C1(rms)
I Q2
t
Ǹ
8
Ǹ2
3
D 2(nom)
p
2
V C1(nom)
V AC(min)
) D 2(nom)
(9)
The ratio of IC1(rms) to IQ2 can be found by using the appropriate graph, Figure 3 for the 1X:1X oscillator option
or Figure 4 for the 1X:2X oscillator option. To use the graphs, locate the ratio of VAC to VC1 along the horizontal
axis then, draw a vertical line to the intersection of the curve for the duty ratio of the PWM stage. Draw a
horizontal line from the intersection to the vertical axis and read the ratio of IC1(rms) to IQ2.
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APPLICATION INFORMATION
IC1(rms)/IQ2
vs
VAC, fPFC:fPWM = 1:1
1.6
DPWM = 0.7
1.4
DPWM = 0.6
1.2
1.0
DPWM = 0.5
0.8
DPWM = 0.4
0.6
DPWM = 0.3
0.4
0.2
0.0
0.0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
VAC/VC1
Figure 3. Graph for Finding IC1(rms) for the 1X:1X Oscillator Option
IC1(rms)/IQ2
vs
VAC, fPFC:fPWM = 1:2
2.0
1.8
1.6
1.4
DPWM = 0.7
1.2
DPWM = 0.6
1.0
DPWM = 0.5
0.8
DPWM = 0.4
0.6
DPWM = 0.3
0.4
0.2
0.0
0.0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
VAC/VC1
Figure 4. Graph for Finding IC1(rms) for the 1X:2X Oscillator Option
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APPLICATION INFORMATION
The current sense resistor for the PWM stage, R5, is selected so that at maximum current, its voltage is the
threshold voltage of the peak current comparator (nominally 1.3 V).
R5 +
V THǒPWM stage
I LIMITǓ
i Q2(peak)
(10)
In many cases, an input line filter will be necessary in order to meet the requirements of an agency or application.
The input line filter design has been omitted from this procedure due to the vast array of requirements and
circumstances. We urge you to refer to Reference [11] for details.
PFC stage control
The PFC stage is designed in a three-step process. First, set the dynamic range of the multiplier, second,
stabilize the average current control loop and third, stabilize the voltage loop that controls the energy storage
capacitor voltage. Use as much of the dynamic range of the multiplier as possible. The current control loop must
have wide bandwidth in order to follow the instantaneous rectified line voltage. The voltage loop must be slower
than twice the ac line frequency so that it will not compromise the power factor.
multiplier
The dynamic range of the multiplier is a function of the currents and/or voltages of the IAC, VAOUT and VFF
pins. Coordinate the selection process to use the full range of the multiplier and obtain the desired power limiting
features. Select the components R1 and R15 to use the iIAC(t) range and the VVFF range under the condition
that the maximum of the VVAOUT range, described in equation 11. The selection process is similar to the
selection process for UC3854, except for the VFF voltage and MOUT current limitations.[12] In this product
series, the divide-by-square function is internally implemented so that it divides by the greater of 1.4 V or VVFF.
If the 1.4-V level controls the divider, power factor correction may still occur if the VAOUT level is within the
functioning range of the multiplier. Power factor correction occurs during that condition because the multiplier
section functions as a two-input multiplier, rather than a three-input multiplier. Notice that the voltage at the VFF
pin will be proportional to the average of the IAC current. Typically, VVFF=1.4 V at low ac line voltage is set as
the design boundary; the upper boundary of VVFF will remain within the range if the functional ac line voltage
range varies by less than 4.3:1.
0 v i IAC(t) v 500 mA,
(11)
0 v V VAOUT(t) v 5 V,
1.4 V v V VFF v V VREF * 1.4 V
www.ti.com
19
SLUS517− DECEMBER 2002
APPLICATION INFORMATION
The selection process begins with the selection of R1 so that the peak IAC current at high ac line is about 500 µA,
see Table 2. Second, select R15 for the minimum VFF voltage, also shown in Table 2. Third, select C8, in Table
2, to average the VFF voltage with sufficiently low ripple to meet a third harmonic distortion budget. For a system
with a 3% THD target, it is typical to allow the feedforward circuit to contribute 1.5% third harmonic distortion
to the input waveform [4]. An attenuation factor of 0.022 will meet the criteria. Finally, select the MOUT resistor
in Table 2, R12, so that the voltage across R12 equals the voltage across sense resistor R2 under the condition
of maximum power, minimum ac line voltage (VVFF, MIN), and VAOUT at its maximum level of 5 V. Experimentally,
the multiplier output resistor, R12, may need to be increased slightly if the energy storage capacitor voltage sags
under maximum load. This would be due to tolerances in the components and the multiplier. In order to minimize
current amplifier offsets, set the value of the resistor on the ISENSE1 pin, R8, equal to the value of R12 as shown
in in Table 2.
Table 2.
REFERENCE
DESIGNATOR
EQUATION
Ǹ2 V
R1
I
R15
AC(max)
p
P
V
R8
V
p
Ǹ2
4
R12
set i
IAC(peak)
+ 500 mA
IAC(peak)
R1
C8
NOTES
IN
VFF(avgmin)
V
AC(min)
f
AC
R15
A
FF(2)
(R1)(R2)k
VAOUT(max)
set V VFF(avgmin) + 1.4 V
*1
ǒ
A
V
VFF(min)
V
AC(min)
Ǔ
2
FF(2)
+ 0.022 for 3% THD
k=1
VVFF(min) = 1.4V
VVAOUT(max) = 5.0V
Always change R8 if R12 is changed
R12
PFC current loop control
This controller uses average current loop control for the PFC stage. The current control loop must typically be
fast enough to track the rectified sinusoidal ac line voltage. There are many ways to design a controller that will
stabilize the PFC current loop. The method that is described here achieves good results for most applications.[5]
This method assumes that both the natural frequency of the system and the zero of the linearized boost PFC
are much lower than both the switching frequency and the desired crossover frequency, fCO(pfc), as described
in equation 12. The left side of the inequality in equation 12 will usually be true since the capacitance of C1 is
quite large.
1 * D PWM(min)
ǸL1
20
C1
2P IN
and
C1
V C1
2
tt 2
p
f CO(pfc) tt 2
p
f S(pfc)
(12)
www.ti.com
SLUS517− DECEMBER 2002
APPLICATION INFORMATION
The left side of the inequality should be at least a factor of 10 lower than the middle term; the right side of the
inequality should be at least five times larger than the middle term. For the purposes of 50 Hz to 60 Hz power
factor correction, good results can be achieved with the crossover frequency set to about 10 kHz. A lower
crossover might be necessary if the switching frequency of the PWM stage is below 100 kHz, or if the
compensator gain at the crossover frequency is large (over ~40 dB).
Upon selecting the crossover frequency, select R13 to set the gain at the crossover frequency, then select C6
to place a zero at the crossover frequency and select C7 to provide a pole at half of the switching frequency.
The equations are in Table 3.
Table 3.
REFERENCE
DESIGNATOR
EQUATION
2
R13
p
f
R12
C6
R13
C7
p
CO(pfc)
L1
V
2
1
p
f
NOTES
V
CT_BUFF(p−p)
V
CT_BUFF(p−p)
+4V
C1
CO(pfc)
1
f
S(pfc)
PFC voltage loop
The voltage loop must crossover at a lower frequency than twice the ac line frequency so that voltage
corrections will not interfere with power factor correction. Second harmonic ripple from the sensed VC1 voltage
directly results in third harmonic distortion on the ac line, similar to ripple on the VFF voltage.
PWM stage control
The control elements of the PWM stage are the same as a typical isolated current program mode converter.
The secondary elements include C12 to C14, D6, R22 to R25, U2 and U3, which perform the error amplifier,
compensation and isolation functions. On the primary side, VERR is connected to the node between the
opto-isolator output, U2, and a pull-up resistor, R17. Resistor R17 represents the gain in the conversion from
the output current of opto-isolator U2 and the VERR input.
Slope compensation is programmed using resistors R18 and R11, which form a summing node at ISENSE2.
The voltage at CT_BUFF is a saw-tooth waveform that swings between 1 V and 5 V.
Many applications require a duty ratio limit for the PWM stage in order to prevent transformer saturation.
Program the maximum duty ratio using the following ratio of resistors R16 to R19.
V VREF
R16 +
*1
R19
1 ) 4 D PWM(max)
(13)
Soft-start
The soft-start capacitor, C5, which is connected to SS2, controls the soft-start ramp of the PWM stage. The
soft-start ramp begins when the VSENSE voltage exceeds 6.75 V. In order to avoid loop saturation, the soft-start
ramp rate must be less than or equal to the open loop response of the PWM stage converter.
www.ti.com
21
SLUS517− DECEMBER 2002
REFERENCE DESIGN
Universal line input 100-W PFC output with 12 V, 8-W bias rail supply design is discussed in UCC28517EVM,
TI literature number SLUU117. The schematic is shown in Figures 5, 6, 7. Please refer to the SLUU117
document on http://www.ti.com for further details.
+
Figure 5. Section A
+
Figure 6. Section B
22
www.ti.com
SLUS517− DECEMBER 2002
REFERENCE DESIGN
Note: D10 and D9 are Schottky diodes
from Vishay, part no. BYS10−25
Figure 7. Section C
www.ti.com
23
SLUS517− DECEMBER 2002
TYPICAL CHARACTERISTICS
STARTUP CURRENT
vs
TEMPERATURE
SUPPLY CURRENT
vs
TEMPERATURE
5.0
4.8
UCC 28510/12/14/16
130
ICC − Supply Current − mA
ICC(off) − Startup Current − µA
140
120
110
100
UCC 28511/13/15/17
4.6
4.4
4.2
4.0
3.8
3.6
3.4
90
3.2
3.0
80
−50
−25
0
25
50
75
100
125
−50
−25
Temperature − °C
50
75
100
125
Figure 8
Figure 9
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
REFERENCE VOLTAGE
vs
TEMPERATURE
7.60
10.2 V UVLO
Turn-On Threshold
4.0
7.58
VREF − Reference Voltage − V
3.5
ICC − Supply Current − mA
25
Temperature − °C
4.5
3.0
2.5
2.0
16 V UVLO
Turn-On Threshold
1.5
1.0
7.56
7.54
7.52
7.50
7.48
7.46
7.44
0.5
7.42
0.0
7.40
4.0
6.0
8.0
10.0
12.0
14.0
16.0
18.0
−50
−25
0
25
50
Temperature − °C
VCC − Supply Voltage − V
Figure 11
Figure 10
24
0
www.ti.com
75
100
125
SLUS517− DECEMBER 2002
TYPICAL CHARACTERISTICS
VREF
vs
LOAD CURRENT
VREF CURRENT LIMIT
7.500
7.0
7.495
6.0
7.490
5.0
VREF − V
VCC = 10 V
VREF − Reference Voltage − V
8.0
7.485
4.0
7.480
3.0
VCC = 12 V
7.475
2.0
7.470
1.0
7.465
0.0
7.460
0
10
20
30
40
VCC = 15 V
0.0
VREF − External Load Current − mA
5.0
10.0
15.0
20.0
IREF − External Load Current − mA
Figure 12
Figure 13
PFC UVLO THRESHOLDS
vs
TEMPERATURE (UCC28510/2/4/6)
PFC UVLO THRESHOLDS
vs
TEMPERATURE (UCC28511/3/5/7)
12
18
UVLO On
UVLO On
16
10
UVLO Threshold − V
UVLO Threshold − V
14
12
UVLO Off
10
8
UVLO Hysteresis
6
UVLO Off
8
6
4
4
2
UVLO Hysteresis
2
0
0
−50
−25
0
25
50
75
100
125
−50
−25
0
25
50
Temperature − °C
Temperature − °C
Figure 14
Figure 15
www.ti.com
75
100
125
25
SLUS517− DECEMBER 2002
TYPICAL CHARACTERISTICS
PWM UVLO2 THRESHOLDS
vs
TEMPERATURE (UCC28512/3/6/7)
PWM UVLO2 THRESHOLDS
vs
TEMPERATURE (UCC28510/1/4/5)
8
8
UVLO2 On
7
6
UVLO On
6
UVLO2 Off
UVLO2 Threshold − V
UVLO2 Threshold − V
7
5
4
3
5
4
UVLO Off
3
UVLO Hysteresis
2
2
UVLO2 Hysteresis
1
1
0
0
−50
−25
0
25
50
75
100
−50
125
−25
0
25
50
Temperature − °C
Temperature − °C
Figure 16
Figure 17
75
100
125
OSCILLATOR FREQUENCY
vs
RT OVER VCC (11 V TO 15 V) (−40°C TO 105°C)
OSCILLATOR FREQUENCY
vs
RT
800
1000
f - Oscillator Frequency − kHz
f - Oscillator Frequency − kHz
700
100
600
500
400
300
200
100
0
0
10
100
1000
100
RT − Timing Resistor − kΩ
RT − Timing Resistor − kΩ
Figure 18
26
10
Figure 19
www.ti.com
1000
SLUS517− DECEMBER 2002
TYPICAL CHARACTERISTICS
GT1 MAXIMUM DUTY CYCLE
vs
PFC SWITCHING FREQUENCY (CAOUT = 0.85 V)
GT1, GT2 PULL-UP, PULL-DOWN RESISTANCE
vs
TEMPERATURE
8
100
RDS(on) − Gate Resistance − Ω
GT1 Maximum Duty Cycle − %
7
99
98
97
96
95
6
Pull-up
5
4
3
2
Pull-down
1
0
94
0
100
300
200
400
−50
500
−25
0
50
75
100
125
Temperature − °C
GT1 Switching Frequency − kHz
Figure 20
Figure 21
GT1 RISE/FALL TIME
vs
CLOAD AND RSERIES (VCC = 12 V)
GT1, GT2 RISE AND FALL TIMES
vs
TEMPERATURE
18
45
tR: RSERIES = 2 Ω
35
16
tR, tF − Rise and Fall Time − ns
40
tR, tF − Rise and Fall Time − ns
25
tF: RSERIES = 2 Ω
30
25
tR: RSERIES = 10 Ω
20
tF: RSERIES = 10 Ω
12
tR
14
12
10
8
6
10
4
5
2
0
0
0
2
4
tF
6
−50
−25
0
25
50
75
100
125
Temperature − °C
CLOAD − nF
Figure 22
Figure 23
www.ti.com
27
SLUS517− DECEMBER 2002
TYPICAL CHARACTERISTICS
GT2 MAXIMUM DUTY CYCLE
vs
D_MAX VOLTAGE
MULTIPLIER OUTPUT CURRENT
vs
VOLTAGE ERROR AMPLIFIER OUTPUT
100
350
100 kHz
80
IMOUT − Multiplier Output Current − µA
GT2 Maximum Duty Cycle − %
90
200 kHz
70
500 kHz
60
50
800 kHz
40
30
20
300
IAC = 150 µA
VFF = 1.4 V
350
IAC = 300 µA
VFF = 2.8 V
200
150
100
IAC = 500 µA
VFF = 4.7 V
50
10
0
0
1.0
1.5
2
2.5
3.0
3.5
4.0
4.5
5.0
5.5
1.0
D_MAX Voltage − V
4.0
2.0
VAOUT − Voltage Error Amplifier Output − V
Figure 24
Figure 25
MULTIPLIER
CONSTANT POWER PERFORMANCE
MULTIPLIER GAIN
vs
VOLTAGE ERROR AMPLIFIER OUTPUT
500
2.0
VAOUT = 5 V
450
1.8
400
IMOUT x VFF − µW
Multiplier Gain − K
6.0
1.6
1.4
IAC = 150 µA, VFF = 1.4 V
1.2
VAOUT = 4 V
350
300
350
VAOUT = 3 V
200
150
VAOUT = 2 V
IAC = 300 µA, VFF = 2.8 V
1.0
50
IAC = 500 µA, VFF = 4.7 V
0
0.8
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
5.5
VAOUT − Voltage Error Amplifier Output − V
0
1
2
3
VFF − Feedforward Voltage − V
Figure 26
28
100
Figure 27
www.ti.com
4
5
SLUS517− DECEMBER 2002
TYPICAL CHARACTERISTICS
CURRENT AMPLIFIER OPEN LOOP
GAIN AND PHASE
120
150
gm − VEA Transconductance − µs
120
Gain − dB, Phase − Degree
Phase
90
60
Gain
30
0
−30
VOLTAGE AMPLIFIER TRANSCONDUCTANCE
vs
TEMPERATURE
100
80
60
40
20
−60
0
−90
104
102
1
106
108
−50
50
Figure 28
Figure 29
40
3
30
2
1
0
−1
−2
0
−10
−20
−4
−40
7.8
125
10
−30
7.6
100
20
−3
7.4
75
VOLTAGE AMPLIFIER OUTPUT CURRENT IN
LINEAR REGION OF OPERATION
VEA − Output Current − µA
Voltage Amplifier Output Current − mA
25
Temperature − °C
4
7.2
0
Frequency − Hz
VOLTAGE AMPLIFIER
OUTPUT CURRENT CAPABILITY
7.0
−25
8.0
VSENSE − Voltage Normalized to VREF − V
Figure 30
7.0
7.2
7.4
7.6
7.8
VSENSE − Voltage Normalized to VREF − V
8.0
Figure 31
www.ti.com
29
SLUS517− DECEMBER 2002
TYPICAL CHARACTERISTICS
VOLTAGE AMPLIFIER VSENSE BIAS CURRENT
vs
TEMPERATURE
VOLTAGE AMPLIFIER
OPEN LOOP GAIN AND PHASE
300
180
VAOUT Load = 15 pF
Gain − dB, Phase − Degree
IIB − VEA VSENSE Bias Current − nA
Phase
150
120
90
Gain
60
30
0
250
200
150
100
50
0
−30
10
107
105
103
−50
−25
25
50
Frequency − Hz
Temperature − °C
Figure 32
Figure 33
VOLTAGE AMPLIFIER SLEW CURRENTS
vs
TEMPERATURE
75
100
125
100
125
SOFTSTART CURRENTS
vs
TEMPERATURE
15
5
4
IDISCHG (mA)
3
ICHG − µA, IDISCHG −mA
VEA ISOURCE, ISINK − mA
0
ISINK
2
1
0
−1
10
5
0
−5
−2
ICHARGE (µA)
−3
ISOURCE
−10
−4
−15
−5
−50
−25
0
25
50
75
100
125
Temperature − °C
−25
0
25
50
Temperature − °C
Figure 34
30
−50
Figure 35
www.ti.com
75
SLUS517− DECEMBER 2002
REFERENCES
1. Evaluation Module and associated User’s Guide, UCC28517EVM, Texas Instruments Literature Number
SLUS419C
2. Datasheet, UCC38500/1/2/3 BiCMOS PFC/PWM Combination Controller, Texas Instruments Literature
Number SLUS419C
3. Power Supply Seminar SEM−600, High Power Factor Preregulator for Off-line Power Supplies, L.H. Dixon,
Texas Instruments Literature Number SLUP087
4. Power Supply Seminar SEM−700, Optimizing the Design of a High Power Factor Switching Preregulator,
L.H. Dixon, Texas Instruments Literature Number SLUP093
5. Power Supply Seminar SEM−1500 Topic 2, Designing High-Power Factor Off−Line Power Supplies, by
James P. Noon
6. Application Note, UC3854 Controlled Power Factor Correction Circuit Design ,Texas Instruments Literature
Number SLUA144
7. Design Note, Optimizing Performance in UC3854 Power Factor Correction, Texas Instruments Literature
Number SLUA172
8. Design Note, UC3854A and UC3854B Advanced Power Factor Correction Control ICs, Texas Instruments
Literature Number SLUA177
9. Design Note, UC3854A/B and UC3855A/B Provide Power Limiting with Sinusoidal Input Current for PFC
Front Ends, Texas Instruments Literature Number SLUA196
10. Laszlo Balogh, A Design and Application Guide for High Speed Power MOSFET Gate Drive Circuits, 2001
Power Supply Design Seminar Manual SEM1400, 2001
11. Bob Mammano and Bruce Carsten, Understanding and Optimizing Electromagnetic Compatibility in
Switchmode Power Supplies, 2002 Power Supply Design Seminar Manual SEM1500, 2002
RELATED PRODUCTS
PART NUMBER
DESCRIPTION
COMMENTS
UCC38500/1/2/3
BiCMOS PFC/PWM combination controller
1:1 leading edge, trailing edge modulation, 50% PWM Max dc
UCC3817/18
BiCMOS power factor preregulator
High PF, UC3854 compatible, leading edge trailing edge modulation
UCC3819
Programmable output power factor preregulator
Tracking boost topology for dynamic output voltage adjustments
UC3854
High Power Factor Preregulator
High PF, industry standard PFC controller; 35 VCC max
UC3854A/B
Enhanced high power factor preregulator
Improved high PF, industry standard PFC controller; 22 VCC max
UC3855A/B
High performance power factor preregulator
ZVT output for lower EMI emission & higher efficiencies
UC3853
High power factor preregulator
8-Pin package; simplified architecture to minimized external components
UCC38050
Transition mode PFC controller
Constant on-time transition mode PFC controller
UC3852
High power factor preregulator
Constant off-time transition mode PFC controller; 30 VCC max
www.ti.com
31
SLUS517− DECEMBER 2002
MECHANICAL DATA
DW (R-PDSO-G**)
PLASTIC SMALL-OUTLINE PACKAGE
16 PINS SHOWN
0.020 (0,51)
0.014 (0,35)
9
0.050 (1,27)
16
0.010 (0,25)
0.419 (10,65)
0.400 (10,15)
0.010 (0,25) NOM
0.299 (7,59)
0.291 (7,39)
Gage Plane
0.010 (0,25)
1
8
0°− 8°
0.050 (1,27)
0.016 (0,40)
A
Seating Plane
0.104 (2,65) MAX
0.012 (0,30)
0.004 (0,10)
PINS **
0.004 (0,10)
16
18
20
24
28
A MAX
0.410
(10,41)
0.462
(11,73)
0.510
(12,95)
0.610
(15,49)
0.710
(18,03)
A MIN
0.400
(10,16)
0.453
(11,51)
0.500
(12,70)
0.600
(15,24)
0.700
(17,78)
DIM
4040000/E 08/01
NOTES: A.
B.
C.
D.
32
All linear dimensions are in inches (millimeters).
This drawing is subject to change without notice.
Body dimensions do not include mold flash or protrusion not to exceed 0.006 (0,15).
Falls within JEDEC MS-013
www.ti.com
SLUS517− DECEMBER 2002
MECHANICAL DATA
N (R-PDIP-T**)
PLASTIC DUAL-IN-LINE PACKAGE
16 PINS SHOWN
PINS **
14
16
18
20
A MAX
0.775
(19,69)
0.775
(19,69)
0.920
(23,37)
1.060
(26,92)
A MIN
0.745
(18,92)
0.745
(18,92)
0.850
(21,59)
0.940
(23,88)
MS-100
VARIATION
AA
BB
AC
DIM
A
16
9
0.260 (6,60)
0.240 (6,10)
1
C
AD
8
0.070 (1,78) MAX
0.035 (0,89) MAX
0.325 (8,26)
0.300 (7,62)
0.020 (0,51) MIN
0.015 (0,38)
Gauge Plane
0.200 (5,08) MAX
Seating Plane
0.010 (0,25) NOM
0.125 (3,18) MIN
0.100 (2,54)
0.430 (10,92) MAX
0.021 (0,53)
0.015 (0,38)
0.010 (0,25) M
14/18 PIN ONLY
20 pin vendor option
D
4040049/E 12/2002
NOTES: A.
B.
C.
D.
All linear dimensions are in inches (millimeters).
This drawing is subject to change without notice.
Falls within JEDEC MS-001, except 18 and 20 pin minimum body lrngth (Dim A).
The 20 pin end lead shoulder width is a vendor option, either half or full width.
www.ti.com
33
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