AD AD549KHZ

Ultralow Input Bias Current
Operational Amplifier
AD549
FEATURES
CONNECTION DIAGRAM
GUARD PIN,
CONNECTED
TO CASE
Ultralow input bias current
60 fA maximum (AD549L)
250 fA maximum (AD549J)
Input bias current guaranteed over the common-mode
voltage range
Low offset voltage
0.25 mV maximum (AD549K)
1.00 mV maximum (AD549J)
Low offset drift
5 μV/°C maximum (AD549K)
20 μV/°C maximum (AD549J)
Low power
700 μA maximum supply current
Low input voltage noise
4 μV p-p over 0.1 Hz to 10 Hz
MIL-STD-883B parts available
OFFSET NULL
INVERTING
INPUT
V+
8
1
AD549
7
2
6
5
3
NONINVERTING
INPUT
4
OUTPUT
OFFSET
NULL
V–
5
4
VOS TRIM
–15V
00511-001
10kΩ
1
Figure 1.
APPLICATIONS
Electrometer amplifier
Photodiode preamp
pH electrode buffer
Vacuum ion gauge measurement
GENERAL DESCRIPTION
1
The AD549 is a monolithic electrometer operational amplifier
with very low input bias current. Input offset voltage and input
offset voltage drift are laser trimmed for precision performance.
The ultralow input current of the part is achieved with Topgate™
JFET technology, a process development exclusive to Analog
Devices, Inc. This technology allows fabrication of extremely
low input current JFETs compatible with a standard junction
isolated bipolar process. The 1015 Ω common-mode impedance,
which results from the bootstrapped input stage, ensures that
the input current is essentially independent of the commonmode voltage.
The AD549 is suited for applications requiring very low input
current and low input offset voltage. It excels as a preamp for a
wide variety of current output transducers, such as photodiodes,
photomultiplier tubes, or oxygen sensors. The AD549 can also
be used as a precision integrator or low droop sample-and-hold.
The AD549 is pin compatible with standard FET and electrometer
op amps, allowing designers to upgrade the performance of
present systems at little additional cost.
The AD549 is available in a TO-99 hermetic package. The case
is connected to Pin 8, thus, the metal case can be independently
connected to a point at the same potential as the input terminals,
minimizing stray leakage to the case. The AD549 is available in
four performance grades. The J, K, and L versions are rated over
the commercial temperature range of 0°C to +70°C. The S grade
is specified over the military temperature range of −55°C to +125°C
and is available processed to MIL-STD-883B, Rev. C. Extended
reliability plus screening is also available. Plus screening includes
168 hour burn-in, as well as other environmental and physical
tests derived from MIL-STD-883B, Rev. C.
PRODUCT HIGHLIGHTS
1.
The AD549 input currents are specified, 100% tested, and
guaranteed after the device is warmed up. They are guaranteed over the entire common-mode input voltage range.
2.
The AD549 input offset voltage and drift are laser trimmed
to 0.25 mV and 5 μV/°C (AD549K), and to 1 mV and
20 μV/°C (AD549J).
3.
A maximum quiescent supply current of 700 μA minimizes
heating effects on input current and offset voltage.
4.
AC specifications include 1 MHz unity-gain bandwidth
and 3 V/μs slew rate. Settling time for a 10 V input step is
5 μs to 0.01%.
1
Protected by U.S. Patent No. 4,639,683.
Rev. H
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113 ©2002–2008 Analog Devices, Inc. All rights reserved.
AD549
TABLE OF CONTENTS
Features .............................................................................................. 1 Offset Nulling ............................................................................. 11 Applications ....................................................................................... 1 Connection Diagram ....................................................................... 1 AC Response with High Value Source and Feedback
Resistance .................................................................................... 12 General Description ......................................................................... 1 Common-Mode Input Voltage Overload ............................... 12 Product Highlights ........................................................................... 1 Differential Input Voltage Overload ........................................ 13 Revision History ............................................................................... 2 Input Protection ......................................................................... 13 Specifications..................................................................................... 3 Sample and Difference Circuit to Measure Electrometer
Leakage Currents ........................................................................ 13 Absolute Maximum Ratings............................................................ 5 ESD Caution .................................................................................. 5 Typical Performance Characteristics ............................................. 6 Functional Description .................................................................. 10 Minimizing Input Current ........................................................ 10 Circuit Board Notes ................................................................... 10 Photodiode Interface ................................................................. 14 Log Ratio Amplifier ................................................................... 15 Temperature Compensated pH Probe Amplifier................... 16 Outline Dimensions ....................................................................... 18 Ordering Guide .......................................................................... 18 REVISION HISTORY
3/08—Rev. G to Rev. H
Changes to Features.......................................................................... 1
Changes to Figure 1 .......................................................................... 1
Deleted Package Option Parameter ............................................... 4
Inserted ESD Caution ...................................................................... 5
Changes to Figure 2, Figure 3, and Figure 7.................................. 6
Changes to Figure 11 ........................................................................ 7
Changes to Figure 17 ........................................................................ 8
Changes to Figure 41 ...................................................................... 14
7/07—Rev. F to Rev. G
Changes to Figure 45 ...................................................................... 16
Changes to Temperature Compensated pH Probe
Amplifier Section ............................................................................ 17
Changes to Figure 46 ...................................................................... 17
Changes to Ordering Guide .......................................................... 18
5/06—Rev. E to Rev. F
Removed ESD Caution .....................................................................5
8/05—Rev. D to Rev. E
Change to Figure 22 ..........................................................................9
5/04—Rev. C to Rev. D
Updated Format .................................................................. Universal
Changes to Features ..........................................................................1
Updated Outline Dimensions ....................................................... 18
Added Ordering Guide .................................................................. 18
10/02—Rev. B to Rev. C
Deleted Product Highlights #5 ........................................................1
Edits to Specifications .......................................................................3
Deleted Metallization Photograph ..................................................3
Updated Outline Dimensions ....................................................... 13
7/02—Rev. A to Rev. B
Edits to Specifications .......................................................................2
Rev. H | Page 2 of 20
AD549
SPECIFICATIONS
@ 25°C and VS = ±15 V dc, unless otherwise noted; all minimum and maximum specifications are guaranteed; specifications in boldface
are tested on all production units at final electrical test, and results from those tests are used to calculate outgoing quality levels.
Table 1.
Parameter
INPUT BIAS CURRENT 1
Either Input, VCM = 0 V
Either Input, VCM = ±10 V
Either Input at TMAX,
VCM = 0 V
Offset Current
Offset Current at TMAX
INPUT OFFSET VOLTAGE 2
Initial Offset
Offset at TMAX
vs. Temperature
vs. Supply
vs. Supply, TMIN to TMAX
Long-Term Offset Stability
INPUT VOLTAGE NOISE
f = 0.1 Hz to 10 Hz
f = 10 Hz
f = 100 Hz
f = 1 kHz
f = 10 kHz
INPUT CURRENT NOISE
f = 0.1 Hz to 10 Hz
f = 1 kHz
INPUT IMPEDANCE
Differential
VDIFF = ±1
Common Mode
VCM = ±10 V
OPEN-LOOP GAIN
VOUT @ ±10 V, RL = 10 kΩ
VOUT @ ±10 V, RL = 10 kΩ,
TMIN to TMAX
VOUT = ±10 V, RL = 2 kΩ
VOUT = ±10 V, RL = 2 kΩ,
TMIN to TMAX
INPUT VOLTAGE RANGE
Differential 3
Common-Mode Voltage
Common-Mode Rejection
Ratio
−10 V ≤ VCM ≤ +10 V
TMIN to TMAX
Min
AD549J
Typ
Max
150
150
11
Min
75
75
4.2
250
250
50
2.2
0.5
10
32
32
15
AD549K
Typ
Max
Min
40
40
2.8
100
100
30
1.3
0.15
1.0
1.9
20
100
100
2
10
10
15
AD549L
Typ
Max
Min
60
60
20
0.85
AD549S
Typ
Max
Unit
75
75
420
fA
fA
pA
100
100
30
125
0.25
0.4
5
32
32
0.3
6
4
90
60
35
35
4
90
60
35
35
μV p-p
nV/√Hz
nV/√Hz
nV/√Hz
nV/√Hz
5
10
10
15
0.3
fA
pA
0.5
0.9
10
32
32
10
10
32
15
0.5
2.0
15
32
50
mV
mV
μV/°C
μV/V
μV/V
μV/month
4
90
60
35
35
4
90
60
35
35
0.7
0.22
0.5
0.16
0.36
0.11
0.5
0.16
fA rms
fA/√Hz
1013||1
1013||1
1013||1
1013||1
Ω||pF
1015||0.8
1015||0.8
1015||0.8
1015||0.8
Ω||pF
300
300
1000
800
300
300
1000
800
300
300
1000
800
300
300
1000
800
V/mV
V/mV
100
80
250
200
100
80
250
200
100
80
250
200
100
25
250
150
V/mV
V/mV
±20
+10
−10
80
76
90
80
±20
+10
−10
90
80
100
90
Rev. H | Page 3 of 20
±20
+10
−10
90
80
100
90
±20
+10
−10
90
80
100
90
V
V
dB
dB
AD549
Parameter
OUTPUT CHARACTERISTICS
VOUT @ RL = 10 kΩ, TMIN to
TMAX
VOUT @ RL = 2 kΩ, TMIN to TMAX
Short-Circuit Current
TMIN to TMAX
Load Capacitance Stability,
G = +1
FREQUENCY RESPONSE
Unity Gain, Small Signal
Full Power Response
Slew Rate
Settling Time, 0.1%
Settling Time, 0.01%
Overload Recovery, 50%
Overdrive, G = −1
POWER SUPPLY
Rated Performance
Operating
Quiescent Current
TEMPERATURE RANGE
Operating, Rated
Performance
Storage
Min
AD549J
Typ
Max
Min
AD549K
Typ
Max
Min
AD549L
Typ
Max
Min
AD549S
Typ
Max
−12
+12
−12
+12
−12
+12
−12
+12
V
+10
35
−10
15
9
+10
35
−10
15
9
+10
35
−10
15
6
+10
35
4000
V
mA
mA
pF
1.0
50
3
4.5
5
2
MHz
kHz
V/μs
μs
μs
μs
−10
15
9
20
4000
0.7
2
4000
1.0
50
3
4.5
5
2
0.7
2
±15
±5
±5
1.0
50
3
4.5
5
2
0.7
2
0
70
−65
+150
1.0
50
3
4.5
5
2
0.7
2
±15
±18
0.70
±5
0
70
−65
+150
0.60
20
4000
±15
±18
0.70
0.60
20
1
±15
±18
0.70
V
V
mA
−55
+125
°C
−65
+150
°C
±18
0.70
±5
0
70
−65
+150
0.60
20
Unit
0.60
Bias current specifications are guaranteed after five minutes of operation at TA = 25°C. Bias current increases by a factor of 2.3 for every 10°C rise in temperature.
Input offset voltage specifications are guaranteed after five minutes of operation at TA = 25°C.
3
Defined as maximum continuous voltage between the inputs, such that neither input exceeds ±10 V from ground.
2
Rev. H | Page 4 of 20
AD549
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
Supply Voltage
Internal Power Dissipation
Input Voltage 1
Output Short-Circuit Duration
Differential Input Voltage
Storage Temperature Range
Operating Temperature Range
AD549J, AD549K, AD549L
AD549S
Lead Temperature (Soldering, 60 sec)
1
Rating
±18 V
500 mW
±18 V
Indefinite
+VS and −VS
−65°C to +125°C
0°C to +70°C
−55°C to +125°C
300°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
ESD CAUTION
For supply voltages less than ±18 V, the absolute maximum input voltage is
equal to the supply voltage.
Rev. H | Page 5 of 20
AD549
TYPICAL PERFORMANCE CHARACTERISTICS
800
INPUT VOLTAGE (V)
15
VIN+
10
VIN–
0
00511-002
5
0
5
10
15
700
600
500
400
20
00511-005
AMPLIFIER QUIESCENT CURRENT (µA)
20
0
5
SUPPLY VOLTAGE (±V)
Figure 2. Input Voltage Range vs. Supply Voltage
15
–VOUT
10
5
0
5
10
15
110
100
90
80
70
–20
20
SUPPLY VOLTAGE (±V)
00511-006
COMMON-MODE REJECTION RATIO (dB)
+VOUT
00511-003
OUTPUT VOLTAGE SWING (V)
20
120
25°C
RL = 10kΩ
–10
0
10
20
INPUT COMMON-MODE VOLTAGE (V)
Figure 3. Output Voltage Swing vs. Supply Voltage
Figure 6. CMRR vs. Input Common-Mode Voltage
30
3000
VS = ±15V
OPEN-LOOP GAIN (V/mV)
25
20
15
10
1000
300
0
10
100
1k
10k
100
100k
LOAD RESISTANCE (Ω)
00511-007
5
00511-004
OUTPUT VOLTAGE SWING (V p-p)
15
Figure 5. Quiescent Current vs. Supply Voltage
20
0
10
SUPPLY VOLTAGE (±V)
0
5
10
15
SUPPLY VOLTAGE (±V)
Figure 4. Output Voltage Swing vs. Load Resistance
Figure 7. Open-Loop Gain vs. Supply Voltage
Rev. H | Page 6 of 20
20
AD549
3000
50
INPUT CURRENT (fA)
OPEN-LOOP GAIN (V/mV)
45
1000
300
40
35
30
–25
5
35
65
95
20
125
00511-011
100
–55
00511-008
25
0
5
TEMPERATURE (°C)
Figure 8. Open-Loop Gain vs. Temperature
15
10
00511-009
5
1
2
3
4
5
6
120
100
80
60
40
20
10
7
100
WARM-UP TIME (Minutes)
Figure 9. Change in Offset Voltage vs. Warm-Up Time
45
10k
INPUT NOISE VOLTAGE (µV p-p)
100k
40
35
30
00511-010
25
–5
0
10k
Figure 12. Input Voltage Noise Spectral Density
50
20
–10
1k
FREQUENCY (Hz)
5
WHENEVER JOHNSON NOISE IS GREATER THAN
AMPLIFIER NOISE, AMPLIFIER NOISE CAN BE
CONSIDERED NEGLIGIBLE FOR THE APPLICATION
1kHz BANDWIDTH
1k
RESISTOR
JOHNSON NOISE
100
10
10Hz BANDWIDTH
1
00511-013
0
140
00511-012
NOISE SPECTRAL DENSITY (nV/ Hz)
20
ΔIVOSI (µV)
20
160
25
INPUT CURRENT (fA)
15
Figure 11. Input Bias Current vs. Power Supply Voltage
30
0
10
POWER SUPPLY VOLTAGE (±V)
AMPLIFIER GENERATED NOISE
0.1
100k
10
COMMON-MODE VOLTAGE (V)
1M
10M
100M
1G
SOURCE RESISTANCE (Ω)
Figure 10. Input Bias Current vs. Common-Mode Voltage
Figure 13. Noise vs. Source Resistance
Rev. H | Page 7 of 20
10G
100G
AD549
60
60
40
40
20
20
0
0
–20
–40
10
100
1k
10k
100k
00511-014
–20
–40
10M
1M
100
80
+PSSR
60
40
–PSSR
20
0
–20
10
00511-017
80
POWER SUPPLY REJECTION RATIO (dB)
80
120
PHASE MARGIN (Degrees)
100
OEPN-LOOP GAIN (dB)
100
100
FREQUENCY (Hz)
1k
10k
100k
1M
10M
FREQUENCY (Hz)
Figure 14. Open-Loop Frequency Response
Figure 17. PSRR vs. Frequency Response
40
10
30
25
20
15
10
0
10
00511-015
5
100
1k
10k
100k
FREQUENCY (Hz)
60
40
20
0
00511-016
COMMON-MODE REJECTION RATIO (dB)
80
1k
10k
100k
10mV
5mV
–5
1mV
0
1
2
3
4
Figure 18. Output Voltage Swing and Error vs. Settling Time
100
100
0
SETTLING TIME (µs)
Figure 15. Large Signal Frequency Response
–20
10
5mV
1mV
–10
1M
10mV
5
00511-018
OUTPUT VOLTAGE SWING (V)
OUTPUT VOLTAGE SWING (V)
35
1M
10M
FREQUENCY (Hz)
Figure 16. CMRR vs. Frequency
Rev. H | Page 8 of 20
5
AD549
10kΩ
+VS
+VS
0.1µF
0.1µF
7
4
0.1µF
RL
10kΩ
CL
100pF
–VS
2
7
AD549
3
4
0.1µF
5
RL
10kΩ
VOUT
CL
100pF
–VS
00511-022
SQUARE
WAVE
INPUT
3
SQUARE
WAVE
INPUT
Figure 22. Unity-Gain Inverter
Figure 19. Unity-Gain Follower
5V
5V
5µs
00511-020
5µs
00511-023
VIN
VIN
VOUT
5
10kΩ
Figure 23. Unity-Gain Inverter Large Signal Pulse Response
Figure 20. Unity-Gain Follower Large Signal Pulse Response
10mV
1µs
00511-021
10mV
1µs
Figure 21. Unity-Gain Follower Small Signal Pulse Response
00511-024
AD549
00511-019
2
Figure 24. Unity-Gain Inverter Small Signal Pulse Response
Rev. H | Page 9 of 20
AD549
FUNCTIONAL DESCRIPTION
MINIMIZING INPUT CURRENT
CIRCUIT BOARD NOTES
The AD549 is optimized for low input current and offset
voltage. Careful attention to how the amplifier is used reduces
input currents in actual applications.
A number of physical phenomena generate spurious currents
that degrade the accuracy of low current measurements. Figure 27
is a schematic of a current to voltage (I-to-V) converter with
these parasitic currents modeled.
Keep the amplifier operating temperature as low as possible to
minimize input current. Like other JFET input amplifiers, the
AD549 input current is sensitive to chip temperature, rising by
a factor of 2.3 for every 10°C. Figure 25 is a plot of the AD549
input current vs. ambient temperature.
CF
RF
2
AD549
1nA
fS
3
8
6
+
VOUT
–
10pA
RP
1pA
VS
V dCP
dV C
II' = R + dT V +
dT P
P
Figure 27. Sources of Parasitic Leakage Currents
100fA
00511-025
10fA
1fA
–55
–25
5
35
65
95
125
TEMPERATURE (°C)
Figure 25. Input Bias Current vs. Ambient Temperature
On-chip power dissipation raises the chip operating temperature, causing an increase in input bias current. Due to the low
quiescent supply current of the AD549, the chip temperature
is less than 3°C higher than its ambient temperature when the
(unloaded) amplifier is operating with 15 V supplies. The
difference in the input current is negligible.
However, heavy output loads can cause a significant increase in
chip temperature and a corresponding increase in the input
current. Maintaining a minimum load resistance of 10 Ω is
recommended. Input current vs. additional power dissipation
due to output drive current is plotted in Figure 26.
6
5
4
BASED ON
TYPICAL IB = 40fA
3
2
1
00511-026
NORMALIZED INPUT BIAS CURRENT
CP
00511-027
INPUT BIAS CURRENT
100pA
0
25
50
75
100
125
150
175
Finite resistance from input lines to voltages on the board,
modeled by Resistor RP, results in parasitic leakage. Insulation
resistance of more than 1015 Ω must be maintained between
the amplifier signal and supply lines to capitalize on the low
input currents of the AD549. Standard PCB material does not
have high enough insulation resistance; therefore, connect the
input leads of the AD549 to standoffs made of insulating
material with adequate volume resistivity (that is, Teflon®). The
surface of the insulator must be kept clean to preserve surface
resistivity. For Teflon, an effective cleaning procedure consists
of swabbing the surface with high grade isopropyl alcohol,
rinsing with deionized water, and baking the board at 80°C for
10 minutes.
In addition to high volume and surface resistivity, other properties are desirable in the insulating material chosen. Resistance
to water absorption is important because surface water films
drastically reduce surface resistivity. The insulator chosen
should also exhibit minimal piezoelectric effects (charge
emission due to mechanical stress) and triboelectric effects
(charge generated by friction). Charge imbalances generated
by these mechanisms can appear as parasitic leakage currents.
These effects are modeled by Variable Capacitor CP in Figure 27.
Table 3 lists various insulators and their properties. 1
Guarding the input lines by completely surrounding them with
a metal conductor biased near the potential of the input lines
has two major benefits. First, parasitic leakage from the signal
line is reduced because the voltage between the input line and
the guard is very low. Second, stray capacitance at the input
node is minimized. Input capacitance can substantially degrade
signal bandwidth and the stability of the I-to-V converter.
200
ADDITIONAL INTERNAL POWER DISSIPATION (mW)
Figure 26. Input Bias Current vs. Additional Power Dissipation
1
Electronic Measurements, pp. 15–17, Keithley Instruments, Inc., Cleveland,
Ohio, 1977.
Rev. H | Page 10 of 20
AD549
The case of the AD549 is connected to Pin 8 so that it can be
bootstrapped near the input potential. This minimizes pin
leakage and input common-mode capacitance due to the case.
Guard schemes for inverting and noninverting amplifier
topologies are illustrated in Figure 28 and Figure 29.
CF
GUARD
RF
OFFSET NULLING
The AD549 input offset voltage can be nulled by using balance
Pin 1 and Pin 5, as shown in Figure 30. Nulling the input offset
voltage in this fashion introduces an added input offset voltage
drift component of 2.4 μV/°C per mV of nulled offset (a maximum additional drift of 0.6 μV/°C for the AD549K, 1.2 μV/°C
for the AD549L, and 2.4 μV/°C for the AD549J).
+VS
2
+
VOUT
–
6
8
7
2
AD549
00511-028
3
5
+
VOUT
–
6
1
3
4
10kΩ
Figure 28. Inverting Amplifier with Guard
00511-030
AD549
–VS
Figure 30. Standard Offset Null Circuit
GUARD
The approach in Figure 31 can be used when the amplifier is
used as an inverter. This method introduces a small voltage
referenced to the power supplies in series with the positive
input terminal of the amplifier. The amplifier input offset
voltage drift with temperature is not affected. However,
variation of the power supply voltages causes offset shifts.
3
AD549
+
VS
2
8
6
VOUT
+
RF
–
00511-029
RI
–
RF
RI
Figure 29. Noninverting Amplifier with Guard
2
AD549
+
Other guidelines include keeping the circuit layout as compact
as possible and keeping the input lines short. Keeping the assembly
rigid and minimizing sources of vibration reduces triboelectric
and piezoelectric effects. All precision, high impedance circuitry
requires shielding against interference noise. Use low noise coaxial
or triaxial cables for remote connections to the input signal lines.
VI
3
+VS
–
499kΩ
200Ω
6
499kΩ
0.1µF
+
VOUT
–
100kΩ
–VS
00511-031
IN
Figure 31. Alternate Offset Null Circuit for Inverter
Table 3. Insulating Materials and Characteristics
Material
Teflon
Kel-F®
Sapphire
Polyethylene
Polystyrene
Ceramic
Glass Epoxy
PVC
Phenolic
1
Volume Resistivity
(V to CM)
1017 to 1018
1017 to 1018
1016 to 1018
1014 to 1018
1012 to 1018
1012 to 1014
1010 to 1017
1010 to 1015
105 to 1012
Minimal
Triboelectric Effect 1
W
W
M
M
W
W
W
G
W
Minimal
Piezoelectric Effect1
W
M
G
G
M
M
M
M
G
G: good with regard to property; M: moderate with regard to property; W: weak with regard to property.
Rev. H | Page 11 of 20
Resistance to
Water Absorption1
G
G
G
M
M
W
W
G
W
AD549
AC RESPONSE WITH HIGH VALUE SOURCE AND
FEEDBACK RESISTANCE
Source and feedback resistances greater than 100 kΩ magnify
the effect of the input capacitances (stray and inherent to
the AD549) on the ac behavior of the circuit. The effects of
common-mode and differential input capacitances should be
taken into account because the circuit bandwidth and stability
can be adversely affected.
10mV
Figure 34. Inverter Pulse Response with 1 MΩ Source
and Feedback Resistance
Figure 32. Follower Pulse Response from 1 MΩ Source Resistance,
Case Not Bootstrapped
10mV
5µs
00511-035
5µs
00511-033
10mV
5µs
00511-034
5µs
00511-032
10mV
In an inverting configuration, the differential input capacitance
forms a pole in the loop transmission of the circuit. This can
create peaking in the ac response and possible instability. A
feedback capacitance can be used to stabilize the circuit. The
inverter pulse response with RF and RS equal to 1 MΩ appears
in Figure 34. Figure 35 shows the response of the same circuit
with a 1 pF feedback capacitance. Typical differential input
capacitance for the AD549 is 1 pF.
Figure 35. Inverter Pulse Response with 1 MΩ Source
and Feedback Resistance, 1 pF Feedback Capacitance
Figure 33. Follower Pulse Response from 1 MΩ Source Resistance,
Case Bootstrapped
In a follower, the source resistance and input common-mode
capacitance form a pole that limits the bandwidth to ½πRSCS.
Bootstrapping the metal case by connecting Pin 8 to the output
minimizes capacitance due to the package. Figure 32 and Figure 33
show the follower pulse response from a 1 MΩ source resistance
with and without the package connected to the output. Typical
common-mode input capacitance for the AD549 is 0.8 pF.
COMMON-MODE INPUT VOLTAGE OVERLOAD
The rated common-mode input voltage range of the AD549 is
from 3 V less than the positive supply voltage to 5 V greater than
the negative supply voltage. Exceeding this range degrades the
CMRR of the amplifier. Driving the common-mode voltage above
the positive supply causes the amplifier output to saturate at the
upper limit of the output voltage. Recovery time is typically 2 μs
after the input has been returned to within the normal operating
range. Driving the input common-mode voltage within 1 V of the
negative supply causes phase reversal of the output signal. In this
case, normal operation typically resumes within 0.5 μs of the
input voltage returning within range.
Rev. H | Page 12 of 20
AD549
RPROTECT
DIFFERENTIAL INPUT VOLTAGE OVERLOAD
100µ
10µ
IIN–
IIN+
100n
10n
1n
3
AD549
6
2
Figure 38. Follower with Input Current Limit
Figure 39 is a schematic of the AD549 as an inverter with an
input voltage clamp. Bootstrapping the clamp diodes at the
inverting input minimizes the voltage across the clamps and
keeps the leakage due to the diodes low. Use low leakage diodes,
such as the FD333s, and shield them from light to prevent photocurrents from being generated. Even with these precautions, the
diodes measurably increase input current and capacitance.
100p
RF
SOURCE
10p
2
AD549
00511-036
1p
100f
10f
–5
–4
–3
–2
–1
0
1
2
3
4
3
PROTECT
DIODES
5
Figure 39. Input Voltage Clamp with Diodes
DIFFERENTIAL INPUT VOLTAGE (V) (VIN+ – VIN–)
SAMPLE-AND-DIFFERENCE CIRCUIT TO MEASURE
ELECTROMETER LEAKAGE CURRENTS
Figure 36. Input Current vs. Differential Input Voltage
INPUT PROTECTION
The AD549 safely handles any input voltage within the supply
voltage range. Subjecting the input terminals to voltages beyond
the power supply can destroy the device or cause shifts in input
current or offset voltage if the amplifier is not protected.
A protection scheme for the amplifier as an inverter is shown
in Figure 37. RP is chosen to limit the current through the
inverting input to 1 mA for expected transient (less than 1 sec)
overvoltage conditions, or to 100 μA for a continuous overload.
Because RP is inside the feedback loop and is much lower in
value than the amplifier input resistance, it does not affect the
dc gain of the inverter. However, the Johnson noise of the
resistor adds root sum of squares to the amplifier input noise.
RF
RPROTECT
CF
2
AD549
6
3
00511-037
SOURCE
6
00511-039
INPUT CURRENT (A)
1µ
SOURCE
00511-038
A plot of the AD549 input currents vs. differential input
voltage (defined as VIN+ − VIN−) appears in Figure 36. The
input current at either terminal stays below a few hundred
femtoamps until one input terminal is forced higher than 1 V
to 1.5 V above the other terminal. Under these conditions, the
input current limits at 30 μA.
There are a number of methods used to test electrometer leakage
currents, including current integration and direct I-to-V conversion. Regardless of the method used, board and interconnect
cleanliness, proper choice of insulating materials (such as Teflon
or Kel-F), correct guarding and shielding techniques, and care
in physical layout are essential to making accurate leakage
measurements.
Figure 40 is a schematic of the sample-and-difference circuit. It
uses two AD549 electrometer amplifiers (A and B) as I-to-V
converters with high value (1010 Ω) sense resistors (RSa and
RSb). R1 and R2 provide for an overall circuit sensitivity of
10 fA/mV (10 pA full scale). CC and CF provide noise suppression
and loop compensation. CC should be a low leakage polystyrene
capacitor. An ultralow leakage Kel-F test socket is used for contacting the device under test. Rigid Teflon coaxial cable is used
to make connections to all high impedance nodes. The use of
rigid coaxial cable affords immunity to error induced by mechanical vibration and provides an outer conductor for shielding. The
entire circuit is enclosed in a grounded metal box.
Figure 37. Inverter with Input Current Limit
In the corresponding version of this scheme for a follower,
shown in Figure 38, RP and the capacitance at the positive input
terminal produce a pole in the signal frequency response at a
f = ½πRC. Again, the Johnson noise, RP, adds to the input
voltage noise of the amplifier.
Rev. H | Page 13 of 20
AD549
The test apparatus is calibrated without a device under test
present. After power is turned on, a 5 minute stabilization
period is required. First, VERR1 and VERR2 are measured. These
voltages are the errors caused by the offset voltages and leakage
currents of the I-to-V converters.
VERR1 = 10 (VOSA – IBA × RSa)
VERR2 = 10 (VOSB – IBB × RSb)
CC
20pF
CF
0.1µF
RSa
1010Ω
R2
9.01kΩ
PHOTODIODE INTERFACE
R1
1kΩ
2
A
6
AD549
8
3
Although a series of devices can be tested after only one calibration measurement, calibration should be updated periodically
to compensate for any thermal drift of the I-to-V converters or
changes in the ambient environment. Laboratory results have
shown that repeatable measurements within 10 fA can be realized
when this apparatus is properly implemented. These results are
achieved in part by the design of the circuit, which eliminates
relays and other parasitic leakage paths in the high impedance
signal lines, and in part by the inherent cancellation of errors
through the calibration and measurement procedure.
CAL/TEST
+
VERR1 /VA
–
The low input current and low input offset voltage of the AD549
make it an excellent choice for very sensitive photodiode preamps
(see Figure 41). The photodiode develops a signal current, IS,
equal to
IS = R × P
GUARD
where P is light power incident on the diode surface, in watts,
and R is the photodiode responsivity in amps/watt. RF converts
the signal current to an output voltage
I (+)
+
DEVICE
UNDER
TEST
VOUT
–
CF
0.1µF
R2
9.01kΩ
VOUT = RF × IS
I (–)
R1
1kΩ
RF
109Ω
VERR2 /VB
8
3
B
6
AD549
CF
10pF
+
2
2
IS
CF
0.1µF
RSb
1010Ω
R2
9.01kΩ
R1
1kΩ
5
+
6
1µF
1
3
VOUT
4
–
–VS
00511-040
CC
20pF
10kΩ
AD549
Figure 40. Sample and Difference Circuit for Measuring
Electrometer Leakage Currents
Once measured, these errors are subtracted from the readings
taken with a device under test present. Amplifier B closes the
feedback loop to the device under testing in addition to providing the I-to-V conversion. The offset error of the device
under testing appears as a common-mode signal and does not
affect the test measurement. As a result, only the leakage
current of the device under testing is measured.
Figure 41. Photodiode Preamp
The dc error sources and an equivalent circuit for a small area
(0.2 mm square) photodiode are indicated in Figure 42.
RF
109Ω
CF
10pF
IS
VA – VERR1 = 10[RSa × IB(+)]
RS
109Ω
CS
20pF
A
IS–
–
VOS
+
Figure 42. Photodiode Preamp DC Error Sources
VX – VERR2 = 10[RSb × IB(–)]
Rev. H | Page 14 of 20
+
VOUT
–
00511-042
–
00511-041
VOS
AD549
Input current, IB, contributes an output voltage error, VE1,
proportional to the feedback resistance
VE1 = IB × RF
The input voltage offset of the op amp causes an error current
through the photodiode shunt resistance, RS
gain that multiplies the op amp input voltage noise contribution. A single-pole filter at the output of the amplifier limits the
op amp output voltage noise bandwidth to 26 Hz, comparable
to the signal bandwidth. This greatly improves the signal-tonoise ratio of the preamplifier (in this case, by a factor of 3).
10µ
IF AND CS, NO FILTERS
VE2 = (1 + RF/RS)VOS
Given typical values of photodiode shunt resistance (on the order
of 109 Ω), RF/RS can easily be greater than 1, especially if a large
feedback resistance is used. Also, RF/RS increases with temperature because photodiode shunt resistance typically drops by a
factor of 2 for every 10°C rise in temperature. An op amp with
low offset voltage and low drift must be used to maintain accuracy.
The AD549K offers a guaranteed maximum 0.25 mV offset
voltage and 5 mV/°C drift for very sensitive applications.
Noise sources associated with the photodiode, amplifier, and
feedback resistance are shown in Figure 43; Figure 44 is the
spectral density vs. frequency plot of the contribution of each of
the noise sources to the output voltage noise (circuit parameters
in Figure 42 are assumed). The rms contribution of each noise
source to the total output voltage noise is obtained by
integrating the square of its spectral density function over
frequency. The rms value of the output voltage noise is the
square root of the sum of all contributions. Minimizing the total
area under these curves optimizes the resolution of the
preamplifier for a given bandwidth.
IF
AD549
OPEN-LOOP GAIN
1µ
100n
EN CONTRIBUTION,
NO FILTER
10n
EN
CONTRIBUTION,
WITH FILTER
1
10
100
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 44. Spectral Density of the Photodiode Preamp Noise
Sources vs. Frequency
Photodiode Preamp Noise
Noise limits the signal resolution obtainable with the preamp.
The output voltage noise divided by the feedback resistance is
the minimum current signal that can be detected. This minimum detectable current divided by the responsivity of the
photodiode represents the lowest light power that is detectable
by the preamp.
IF AND CS, WITH FILTERS
00511-044
The error current results in an error voltage (VE2) at the
amplifier output equal to
VOLTAGE NOISE CONTRIBUTIONS
NOISE SPECTRAL DENSITY (nV/ Hz)
I = VOS/RS
LOG RATIO AMPLIFIER
Logarithmic ratio circuits are useful for processing signals with
wide dynamic range. The 60 fA maximum input current of the
AD549L makes it possible to build a log ratio amplifier with
1% log conformance for input currents ranging from 10 pA to
1 mA, a dynamic range of 160 dB.
The log ratio amplifier in Figure 45 provides an output voltage
proportional to the log base 10 of the ratio of Input Current I1
and Input Current I2. Resistor R1 and Resistor R2 are provided
for voltage inputs. Because NPN devices are used in the feedback
loop of the front-end amplifiers that provide the log transfer
function, the output is valid only for positive input voltages and
input currents. The input currents set the Collector Current IC1
and Collector Current IC2 of a matched pair of log transistors,
Q1 and Q2, to develop Voltage VA and Voltage VB
VA, VB = –(kT/q)ln IC/IES
where IES is the saturation current of the transistor.
RF
The difference of VA and VB is taken by the subtractor section to
obtain
CF
VC = (kT/q)ln(IC2/IC1)
RS
CS
A
IN
EN
VC is scaled up by the ratio of (R9 + R10)/R8, which is equal to
approximately 16 at room temperature, resulting in the output
voltage
00511-043
IS
Figure 43. Photodiode Preamp Noise Sources
The photodiode preamp in Figure 41 can detect a signal current
of 26 fA rms at a bandwidth of 16 Hz, which, assuming a
photodiode responsivity of 0.5 A/W, translates to a 52 fW rms
minimum detectable power. The photodiode used has a high
source resistance and low junction capacitance. CF sets the
signal bandwidth with RF and also limits the peak in the noise
VOUT = 1 V × log(IC2/IC1)
R8 is a resistor with a positive 3500 ppm/°C temperature coefficient to provide the necessary temperature compensation. The
parallel combination of R15 and R7 is provided to keep the gain
of the subtractor section for positive and negative inputs matched
over temperature.
Rev. H | Page 15 of 20
AD549
composed of R13, D1, R16, R14, D2, and R17 compensate for
these errors, so that this circuit has less than a 1% log conformance error at 1 mA input currents. The correct value for R13
and R14 depends on the type of log transistors used. The 49.9 kΩ
resistors were chosen for use with LM394 transistors. Smaller
resistance values are needed for smaller log transistors.
Frequency compensation is provided by R11, R12, C1, and C2.
The bandwidth of the circuit is 300 kHz at input signals greater
than 50 μA; bandwidth decreases smoothly with decreasing
signal levels.
To trim the circuit, set the input currents to 10 μA and trim the
A3 offset using the trim potentiometer of the amplifier for the
output to equal 0. Next, set I1 to 1 μA and adjust the output to
equal 1 V by trimming R10. Additional offset trims on Amplifier A1 and Amplifier A2 can be used to increase the voltage
input accuracy and dynamic range.
TEMPERATURE COMPENSATED pH PROBE
AMPLIFIER
A pH probe can be modeled as an mV-level voltage source
with a series source resistance dependent on the electrode
composition and configuration. The glass bulb resistance of a
typical pH electrode pair falls between 106 Ω and 109 Ω. It is
therefore important to select an amplifier with low enough
input currents such that the voltage drop produced by the
amplifier input bias current and the electrode resistance does
not become an appreciable percentage of a pH unit.
The very low input current of the AD549 makes this circuit
useful over a very wide range of signal currents. The total input
current (which determines the low level accuracy of the circuit)
is the sum of the amplifier input current, the leakage across the
compensating capacitor (negligible if a polystyrene or Teflon
capacitor is used), and the collector-to-collector and collectorto-base leakages of one side of the dual log transistors. The
magnitudes of these last two leakages depend on the amplifier
input offset voltage and are typically less than 10 fA with 1 mV
offsets. The low level accuracy is limited primarily by the
amplifier input current, only 60 fA maximum when the
AD549L is used.
The circuit in Figure 46 illustrates the use of the AD549 as a pH
probe amplifier. As with other electrometer applications, the use of
guarding, shielding, and Teflon standoffs is necessary to capitalize
on the AD549 low input current. If an AD549L (60 fA maximum
input current) is used, the error contributed by the input current is
held below 60 μV for pH electrode source impedances up to 109 Ω.
Input offset voltages (which can be trimmed) are below 0.5 mV.
The effects of the emitter resistance of Q1 and Q2 can degrade
circuit accuracy at input currents above 100 μA. The networks
FOR EACH AMPLIFIER
1
5
A1
D3
6
AD549
I1 IN
2
R1
10kΩ
0.1µF
0.1µF
PIN 4
R11
4.99kΩ
Q1
R15
1kΩ
R3
20kΩ
R5
20kΩ
R7
15kΩ
3
D1
VOUT
A3
AD549
D2
R17
10Ω
V2 IN
R13
49.9kΩ
R2
10kΩ
C2
100pF
Q2
R4
20kΩ
B
1
2
4
R6
20kΩ
*
2
A2
D4
AD549
5
6
1
3
4
10kΩ
V2
OFFSET
5
6
R10
10kΩ 2kΩ
OUTPUT
OFFSET
SCALE
FACTOR
ADJ
R9
R8 14.3kΩ
1kΩ
VOUT = 1V × LOG10
4.99kΩ
I2 IN
–VS
*
A
R16
10Ω
R14
49.9kΩ
+VS
Q1, Q2 = LM394
DUAL LOG TRANSISTORS
C1
100pF
V1 IN
PIN 7
V2
V1
I2
VOUT = 1V × LOG10
I1
D1, D4 1N4148 DIODES
R8, R15 1kΩ + 350 ppm/°C TC RESISTOR
*TELLAB QB1 OR PRECISION RESISTOR PT146
ALL OTHER RESISTORS ARE 1% METAL FILM
Figure 45. Log Ratio Amplifier
Rev. H | Page 16 of 20
00511-045
4
3
10kΩ
V1
OFFSET
AD549
the compensation. The AD549 is set for a noninverting gain of
13.51. The output of the AD590 circuitry (Point C) is equal to
10 V at 100°C and decreases by 26.8 mV/°C. The output of the
AD534 analog divider (Point D) is a temperature-compensated
output voltage centered at 0 V for a pH of 7 and has a transfer
function of –1.00 V/pH unit. The output range spans from
−7.00 V (pH = 14) to +7.00 V (pH = 0).
The pH probe output is ideally 0 V at a pH of 7, independent
of temperature. The slope of the transfer function of the probe,
though predictable, is temperature dependent (−54.2 mV/pH at
0°C and −74.04 mV/pH at 100°C). By using an AD590 temperature sensor and an AD534 analog divider, an accurate temperature
compensation network can be added to the basic pH probe amplifier. Table 4 shows voltages at various points, thereby illustrating
+15V
0.1µF
0.1µF
pH
PROBE
OUTPUT
14
(A)
3
7
AD549
2
(B)
6
Z2
11
Z1
1
X1
2
X2
0.1µF
12kΩ
(D)
OUTPUT
Y2 7
Y1 6
8
1kΩ
SCALE FACTOR
ADJUST
0.1µF
+15V
AD590
IN STAINLESS
STEEL PROBE
OR AC2626
OUT 12
4
8
(C)
1kΩ
AD534
10
–15V
+
–
00511-046
26.6kΩ
Figure 46. Temperature Compensated pH Amplifier
Table 4. Illustration of Temperature Compensation
Point
Probe Temperature (°C)
0
25
37
60
100
A (Probe Output) (mV)
54.20
59.16
61.54
66.10
74.04
B (A × 13.51) (V)
0.732
0.799
0.831
0.893
1.000
Rev. H | Page 17 of 20
C (AD590 Output) (V)
7.32
7.99
8.31
8.93
10.00
D (10 × (B ÷ C)) (V)
1.00
1.00
1.00
1.00
1.00
AD549
OUTLINE DIMENSIONS
REFERENCE PLANE
0.1850 (4.70)
0.1650 (4.19)
0.5000 (12.70)
MIN
0.2500 (6.35) MIN
0.0500 (1.27) MAX
0.1000 (2.54)
BSC
0.1600 (4.06)
0.1400 (3.56)
0.3350 (8.51)
0.3050 (7.75)
0.3700 (9.40)
0.3350 (8.51)
5
6
4
0.2000
(5.08)
BSC
3
7
2
0.0400 (1.02) MAX
0.0400 (1.02)
0.0100 (0.25)
0.1000
(2.54)
BSC
0.0190 (0.48)
0.0160 (0.41)
0.0210 (0.53)
0.0160 (0.41)
8
1
0.0450 (1.14)
0.0270 (0.69)
0.0340 (0.86)
0.0280 (0.71)
45° BSC
COMPLIANT TO JEDEC STANDARDS MO-002-AK
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
022306-A
BASE & SEATING PLANE
Figure 47. 8-Lead Metal Can [TO-99]
(H-08)
Dimensions shown in inches and (millimeters)
ORDERING GUIDE
Model
AD549JH
AD549JHZ 1
AD549KH
AD549KHZ1
AD549LH
AD549LHZ1
AD549SH/883B
1
Temperature Range
0°C to +70°C
0°C to +70°C
0°C to +70°C
0°C to +70°C
0°C to +70°C
0°C to +70°C
−55°C to +125°C
Package Description
8-Lead Metal Can (TO-99)
8-Lead Metal Can (TO-99)
8-Lead Metal Can (TO-99)
8-Lead Metal Can (TO-99)
8-Lead Metal Can (TO-99)
8-Lead Metal Can (TO-99)
8-Lead Metal Can (TO-99)
Z = RoHS Compliant Part.
Rev. H | Page 18 of 20
Package Option
H-08
H-08
H-08
H-08
H-08
H-08
H-08
AD549
NOTES
Rev. H | Page 19 of 20
AD549
NOTES
©2002–2008 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D00511-0-3/08(H)
Rev. H | Page 20 of 20