L6911C 5 BIT PROGRAMMABLE STEP DOWN CONTROLLER WITH SYNCHRONOUS RECTIFICATION ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ OPERATING SUPPLY IC VOLTAGE FROM 5V TO 12V BUSES UP TO 1.3A GATE CURRENT CAPABILITY TTL-COMPATIBLE 5 BIT PROGRAMMABLE OUTPUT COMPLIANT WITH VRM 8.4 : 1.3V TO 2.05V WITH 0.05V BINARY STEPS 2.1V TO 3.5V WITH 0.1V BINARY STEPS VOLTAGE MODE PWM CONTROL EXCELLENT OUTPUT ACCURACY: ±1% OVER LINE AND TEMPERATURE VARIATIONS VERY FAST LOAD TRANSIENT RESPONSE: FROM 0% TO 100% DUTY CYCLE POWER GOOD OUTPUT VOLTAGE OVERVOLTAGE PROTECTION AND MONITOR OVERCURRENT PROTECTION REALIZED USING THE UPPER MOSFET’S RdsON 200KHz INTERNAL OSCILLATOR OSCILLATOR EXTERNALLY ADJUSTABLE FROM 50KHz TO 1MHz SOFT START AND INHIBIT FUNCTIONS APPLICATIONS ■ POWER SUPPLY FOR ADVANCED MICROPROCESSOR CORE ■ DISTRIBUTED POWER SUPPLY ■ HIGH POWER DC-DC REGULATORS SO-20 ORDERING NUMBERS: L6911C L6911CTR (Tape and Reel) DESCRIPTION The device is a power supply controller specifica lly designed to provide a high performance DC/DC conversion for high current microprocessors. A precise 5-bit digital to analog converter (DAC) allows adjusting the output voltage from 1.30V to 2.05V with 50mV binary steps and from 2.10V to 3.50V with 100mV binary steps. The high precision internal reference assures the selected output voltage to be within ±1%. The high peak current gate drive affords to have fast switching to the external power mos providing low switching losses. The device assures a fast protection against load overcurrent and load overvoltage. An external SCR is triggered to crowbar the input supply in case of hard over-voltage. An internal crowbar is also provided turning on the low side mosfet as long as the overvoltage is detected. In case of over-current detection, the soft start capacitor is discharged and the system works in HICCUP mode. BLOCK DIAGRAM Vcc 5 to 12V Vin 5V to12V VCC PGOOD OCSET BOOT SS MONITOR and PROTECTION UGATE OVP RT PHASE Vo 1.300V to 3.500V OSC LGATE VD0 VD1 VD2 VD3 VD4 PGND - D/A + + - D01IN1260 November 2001 GND PWM VSEN E/A VFB COMP 1/20 L6911C ABSOLUTE MAXIMUM RATINGS Symbol Value Unit VCC to GND, PGND 15 V Boot Voltage 15 V 15 V -0.3 to Vcc+0.3 V 7 V 6.5 V Value Unit Thermal Resistance Junction to Ambient 110 °C/W Maximum junction temperature 150 °C VCC VBOOT-V PHASE Parameter VHGATE-VPHASE OCSET, LGATE, PHASE RT, SS, FB, PGOOD, VSEN, VID0-4 OVP, COMP THERMAL DATA Symbol R th j-amb Tj Parameter T stg Storage temperature range -40 to 150 °C TJ Junction temperature range 0 to 125 °C PIN CONNECTION (Top view) VSEN 1 20 RT OCSET 2 19 OVP SS/INH 3 18 VCC VID0 4 17 LGATE VID1 5 16 PGND VID2 6 15 BOOT VID3 7 14 UGATE VID4 8 13 PHASE COMP 9 12 PGOOD 10 11 GND FB D98IN958 2/20 L6911C PIN FUNCTION Pin Num. Name 1 VSEN 2 OCSET A resistor connected from this pin and the upper Mos Drain sets the current limit protection. The internal 200µA current generator sinks a current from the drain through the external resistor. The Over-Current threshold is due to the following equation: I OCSE T ⋅ R O CSET IP = ---------------------------------------------R DSon 3 SS/INH The soft start time is programmed connecting an external capacitor from this pin and GND. The internal current generator forces through the capacitor 10µA. This pin can be used to disable the device forcing a voltage lower than 0.4V 4-8 VID0 - 4 Voltage Identification Code pins. These input are internally pulled-up and TTL compatible. They are used to program the output voltage as specified in Table 1 and to set the overvoltage and power good thresholds. Connect to GND to program a ‘0’ while leave floating to program a ‘1’. 9 COMP This pin is connected to the error amplifier output and is used to compensate the voltage control feedback loop. 10 FB This pin is connected to the error amplifier inverting input and is used to compensate the voltage control feedback loop. 11 GND 12 PGOOD This pin is an open collector output and is pulled low if the output voltage is not within the above specified thresholds. If not used may be left floating. 13 PHASE This pin is connected to the source of the upper mosfet and provides the return path for the high side driver. This pin monitors the drop across the upper mosfet for the current limit 14 UGATE High side gate driver output. 15 BOOT Bootstrap capacitor pin. Through this pin is supplied the high side driver and the upper mosfet. Connect through a capacitor to the PHASE pin and through a diode to Vcc (cathode vs. boot). 16 PGND Power ground pin. This pin has to be connected closely to the low side mosfet source in order to reduce the noise injection into the device 17 LGATE This pin is the lower mosfet gate driver output 18 VCC Device supply voltage. The operative nominal supply voltage ranges from 5 to 12V. DO NOT CONNECT VIN TO A VOLTAGE GREATER THAN VCC. 19 OVP Over voltage protection. If the output voltage reaches the 17% above the programmed voltage this pin is driven high and can be used to drive an external SCR that crowbar the supply voltage. If not used, it may be left floating. 20 RT Description Connected to the output voltage is able to manage over-voltage conditions and the PGOOD signal. All the internal references are referred to this pin. Connect it to the PCB signal ground. Oscillator switching frequency pin. Connecting an external resistor from this pin to GND, the external frequency is increased according to the equation: 6 4.94 ⋅ 10 f S = 200kHz + ------------------------R T ( kΩ ) Connecting a resistor from this pin to Vcc (12V), the switching frequency is reduced according to the equation: 7 4.306 ⋅ 10 f S = 200kHz – ----------------------------R T ( kΩ ) If the pin is not connected, the switching frequency is 200KHz. The voltage at this pin is fixed at 1.23V (typ). Forcing a 50µA current into this pin, the built in oscillator stops to switch. 3/20 L6911C ELECTRICAL CHARACTERISTCS (VCC = 12V, Tamb = 25°C unless otherwise specified) Symbol Parameter Test Condition Min. Typ. Max. Unit V CC SUPPLY CURRENT Icc Vcc Supply current UGATE and LGATE open 5 mA POWER-ON Turn-On Vcc threshold VOCSET=4.5V Turn-Off Vcc threshold VOCSET=4.5V 4.6 3.6 Rising VOCSET threshold ISS Soft start Current V V 1.24 V 10 µA OSCILLATOR ∆Vosc Free running frequency RT = OPEN 180 Total Variation 6 KΩ < RT to GND < 200 KΩ -15 Ramp amplitude RT = OPEN 200 220 KHz 15 % 1.9 Vp-p REFERENCE AND DAC DACOUT Voltage Accuracy VID0, VID1, VID2, VID3, VID4 see Table1; Tamb = 0 to 70°C -1 VID Pull-Up voltage 1 % 4 V 88 dB ERROR AMPLIFI ER DC Gain GBWP SR Gain-Bandwidth Product Slew-Rate COMP=10pF 10 MHz 10 V/µS 1.3 A GATE DRIVERS 1 IUGATE High Side Source Current V BOOT - VPHASE=12V, V UGATE - VPHASE= 6V R UGATE High Side Sink Resistance V BOOT-VPHASE=12V, IUGATE = 300mA ILGATE Low Side Source Current Vcc=12V, VLGATE = 6V R LGATE Low Side Sink Resistance Vcc=12V, ILGATE = 300mA 1.5 Output Driver Dead Time PHASE connected to GND 120 Over Voltage Trip (VSEN/DACOUT) V SEN Rising 117 120 % OCSET Current Source VOCSET = 4.5V 170 200 230 µA OVP Sourcing Current V SEN > OVP Trip, VOVP=0V 60 Upper Threshold (VSEN/DACOUT) V SEN Rising 110 112 114 % Lower Threshold (VSEN/DACOUT) V SEN Falling 86 88 90 % Hysteresis (VSEN/DACOUT) Upper and Lower threshold PGOOD Voltage Low IPGOOD = -5mA 2 0.9 4 1.1 Ω A 3 Ω ns PROTECTIONS IOCSET I OVP mA POWER GOOD VPGOOD 4/20 2 % 0.5 V L6911C Table 1. VID Settings VID4 VID3 VID2 VID1 VID0 Output Voltage (V) VID4 VID3 VID2 VID1 VID0 Output Voltage (V) 0 1 1 1 1 1.30 1 1 1 1 1 Output Off 0 1 1 1 0 1.35 1 1 1 1 0 2.1 0 1 1 0 1 1.40 1 1 1 0 1 2.2 0 1 1 0 0 1.45 1 1 1 0 0 2.3 0 1 0 1 1 1.50 1 1 0 1 1 2.4 0 1 0 1 0 1.55 1 1 0 1 0 2.5 0 1 0 0 1 1.60 1 1 0 0 1 2.6 0 1 0 0 0 1.65 1 1 0 0 0 2.7 0 0 1 1 1 1.70 1 0 1 1 1 2.8 0 0 1 1 0 1.75 1 0 1 1 0 2.9 0 0 1 0 1 1.80 1 0 1 0 1 3.0 0 0 1 0 0 1.85 1 0 1 0 0 3.1 0 0 0 1 1 1.90 1 0 0 1 1 3.2 0 0 0 1 0 1.95 1 0 0 1 0 3.3 0 0 0 0 1 2.00 1 0 0 0 1 3.4 0 0 0 0 0 2.05 1 0 0 0 0 3.5 Device Description The device is an integrated circuit realized in BCD technology. It provides complete control logic and protections for a high performance step-down DC-DC converter optimized for microprocessor power supply. It is designed to drive N-Channel Mosfets in a synchronous-rectified buck topology. The device works properly with Vcc ranging from 5V to 12V and regulates the output voltage starting from a 1.26V power stage supply voltage (Vin). The output voltage of the converter can be precisely regulated, programming the VID pins, from 1.3V to 2.05V with 50mV binary steps and from 2.1V to 3.5V with 100mV binary steps, with a maximum tolerance of ±1% over temperature and line voltage variations. The device provides voltage-mode control with fast transient response. It includes a 200kHz free-running oscillator that is adjustable from 50kHz to 1MHz. The error amplifier features a 15MHz gain-bandwidth product and 10V/µsec slew rate which permits high converter bandwidth for fast transient performance. The resulting PWM duty cycle ranges from 0% to 100%. The device protects against overcurrent conditions entering in HICCUP mode. The device monitors the current by using the rDS(ON) of the upper MOSFET which eliminates the need for a current sensing resistor. The device is available in SO20 package. Oscillator The switching frequency is internally fixed to 200kHz. The internal oscillator generates the triangular waveform for the PWM charging and discharging with a constant current an internal capacitor. The current delivered to the oscillator is typically 50µA (Fsw=200KHz) and may be varied using an external resistor (RT) connected between RT pin and GND or VCC. Since the RT pin is maintained at fixed voltage (typ. 1.235V), the frequency is varied proportionally to the current sunk (forced) from (into) the pin. In particular connecting it to GND the frequency is increased (current is sunk from the pin), according to the following relationship: 6 4.94 ⋅ 10 f S = 200kH z + ------------------------R T ( kΩ ) Connecting RT to VCC=12V or to VCC=5V the frequency is reduced (current is forced into the pin), according to the following relationships: 5/20 L6911C 7 4.306 ⋅ 10 f S = 200kH z + ----------------------------R T ( kΩ ) VCC = 12V 7 15 ⋅ 10 f S = 200k H z + -------------------R T ( kΩ ) VCC = 5V Switching frequency variations vs. RT are reported in Fig.1. Note that forcing a 50µA current into this pin, the device stops switching because no current is delivered to the oscillator. Figure 1. 1 0 00 0 R e s is ta n c e [kO h m ] 1 00 0 10 0 RT to G N D 10 RT to V CC =1 2 V RT to V CC =5 V 10 100 1 000 F re qu e nc y [k H z ] Digital to Analog Converter The built-in digital to analog converter allows the adjustment of the output voltage from 1.30V to 2.05V with 50mV binary steps and from 2.10V to 3.50V with 100mV binary steps as shown in the previous table 1. The internal reference is trimmed to ensure the precision of 1%. The internal reference voltage for the regulation is programmed by the voltage identification (VID) pins. These are TTL compatible inputs of an internal DAC that is realized by means of a series of resistors providing a partition of the internal voltage reference. The VID code drives a multiplexer that selects a voltage on a precise point of the divider. The DAC output is delivered to an amplifier obtaining the VPROG voltage reference (i.e. the set-point of the error amplifier). Internal pull-ups are provided (realized with a 5µA current generator); in this way, to program a logic ”1” it is enough to leave the pin floating, while to program a logic ”0” it is enough to short the pin to GND. The voltage identification (VID) pin configuration also sets the power-good thresholds (PGOOD) and the overvoltage protection (OVP) thresholds. The VID code ”11111” disable the device (as a short on the SS pin) and no output voltage is regulated. Soft Start and Inhibit At start-up a ramp is generated charging the external capacitor CSS by means of a 10µA constant current, as shown in figure 1. When the voltage across the soft start capacitor (VSS) reaches 0.5V the lower power MOS is turned on to dis- 6/20 L6911C charge the output capacitor. As VSS reaches 1V (i.e. the oscillator triangular wave inferior limit) also the upper MOS begins to switch and the output voltage starts to increase. The VSS growing voltage initially clamps the output of the error amplifier, and consequently VOUT linearly increases, as shown in figure 2. In this phase the system works in open loop. When VSS is equal to VCOMP the clamp on the output of the error amplifier is released. In any case another clamp on the input of the error amplifier remains active, allowing to VOUT to grow with a lower slope (i.e. the slope of the VSS voltage, see figure 2). In this second phase the system works in closed loop with a growing reference. As the output voltage reaches the desired value VPROG, also the clamp on the error amplifier input is removed, and the soft start finishes. Vss increases until a maximum value of about 4V. The Soft-Start will not take place, and the relative pin is internally shorted to GND, if both VCC and OCSET pins are not above their own turn-on thresholds. During normal operation, if any under-voltage is detected on one of the two supplies, the SS pin is internally shorted to GND and so the SS capacitor is rapidly discharged. The device goes in INHIBIT state forcing SS pin below 0.4V. In this condition both external MOSFETS are kept off. Figure 2. Soft Start V cc T urn-on thresh old V cc Vin V in Turn-on threshold 1V Vss to G ND 0.5V LGAT E Vout Timing Diagram Aquisition: CH1 = PHASE; CH2 = VOUT; CH3 = PGOOD; CH4 = VSS Driver Section The driver capability on the high and low side drivers allows using different types of power MOS (also multiple MOS to reduce the RDSON), maintaining fast switching transition. The low-side mos driver is supplied directly by Vcc while the high-side driver is supplied by the BOOT pin. Adaptative dead time control is implemented to prevent cross-conduction and allow to use several kinds of mosfets. The upper mos turn-on is avoided if the lower gate is over about 200mV while the lower mos turn-on is avoided if the PHASE pin is over about 500mV. The upper mos is in any case turned-on after 200nS from the low side turn-off. The peak current is shown for both the upper (fig. 3) and the lower (fig. 4) driver at 5V and 12V. A 4nF capacitive load has been used in these measurements. For the lower driver, the source peak current is 1.1A @ Vcc=12V and 500mA @ Vcc=5V, and the sink peak current is 1.3A @ Vcc=12V and 500mA @ Vcc=5V. Similarly, for the upper driver, the source peak current is 1.3A @ Vboot-Vphase=12V and 600mA @ VbootVphase =5V, and the sink peak current is 1.3A @ Vboot-Vphase =12V and 550mA @ Vboot-Vphase = 5V. 7/20 L6911C Figure 3. High Side driver peak current. Vboot-Vphase=12V (left) Vboot-Vphase=5V (right) CH1 = High Side Gate CH4 = Gate Current Figure 4. Low Side driver peak current. Vcc=12V (left) Vcc=5V (right) CH1 = Low Side Gate CH4 = Gate Current Monitoring and Protections The output voltage is monitored by means of pin 1 (VSEN). If it is not within ±12% (typ.) of the programmed value, the powergood output is forced low. The device provides overvoltage protection, when the output voltage reaches a value 17% (typ.) grater than the nominal one. If the output voltage exceeds this threshold, the OVP pin is forced high, triggering an external SCR to shuts the supply (VIN) down, and also the lower driver is turned on as long as the over-voltage is detected. To perform the overcurrent protection the device compares the drop across the high side MOS, due to the RDSON, with the voltage across the external resistor (ROCS) connected between the OCSET pin and drain of the upper MOS. Thus the overcurrent threshold (IP) can be calculated with the following relationship: I OCS ⋅ R OCS I P = --------------------------------R DSON Where the typical value of IOCS is 200µA. To calculate the ROCS value it must be considered the maximum RDSON (also the variation with temperature) and the minimum value of IOCS. To avoid undesirable trigger of 8/20 L6911C overcurrent protection this relationship must be satisfied: ∆l I P ≥ IO UT MAX + ----- = I PEAK 2 Where ∆I is the inductance ripple current and IOUTMAX is the maximum output current. In case of output short circuit the soft start capacitor is discharged with constant current (10µA typ.) and when the SS pin reaches 0.5V the soft start phase is restarted. During the soft start the over-current protection is always active and if such kind of event occurs, the device turns off both mosfets, and the SS capacitor is discharged again (after reaching the upper threshold of about 4V). The system is now working in HICCUP mode, as shown in figure 5a. After removing the cause of the over-current, the device restart working normally without power supplies turn off and on. Figure 5. 9 L=1.5µH, Vin=12V Inductor Ripple [A] 8 7 L=2µH, Vin=12V 6 L=3µH, Vin=12V 5 4 L=1.5µH, Vin=5V 3 L=2µH, Vin=5V 2 L=3µH, Vin=5V 1 0 0 .5 1 .5 2 .5 3 .5 O utput Voltage [V] a: Hiccup Mode b: Inductor Ripple Current vs. Vout Inductor design The inductance value is defined by a compromise between the transient response time, the efficiency, the cost and the size. The inductor has to be calculated to sustain the output and the input voltage variation to maintain the ripple current ∆ IL between 20% and 30% of the maximum output current. The inductance value can be calculated with this relationship: VIN – V O UT V OUT L = ------------------------------ ⋅ --------------fS ⋅ ∆I L V IN Where fSW is the switching frequency, VIN is the input voltage and VOUT is the output voltage. Figure 5b shows the ripple current vs. the output voltage for different values of the inductor, with VIN = 5V and VIN = 12V. Increasing the value of the inductance reduces the ripple current but, at the same time, reduces the converter response time to a load transient. If the compensation network is well designed, the device is able to open or close the duty cycle up to 100% or down to 0%. The response time is now the time required by the inductor to change its current from initial to final value. Since the inductor has not finished its charging time, the output current is supplied by the output capacitors. Minimizing the response time can minimize the output capacitance required. The response time to a load transient is different for the application or the removal of the load: if during the application of the load the inductor is charged by a voltage equal to the difference between the input and the output voltage, during the removal it is discharged only by the output voltage. The following expressions give approximate response time for ∆I load transient in case of enough fast compensation network response: 9/20 L6911C L ⋅ ∆I ta pplicatio n = -----------------------------V IN – VOU T L ⋅ ∆I t removal = --------------VOUT The worst condition depends on the input voltage available and the output voltage selected. Anyway the worst case is the response time after removal of the load with the minimum output voltage programmed and the maximum input voltage available. Output Capacitor Since the microprocessors require a current variation beyond 10A doing load transients, with a slope in the range of tenth A/µsec, the output capacitor is a basic component for the fast response of the power supply. In fact for first few microseconds they supply the current to the load. The controller recognizes immediately the load transient and sets the duty cycle at 100%, but the current slope is limited by the inductor value. The output voltage has a first drop due to the current variation inside the capacitor (neglecting the effect of the ESL): ∆VOUT = ∆IOUT · ESR A minimum capacitor value is required to sustain the current during the load transient without discharge it. The voltage drop due to the output capacitor discharge is given by the following equation: 2 ∆VOU T ∆I OU T L = -------------------------------------------------------------------------------------------2 ⋅ C OU T ⋅ ( VINM IN ⋅ D M AX – V OUT ) Where DMAX is the maximum duty cycle value that is 100%. The lower is the ESR, the lower is the output drop during load transient and the lower is the output voltage static ripple. Input Capacitor The input capacitor has to sustain the ripple current produced during the on time of the upper MOS, so it must have a low ESR to minimize the losses. The rms value of this ripple is: I rm s = IO UT D ⋅ ( 1 – D ) Where D is the duty cycle. The equation reaches its maximum value with D=0.5. The losses in worst case are: 2 P = ESR ⋅ Irm s Compensation network design The control loop is a voltage mode (figure 7) that uses a droop function to satisfy the requirements for a VRM module, reducing the size and the cost of the output capacitor. This method ”recovers” part of the drop due to the output capacitor ESR in the load transient, introducing a dependence of the output voltage on the load current: at light load the output voltage will be higher than the nominal level, while at high load the output voltage will be lower than the nominal value. 10/20 L6911C Figure 6. Output transient response without (a) and with (b) the droop function ESR DROP ESR DROP V MAX VDROOP VNOM V MIN (a) (b) As shown in figure 6, the ESR drop is present in any case, but using the droop function the total deviation of the output voltage is minimized. In practice the droop function introduces a static error (Vdroop in figure 6) proportional to the output current. Since a sense resistor is not present, the output DC current is measured by using the intrinsic resistance of the inductance (a few mΩ). So the low-pass filtered inductor voltage (that is the inductor current) is added to the feedback signal, implementing the droop function in a simple way. Referring to the schematic in figure 7, the static characteristic of the closed loop system is: R 3 + R 8 // R9 R L ⋅ R8 // R9 V OU T = V PROG + V PROG ⋅ ------------------------------------- – -----------------------------------⋅ I OUT R2 R8 Where VPROG is the output voltage of the digital to analog converter (i.e. the set point) and RL is the inductance resistance. The second term of the equation allows a positive offset at zero load (∆V+); the third term introduces the droop effect (∆VDROOP). Note that the droop effect is equal the ESR drop if: R L ⋅ R 8 // R9 ----------------------------------- = ESR R8 Figure 7. Compensation network VIN VC VP OM P R L2 HA SE L V O UT PW M E SR R 8 C 18 Z C 6 -1 5 F C20 R4 R 9 C 25 R3 V ZI P RO G R2 Considering the previous relationships R2, R3, R8 and R9 may be determined in order to obtain the desired droop effect as follow: ■ Choose a value for R2 in the range of hundreds of KΩ to obtain realistic values for the other components. 11/20 L6911C ■ From the above equations, it results: + ∆ V ⋅ R 2 R L ⋅ I MAX R8 = ----------------------- ⋅ --------------------------- ; V PROG ∆V DROOP ∆V DRO OP 1 R9 = R 8 ⋅ --------------------------- ⋅ -------------------------------------; R L ⋅ I M AX ∆VD ROOP 1 + --------------------------R L ⋅ IM AX Where IMAX is the maximum output current. ■ The component R3 must be chosen in order to obtain R3<<R8//R9 to permit these and successive simplifications. Therefore, with the droop function the output voltage decreases as the load current increases, so the DC output impedance is equal to a resistance ROUT. It is easy to verify that the output voltage deviation under load transient is minimum when the output impedance is constant with frequency. To choose the other components of the compensation network, the transfer function of the voltage loop is considered. To simplify the analysis is supposed that R3 << Rd, where Rd = (R8//R9). Figure 8. Compensation network definition |A v | 2 f LC fC E f2 f1 fE C fC C f |R | R0 fD f3 f |G lo o p| G0 fc f ConverterS ingularity f LC = 1 / 2 π ⋅ LC fCE = 1 / 2 π ⋅ ESR ⋅ C OUT f = 1 / 2 π ⋅ ESR ⋅ Cceramic EC f = 1 / 2 π ⋅ Rceramic ⋅ Cceramic CC Introduced by f1 = 1 / 2π ⋅ R 4 ⋅ C 20 f2 = 1 / 2 π ⋅ ( R 3 + R 4 ) ⋅ C 20 f3 = 1 / 2 π ⋅ R 3 ⋅ C 25 CeramicCap acitor fd = 1 / 2 π ⋅ Rd ⋅ C 25 doublepole ESRzero Compensati onNetworkS ingularity The transfer function may be evaluated neglecting the connection of R8 to PHASE because, as will see later, this connection is important only at low frequencies. So R4 is considered connected to VOUT. Under this assumption, the voltage loop has the following transfer function: 12/20 L6911C ZC ( s ) Zf ( s ) Vin G loop ( s) = Av ( s) ⋅ R ( s ) = Av ( s ) ⋅ -------------- Where Av ( s ) = ---------------- ⋅ ------------------------------------∆Vo sc Z C ( s ) + Z L ( s ) Z i ( s) Where ZC(s) and ZL(s) are the output capacitor and inductor impedance respectively. The expression of ZI(s) may be simplified as follow: 2 R3 1 1 Rd 1 + s ⋅ ( τ1 + τd ) + s ⋅ -------- ⋅ τ 1 ⋅ τd - ⋅ C 20 ⋅ R 3 Rd ⋅ --- ⋅ C 25 R4 + -Rd s s Z I ( s ) = ---------------------------------- + ----------------------------------------------------- - = -------------------------------------------------------------------------------------------------- -= ( 1 + s ⋅ τ 2 ) ⋅ ( 1 + s ⋅ τd ) 1 R d + --- ⋅ C25 R 4 + 1 --- ⋅ C 20 + R3 s s 1 + s R3 -------- ⋅ τ d ⋅ ( 1 + s ⋅ τ1 ) Rd = R d -------------------------------------------------------------------- ( 1 + s ⋅ τ 2) ⋅ ( 1 + s ⋅ τd ) Where: τ1 = R4×C20, τ2 = (R4+R3)×C20 and τd = Rd×C25. The regulator transfer function became now: ( 1 + s ⋅ τ2 ) ⋅ ( 1 + s ⋅ τd ) R ( s ) ≈ -----------------------------------------------------------------------------------------------------R3 s ⋅ C 18 ⋅ Rd ⋅ 1 + s -------- ⋅ τd ⋅ ( 1 + s ⋅ τ1 ) Rd Figure 8 shows a method to select the regulator components (please note that the frequencies fEC and fCC corresponds to the singularities introduced by additional ceramic capacitors in parallel to the output main electrolytic capacitor). ■ To obtain a flat frequency response of the output impedance, the droop time constantτd has to be equal to the inductor time constant (see the note at the end of the section): L τ d = Rd ⋅ C 25 = ------- = τ L RL ■ To obtain a constant -20dB/dec Gloop(s) shape the singularity 1f and f2 are placed in proximity of fC E and fLC respectively. This implies that: f2 f LC ---- = --------fC E f1 f 1 = fCE ■ L ⇒ C25 = ----------------------( RL ⋅ Rd ) ⇒ ⇒ fLC R4 = R 3 ⋅ --------- – 1 f CE 1 C20 = --- ⋅ π ⋅ R 4 ⋅ fCE 2 To obtain a Gloop bandwidth of fC, results: G0 ⋅ f LC = 1 ⋅ fC ⇒ fC VIN C 20 // C25 G 0 = A 0 ⋅ R 0 = ------------------ ⋅ ----------------------------- = -------fLC ∆ Vosc C18 VIN C 20 ⋅ C25 f LC ⇒ C 18 = ------------------ ⋅ ----------------------------- ⋅ -------∆Vosc C20 + C25 f C Note. To understand the reason of the previous assumption, the scheme in figure 9 must be considered. In this scheme, the inductor current has been substituted by the load current, because in the frequencies range of interest for the Droop function these current are substantially the same and it was supposed that the droop network don’t represent a charge for the inductor. 13/20 L6911C Figure 9. Voltage regulation with droop function block scheme V com p Vout A v(s) R (s) It results: R OUT ⋅ Io ut 1 + s ⋅ τL 1 + s ⋅τd G LOOP 1 + sτL 1 + sτL Vo Z OU T = ---------------- = R d ⋅ ------------------ ⋅ ----------------------------- = R OUT ⋅ -----------------ILOAD 1 + sτd 1 + GL OOP 1 + sτd Because in the interested range |Gloop|>>1. To obtain a flat shape, the relationship considered will naturally follow. Demo Board Description The L6911C demo board shows the operation of the device in a standard VRM 8.4 application. This evaluation board allows voltage adjustability (1.3V - 3.5V) through the switches S1-S5 and high output current capability (up to 14A). The device is supplied by the 12V input rail while the power conversion starts from the 5V input rail. The device is also able to operate with a 5V supply voltage; in this case 12V input can be directly connected to the 5V power source. The four layers demo board’s copper thickness is of 70µm in order to minimize conduction losses considering the high current that the circuit is able to deliver. Figure 10 shows the demo board’s schematic circuit. Figure 10. Demo Board Schematic F1 L1 +5 VIN C21-22 G1 15 VCC +12Vcc C24 19 OCSET 2 18 GND C17 GND12 OVP D1 BOOT GNDIN 11 S1 UGATE 4 VID1 S3 VID2 S4 VID3 S5 VID4 R7 R13 Q1-2 L2 13 U1 17 L6911C16 6 7 8 VOUTCORE LGATE PGND R14 Q4-5 D2 C6-15 R12 GNDCORE PGOOD 12 SS 3 OSC PWRGD VSEN 20 1 10 C18 C19 R8 VFB COMP 9 R1 C23 PHASE 5 C16 C1-5 14 VID0 S2 Q8 R5 R3 C20 R4 R9 C25 R2 L6911-L6912 EVALUATIONKIT REV. 14/20 L6911C Efficiency Figure 11 shows the measured efficiency versus load current for different values of output voltage. The measure was done at Vin=5V for different values of the output voltage (2.05V and 2.75V). Two different measurements were done using IC supply voltage of 5V and 12V. In the application two mosfets STS12NF30L (30V, 10mΩ typ @ Vgs=4.5V) connected in parallel are used for both the low and the high side. The board has been layed out with the possibility to use up to three SO8 mosfets for both high and low side switch. Two D2PACK mosfets (one for each high and low side) may also be used in order to allow the maximum flexibility in meeting different requirements. 95 95 90 90 85 85 Efficiency [%] Efficiency [%] Figure 11. Efficiency vs. load 80 75 70 Vout = 1.7V Vout = 2.0V Vout = 2.5V 65 60 80 75 70 V out = 1.7V V out = 2.0V V out = 2.5V 65 60 55 55 0 2 4 6 8 10 12 14 16 Output Cur r ent [A ] Vcc = 12V; Vin = 5V 0 2 4 6 8 10 12 14 16 Out put C ur r en t [ A ] Vcc = Vin = 5V Figure 13. Load Transient Response Details Load Transient Response Figure 12 shows the demo board response to a load transient application. The load transient applied changes from 0A to 14A on the output current (Channel 4). It may be observed that output voltage (Channel 1) remains within the 100mV tolerance across the regulated voltage. Figure 13 shows details about the the circuit response during current rising and falling edge; it is possible to observe that the duty cycle of the Phase signal (Channel 2) goes up to 100% or down to 0% if necessary. Figure 12. Load Transient Response 15/20 L6911C Inductor selection Since the maximum output current is equal to 14A, to have a 30% ripple (4A) in worst case a 3µH inductor has been chosen. So the ripple is 4.1A @ 3.5V with VIN=12V and 1.7A @ 3.5V with VIN = 5V. In worst case the peak is equal to 18.1A. Output Capacitor In the demo ten Sanyo capacitors, model 6MV1000GX are used, with a maximum ESR equal to 69mΩ. Therefore, the resultant ESR is 69mΩ/10 = 1.9mΩ. For a load transient of 14A in worst case the drop results: ∆Vout = 14 * 0.00069 = 96.6mV The voltage drop due to the capacitor discharge during load transient, considering that the maximum duty cycle is equal to 100% results in 13mV with 2.5V of programmed output. Input Capacitor For IOUT = 14A and D = 0.5 (worst case for input ripple current), Irms is equal to 7A. Five Sanyo electrolytic capacitors 25MV330GX, with a maximum ESR equal to 69mΩ, are chosen to sustain the ripple. Therefore, the resultant ESR is equal to 69mΩ/5 = 13.8mΩ. So the losses in worst case are: 2 P = ESR ⋅ Irm s = 670mW Over-Current Protection Substituting the demo board parameters in the relationship reported in the relative section, (IOCSMIN = 170µA; IP = 19A; RDSONMAX = 9mΩ) it results that ROCS = 1kΩ. Part List R2 499k 1% R3, R7 1k 1% R4 20 SMD 0805 SMD 0805 SMD 0805 R5, R8 20k SMD 0805 R9 15k SMD 0805 R12 1K SMD 0805 R13, R14 0 C1, C2…C5 330µ SANYO – 25MV330GX C6, C7…C15 1000µ SANYO – 6MV1000GX Radial 8x20mm C16, C17, C24, C25 100n Ceramic SMD 0805 C18 2.2n Ceramic SMD 0805 C19 8.2n Ceramic SMD 0805 C20 82n Ceramic SMD 0805 SMD 0805 Radial 8x20mm C21, C22 1µ Ceramic SMD 1206 C23 1n Ceramic SMD 0805 L1 1.5µ T44-52 Core, 7T-18AWG L2 3µ T50-52B Core, 10T-16AWG U1 L6911C STMicroelectronics SO20 Q1, Q2…Q6 STS12NF30L STMicroelectronics SO8 D1 1N4148 STMicroelectronics SOT23 D2 STPS3L25U STMicroelectronics SMB F1 251015A-15° Littlefuse AXIAL 16/20 L6911C PCB AND COMPONENTS LAYOUTS Figure 14. PCB and Components Layouts Component Side Internal Ground Plane Figure 15. PCB and Components Layouts Internal Layer Solder Side 17/20 L6911C Application Circuit Examples Figure 16 reports the schematic circuit for a motherboard chipset power supply. This application works from a single 5V power supply and is able to deliver up to 10A with a 300KHz switching frequency. Figure 16. Motherboard chipset power supply; 2.5Vout, 10A C oilC raft D O3316P 1uH 10A fuse +5 VIN 1N4 148 1n OVP BOOT GN D IN 15 V CC 100nF G ND V ID0 V ID1 V ID2 V ID3 V ID4 SS O SC 19 2 18 11 14 U1 17 L6911C 6 7 16 12 3 20 C oilC raft D O5022P 1.5uH P HAS E VOU TC OR E LG AT E STS 12N F30L P GN D 1 P GO OD V S EN VFB 10 220pF 6 80pF 100k 750k 18/20 UGA TE STP S3L25U 10k 5x 47 0uF Sanyo TPB GN D C OR E 8 COMP 43k 3x 220uF Sanyo 1k STS 12N F30L 13 9 1 00n O CS E T 4 5 2 x1uF c eramic 100n 10k 5.6n 220 PW R GD L6911C mm DIM. MIN. TYP. inch MAX. MIN. TYP. MAX. A 2.35 2.65 0.093 0.104 A1 0.1 0.3 0.004 0.012 B 0.33 0.51 0.013 0.020 C 0.23 0.32 0.009 0.013 D 12.6 13 0.496 0.512 E 7.4 7.6 0.291 0.299 e 1.27 OUTLINE AND MECHANICAL DATA 0.050 H 10 10.65 0.394 0.419 h 0.25 0.75 0.010 0.030 L 0.4 1.27 0.016 0.050 SO20 K 0° (min.)8° (max.) L h x 45° A B e A1 K C H D 20 11 E 1 1 0 SO20MEC 19/20 L6911C Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. N o license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. 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