AD AD9752

a
12-Bit, 125 MSPS High Performance
TxDAC® D/A Converter
AD9752*
FEATURES
High Performance Member of Pin-Compatible
TxDAC Product Family
125 MSPS Update Rate
12-Bit Resolution
Excellent Spurious Free Dynamic Range Performance
SFDR to Nyquist @ 5 MHz Output: 79 dBc
Differential Current Outputs: 2 mA to 20 mA
Power Dissipation: 185 mW @ 5 V
Power-Down Mode: 20 mW @ 5 V
On-Chip 1.20 V Reference
CMOS-Compatible +2.7 V to +5.5 V Digital Interface
Package: 28-Lead SOIC and TSSOP
Edge-Triggered Latches
APPLICATIONS
Wideband Communication Transmit Channel:
Direct IF
Basestations
Wireless Local Loop
Digital Radio Link
Direct Digital Synthesis (DDS)
Instrumentation
PRODUCT DESCRIPTION
The AD9752 is a 12-bit resolution, wideband, second generation
member of the TxDAC series of high performance, low power
CMOS digital-to-analog-converters (DACs). The TxDAC family,
which consists of pin compatible 8-, 10-, 12-, and 14-bit DACs, is
specifically optimized for the transmit signal path of communication systems. All of the devices share the same interface options,
small outline package and pinout, thus providing an upward or
downward component selection path based on performance,
resolution and cost. The AD9752 offers exceptional ac and dc
performance while supporting update rates up to 125 MSPS.
The AD9752’s flexible single-supply operating range of 4.5 V to
5.5 V and low power dissipation are well suited for portable and
low power applications. Its power dissipation can be further
reduced to a mere 65 mW, without a significant degradation in
performance, by lowering the full-scale current output. Also, a
power-down mode reduces the standby power dissipation to
approximately 20 mW.
The AD9752 is manufactured on an advanced CMOS process.
A segmented current source architecture is combined with a
proprietary switching technique to reduce spurious components
and enhance dynamic performance. Edge-triggered input latches
and a 1.2 V temperature compensated bandgap reference have
been integrated to provide a complete monolithic DAC solution.
The digital inputs support +2.7 V to +5 V CMOS logic families.
TxDAC is a registered trademark of Analog Devices, Inc.
*Protected by U.S. Patents Numbers 5450084, 5568145, 5689257, 5612697 and
5703519. Other patents pending.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
FUNCTIONAL BLOCK DIAGRAM
+5V
REFLO
0.1mF
RSET
+5V
ACOM
AD9752
CURRENT
SOURCE
ARRAY
DVDD
DCOM
CLOCK
AVDD
150pF
+1.20V REF
REFIO
FS ADJ
SEGMENTED
SWITCHES
CLOCK
LSB
SWITCHES
ICOMP
0.1mF
IOUTA
IOUTB
LATCHES
SLEEP
DIGITAL DATA INPUTS (DB11–DB0)
The AD9752 is a current-output DAC with a nominal full-scale
output current of 20 mA and > 100 kΩ output impedance.
Differential current outputs are provided to support singleended or differential applications. Matching between the two
current outputs ensures enhanced dynamic performance in a
differential output configuration. The current outputs may be
tied directly to an output resistor to provide two complementary, single-ended voltage outputs or fed directly into a transformer. The output voltage compliance range is 1.25 V.
The on-chip reference and control amplifier are configured for
maximum accuracy and flexibility. The AD9752 can be driven
by the on-chip reference or by a variety of external reference
voltages. The internal control amplifier, which provides a wide
(>10:1) adjustment span, allows the AD9752 full-scale current
to be adjusted over a 2 mA to 20 mA range while maintaining
excellent dynamic performance. Thus, the AD9752 may operate at reduced power levels or be adjusted over a 20 dB range to
provide additional gain ranging capabilities.
The AD9752 is available in 28-lead SOIC and TSSOP packages.
It is specified for operation over the industrial temperature range.
PRODUCT HIGHLIGHTS
1. The AD9752 is a member of the wideband TxDAC product
family that provides an upward or downward component selection path based on resolution (8 to 14 bits), performance and
cost. The entire family of TxDACs is available in industry
standard pinouts.
2. Manufactured on a CMOS process, the AD9752 uses a
proprietary switching technique that enhances dynamic
performance beyond that previously attainable by higher
power/cost bipolar or BiCMOS devices.
3. On-chip, edge-triggered input CMOS latches interface readily
to +2.7 V to +5 V CMOS logic families. The AD9752 can
support update rates up to 125 MSPS.
4. A flexible single-supply operating range of 4.5 V to 5.5 V and
a wide full-scale current adjustment span of 2 mA to 20 mA
allow the AD9752 to operate at reduced power levels.
5. The current output(s) of the AD9752 can be easily configured for various single-ended or differential circuit topologies.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
© Analog Devices, Inc., 1999
AD9752–SPECIFICATIONS
DC SPECIFICATIONS (T
MIN
to TMAX, AVDD = +5 V, DVDD = +5 V, IOUTFS = 20 mA, unless otherwise noted)
Parameter
Min
RESOLUTION
Typ
Max
12
Units
Bits
1
DC ACCURACY
Integral Linearity Error (INL)
TA = +25°C
TMIN to TMAX
Differential Nonlinearity (DNL)
TA = +25°C
TMIN to TMAX
ANALOG OUTPUT
Offset Error
Gain Error (Without Internal Reference)
Gain Error (With Internal Reference)
Full-Scale Output Current2
Output Compliance Range
Output Resistance
Output Capacitance
REFERENCE OUTPUT
Reference Voltage
Reference Output Current3
REFERENCE INPUT
Input Compliance Range
Reference Input Resistance
Small Signal Bandwidth
–1.5
–2.0
± 0.5
+1.5
+2.0
LSB
LSB
–0.75
–1.0
± 0.25
+0.75
+1.0
LSB
LSB
+0.02
+2
+5
20.0
1.25
% of FSR
% of FSR
% of FSR
mA
V
kΩ
pF
1.26
V
nA
1.25
1
0.5
V
MΩ
MHz
0
± 50
± 100
± 50
ppm of FSR/°C
ppm of FSR/°C
ppm of FSR/°C
ppm/°C
–0.02
–2
–5
2.0
–1.0
± 0.5
± 1.5
100
5
1.14
1.20
100
0.1
TEMPERATURE COEFFICIENTS
Offset Drift
Gain Drift (Without Internal Reference)
Gain Drift (With Internal Reference)
Reference Voltage Drift
POWER SUPPLY
Supply Voltages
AVDD
DVDD
Analog Supply Current (IAVDD)4
Digital Supply Current (IDVDD)5
Supply Current Sleep Mode (IAVDD)6
Power Dissipation5 (5 V, IOUTFS = 20 mA)
Power Supply Rejection Ratio7—AVDD
Power Supply Rejection Ratio7—DVDD
OPERATING RANGE
4.5
2.7
5.0
5.0
34
3
4
185
–0.4
–0.025
5.5
5.5
39
5
8
220
+0.4
+0.025
V
V
mA
mA
mA
mW
% of FSR/V
% of FSR/V
–40
+85
°C
NOTES
1
Measured at IOUTA, driving a virtual ground.
2
Nominal full-scale current, I OUTFS, is 32 × the IREF current.
3
Use an external buffer amplifier to drive any external load.
4
Requires +5 V supply.
5
Measured at fCLOCK = 25 MSPS and IOUT = static full scale (20 mA).
6
Logic level for SLEEP pin must be referenced to AVDD. Min V IH = 3.5 V.
7
± 5% Power supply variation.
Specifications subject to change without notice.
–2–
REV. 0
AD9752
DYNAMIC SPECIFICATIONS
(TMIN to TMAX, AVDD = +5 V, DVDD = +5 V, IOUTFS = 20 mA, Differential Transformer Coupled Output,
50 ⍀ Doubly Terminated, unless otherwise noted)
Parameter
Min
DYNAMIC PERFORMANCE
Maximum Output Update Rate (fCLOCK)
Output Settling Time (tST) (to 0.1%)1
Output Propagation Delay (tPD)
Glitch Impulse
Output Rise Time (10% to 90%)1
Output Fall Time (10% to 90%)1
Output Noise (IOUTFS = 20 mA)
Output Noise (IOUTFS = 2 mA)
Max
125
AC LINEARITY
Spurious-Free Dynamic Range to Nyquist
fCLOCK = 25 MSPS; fOUT = 1.00 MHz
0 dBFS Output
TA = +25°C
–6 dBFS Output
–12 dBFS Output
fCLOCK = 50 MSPS; fOUT = 1.00 MHz
fCLOCK = 50 MSPS; fOUT = 2.51 MHz
fCLOCK = 50 MSPS; fOUT = 5.02 MHz
fCLOCK = 50 MSPS; fOUT = 14.02 MHz
fCLOCK = 50 MSPS; fOUT = 20.2 MHz
fCLOCK = 100 MSPS; fOUT = 2.5 MHz
fCLOCK = 100 MSPS; fOUT = 5 MHz
fCLOCK = 100 MSPS; fOUT = 20 MHz
fCLOCK = 100 MSPS; fOUT = 40 MHz
Spurious-Free Dynamic Range within a Window
fCLOCK = 25 MSPS; fOUT = 1.00 MHz
fCLOCK = 50 MSPS; fOUT = 5.02 MHz; 2 MHz Span
fCLOCK = 100 MSPS; fOUT = 5.04 MHz; 4 MHz Span
Total Harmonic Distortion
fCLOCK = 25 MSPS; fOUT = 1.00 MHz
TA = +25°C
fCLOCK = 50 MHz; fOUT = 2.00 MHz
fCLOCK = 100 MHz; fOUT = 2.00 MHz
Multitone Power Ratio (8 Tones at 110 kHz Spacing)
fCLOCK = 20 MSPS; fOUT = 2.00 MHz to 2.99 MHz
0 dBFS Output
–6 dBFS Output
–12 dBFS Output
–18 dBFS Output
MSPS
ns
ns
pV-s
ns
ns
pA/√Hz
pA/√Hz
75
84
76
81
81
81
76
62
60
78
76
63
55
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
84
93
86
86
dBc
dBc
dBc
81
81
85
86
Specifications subject to change without notice.
–3–
Units
35
1
5
2.5
2.5
50
30
–82
–76
–76
NOTES
1
Measured single ended into 50 Ω load.
REV. 0
Typ
–74
dBc
dBc
dBc
dBc
dBc
dBc
dBc
AD9752
DIGITAL SPECIFICATIONS (T
MIN
to TMAX, AVDD = +5 V, DVDD = +5 V, IOUTFS = 20 mA, unless otherwise noted)
Parameter
DIGITAL INPUTS
Logic “1” Voltage @ DVDD = +5 V1
Logic “1” Voltage @ DVDD = +3 V
Logic “0” Voltage @ DVDD = +5 V1
Logic “0” Voltage @ DVDD = +3 V
Logic “1” Current
Logic “0” Current
Input Capacitance
Input Setup Time (tS)
Input Hold Time (tH)
Latch Pulsewidth (tLPW)
Min
Typ
3.5
2.1
5
3
0
0
Max
Units
V
V
V
V
µA
µA
pF
ns
ns
ns
1.3
0.9
+10
+10
–10
–10
5
2.0
1.5
3.5
NOTES
1
When DVDD = +5 V and Logic 1 voltage ≈3.5 V and Logic 0 voltage ≈1.3 V. IVDD can increase by up to 10 mA, depending on f CLOCK.
Specifications subject to change without notice.
DB0–DB11
tS
tH
CLOCK
tLPW
tPD
tST
IOUTA
OR
IOUTB
0.1%
0.1%
Figure 1. Timing Diagram
ABSOLUTE MAXIMUM RATINGS*
Parameter
AVDD
DVDD
ACOM
AVDD
CLOCK, SLEEP
Digital Inputs
IOUTA, IOUTB
ICOMP
REFIO, FSADJ
REFLO
Junction Temperature
Storage Temperature
Lead Temperature
(10 sec)
ORDERING GUIDE
With
Respect to
Min
Max
Units
ACOM
DCOM
DCOM
DVDD
DCOM
DCOM
ACOM
ACOM
ACOM
ACOM
–0.3
–0.3
–0.3
–6.5
–0.3
–0.3
–1.0
–0.3
–0.3
–0.3
+6.5
+6.5
+0.3
+6.5
DVDD + 0.3
DVDD + 0.3
AVDD + 0.3
AVDD + 0.3
AVDD + 0.3
+0.3
+150
+150
V
V
V
V
V
V
V
V
V
V
°C
°C
+300
°C
–65
Model
Temperature
Range
Package
Description
Package
Options*
AD9752AR –40°C to +85°C 28-Lead 300 Mil SOIC R-28
AD9752ARU –40°C to +85°C 28-Lead TSSOP
RU-28
AD9752-EB
Evaluation Board
*R = Small Outline IC; RU = Thin Shrink Small Outline Package.
THERMAL CHARACTERISTICS
Thermal Resistance
28-Lead 300 Mil SOIC
θJA = 71.4°C/W
θJC = 23°C/W
28-Lead TSSOP
θJA = 97.9°C/W
θJC = 14.0°C/W
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
sections of this specification is not implied. Exposure to absolute maximum
ratings for extended periods may effect device reliability.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD9752 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
–4–
WARNING!
ESD SENSITIVE DEVICE
REV. 0
AD9752
PIN CONFIGURATION
(MSB) DB11 1
28 CLOCK
DB10 2
27 DVDD
DB9 3
26 DCOM
DB8 4
25 NC
DB7 5
24 AVDD
AD9752
DB6 6
TOP VIEW 23 ICOMP
DB5 7 (Not to Scale) 22 IOUTA
DB4 8
21 IOUTB
DB3 9
20 ACOM
DB2 10
19 NC
DB1 11
18 FS ADJ
DB0 12
17 REFIO
NC 13
16 REFLO
NC 14
15 SLEEP
NC = NO CONNECT
PIN FUNCTION DESCRIPTIONS
Pin No.
Name
1
2–11
12
13, 14,
19, 25
15
DB11
Most Significant Data Bit (MSB).
DB10–DB1 Data Bits 1–10.
DB0
Least Significant Data Bit (LSB).
16
17
REFLO
REFIO
18
19
20
21
22
23
24
26
27
28
FS ADJ
NC
ACOM
IOUTB
IOUTA
ICOMP
AVDD
DCOM
DVDD
CLOCK
REV. 0
NC
SLEEP
Description
No Internal Connection.
Power-Down Control Input. Active High. Contains active pull-down circuit, thus may be left unterminated
if not used.
Reference Ground when Internal 1.2 V Reference Used. Connect to AVDD to disable internal reference.
Reference Input/Output. Serves as reference input when internal reference disabled (i.e., Tie REFLO to
AVDD). Serves as 1.2 V reference output when internal reference activated (i.e., Tie REFLO to ACOM).
Requires 0.1 µF capacitor to ACOM when internal reference activated.
Full-Scale Current Output Adjust.
No Connect.
Analog Common.
Complementary DAC Current Output. Full-scale current when all data bits are 0s.
DAC Current Output. Full-scale current when all data bits are 1s.
Internal Bias Node for Switch Driver Circuitry. Decouple to ACOM with 0.1 µF capacitor.
Analog Supply Voltage (+4.5 V to +5.5 V).
Digital Common.
Digital Supply Voltage (+2.7 V to +5.5 V).
Clock Input. Data latched on positive edge of clock.
–5–
AD9752
DEFINITIONS OF SPECIFICATIONS
Linearity Error (Also Called Integral Nonlinearity or INL)
Power Supply Rejection
The maximum change in the full-scale output as the supplies
are varied from nominal to minimum and maximum specified
voltages.
Linearity error is defined as the maximum deviation of the
actual analog output from the ideal output, determined by a
straight line drawn from zero to full scale.
Settling Time
DNL is the measure of the variation in analog value, normalized
to full scale, associated with a 1 LSB change in digital input code.
The time required for the output to reach and remain within a
specified error band about its final value, measured from the
start of the output transition.
Monotonicity
Glitch Impulse
A D/A converter is monotonic if the output either increases or
remains constant as the digital input increases.
Asymmetrical switching times in a DAC give rise to undesired
output transients that are quantified by a glitch impulse. It is
specified as the net area of the glitch in pV-s.
Differential Nonlinearity (or DNL)
Offset Error
The deviation of the output current from the ideal of zero is
called offset error. For IOUTA, 0 mA output is expected when
the inputs are all 0s. For IOUTB, 0 mA output is expected
when all inputs are set to 1s.
Spurious-Free Dynamic Range
Gain Error
THD is the ratio of the rms sum of the first six harmonic
components to the rms value of the measured input signal. It is
expressed as a percentage or in decibels (dB).
The difference, in dB, between the rms amplitude of the output
signal and the peak spurious signal over the specified bandwidth.
Total Harmonic Distortion
The difference between the actual and ideal output span. The
actual span is determined by the output when all inputs are set
to 1s minus the output when all inputs are set to 0s.
Multitone Power Ratio
The spurious-free dynamic range for an output containing multiple carrier tones of equal amplitude. It is measured as the
difference between the rms amplitude of a carrier tone to the
peak spurious signal in the region of a removed tone.
Output Compliance Range
The range of allowable voltage at the output of a current-output
DAC. Operation beyond the maximum compliance limits may
cause either output stage saturation or breakdown resulting in
nonlinear performance.
Temperature Drift
Temperature drift is specified as the maximum change from the
ambient (+25°C) value to the value at either TMIN or TMAX. For
offset and gain drift, the drift is reported in ppm of full-scale
range (FSR) per °C. For reference drift, the drift is reported
in ppm per °C.
+5V
REFLO
AVDD
150pF
0.1mF
REFIO
PMOS
CURRENT SOURCE
ARRAY
FS ADJ
RSET
2kV
+5V
DVDD
CLOCK
50V
0.1mF
MINI-CIRCUITS
T1-1T
LSB
SWITCHES
100V
IOUTB
LATCHES
50V
SLEEP
RETIMED
CLOCK
OUTPUT*
LECROY 9210
PULSE GENERATOR
ICOMP
IOUTA
SEGMENTED SWITCHES
FOR DB11–DB3
DCOM
DVDD
DCOM
ACOM
AD9752
+1.20V REF
50V
CLOCK
OUTPUT
TO HP3589A
SPECTRUM/
NETWORK
ANALYZER
50V INPUT
20pF
20pF
DIGITAL
DATA
* AWG2021 CLOCK RETIMED
SUCH THAT DIGITAL DATA
TRANSITIONS ON FALLING EDGE
OF 50% DUTY CYCLE CLOCK.
TEKTRONIX
AWG-2021
W/OPTION 4
Figure 2. Basic AC Characterization Test Setup
–6–
REV. 0
AD9752
Typical AC Characterization Curves @ +5 V Supplies
(AVDD = +5 V, DVDD = +5 V, IOUTFS = 20 mA, 50 ⍀ Doubly Terminated Load, Differential Output, TA = +25ⴗC, SFDR up to Nyquist, unless otherwise noted)
90
–12dBFS
70
SFDR – dB
80
125MSPS
65MSPS
60
80
–6dBFS
SFDR – dBc
80
SFDR – dB
90
90
50MSPS
25MSPS
70
0dBFS
60
0dBFS
–6dBFS
70
60
–12dBFS
50
40
50
50
0
10
1
40
100
0
2
4
6
8
fOUT – MHz
fOUT – MHz
Figure 3. SFDR vs. fOUT @ 0 dBFS
10
12
90
80
80
0
10
5
15
20
25
fOUT – MHz
Figure 4. SFDR vs. fOUT @ 25 MSPS
90
40
14
Figure 5. SFDR vs. fOUT @ 50 MSPS
90
–12dBFS
70
0dBFS
60
–6dBFS
70
80
SFDR – dBc
–6dBFS
SFDR – dB
SFDR – dBc
10mA FS
0dBFS
60
20mA FS
5mA FS
70
–12dBFS
40
60
50
50
40
0
10
5
15
20
25
30
0
10
20
30
40
50
60
50
0
4
2
6
fOUT – MHz
fOUT – MHz
Figure 6. SFDR vs. fOUT @ 65 MSPS
Figure 7. SFDR vs. fOUT @ 125 MSPS
10
12
Figure 8. SFDR vs. fOUT and IOUTFS
@ 25 MSPS and 0 dBFS
90
90
8
fOUT – MHz
80
[email protected]
[email protected]
[email protected]
80
80
20mA FS
[email protected]
70
[email protected]
60
70
SNR – dB
SFDR – dB
SFDR – dB
70
60
5mA FS
10mA FS
60
[email protected]
[email protected]
50
50
[email protected]
40
–30
–25
–20
–15
–10
AOUT – dBFS
–5
0
Figure 9. Single-Tone SFDR vs. AOUT
@ fOUT = fCLOCK/11
REV. 0
40
–30
–25
–20
–15
–10
AOUT – dBFS
–5
Figure 10. Single-Tone SFDR vs.
AOUT @ fOUT = fCLOCK/5
–7–
0
50
0
20
40
60
80
100
120
fCLOCK – MSPS
Figure 11. SNR vs. fCLOCK and IOUTFS
@ fOUT = 2 MHz and 0 dBFS
AD9752
0.1
0.7
90
0.6
0.0
0.5
0.3
0.2
0.1
0.0
–0.1
SFDR – dBc
ERROR – LSB
ERROR – LSB
fOUT = 4MHz
80
0.4
–0.2
fOUT = 10MHz
70
fOUT = 29MHz
–0.3
–0.1
60
–0.2
–0.4
fOUT = 40MHz
–0.3
–0.4
–0.5
0
1000
2000
CODE
4000
3000
0
2000
CODE
3000
4000
Figure 13. Typical DNL
Figure 12. Typical INL
50
–55
–30
–5
20
45
TEMPERATURE – 8C
70
95
Figure 14. SFDR vs. Temperature @
125 MSPS, 0 dBFS
0
0
fCLK = 125MSPS
fOUT1 = 13.5MHz
fOUT2 = 14.5MHz
–20
AOUT = 0dBFS
SFDR = 68.4dBc
–30
–30
fCLK = 65MSPS
fOUT1 = 6.25MHz
fOUT2 = 6.75MHz
fOUT3 = 7.25MHz
fOUT4 = 7.75MHz
–40
SFDR = 69dBc
AMPLITUDE = 0dBFS
–10
SIGNAL AMPLITUDE – dBm
–10
SIGNAL AMPLITUDE – dBm
1000
–40
–50
–60
–70
–80
–20
–50
–60
–70
–80
–90
–90
–100
–100
0
10
20
30
40
50
fOUT – MHz
Figure 15. Dual-Tone SFDR
60
0
5.0
10.0
15.0
20.0
25.0
30.0
fOUT – MHz
Figure 16. Four-Tone SFDR
–8–
REV. 0
AD9752
+5V
REFLO
AVDD
150pF
+1.20V REF
VREFIO
0.1mF
RSET
2kV
AD9752
REFIO
IREF
FS ADJ
+5V
DVDD
DCOM
CLOCK
CLOCK
ACOM
PMOS
CURRENT SOURCE
ARRAY
ICOMP
0.1mF
VDIFF = VOUTA – VOUTB
IOUTA
IOUTA
SEGMENTED SWITCHES
FOR DB11–DB3
LSB
SWITCHES
IOUTB
IOUTB
VOUTA
VOUTB
RLOAD
50V
LATCHES
SLEEP
RLOAD
50V
DIGITAL DATA INPUTS (DB11–DB0)
Figure 17. Functional Block Diagram
IOUTB = (4095 – DAC CODE)/4096 × IOUTFS
FUNCTIONAL DESCRIPTION
Figure 17 shows a simplified block diagram of the AD9752.
The AD9752 consists of a large PMOS current source array that
is capable of providing up to 20 mA of total current. The array
is divided into 31 equal currents that make up the five most
significant bits (MSBs). The next four bits or middle bits consist
of 15 equal current sources whose value is 1/16th of an MSB
current source. The remaining LSBs are binary weighted fractions of the middle-bits current sources. Implementing the
middle and lower bits with current sources, instead of an R-2R
ladder, enhances its dynamic performance for multitone or low
amplitude signals and helps maintain the DAC’s high output
impedance (i.e., >100 kΩ).
where DAC CODE = 0 to 4095 (i.e., Decimal Representation).
As mentioned previously, IOUTFS is a function of the reference
current IREF, which is nominally set by a reference voltage
VREFIO and external resistor RSET. It can be expressed as:
IOUTFS = 32 × IREF
where IREF = VREFIO/RSET
(4)
VOUTA = IOUTA × RLOAD
(5)
VOUTB = IOUTB × RLOAD
(6)
Note the full-scale value of VOUTA and VOUTB should not exceed
the specified output compliance range to maintain specified
distortion and linearity performance.
The analog and digital sections of the AD9752 have separate
power supply inputs (i.e., AVDD and DVDD). The digital
section, which is capable of operating up to a 125 MSPS clock
rate and over a +2.7 V to +5.5 V operating range, consists of
edge-triggered latches and segment decoding logic circuitry.
The analog section, which can operate over a +4.5 V to +5.5 V
range, includes the PMOS current sources, the associated differential switches, a 1.20 V bandgap voltage reference and a reference control amplifier.
The differential voltage, VDIFF, appearing across IOUTA and
IOUTB is:
VDIFF = (IOUTA – IOUTB) × RLOAD
(7)
Substituting the values of IOUTA, IOUTB, and IREF; VDIFF can be
expressed as:
VDIFF = {(2 DAC CODE – 4095)/4096} ×
(32 RLOAD/RSET) × VREFIO
The full-scale output current is regulated by the reference control amplifier and can be set from 2 mA to 20 mA via an external resistor, RSET. The external resistor, in combination with
both the reference control amplifier and voltage reference VREFIO,
sets the reference current IREF, which is mirrored over to the
segmented current sources with the proper scaling factor. The
full-scale current, IOUTFS, is thirty-two times the value of IREF.
(8)
These last two equations highlight some of the advantages of
operating the AD9752 differentially. First, the differential operation will help cancel common-mode error sources associated
with IOUTA and IOUTB such as noise, distortion and dc offsets.
Second, the differential code dependent current and subsequent
voltage, VDIFF, is twice the value of the single-ended voltage
output (i.e., VOUTA or VOUTB), thus providing twice the signal
power to the load.
DAC TRANSFER FUNCTION
The AD9752 provides complementary current outputs, IOUTA
and IOUTB. IOUTA will provide a near full-scale current output,
IOUTFS, when all bits are high (i.e., DAC CODE = 4095) while
IOUTB, the complementary output, provides no current. The
current output appearing at IOUTA and IOUTB is a function
of both the input code and IOUTFS and can be expressed as:
REV. 0
(3)
The two current outputs will typically drive a resistive load
directly or via a transformer. If dc coupling is required, IOUTA
and IOUTB should be directly connected to matching resistive
loads, RLOAD, which are tied to analog common, ACOM. Note,
RLOAD may represent the equivalent load resistance seen by
IOUTA or IOUTB as would be the case in a doubly terminated
50 Ω or 75 Ω cable. The single-ended voltage output appearing
at the IOUTA and IOUTB nodes is simply :
All of these current sources are switched to one or the other of
the two output nodes (i.e., IOUTA or IOUTB) via PMOS
differential current switches. The switches are based on a new
architecture that drastically improves distortion performance.
This new switch architecture reduces various timing errors and
provides matching complementary drive signals to the inputs of
the differential current switches.
IOUTA = (DAC CODE/4096) × IOUTFS
(2)
Note, the gain drift temperature performance for a single-ended
(VOUTA and VOUTB) or differential output (VDIFF) of the AD9752
can be enhanced by selecting temperature tracking resistors for
RLOAD and RSET due to their ratiometric relationship as shown
in Equation 8.
(1)
–9–
AD9752
REFERENCE OPERATION
REFERENCE CONTROL AMPLIFIER
The AD9752 contains an internal 1.20 V bandgap reference
that can easily be disabled and overridden by an external reference. REFIO serves as either an input or output depending on
whether the internal or an external reference is selected. If
REFLO is tied to ACOM, as shown in Figure 18, the internal
reference is activated and REFIO provides a 1.20 V output. In
this case, the internal reference must be compensated externally
with a ceramic chip capacitor of 0.1 µF or greater from REFIO
to REFLO. Also, REFIO should be buffered with an external
amplifier having an input bias current less than 100 nA if any
additional loading is required.
The AD9752 also contains an internal control amplifier that is
used to regulate the DAC’s full-scale output current, IOUTFS.
The control amplifier is configured as a V-I converter as shown
in Figure 19, such that its current output, IREF, is determined by
the ratio of the VREFIO and an external resistor, RSET, as stated
in Equation 4. IREF is copied over to the segmented current
sources with the proper scaling factor to set IOUTFS as stated in
Equation 3.
The control amplifier allows a wide (10:1) adjustment span of
IOUTFS over a 2 mA to 20 mA range by setting IREF between
62.5 µA and 625 µA. The wide adjustment span of IOUTFS
provides several application benefits. The first benefit relates
directly to the power dissipation of the AD9752, which is
proportional to IOUTFS (refer to the Power Dissipation section).
The second benefit relates to the 20 dB adjustment, which is
useful for system gain control purposes.
+5V
OPTIONAL
EXTERNAL
REF BUFFER
REFLO
AVDD
150pF
+1.2V REF
REFIO
ADDITIONAL
LOAD
0.1mF
CURRENT
SOURCE
ARRAY
FS ADJ
2kV
AD9752
Figure 18. Internal Reference Configuration
The internal reference can be disabled by connecting REFLO to
AVDD. In this case, an external reference may then be applied
to REFIO as shown in Figure 19. The external reference may
provide either a fixed reference voltage to enhance accuracy and
drift performance or a varying reference voltage for gain control.
Note that the 0.1 µF compensation capacitor is not required
since the internal reference is disabled, and the high input impedance (i.e., 1 MΩ) of REFIO minimizes any loading of the
external reference.
AVDD
REFLO
AVDD
150pF
AVDD
The small signal bandwidth of the reference control amplifier is
approximately 0.5 MHz. The output of the control amplifier is
internally compensated via a 150 pF capacitor that limits the
control amplifier small-signal bandwidth and reduces its
output impedance. Since the –3 dB bandwidth corresponds to
the dominant pole, and hence the time constant, the settling
time of the control amplifier to a stepped reference input response can be approximated. In this case, the time constant can
be approximated to be 320 ns.
There are two methods in which IREF can be varied for a fixed
RSET. The first method is suitable for a single-supply system in
which the internal reference is disabled, and the common-mode
voltage of REFIO is varied over its compliance range of 1.25 V
to 0.10 V. REFIO can be driven by a single-supply amplifier or
DAC, thus allowing IREF to be varied for a fixed RSET. Since the
input impedance of REFIO is approximately 1 MΩ, a simple,
low cost R-2R ladder DAC configured in the voltage mode
topology may be used to control the gain. This circuit is shown
in Figure 20 using the AD7524 and an external 1.2 V reference,
the AD1580.
+1.2V REF
VREFIO
EXTERNAL
REF
REFIO
CURRENT
SOURCE
ARRAY
FS ADJ
RSET
IREF =
VREFIO/RSET
REFERENCE
CONTROL
AMPLIFIER
AD9752
Figure 19. External Reference Configuration
AVDD
AVDD
REFLO
RFB
1.2V
AD1580
OUT1
VREF
0.1V TO 1.2V
REFIO
FS ADJ
OUT2
AGND
AVDD
+1.2V REF
VDD
AD7524
150pF
RSET
IREF =
VREF/RSET
CURRENT
SOURCE
ARRAY
AD9752
DB7–DB0
Figure 20. Single-Supply Gain Control Circuit
–10–
REV. 0
AD9752
The second method may be used in a dual-supply system in
which the common-mode voltage of REFIO is fixed and IREF is
varied by an external voltage, VGC, applied to RSET via an amplifier. An example of this method is shown in Figure 21, in which
the internal reference is used to set the common-mode voltage
of the control amplifier to 1.20 V. The external voltage, VGC, is
referenced to ACOM and should not exceed 1.2 V. The value
of RSET is such that IREFMAX and IREFMIN do not exceed 62.5 µA
and 625 µA, respectively. The associated equations in Figure 21
can be used to determine the value of RSET.
AVDD
REFLO
For applications requiring the optimum dc linearity, IOUTA
and/or IOUTB should be maintained at a virtual ground via an
I-V op amp configuration. Maintaining IOUTA and/or IOUTB
at a virtual ground keeps the output impedance of the AD9752
fixed, significantly reducing its effect on linearity. However,
it does not necessarily lead to the optimum distortion performance due to limitations of the I-V op amp. Note that the
INL/DNL specifications for the AD9752 are measured in
this manner using IOUTA. In addition, these dc linearity
specifications remain virtually unaffected over the specified
power supply range of 4.5 V to 5.5 V.
AVDD
150pF
+1.2V REF
REFIO
RSET
VGC
CURRENT
SOURCE
ARRAY
FS ADJ
1mF
IREF
AD9752
IREF = (1.2–VGC)/RSET
WITH VGC < VREFIO AND 62.5mA # IREF # 625A
Figure 21. Dual-Supply Gain Control Circuit
ANALOG OUTPUTS
The AD9752 produces two complementary current outputs,
IOUTA and IOUTB, which may be configured for single-end
or differential operation. IOUTA and IOUTB can be converted
into complementary single-ended voltage outputs, VOUTA and
VOUTB, via a load resistor, RLOAD, as described in the DAC
Transfer Function section by Equations 5 through 8. The
differential voltage, VDIFF, existing between VOUTA and VOUTB
can also be converted to a single-ended voltage via a transformer
or differential amplifier configuration.
Figure 22 shows the equivalent analog output circuit of the
AD9752 consisting of a parallel combination of PMOS differential current switches associated with each segmented current
source. The output impedance of IOUTA and IOUTB is determined by the equivalent parallel combination of the PMOS
switches and is typically 100 kΩ in parallel with 5 pF. Due to
the nature of a PMOS device, the output impedance is also
slightly dependent on the output voltage (i.e., VOUTA and VOUTB)
and, to a lesser extent, the analog supply voltage, AVDD, and
full-scale current, IOUTFS. Although the output impedance’s
signal dependency can be a source of dc nonlinearity and ac linearity (i.e., distortion), its effects can be limited if certain precautions are noted.
AVDD
IOUTA
IOUTB
RLOAD
RLOAD
Figure 22. Equivalent Analog Output
REV. 0
IOUTA and IOUTB also have a negative and positive voltage
compliance range. The negative output compliance range of
–1.0 V is set by the breakdown limits of the CMOS process.
Operation beyond this maximum limit may result in a breakdown of the output stage and affect the reliability of the AD9752.
The positive output compliance range is slightly dependent on
the full-scale output current, IOUTFS. It degrades slightly from its
nominal 1.25 V for an IOUTFS = 20 mA to 1.00 V for an IOUTFS =
2 mA. Operation beyond the positive compliance range will
induce clipping of the output signal which severely degrades
the AD9752’s linearity and distortion performance.
Operating the AD9752 with reduced voltage output swings at
IOUTA and IOUTB in a differential or single-ended output
configuration reduces the signal dependency of its output
impedance thus enhancing distortion performance. Although
the voltage compliance range of IOUTA and IOUTB extends
from –1.0 V to +1.25 V, optimum distortion performance is
achieved when the maximum full-scale signal at IOUTA and
IOUTB does not exceed approximately 0.5 V. A properly selected transformer with a grounded center-tap will allow the
AD9752 to provide the required power and voltage levels to
different loads while maintaining reduced voltage swings at
IOUTA and IOUTB. DC-coupled applications requiring a
differential or single-ended output configuration should size
RLOAD accordingly. Refer to Applying the AD9752 section for
examples of various output configurations.
The most significant improvement in the AD9752’s distortion
and noise performance is realized using a differential output
configuration. The common-mode error sources of both
IOUTA and IOUTB can be substantially reduced by the
common-mode rejection of a transformer or differential amplifier. These common-mode error sources include evenorder distortion products and noise. The enhancement in
distortion performance becomes more significant as the reconstructed waveform’s frequency content increases and/or its
amplitude decreases.
The distortion and noise performance of the AD9752 is also
slightly dependent on the analog and digital supply as well as the
full-scale current setting, IOUTFS. Operating the analog supply at
5.0 V ensures maximum headroom for its internal PMOS current
sources and differential switches leading to improved distortion
performance. Although IOUTFS can be set between 2 mA and
20 mA, selecting an IOUTFS of 20 mA will provide the best distortion and noise performance also shown in Figure 8. The
noise performance of the AD9752 is affected by the digital supply (DVDD), output frequency, and increases with increasing
clock rate as shown in Figure 11. Operating the AD9752 with
low voltage logic levels between 3 V and 3.3 V will slightly reduce the amount of on-chip digital noise.
–11–
AD9752
data interface circuitry should be specified to meet the minimum setup and hold times of the AD9752 as well as its required min/max input logic level thresholds. Typically, the
selection of the slowest logic family that satisfies the above conditions will result in the lowest data feedthrough and noise.
In summary, the AD9752 achieves the optimum distortion and
noise performance under the following conditions:
(1) Differential Operation.
(2) Positive voltage swing at IOUTA and IOUTB limited to
+0.5 V.
Digital signal paths should be kept short and run lengths
matched to avoid propagation delay mismatch. The insertion of
a low value resistor network (i.e., 20 Ω to 100 Ω) between the
AD9752 digital inputs and driver outputs may be helpful in reducing any overshooting and ringing at the digital inputs that contribute to data feedthrough. For longer run lengths and high data
update rates, strip line techniques with proper termination resistors should be considered to maintain “clean” digital inputs. Also,
operating the AD9752 with reduced logic swings and a corresponding digital supply (DVDD) will also reduce data feedthrough.
(3) IOUTFS set to 20 mA.
(4) Analog Supply (AVDD) set at 5.0 V.
(5) Digital Supply (DVDD) set at 3.0 V to 3.3 V with appropriate logic levels.
Note that the ac performance of the AD9752 is characterized
under the above mentioned operating conditions.
DIGITAL INPUTS
The AD9752’s digital input consists of 12 data input pins and a
clock input pin. The 12-bit parallel data inputs follow standard
positive binary coding where DB11 is the most significant bit
(MSB) and DB0 is the least significant bit (LSB). IOUTA
produces a full-scale output current when all data bits are at
Logic 1. IOUTB produces a complementary output with the
full-scale current split between the two outputs as a function of
the input code.
The external clock driver circuitry should provide the AD9752
with a low jitter clock input meeting the min/max logic levels
while providing fast edges. Fast clock edges will help minimize
any jitter that will manifest itself as phase noise on a reconstructed waveform. Thus, the clock input should be driven by
the fastest logic family suitable for the application.
Note, the clock input could also be driven via a sine wave, which is
centered around the digital threshold (i.e., DVDD/2), and meets
the min/max logic threshold. This will typically result in a slight
degradation in the phase noise, which becomes more noticeable
at higher sampling rates and output frequencies. Also, at higher
sampling rates, the 20% tolerance of the digital logic threshold
should be considered since it will affect the effective clock duty
cycle and subsequently cut into the required data setup and
hold times.
The digital interface is implemented using an edge-triggered
master slave latch. The DAC output is updated following the
rising edge of the clock as shown in Figure 1 and is designed to
support a clock rate as high as 125 MSPS. The clock can be
operated at any duty cycle that meets the specified latch pulsewidth. The setup and hold times can also be varied within the
clock cycle as long as the specified minimum times are met;
although the location of these transition edges may affect digital
feedthrough and distortion performance. Best performance is
typically achieved when the input data transitions on the falling edge
of a 50% duty cycle clock.
INPUT CLOCK/DATA TIMING RELATIONSHIP
The digital inputs are CMOS compatible with logic thresholds,
VTHRESHOLD set to approximately half the digital positive supply
(DVDD) or
VTHRESHOLD = DVDD/2 (± 20%)
68
64
FS = 65MSPS
60
SNR – dB
The internal digital circuitry of the AD9752 is capable of operating
over a digital supply range of 2.7 V to 5.5 V. As a result, the
digital inputs can also accommodate TTL levels when DVDD is
set to accommodate the maximum high level voltage of the TTL
drivers VOH(MAX). A DVDD of 3 V to 3.3 V will typically ensure
proper compatibility with most TTL logic families. Figure 23
shows the equivalent digital input circuit for the data and clock
inputs. The sleep mode input is similar with the exception that
it contains an active pull-down circuit, thus ensuring that the
AD9752 remains enabled if this input is left disconnected.
SNR in a DAC is dependent on the relationship between the
position of the clock edges and the point in time at which the
input data changes. The AD9752 is positive edge triggered, and
so exhibits SNR sensitivity when the data transition is close to
this edge. In general, the goal when applying the AD9752 is to
make the data transitions shortly after the positive clock edge.
This becomes more important as the sample rate increases. Figure
24 shows the relationship of SNR to clock placement with different sample rates and different frequencies out. Note that at
the lower sample rates, much more tolerance is allowed in clock
placement, while at higher rates, much more care must be taken.
DVDD
56
52
FS = 125MSPS
DIGITAL
INPUT
48
44
40
Figure 23. Equivalent Digital Input
Since the AD9752 is capable of being updated up to 125 MSPS,
the quality of the clock and data input signals are important in
achieving the optimum performance. The drivers of the digital
–12–
–8
–6
–4
–2
0
2
4
6
TIME OF DATA CHANGE RELATIVE TO
RISING CLOCK EDGE – ns
8
10
Figure 24. SNR vs. Clock Placement @ fOUT = 10 MHz
REV. 0
AD9752
SLEEP MODE OPERATION
8
125MSPS
The AD9752 has a power-down function which turns off the
output current and reduces the supply current to less than
8.5 mA over the specified supply range of 2.7 V to 5.5 V and
temperature range. This mode can be activated by applying a
logic level “1” to the SLEEP pin. This digital input also contains an active pull-down circuit that ensures the AD9752 remains enabled if this input is left disconnected. The AD9752
takes less than 50 ns to power down and approximately 5 µs to
power back up.
IDVDD – mA
6
100MSPS
4
50MSPS
2
25MSPS
POWER DISSIPATION
The power dissipation, PD, of the AD9752 is dependent on
several factors which include: (1) AVDD and DVDD, the
power supply voltages; (2) IOUTFS, the full-scale current output;
(3) fCLOCK, the update rate; (4) and the reconstructed digital
input waveform. The power dissipation is directly proportional
to the analog supply current, IAVDD, and the digital supply current, IDVDD. IAVDD is directly proportional to IOUTFS as shown in
Figure 25 and is insensitive to fCLOCK.
Conversely, IDVDD is dependent on both the digital input waveform, fCLOCK, and digital supply DVDD. Figures 26 and 27
show IDVDD as a function of full-scale sine wave output ratios
(fOUT/fCLOCK) for various update rates with DVDD = 5 V and
DVDD = 3 V, respectively. Note, how IDVDD is reduced by more
than a factor of 2 when DVDD is reduced from 5 V to 3 V.
35
30
IAVDD – mA
25
20
15
10
5
2
4
6
8
10
12
IOUTFS – mA
14
16
18
20
Figure 25. IAVDD vs. IOUTFS
125MSPS
14
100MSPS
IDVDD – mA
12
10
8
Figure 27. IDVDD vs. Ratio @ DVDD = 3 V
APPLYING THE AD9752
OUTPUT CONFIGURATIONS
The following sections illustrate some typical output configurations for the AD9752. Unless otherwise noted, it is assumed
that IOUTFS is set to a nominal 20 mA. For applications requiring the optimum dynamic performance, a differential output
configuration is suggested. A differential output configuration
may consist of either an RF transformer or a differential op amp
configuration. The transformer configuration provides the optimum high frequency performance and is recommended for any
application allowing for ac coupling. The differential op amp
configuration is suitable for applications requiring dc coupling, a
bipolar output, signal gain and/or level shifting.
A single-ended output is suitable for applications requiring a
unipolar voltage output. A positive unipolar output voltage will
result if IOUTA and/or IOUTB is connected to an appropriately sized load resistor, RLOAD, referred to ACOM. This configuration may be more suitable for a single-supply system
requiring a dc coupled, ground referred output voltage. Alternatively, an amplifier could be configured as an I-V converter thus
converting IOUTA or IOUTB into a negative unipolar voltage.
This configuration provides the best dc linearity since IOUTA
or IOUTB is maintained at a virtual ground. Note, IOUTA
provides slightly better performance than IOUTB.
50MSPS
25MSPS
2
5MSPS
0
0.01
0.1
1
RATIO (fCLOCK/fOUT)
Figure 26. IDVDD vs. Ratio @ DVDD = 5 V
REV. 0
1
An RF transformer can be used to perform a differential-tosingle-ended signal conversion as shown in Figure 28. A
differentially coupled transformer output provides the optimum
distortion performance for output signals whose spectral content
lies within the transformer’s passband. An RF transformer such
as the Mini-Circuits T1-1T provides excellent rejection of
common-mode distortion (i.e., even-order harmonics) and noise
over a wide frequency range. It also provides electrical isolation
and the ability to deliver twice the power to the load. Transformers with different impedance ratios may also be used for
impedance matching purposes. Note that the transformer
provides ac coupling only.
16
4
5MSPS
0.1
RATIO (fCLOCK/fOUT)
DIFFERENTIAL COUPLING USING A TRANSFORMER
18
6
0
0.01
–13–
AD9752
MINI-CIRCUITS
T1-1T
500V
AD9752
IOUTA
225V
AD9752
IOUTA
RLOAD
IOUTB
COPT
OPTIONAL RDIFF
25V
Figure 28. Differential Output Using a Transformer
The center tap on the primary side of the transformer must be
connected to ACOM to provide the necessary dc current path
for both IOUTA and IOUTB. The complementary voltages
appearing at IOUTA and IOUTB (i.e., VOUTA and VOUTB)
swing symmetrically around ACOM and should be maintained
with the specified output compliance range of the AD9752. A
differential resistor, RDIFF, may be inserted in applications in
which the output of the transformer is connected to the load,
RLOAD, via a passive reconstruction filter or cable. RDIFF is determined by the transformer’s impedance ratio and provides the
proper source termination which results in a low VSWR. Note
that approximately half the signal power will be dissipated across
RDIFF.
DIFFERENTIAL USING AN OP AMP
An op amp can also be used to perform a differential to singleended conversion as shown in Figure 29. The AD9752 is configured with two equal load resistors, RLOAD, of 25 Ω. The
differential voltage developed across IOUTA and IOUTB is
converted to a single-ended signal via the differential op amp
configuration. An optional capacitor can be installed across
IOUTA and IOUTB forming a real pole in a low-pass filter.
The addition of this capacitor also enhances the op amps distortion performance by preventing the DACs high slewing output
from overloading the op amp’s input.
AVDD
1kV
Figure 31 shows the AD9752 configured to provide a unipolar
output range of approximately 0 V to +0.5 V for a doubly terminated 50 Ω cable since the nominal full-scale current, IOUTFS, of
20 mA flows through the equivalent RLOAD of 25 Ω. In this
case, RLOAD represents the equivalent load resistance seen by
IOUTA or IOUTB. The unused output (IOUTA or IOUTB)
can be connected to ACOM directly or via a matching RLOAD.
Different values of IOUTFS and RLOAD can be selected as long as
the positive compliance range is adhered to. One additional
consideration in this mode is the integral nonlinearity (INL) as
discussed in the ANALOG OUTPUT section of this data sheet.
For optimum INL performance, the single-ended, buffered
voltage output configuration is suggested.
IOUTFS = 20mA
VOUTA = 0 TO +0.5V
IOUTA
50V
50V
IOUTB
25V
Figure 31. 0 V to +0.5 V Unbuffered Voltage Output
SINGLE-ENDED, BUFFERED VOLTAGE OUTPUT
CONFIGURATION
225V
IOUTA
AD8055
Figure 32 shows a buffered single-ended output configuration in
which the op amp U1 performs an I-V conversion on the AD9752
output current. U1 maintains IOUTA (or IOUTB) at a virtual
ground, thus minimizing the nonlinear output impedance effect
on the DAC’s INL performance as discussed in the ANALOG
OUTPUT section. Although this single-ended configuration
typically provides the best dc linearity performance, its ac distortion performance at higher DAC update rates may be limited by
U1’s slewing capabilities. U1 provides a negative unipolar output voltage and its full-scale output voltage is simply the
product of RFB and IOUTFS. The full-scale output should be set
within U1’s voltage output swing capabilities by scaling IOUTFS
and/or RFB. An improvement in ac distortion performance may
result with a reduced IOUTFS since the signal current U1 will be
required to sink will be subsequently reduced.
COPT
500V
25V
25V
SINGLE-ENDED UNBUFFERED VOLTAGE OUTPUT
AD9752
AD9752
225V
1kV
Figure 30. Single-Supply DC Differential Coupled Circuit
500V
IOUTB
AD8041
225V
IOUTB
25V
Figure 29. DC Differential Coupling Using an Op Amp
The common-mode rejection of this configuration is typically
determined by the resistor matching. In this circuit, the differential op amp circuit is configured to provide some additional
signal gain. The op amp must operate off of a dual supply since
its output is approximately ± 1.0 V. A high speed amplifier such
as the AD8055 or AD9632 capable of preserving the differential
performance of the AD9752 while meeting other system level
objectives (i.e., cost, power) should be selected. The op amps
differential gain, its gain setting resistor values, and full-scale
output swing capabilities should all be considered when optimizing this circuit.
The differential circuit shown in Figure 30 provides the necessary level-shifting required in a single supply system. In this
case, AVDD which is the positive analog supply for both the
AD9752 and the op amp is also used to level-shift the differential output of the AD9752 to midsupply (i.e., AVDD/2). The
AD8041 is a suitable op amp for this application.
–14–
REV. 0
AD9752
frequency power supply noise to higher frequencies. Worst case
PSRR for either one of the differential DAC outputs will occur
when the full-scale current is directed towards that output. As a
result, the PSRR measurement in Figure 33 represents a worst
case condition in which the digital inputs remain static and the
full scale output current of 20 mA is directed to the DAC output being measured.
COPT
RFB
200V
IOUTFS = 10mA
AD9752
IOUTA
U1
VOUT = IOUTFS 3 RFB
IOUTB
200V
Figure 32. Unipolar Buffered Voltage Output
POWER AND GROUNDING CONSIDERATIONS, POWER
SUPPLY REJECTION
Many applications seek high speed and high performance under
less than ideal operating conditions. In these circuits, the implementation and construction of the printed circuit board design
is as important as the circuit design. Proper RF techniques must
be used for device selection, placement and routing as well as
power supply bypassing and grounding to ensure optimum
performance. Figures 42-47 illustrate the recommended printed
circuit board ground, power and signal plane layouts which are
implemented on the AD9752 evaluation board.
One factor that can measurably affect system performance is the
ability of the DAC output to reject dc variations or ac noise
superimposed on the analog or digital dc power distribution
(i.e., AVDD, DVDD). This is referred to as Power Supply
Rejection Ratio (PSRR). For dc variations of the power supply,
the resulting performance of the DAC directly corresponds to a
gain error associated with the DAC’s full-scale current, IOUTFS.
AC noise on the dc supplies is common in applications where
the power distribution is generated by a switching power supply.
Typically, switching power supply noise will occur over the
spectrum from tens of kHz to several MHz. PSRR vs. frequency
of the AD9752 AVDD supply, over this frequency range, is
given in Figure 33.
An example serves to illustrate the effect of supply noise on the
analog supply. Suppose a switching regulator with a switching
frequency of 250 kHz produces 10 mV rms of noise and for
simplicity sake (i.e., ignore harmonics), all of this noise is concentrated at 250 kHz. To calculate how much of this undesired
noise will appear as current noise super imposed on the DAC’s
full-scale current, IOUTFS, one must determine the PSRR in dB
using Figure 33 at 250 kHz. To calculate the PSRR for a given
RLOAD, such that the units of PSRR are converted from A/V to
V/V, adjust the curve in Figure 33 by the scaling factor 20 × Log
(RLOAD). For instance, if RLOAD is 50 Ω, the PSRR is reduced
by 34 dB (i.e., PSRR of the DAC at 1 MHz which is 74 dB in
Figure 33 becomes 40 dB VOUT/VIN).
Proper grounding and decoupling should be a primary objective
in any high speed, high resolution system. The AD9752 features
separate analog and digital supply and ground pins to optimize
the management of analog and digital ground currents in a
system. In general, AVDD, the analog supply, should be decoupled to ACOM, the analog common, as close to the chip as
physically possible. Similarly, DVDD, the digital supply, should
be decoupled to DCOM as close as physically as possible.
For those applications that require a single +5 V or +3 V supply
for both the analog and digital supply, a clean analog supply
may be generated using the circuit shown in Figure 34. The
circuit consists of a differential LC filter with separate power
supply and return lines. Lower noise can be attained using low
ESR type electrolytic and tantalum capacitors.
FERRITE
BEADS
90
TTL/CMOS
LOGIC
CIRCUITS
AVDD
100mF
ELECT.
10-22mF
TANT.
0.1mF
CER.
ACOM
PSRR – dB
80
+5V OR +3V
POWER SUPPLY
Figure 34. Differential LC Filter for Single +5 V or +3 V
Applications
70
60
0.26
0.5
0.75
FREQUENCY – MHz
1.0
Figure 33. Power Supply Rejection Ratio of AD9752
Note that the units in Figure 33 are given in units of (amps out)/
(volts in). Noise on the analog power supply has the effect of
modulating the internal switches, and therefore the output
current. The voltage noise on the dc power, therefore, will be
added in a nonlinear manner to the desired IOUT. Due to the
relative different sizes of these switches, PSRR is very code dependent. This can produce a mixing effect which can modulate low
REV. 0
Maintaining low noise on power supplies and ground is critical
to obtaining optimum results from the AD9752. If properly
implemented, ground planes can perform a host of functions on
high speed circuit boards: bypassing, shielding, current transport, etc. In mixed signal design, the analog and digital portions
of the board should be distinct from each other, with the analog
ground plane confined to the areas covering the analog signal
traces, and the digital ground plane confined to areas covering
the digital interconnects.
All analog ground pins of the DAC, reference and other analog
components should be tied directly to the analog ground plane.
The two ground planes should be connected by a path 1/8
to 1/4 inch wide underneath or within 1/2 inch of the DAC to
–15–
AD9752
maintain optimum performance. Care should be taken to ensure
that the ground plane is uninterrupted over crucial signal paths.
On the digital side, this includes the digital input lines running
to the DAC as well as any clock signals. On the analog side, this
includes the DAC output signal, reference signal and the supply
feeders.
The use of wide runs or planes in the routing of power lines is
also recommended. This serves the dual role of providing a low
series impedance power supply to the part, as well as providing
some “free” capacitive decoupling to the appropriate ground
plane. It is essential that care be taken in the layout of signal and
power ground interconnects to avoid inducing extraneous voltage drops in the signal ground paths. It is recommended that all
connections be short, direct and as physically close to the package as possible in order to minimize the sharing of conduction
paths between different currents. When runs exceed an inch in
length, strip line techniques with proper termination resistor
should be considered. The necessity and value of this resistor
will be dependent upon the logic family used.
For a more detailed discussion of the implementation and
construction of high speed, mixed signal printed circuit boards,
refer to Analog Devices’ application notes AN-280 and
AN-333.
–30
AMPLITUDE – dBm
–40
–60
–70
–80
–90
800k
FREQUENCY – Hz
1M
Figure 35a. Notch in Missing Bin at 750 kHz is Down
>60 dB. (Peak Amplitude + 0 dBm).
–30
–40
AMPLITUDE – dBm
–50
–60
–70
–80
–90
–100
–110
4.85
Very High Frequency Digital Subscriber Line (VDSL) technology is growing rapidly in applications requiring data transfer
over relatively short distances. By using QAM modulation and
transmitting the data in multiple discrete tones, high data rates
can be achieved.
As with other multitone applications, each VDSL tone is capable of transmitting a given number of bits, depending on the
signal-to-noise ratio (SNR) in a narrow band around that tone.
The tones are evenly spaced over the range of several kHz to
10 MHz. At the high frequency end of this range, performance
is generally limited by cable characteristics and environmental
factors, such as external interferers. Performance at the lower
frequencies is much more dependent on the performance of the
components in the signal chain. In addition to in-band noise,
intermodulation from other tones can also potentially interfere
with the recovery of data for a given tone. The two graphs in
Figure 35 represent a 500 tone missing bin test vector, with
frequencies evenly spaced from 400 Hz to 10 MHz. This test is
very commonly done to determine if distortion will limit the
number of bits which can be transmitted in a tone. The test
vector has a series of missing tones around 750 kHz, which is
represented in Figure 35a and a series of missing tones around
5 MHz which is represented in Figure 35b. In both cases, the
spurious free range between the transmitted tones and the empty
bins is greater than 60 dB.
Using the AD9752 for Quadrature Amplitude Modulation
(QAM)
–50
–100
600k
APPLICATIONS
VDSL Applications Using the AD9752
5
FREQUENCY – MHz
5.15
QAM is one of the most widely used digital modulation
schemes in digital communication systems. This modulation
technique can be found in FDM as well as spreadspectrum (i.e.,
CDMA) based systems. A QAM signal is a carrier frequency
that is modulated in both amplitude (i.e., AM modulation) and
phase (i.e., PM modulation). It can be generated by independently modulating two carriers of identical frequency but with a
90° phase difference. This results in an in-phase (I) carrier component and a quadrature (Q) carrier component at a 90° phase
shift with respect to the I component. The I and Q components
are then summed to provide a QAM signal at the specified carrier frequency.
A common and traditional implementation of a QAM modulator is shown in Figure 36. The modulation is performed in the
analog domain in which two DACs are used to generate the
baseband I and Q components, respectively. Each component is
then typically applied to a Nyquist filter before being applied to
a quadrature mixer. The matching Nyquist filters shape and
limit each component’s spectral envelope while minimizing
intersymbol interference. The DAC is typically updated at the
QAM symbol rate or possibly a multiple of it if an interpolating
filter precedes the DAC. The use of an interpolating filter typically eases the implementation and complexity of the analog
filter, which can be a significant contributor to mismatches in
gain and phase between the two baseband channels. A quadrature mixer modulates the I and Q components with in-phase
and quadrature phase carrier frequency and then sums the two
outputs to provide the QAM signal.
Figure 35b. Notch in Missing Bin at 5 MHz is Down
>60 dB. (Peak Amplitude + 0 dBm).
–16–
REV. 0
AD9752
subsystem. The AD6122 has functions, such as external gain
control and low distortion characteristics, needed for the superior Adjacent Channel Power (ACP) requirements of W-CDMA.
12
AD9752
DSP
OR
ASIC
0
CARRIER
FREQUENCY
TO
MIXER
S
90
CDMA
12
Carrier Division Multiple Access, or CDMA, is an air transmit/
receive scheme where the signal in the transmit path is modulated with a pseudorandom digital code (sometimes referred to
as the spreading code). The effect of this is to spread the transmitted signal across a wide spectrum. Similar to a DMT waveform, a CDMA waveform containing multiple subscribers can
be characterized as having a high peak to average ratio (i.e.,
crest factor), thus demanding highly linear components in the
transmit signal path. The bandwidth of the spectrum is defined
by the CDMA standard being used, and in operation is implemented by using a spreading code with particular characteristics.
AD9752
NYQUIST
FILTERS
QUADRATURE
MODULATOR
Figure 36. Typical Analog QAM Architecture
In this implementation, it is much more difficult to maintain
proper gain and phase matching between the I and Q channels.
The circuit implementation shown in Figure 37 helps improve
upon the matching and temperature stability characteristics
between the I and Q channels, as well as showing a path for upconversion using the AD8346 quadrature modulator. Using a
single voltage reference derived from U1 to set the gain for both
the I and Q channels will improve the gain matching and stability. RCAL can be used to compensate for any mismatch in gain
between the two channels. This mismatch may be attributed to
the mismatch between RSET1 and RSET2, effective load resistance
of each channel, and/or the voltage offset of the control amplifier in each DAC. The differential voltage outputs of U1 and U2
are fed into the respective differential inputs of the AD8346 via
matching networks.
Distortion in the transmit path can lead to power being transmitted out of the defined band. The ratio of power transmitted
in-band to out-of-band is often referred to as Adjacent Channel
Power (ACP). This is a regulatory issue due to the possibility of
interference with other signals being transmitted by air. Regulatory bodies define a spectral mask outside of the transmit band,
and the ACP must fall under this mask. If distortion in the
transmit path cause the ACP to be above the spectral mask,
then filtering, or different component selection is needed to
meet the mask requirements.
Using the same matching techniques described above, Figure 38
shows an example of the AD9752 used in a W-CDMA transmitter application using the AD6122 CDMA 3 V transmitter IF
+5V
1.82V
634V
DVDD
100W
REFLO
REFIO
FSADJ
RSET1
2kV
0.1mF
VPBF
AD9752
(“I DAC”)
U1
DAC
BBIP
500V
IOUTA
CFILTER
LATCHES
IOUTB
I DATA
INPUT
500V
+
BBIN
VOUT
100V
AVDD
CLK
LOIP
AVDD REFLO
QOUTA
LATCHES
Q DATA
INPUT
REFIO
500V
500V
PHASE
SPLITTER
BBQP
LOIN
CFILTER
500V
QOUTB
FSADJ
500V
100V
U2
DAC
AD9752
(“Q DAC”)
0.1mF
500V
500V
AVDD
BBQN
AD8346
100V
SLEEP
500mV p-p WITH
VCM=1.2V
RSET2
1.9kV
RCAL
220V
DCOM
NOTE: 500V RESISTOR NETWORK - OHMTEK ORN5000D
100V RESISTOR NETWORK - TOMC1603-100D
ACOM
Figure 37. Baseband QAM Implementation Using Two AD9752s
REV. 0
–17–
AD9752
+3V
634V
DVDD
100W
REFLO
REFIO
FSADJ
RSET1
2kV
AVDD
500V
500V
AD9752
(“I DAC”)
AD6122
IIPP
500V
IOUTA
U1
DAC
CFILTER
LATCHES
IOUTB
I DATA
INPUT
IIPN
500V
100V
AVDD
CLK
LOIPP
LOIPN
AVDD REFLO
Q DATA
INPUT
500V
500V
QOUTA
LATCHES
PHASE
SPLITTER
42
IIQP
500V
U2
DAC
AD9752
(“Q DAC”)
REFIO
100V
500V
QOUTB
FSADJ
IIQN
TEMPERATURE
COMPENSATION
100V
SLEEP
RSET2
1.9kV
0.1mF
REFIN
RCAL
220V
VCC
GAIN
CONTROL
SCALE
FACTOR
DCOM
ACOM
GAIN
CONTROL
VCC
VGAIN
TXOPP
TXOPN
Figure 38. CDMA Transmit Application Using AD9752
Figure 39 shows the AD9752 reconstructing a wideband, or
W-CDMA test vector with a bandwidth of 5 MHz, centered at
15.625 MHz and being sampled at 62.5 MSPS. ACP for the given
test vector is measured at 70 dB.
QAM carrier frequency. Figure 40 shows a block diagram of
such an implementation using the AD9752.
I DATA
–20
Q DATA
12
12
STEL-1130
QAM
12
LPF
AD9752
50V
TO
MIXER
50V
–30
12
SIN
–40
–50
CARRIER 12
FREQUENCY
–60
–70
12
COS
STEL-1177
NCO
CLOCK
–80
Figure 40. Digital QAM Architecture
–90
CL1
CO
CO
CU1
AD9752 EVALUATION BOARD
General Description
–100
–110
The AD9752-EB is an evaluation board for the AD9752 12-bit
D/A converter. Careful attention to layout and circuit design
combined with a prototyping area allow the user to easily and
effectively evaluate the AD9752 in any application where high
resolution, high speed conversion is required.
–120
CENTER 16.384MHz
1.4096MHz
SPAN 14.096MHz
Figure 39. CDMA Signal, Sampled at 65 MSPS, Adjacent
Channel Power >70 dBm
It is also possible to generate a QAM signal completely in the
digital domain via a DSP or ASIC, in which case only a single
DAC of sufficient resolution and performance is required to
reconstruct the QAM signal. Also available from several vendors
are Digital ASICs which implement other digital modulation
schemes such as PSK and FSK. This digital implementation has
the benefit of generating perfectly matched I and Q components
in terms of gain and phase, which is essential in maintaining
optimum performance in a communication system. In this implementation, the reconstruction DAC must be operating at a
sufficiently high clock rate to accommodate the highest specified
This board allows the user the flexibility to operate the AD9752
in various configurations. Possible output configurations include
transformer coupled, resistor terminated, inverting/noninverting
and differential amplifier outputs. The digital inputs are designed
to be driven directly from various word generators, with the
on-board option to add a resistor network for proper load
termination. Provisions are also made to operate the AD9752
with either the internal or external reference, or to exercise the
power-down feature.
Refer to the application note AN-420 for a thorough description
and operating instructions for the AD9752 evaluation board.
–18–
REV. 0
REV. 0
2
4
6
8
10
12
14
16
18
20
22
24
26
28
30
32
34
36
38
40
Figure 41. Evaluation Board Schematic
–19–
A
J4
A
J3
P1
R1
A
R38
49.9V
A
OUT2
R20
49.9V
OUT1
R2
C13
22pF
R14
0
C12
22pF
A
6
5
4
T1
1
2 3 4 5 6 7 8 9 10
10 9 8 7 6 5 4 3 2
1
3
5
7
9
11
13
15
17
19
21
23
25
27
29
31
33
35
37
39
1
1
3
R5
C4
10mF
TP4
B3
R6
A
J7
JP6B
C20
0
JP6A
A
C30
C31
C32
C33
C34
C35
C36
C19
C1
C2
C25
C26
C27
C28
C29
A
AGND
R13
OPEN
A
R12
OPEN
DVDD
1
2 3 4 5 6 7 8 9 10
AVDD
10 9 8 7 6 5 4 3 2
1
DVDD
TP2
TP3
C3
10mF
B2
DGND
B1
DVDD
16
15
14
13
12
11
10
9
A
B
JP7B
1
2
3
4
5
6
7
R9
1kV
16
15
14
13
12
11
10
16 PINDIP
RES PK
1
2
3
4
5
6
7
8
JP9
A
R3
C6
10mF
TP7
B6
R4
B
A
A
B
R10
1kV
A
R35
1kV
A
B
JP8
R18
1kV
2
3
R7
U4
7
AVCC
R8
AVEE
4
A
A
C23
0.1mF
R36
1kV
6
C21
0.1mF
C24
1mF
R37
49.9V
C22
1mF
A
J6
DVDD
1
1
2
3
4
5
6
7
8
9
10
11
12
13
14
J1
2 3 4 5 6 7 8 9 10
EXTCLK
10 9 8 7 6 5 4 3 2
1
DVDD
A
AD8047
1
2 3 4 5 6 7 8 9 10
A
AVCC
10 9 8 7 6 5 4 3 2
1
C5
10mF
TP6
B5
JP7A
AVEE
16 PINDIP
RES PK
TP5
TP18
TP19
B4
A
C18
0.1mF
TP12
A
28
27
26
25
24
23
22
21
20
19
18
17
16
15
2
CLK
JP1
A
C16
1mF
TP11
U7
A
A
4
GND
R44
50V
VOUT
REF43
A
3
2
1
AVDD
JP2
VIN
A
C7
1mF
AVDD
EXTREFIN
J5
2
PDIN
J2
A
3
B
AVCC
R17
49.9V
CLOCK
DVDD
DCOM
NC
AVDD
COMP2
IOUTA
IOUTB
ACOM
COMP1
FS ADJ
REFIO
REFLO
SLEEP
CT1
DB13
DB12
DB11
DB10
DB9
DB8
DB7
DB6
DB5
DB4
DB3
DB2
DB1
DB0
AD975x
U1
1
R15
49.9V
TP1
C14
1mF
6
A
R45
1kV
R43
5kV
R42
1kV
A
A
JP4
A
R46
1kV
2
3
7
A
3
4
A
AD8047
U6
JP3
1
AVEE
A
C15
0.1mF
2
JP5
6
C17
0.1mF
AVCC
C10
0.1mF
TP9
OUT 2
TP8
OUT 1
C9
0.1mF
AVDD
CW
TP14
R16
2kV
TP10
AVDD
C11
0.1mF
C8
0.1mF
1
2
3
A
B
TP13
AD9752
AD9752
Figure 42. Silkscreen Layer—Top
Figure 43. Component Side PCB Layout (Layer 1)
–20–
REV. 0
AD9752
Figure 44. Ground Plane PCB Layout (Layer 2)
Figure 45. Power Plane PCB Layout (Layer 3)
REV. 0
–21–
AD9752
Figure 46. Solder Side PCB Layout (Layer 4)
Figure 47. Silkscreen Layer—Bottom
–22–
REV. 0
AD9752
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
28-Lead, 300 Mil SOIC
(R-28)
28
C3332–8–1/99
0.7125 (18.10)
0.6969 (17.70)
15
0.2992 (7.60)
0.2914 (7.40)
0.4193 (10.65)
0.3937 (10.00)
14
1
PIN 1
0.1043 (2.65)
0.0926 (2.35)
0.0500
(1.27)
BSC
0.0118 (0.30)
0.0040 (0.10)
0.0291 (0.74)
3 458
0.0098 (0.25)
88 0.0500 (1.27)
0.0192 (0.49)
08 0.0157 (0.40)
SEATING 0.0125 (0.32)
0.0138 (0.35)
PLANE 0.0091 (0.23)
28-Lead TSSOP
(RU-28)
0.386 (9.80)
0.378 (9.60)
15
0.256 (6.50)
0.246 (6.25)
0.177 (4.50)
0.169 (4.30)
28
1
14
PIN 1
0.006 (0.15)
0.002 (0.05)
0.0256 (0.65)
BSC
0.0118 (0.30)
0.0075 (0.19)
0.0079 (0.20)
0.0035 (0.090)
88
08
0.028 (0.70)
0.020 (0.50)
PRINTED IN U.S.A.
SEATING
PLANE
0.0433
(1.10)
MAX
REV. 0
–23–