AD AD9042

a
FUNCTIONAL BLOCK DIAGRAM
AVCC
AIN
TH2
VREF
INTERNAL
TIMING
DIGITAL ERROR CORRECTION LOGIC
MSB
27 D10
LSB
PRODUCT HIGHLIGHTS
1. Guaranteed sample rate is 41 MSPS.
2. Dynamic performance specified over entire Nyquist band;
spurious signals typ. 80 dBc for –1 dBFS input signals.
3. Low power dissipation: 595 mW off a single +5 V supply.
4. Reference and track-and-hold included on chip.
5. Packaged in 28-pin ceramic DIP and 44-pin TQFP.
GND
GND
DVCC
DVCC
GND
DVCC
GND
D11 (MSB)
D9
AD9042AST PIN DESIGNATIONS
AD9042AD PIN DESIGNATIONS
DVCC 2
7
AD9042
6
cofired ceramic package forms a multilayer substrate to which
internal bypass capacitors and the 9042 die are attached and a
44-pin TQFP low profile surface mount package. The AD9042
industrial grade is specified from –40°C to +85°C. However,
the AD9042 was designed to perform over the full military
temperature range (–55°C to +125°C); consult factory for
military grade product options.
The AD9042 is built on Analog Devices’ high speed complementary bipolar process (XFCB) and uses an innovative multipass
architecture. Units are packaged in a 28-pin DIP; this custom
28 D11 (MSB)
A2
D11 D10 D9 D8 D7 D6 D5 D4 D3 D2 D1 D0
D10
Designed specifically to address the needs of wideband,
multichannel receivers, the AD9042 maintains 80 dB
spurious-free dynamic range (SFDR) over a bandwidth of
20 MHz. Noise performance is also exceptional; typical
signal-to-noise ratio is 68 dB.
DAC
ADC
+2.4V
REFERENCE
GND
The AD9042 is a high speed, high performance, low power,
monolithic 12-bit analog-to-digital converter. All necessary
functions, including track-and-hold (T/H) and reference are
included on chip to provide a complete conversion solution.
The AD9042 runs off of a single +5 V supply and provides
CMOS-compatible digital outputs at 41 MSPS.
TH3
ADC
ENCODE
PRODUCT DESCRIPTION
GND 1
TH1
VOFFSET
ENCODE
APPLICATIONS
Cellular/PCS Base Stations
GPS Anti-Jamming Receivers
Communications Receivers
Spectrum Analyzers
Electro-Optics
Medical Imaging
ATE
A1
DVCC
DVCC
FEATURES
41 MSPS Minimum Sample Rate
80 dB Spurious-Free Dynamic Range
595 mW Power Dissipation
Single +5 V Supply
On-Chip T/H and Reference
Twos Complement Output Format
CMOS-Compatible Output Levels
12-Bit, 41 MSPS
Monolithic A/D Converter
AD9042
44 43 42 41 40 39 38 37 36 35 34
DVCC 1
DVCC 2
33 D8
32 D7
PIN 1
31 D6
ENCODE 3
GND 3
26 D9
ENCODE 4
25 D8
ENCODE 5
24 D7
GND 5
AD9042
29 D4
TOP VIEW 23 D6
GND 7 (Not to Scale) 22 D5
GND 6
TOP VIEW
(Not to Scale)
28 D3
GND 6
AIN 7
AIN 8
21 D4
VOFFSET 8
VOFFSET 9
20 D3
VREF 9
VREF 10
19 D2
C1 10
GND 11
18 D1
AVCC 11
27 D2
26 D1
25 D0 (LSB)
24 GND
23 NC
GND
GND
AVCC
GND
AVCC
15 NC
NC = NO CONNECT
16 17 18 19 20 21 22
GND
AVCC 14
12 13 14 15
GND
16 NC
AVCC
GND 13
AVCC
17 D0 (LSB)
GND
AVCC 12
30 D5
AVCC
AD9042
ENCODE 4
NC = NO CONNECT
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
© Analog Devices, Inc., 1996
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
Fax: 617/326-8703
AD9042–SPECIFICATIONS
DC SPECIFICATIONS (AV
Parameter
CC
= DVCC = +5 V; VREF tied to VOFFSET through 50 Ω; TMIN = –408C, TMAX = +858C)1
Temp
Test
Level
AD9042AST
Min Typ Max
RESOLUTION
Full
Full
Full
Full
Full
VI
VI
V
VI
V
REFERENCE OUT (VREF)2
+25°C
V
ANALOG INPUT (AIN)
Input Voltage Range
Input Resistance
Input Capacitance
Full
+25°C
IV
V
ENCODE INPUT3
Logic Compatibility4
Logic “1” Voltage
Logic “0” Voltage
Logic “1” Current (VINH = 5 V)
Logic “0” Current (VINL = 0 V)
Input Capacitance
Full
Full
Full
Full
+25°C
VI
VI
VI
VI
V
+25°C
Full
+25°C
Full
I
IV
I
IV
Full
Full
Full
Full
Full
Full
+25°C
Full
VI
V
VI
V
VI
VI
I
V
Logic “0” Voltage (IOL = 10 µA)
Output Coding
POWER SUPPLY
AVCC Supply Voltage
I (AVCC) Current
DVCC Supply Voltage
I (DVCC) Current
ICC (Total) Supply Current
Power Dissipation
Power Supply Rejection
(PSRR)
Min
AD9042AD
Typ Max
12
DC ACCURACY
No Missing Codes
Offset Error
Offset Tempco
Gain Error
Gain Tempco
DIGITAL OUTPUTS
Logic Compatibility
Logic “1” Voltage (IOH = 10 µA)
Test
Level
12
Guaranteed
±3
+10
25
–6.5 0
+6.5
–50
–10
2.4
200
TTL/CMOS
2.0
5.0
0
0.8
450
625
800
–400 –300 –200
2
CMOS
4.2
0.75
0.80
0.85
Twos Complement
–20
5.0
109
5.0
10
119
595
±1
±5
147
735
+20
IV
V
VI
VI
VI
VI
V
I
IV
I
IV
VI
V
VI
V
VI
VI
I
V
Bits
Guaranteed
±3
+10
25
–6.5 0
+6.5
–50
–10
V
VREF ± 0.500
250
300
5.5
3.5
3.5
VI
VI
V
VI
V
2.4
200
VREF ± 0.500
250
300
7
CMOS
4.2
0.75
0.80
0.85
Twos Complement
–20
5.0
109
5.0
10
119
595
±1
±5
mV
ppm/°C
% FS
ppm/°C
V
TTL/CMOS
2.0
5.0
0
0.8
450
625
800
–400 –300 –200
2.5
3.5
3.5
Units
147
735
+20
V
Ω
pF
V
V
µA
µA
pF
V
V
V
V
V
mA
V
mA
mA
mW
mV/V
mV/V
NOTES
1
C1 (Pin 10 on AD9042AST only) tied to GND through 0.01 µF capacitor.
2
VREF is normally tied to V OFFSET through 50 Ω. If VREF is used to provide dc offset to other circuits, it should first be buffered.
3
ENCODE driven by single-ended source; ENCODE bypassed to ground through 0.01 µF capacitor.
4
ENCODE may also be driven differentially in conjunction with ENCODE; see “Encoding the AD9042” for details.
Specifications subject to change without notice.
SWITCHING SPECIFICATIONS
(AVCC = DVCC = +5 V; ENCODE & ENCODE = 41 MSPS;
VREF tied to VOFFSET through 50 Ω; TMIN = –408C, TMAX = +858C)1
Parameter (Conditions)
Temp
Test
Level
AD9042AST
Min Typ Max
Test
Level
AD9042AD
Min Typ Max
Maximum Conversion Rate
Minimum Conversion Rate
Aperture Delay (tA)
Aperture Uncertainty (Jitter)
ENCODE Pulse Width High
ENCODE Pulse Width Low
Output Delay (tOD)
Full
Full
+25°C
+25°C
+25°C
+25°C
Full
VI
IV
V
V
IV
IV
IV
41
VI
IV
V
V
IV
IV
IV
41
5
–250
0.7
10
10
5
9
14
5
–250
0.7
10
10
5
9
14
Units
MSPS
MSPS
ps
ps rms
ns
ns
ns
NOTE
1
C1 (Pin 10 on AD9042AST only) tied to GND through 0.01 µF capacitor.
–2–
REV. A
AC SPECIFICATIONS1
(AVCC = DVCC = +5 V; ENCODE & ENCODE = 41 MSPS;
VREF tied to VOFFSET through 50 Ω; TMIN = –408C, TMAX = +858C)2
Temp
Test
Level
+25°C
Full
+25°C
Full
+25°C
Full
V
V
V
V
I
V
+25°C
Full
+25°C
Full
+25°C
Full
V
V
V
V
I
V
+25°C
Full
+25°C
Full
+25°C
Full
V
V
V
V
I
V
Small Signal SFDR (w/Dither)6
Analog Input @1.2 MHz
9.6 MHz
19.5 MHz
Full
Full
Full
Two-Tone IMD Rejection7
F1, F2 @ –7 dBFS
Parameter (Conditions)
SNR3
Analog Input
@ –1 dBFS
1.2 MHz
AD9042AST
Min Typ Max
Test
Level
AD9042
Min
68
67.5
67.5
67
67
66.5
I
V
I
V
I
V
67.5
67
67.5
67
67
66.5
I
V
I
V
I
V
64
80
78
80
78
80
78
I
V
I
V
I
V
74
V
V
V
90
90
90
Full
V
Two-Tone SFDR (w/Dither)8
Full
Thermal Noise
Units
68
67.5
67.5
67
67
66.5
dB
dB
dB
dB
dB
dB
67.5
67
67.5
67
67
66.5
dB
dB
dB
dB
dB
dB
80
78
80
78
80
78
dBc
dBc
dBc
dBc
dBc
dBc
V
V
V
90
90
90
dBFS
dBFS
dBFS
80
V
80
dBc
V
90
V
90
dBFS
+25°C
V
0.33
V
0.33
LSB rms
Differential Nonlinearity
(ENCODE = 20 MSPS)
+25°C
Full
I
V
Integral Nonlinearity
(ENCODE = 20 MSPS)
Full
V
± 0.75
V
± 0.75
LSB
Analog Input Bandwidth
+25°C
V
100
V
100
MHz
Transient Response
+25°C
V
10
V
10
ns
Overvoltage Recovery Time
+25°C
V
25
V
25
ns
9.6 MHz
19.5 MHz
SINAD4
Analog Input
@ –1 dBFS
1.2 MHz
9.6 MHz
19.5 MHz
Worst Spur5
Analog Input
@ –1 dBFS
1.2 MHz
9.6 MHz
19.5 MHz
64
64
73
–1.0
± 0.3
± 0.4
+1.0
I
VI
65
AD9042AD
Typ Max
64.5
64
64
64
74
73
–1.0
–1.0
± 0.3
+1.0
+1.25
LSB
LSB
NOTES
1
All ac specifications tested by driving ENCODE and ENCODE differentially; see “ENCODING the AD9042” for details.
2
C1 (Pin 10 on AD9042AST only) tied to GND through 0.01 µF capacitor.
3
Analog input signal power at –1 dBFS; signal-to-noise ratio (SNR) is the ratio of signal level to total noise (first five harmonics removed).
4
Analog input signal power at –1 dBFS; signal-to-noise and distortion (SINAD ) is the ratio of signal level to total noise + harmonics.
5
Analog input signal power at –1 dBFS; worst spur is the ratio of the signal level to worst spur, usually limited by harmonics.
6
Analog input signal power swept from –20 dBFS to –95 dBFS; dither power = –32.5 dBm; dither circuit used on input signal (see “Overcoming Static Nonlinearities
with Dither”); SFDR is ratio of converter full scale to worst spur.
7
Tones at –7 dBFS (F1 = 15.3 MHz, F2 = 19.5 MHz); two tone intermodulation distortion (IMD) rejection is ratio of either tone to worst third order intermod product.
8
Both input tones swept from –20 to –95 dBFS; Dither power = –32.5 dBm; dither circuit used on input signal (see “Overcoming Static Nonlinearities with Dither);
two tone spurious-free dynamic range (SFDR) is the ratio of converter full scale to worst spur.
Specifications subject to change without notice.
REV. A
–3–
AD9042
WAFER TEST LIMITS1 (AV
CC
= DVCC = +5 V; ENCODE = 10.3 MSPS unless otherwise noted)
Parameter
Temp
AD9042CHIPS
Min
Max
Units
POWER SUPPLY
ICC Supply Current
+25°C
90
147
mA
ENCODE Input
Logic “1” Current
Logic “0” Current
+25°C
+25°C
450
–400
800
–200
µA
µA
DC ACCURACY
Offset Error
Gain Error
No Missing Codes
Differential Nonlinearity @ 5.3 MSPS
+25°C
+25°C
+25°C
+25°C
–8
–6
8
6
Guaranteed
–0.995
mV
% FS
LSB
NOTES
1
Electrical test is performed at wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after
packaging is not guaranteed for standard product dice.
2
Die substrate is connected to 0 V.
ABSOLUTE MAXIMUM RATINGS 1
Parameter
Min
ELECTRICAL
AVCC Voltage
DVCC Voltage
Analog Input Voltage
Analog Input Current
Digital Input Voltage (ENCODE)
ENCODE, ENCODE Differential
Voltage
Digital Output Current
0
0
0.5
Max
Units
0
7
7
4.5
20
AVCC
V
V
V
mA
V
–40
4
40
V
mA
+85
°C
+175
+150
+300
+150
°C
°C
°C
°C
ENVIRONMENTAL2
Operating Temperature Range
(Ambient)
–40
Maximum Junction Temperature
AD9042AD
AD9042AST
Lead Temperature (Soldering, 10 sec)
Storage Temperature Range (Ambient) –65
EXPLANATION OF TEST LEVELS
Test Level
I
II
–
–
III –
IV –
V –
VI –
100% production tested.
100% production tested at +25°C, and sample tested at
specified temperatures. AC testing done on sample
basis.
Sample tested only.
Parameter is guaranteed by design and characterization
testing.
Parameter is a typical value only.
All devices are 100% production tested at +25°C;
sample tested at temperature extremes.
NOTES
1
Absolute maximum ratings are limiting values to be applied individually, and
beyond which the serviceability of the circuit may be impaired. Functional
operability is not necessarily implied. Exposure to absolute maximum rating
conditions for an extended period of time may affect device reliability.
2
Typical thermal impedances for “D” package (custom ceramic 28-pin DIP):
θJC = 14°C/W; θJA = 34°C/W. For “ST” package (44-pin TQFP) ; θJA = 55°C/W.
ORDERING GUIDE
Model
Temperature Range
Package Description
Package Option
AD9042AST
AD9042AD
AD9042CHIPS
AD9042ST/PCB
AD9042D/PCB
–40°C to +85°C (Ambient)
–40°C to +85°C (Ambient)
–40°C to +85°C (Ambient)
44-Pin TQFP (Thin Quad Plastic Flatpack)
28-Pin 600 Mil Hermetic Ceramic DIP (DH-28)
Unpackaged Die
Evaluation Board with AD9042AST
Evaluation Board with AD9042AD
ST-44
DH-28
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD9042 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
–4–
WARNING!
ESD SENSITIVE DEVICE
REV. A
AD9042
AD9042AST PIN DESCRIPTIONS
AD9042AD PIN DESCRIPTIONS
Pin No. Name
Function
Pin No.
Name
Function
1, 2
DVCC
1
2
GND
DVCC
3
ENCODE
+5 V Power Supply (Digital).
Powers output stage only.
Encode input. Data conversion
initiated on rising edge.
Complement of ENCODE. Drive
differentially with ENCODE or
bypass to Ground for single-ended
clock mode.
Ground.
Analog Input.
Voltage Offset Input. Sets midpoint of analog input range.
Normally tied to VREF through
50 Ω resistor.
Internal Voltage Reference.
Nominally +2.4 V; normally tied
to VOFFSET through 50 Ω resistor.
Bypass to Ground with 0.1 µF +
0.01 µF microwave chip cap.
Internal Bias Point. Bypass to
ground with 0.01 µF cap.
+5 V Power Supply (Analog).
Ground.
+5 V Power Supply (Analog).
Ground.
+5 V Power Supply (Analog).
Ground.
Ground.
No Connects.
Ground.
Digital Output Bit
(Least Significant Bit)
Digital Output Bits
Ground.
+5 V Power Supply (Digital).
Powers output stage only.
Ground.
+5 V Power Supply (Digital).
Powers Output Stage only.
Digital Output Bits.
Digital Output Bit
(Most Significant Bit).
3
4
GND
ENCODE
5
ENCODE
6, 7
8
9
GND
AIN
VOFFSET
10
VREF
11
12
13
14
15, 16
17
GND
AVCC
GND
AVCC
NC
D0 (LSB)
18–27
28
D1–D10
D11 (MSB)1
Ground.
+5 V Power Supply (Digital).
Powers output stage only.
Ground.
Encode input. Data conversion
initiated on rising edge.
Complement of ENCODE. Drive
differentially with ENCODE or
bypass to Ground for single-ended
clock mode.
Ground.
Analog Input.
Voltage Offset Input. Sets midpoint of analog input range.
Normally tied to VREF through
50 Ω resistor.
Internal Voltage Reference.
Nominally +2.4 V; normally tied
to VOFFSET through 50 Ω resistor.
Bypass to Ground with 0.1 µF cap.
Ground.
+5 V Power Supply (Analog).
Ground.
+5 V Power Supply (Analog).
No Connects.
Digital Output Bit.
(Least Significant Bit).
Digital Output Bits.
Digital Output Bit
(Most Significant Bit).
4
ENCODE
5, 6
7
8
GND
AIN
VOFFSET
9
VREF
10
C1
11, 12
13, 14
15, 16
17, 18
19, 20
21
22
23
24
25
AVCC
GND
AVCC
GND
AVCC
GND
GND
NC
GND
D0 (LSB)
26–33
34, 35
36, 37
D1–D8
GND
DVCC
38, 39
40, 41
GND
DVCC
42, 43
44
D9–D10
D11 (MSB)1
NOTE
1
Output coded as twos complement.
AD9042 CUSTOM 28-PIN DIP PACKAGE
NOTE
1
Output coded as twos complement.
REV. A
–5–
AD9042
DIE LAYOUT AND MECHANICAL INFORMATION
Die Dimensions . . . . . . . . . . . . . . . . 155 × 168 × 21 (± 1) mils
Pad Dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 × 4 mils
Metalization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Aluminum
Backing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . None
Substrate Potential . . . . . . . . . . . . . . . . . . . . . . . . . . . . GND
Transistor Count . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2,605
Passivation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Oxynitride
Die Attach . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Silver Filled
Bond Wire . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Gold
DIE LAYOUT W/PAD LABELS
Harmonic Distortion
The ratio of the rms signal amplitude to the rms value of the
worst harmonic component, reported in dBc.
Integral Nonlinearity
The deviation of the transfer function from a reference line
measured in fractions of 1 LSB using a “best straight line”
determined by a least square curve fit.
Minimum Conversion Rate
The encode rate at which the SNR of the lowest analog signal
frequency drops by no more than 3 dB below the guaranteed
limit.
Maximum Conversion Rate
The encode rate at which parametric testing is performed.
Output Propagation Delay
The delay between the 50% point of the rising edge of ENCODE
command and the time when all output data bits are within
valid logic levels.
Overvoltage Recovery Time
The amount of time required for the converter to recover to
0.02% accuracy after an analog input signal 150% of full scale is
reduced to midscale.
Power Supply Rejection Ratio
The ratio of a change in input offset voltage to a change in
power supply voltage.
Signal-to-Noise-and-Distortion (SINAD)
The ratio of the rms signal amplitude (set at 1 dB below full
scale) to the rms value of the sum of all other spectral
components, including harmonics but excluding dc.
Signal-to-Noise Ratio (without Harmonics)
The ratio of the rms signal amplitude (set at 1 dB below full
scale) to the rms value of the sum of all other spectral
components, excluding the first five harmonics and dc.
DEFINITION OF SPECIFICATIONS
Analog Bandwidth
The analog input frequency at which the spectral power of the
fundamental frequency (as determined by the FFT analysis) is
reduced by 3 dB.
Aperture Delay
The delay between the 50% point of the rising edge of the
ENCODE command and the instant at which the analog input
is sampled.
Spurious-Free Dynamic Range
The ratio of the rms signal amplitude to the rms value of the
peak spurious spectral component. The peak spurious
component may or may not be a harmonic. May be reported in
dBc (i.e., degrades as signal levels is lowered), or in dBFS
(always related back to converter full scale).
Transient Response
Aperture Uncertainty (Jitter)
The time required for the converter to achieve 0.02%
accuracy when a one-half full-scale step function is applied to
the analog input.
The sample-to-sample variation in aperture delay.
Two-Tone Intermodulation Distortion Rejection
Differential Nonlinearity
The ratio of the rms value of either input tone to the rms
value of the worst third order intermodulation product;
reported in dBc.
The deviation of any code from an ideal 1 LSB step.
Encode Pulse Width/Duty Cycle
Pulse width high is the minimum amount of time that the
ENCODE pulse should be left in logic “1” state to achieve
rated performance; pulse width low is the minimum time
ENCODE pulse should be left in low state. At a given clock
rate, these specs define an acceptable Encode duty cycle.
Two-Tone SFDR
The ratio of the rms value of either input tone to the rms value
of the peak spurious component. The peak spurious component
may or may not be an IMD product. May be reported in dBc
(i.e., degrades as signal levels is lowered), or in dBFS (always
related back to converter full scale).
–6–
REV. A
Equivalent Circuits–AD9042
N
tA = –250 PS TYP
ANALOG
INPUT
(AIN)
N+1
ENCODE
INPUTS
(ENCODE)
DIGITAL
OUTPUTS
(D11–D0)
N–2
N
N–1
tOD = 9ns TYP
Figure 1. Timing Diagram
DVCC
AVCC
+3.5V
CURRENT
MIRROR
AVCC
250µA
250Ω
250Ω
AIN
AVCC
250µA
DVCC
VOFFSET
VREF
200Ω
D0–D11
+1.5V
6pF
Figure 2. Analog Input Stage
CURRENT
MIRROR
AVCC
AVCC
R1
17kΩ
R1
17kΩ
AVCC
Figure 5. Digital Output Stage
ENCODE
ENCODE
TIMING
CIRCUITS
R2
8kΩ
AVCC
R2
8kΩ
AVCC
2.4V
VREF
0.5mA
Figure 3. Encode Inputs
Figure 6. 2.4 V Reference
AVCC
+5V
VREF
+5V
2,12,14
AVCC
AVCC
0.1µF
200kHz
SINEWAVE
8
9
49.9Ω
CURRENT
MIRROR
C1
(PIN 10*)
TTL CLOCK OSC.
*AD9042AST ONLY
NC
INTERNAL NODE ON AD9042AD
10
4
5
D11
AIN
28
10kΩ
VOFFSET
VREF
ENCODE
ENCODE
D0
17
1,3,6,7,11,13
Figure 4. Compensation Pin, C1
NOTE: ALL +5V SUPPLY PINS & VREF PIN BYPASSED TO GND
WITH A 0.1µF CAPACITOR. PINS 15,16 ARE NOT CONNECTED.
Figure 7. AD9042AD Burn-In Diagram
REV. A
–7–
AD9042–Typical Performance Characteristics
ENCODE = 41 MSPS
AIN = 1.2MHz
81
–20
T = +25°C
WORST CASE HARMONIC – dBc
POWER RELATIVE TO ADC FULL SCALE – dB
0
–40
–60
2
3
4
5
6
7
8
9
–80
–100
80
ENCODE = 41 MSPS
TEMP = –40°C, +25°C, & +85°C
T = –40°C
T = +85°C
79
78
77
–120
dc
8.2
12.3
FREQUENCY – MHz
4.1
16.4
20.5
0
Figure 8. Single Tone at 1.2 MHz
4
2
6
8
10
12
14
16
ANALOG INPUT FREQUENCY – MHz
18
20
Figure 11. Harmonics vs. AIN
ENCODE = 41 MSPS
AIN = 9.6MHz
ENCODE = 41 MSPS
TEMP = –40°C, +25°C, & +85°C
70
–20
69
–40
SNR – dB
POWER RELATIVE TO ADC FULL SCALE – dB
0
–60
4
8 8
5
3
7
2
6
–80
T = –40°C
68
T = +25°C
T = +85°C
67
–100
66
–120
dc
8.2
12.3
FREQUENCY – MHz
4.1
16.4
0
20.5
6
8
10
12
14
16
ANALOG INPUT FREQUENCY – MHz
18
20
Figure 12. Noise vs. AIN
Figure 9. Single Tone at 9.6 MHz
90
0
ENCODE = 41 MSPS
ENCODE = 41 MSPS
AIN = 19.5MHz
–20
80
WORST HARMONIC – dBc
POWER RELATIVE TO ADC FULL SCALE – dB
4
2
–40
–60
2
4
6
8
9
7
5
3
–80
70
60
50
40
–100
–120
dc
30
4.1
8.2
12.3
FREQUENCY – MHz
16.4
20.5
Figure 10. Single Tone at 19.5 MHz
1
2
10
4
20
40
ANALOG INPUT FREQUENCY – MHz
100
Figure 13. Harmonics vs. AIN
–8–
REV. A
AD9042
–20
85
SNR, WORST CASE SPURIOUS – dB, dBc
POWER RELATIVE TO ADC FULL SCALE – dB
0
ENCODE = 41 MSPS
AIN = 15.3, 19.5MHz
–40
–60
–80
–100
–120
dc
8.2
12.3
FREQUENCY – MHz
4.1
16.4
SNR, WORST FULL SCALE SPURIOUS – dBc
WORST CASE SPURIOUS – dBc AND dBFS
dBFS
80
70
ENCODE = 41 MSPS
AIN = 19.5MHz
50
40
dBc
SFDR = 80dB
REFERENCE LINE
30
20
10
–70
–60
–50
–40
–30
–20
ANALOG INPUT POWER LEVEL – dBFS
–10
POWER RELATIVE TO ADC FULL SCALE – dB
WORST CASE SPURIOUS – dBc AND dBFS
5
10
15
20
25
30
35
SAMPLE RATE – MSPS
40
50
45
ENCODE = 41 MSPS
AIN = 19.5MHz
85
80
WORST SPUR
75
70
SNR
65
60
55
50
45
40
35
30
35
40
45
50
55
60
ENCODE DUTY CYCLE – %
65
75
70
0
90
80
70
ENCODE = 41 MSPS
F1 = 19.3MHz
F2 = 19.51MHz
SFDR = 80dB
REFERENCE LINE
40
30
20
10
–70
–60
–50
–40
–30
–20
INPUT POWER LEVEL (F1 = F2) – dBFS
–10
0
Figure 16. AD9042AD Two Tone SFDR
REV. A
65
Figure 18. SNR, Worst Spurious vs. Duty Cycle
100
0
–80
SNR
30
25
0
Figure 15. AD9042AD Single Tone SFDR
50
70
90
90
60
75
Figure 17. SNR, Worst Harmonic vs. Encode
100
0
–80
80
60
dc
20.5
Figure 14. Two Tones at 15.3 MHz & 19.5 MHz
60
AIN = 4.3MHz
WORST SPUR
ENCODE = 41 MSPS
AIN = BROADBAND_NOISE
–20
–40
–60
4
–80
–100
–120
dc
4.1
8.2
12.3
FREQUENCY – MHz
16.4
Figure 19. NPR Output Spectrum
–9–
20.5
AD9042
0
ENCODE = 41 MSPS
AIN = 19.5MHz @ –29 dBFS
NO DITHER
–20
POWER RELATIVE TO ADC FULL SCALE – dB
POWER RELATIVE TO ADC FULL SCALE – dB
0
–40
–60
2
–80
6
4
8
8
7
5
3
–100
–120
dc
4.1
8.2
12.3
FREQUENCY – MHz
16.4
–20
–40
–60
–80
–120
dc
20.5
4
6
8
8
7
5
3
4.1
8.2
12.3
FREQUENCY – MHz
16.4
20.5
Figure 23. 4K FFT with Dither
100
100
90
90
ENCODE = 41 MSPS
AIN = 19.5MHz
NO DITHER
80
WORST CASE SPURIOUS – dBc
WORST CASE SPURIOUS – dBc
2
–100
Figure 20. 4K FFT without Dither
70
60
50
40
30
20
SFDR = 80dB
REFERENCE LINE
10
0
–80
–70
80
ENCODE = 41 MSPS
AIN = 19.5MHz
DITHER = –32.5dBm
70
60
50
40
30
20
SFDR = 80dB
REFERENCE LINE
10
–60
–50
–40
–30
–20
–10
ANALOG INPUT POWER LEVEL – dBFS
0
–80
0
Figure 21. SFDR without Dither
–70
–60
–50
–40
–30
–20
–10
ANALOG INPUT POWER LEVEL – dBFS
0
Figure 24. SFDR with Dither
0
0
ENCODE = 41 MSPS
AIN = 2.5MHz @ –26 dBFS
NO DITHER
–20
POWER RELATIVE TO ADC FULL SCALE – dB
POWER RELATIVE TO ADC FULL SCALE – dB
ENCODE = 41 MSPS
AIN = 19.5MHz @ –29 dBFS
DITHER = –32.5dBm
–40
–60
–80
–100
–120
dc
4.1
8.2
12.3
FREQUENCY – MHz
16.4
–40
–60
–80
–100
–120
dc
20.5
Figure 22. 128K FFT without Dither
ENCODE = 41 MSPS
AIN = [email protected]
DITHER = –32.5dBm
–20
4.1
8.2
12.3
FREQUENCY – MHz
16.4
20.5
Figure 25. 128K FFT with Dither
–10–
REV. A
AD9042
THEORY OF OPERATION
V1 =
The AD9042 analog-to-digital converter (ADC) employs a twostage subrange architecture. This design approach ensures
12-bit accuracy, without the need for laser trim, at low power.
As shown in the functional block diagram, the 1 V p-p singleended analog input, centered at 2.4 V, drives a single-in to
differential-out amplifier, A1. The output of A1 drives the first
track-and-hold, TH1. The high state of the ENCODE pulse
places TH1 in hold mode. The held value of TH1 is applied to
the input of the 6-bit coarse ADC. The digital output of the
coarse ADC drives a 6-bit DAC; the DAC is 12 bits accurate.
The output of the 6-bit DAC is subtracted from the delayed
analog signal at the input to TH3 to generate a residue signal.
TH2 is used as an analog pipeline to null out the digital delay of
the coarse ADC.
ENCODE
SOURCE
ENCODE
Vl
0.01µF
V1 =
R1
ENCODE
R2
RX
AD9042
5R2
RR
R2 + 1 X
R1 + RX
to raise logic threshold.
AVCC
RX
+5V
ENCODE
SOURCE
The 6-bit coarse ADC word and 7-bit residue word are added
together and corrected in the digital error correction logic to
generate the output word. The result is a 12-bit parallel digital
word which is CMOS-compatible, coded as twos complement.
ENCODE
R1
Vl
ENCODE
R2
0.01µF
APPLYING THE AD9042
Encoding the AD9042
+5V
Figure 27. Lower Logic Threshold for Encode
The residue signal is passed to TH3 on a subsequent clock cycle
where the signal is amplified by the residue amplifier, A2, and
converted to a digital word by the 7-bit residue ADC. One bit
of overlap is used to accommodate any linearity errors in the
coarse ADC.
AD9042
Figure 28. Raise Logic Threshold for Encode
While the single-ended encode will work well for many
applications, driving the encode differentially will provide
increased performance. Depending on circuit layout and system
noise, a 1 dB to 3 dB improvement in SNR can be realized. It is
not recommended that differential TTL logic be used however,
because most TTL families that support complementary
outputs are not delay or slew rate matched. Instead, it is
recommended that the encode signal be ac-coupled into the
ENCODE and ENCODE pins.
The AD9042 is designed to interface with TTL and CMOS
logic families. The source used to drive the ENCODE pin(s)
must be clean and free from jitter. Sources with excessive jitter
will limit SNR (ref. Equation 1 under “Noise Floor and SNR”).
AD9042
TTL OR CMOS
SOURCE
5R2 RX
to lower logic threshold.
R1R2 + R1RX + R2 RX
ENCODE
ENCODE
0.01µF
Figure 26. Single-Ended TTL /CMOS Encode
The AD9042 encode inputs are connected to a differential input
stage (see Figure 3 under EQUIVALENT CIRCUITS). With
no input connected to either the ENCODE or input, the voltage
dividers bias the inputs to 1.6 volts. For TTL or CMOS usage,
the encode source should be connected to ENCODE.
ENCODE should be decoupled using a low inductance or
microwave chip capacitor to ground. Devices such as AVX
05085C103MA15, a 0.01 µF capacitor, work well.
The simplest option is shown below. The low jitter TTL signal
is coupled with a limiting resistor, typically 100 ohms, to the
primary side of an RF transformer (these transformers are
inexpensive and readily available; part# in Figure 29 is from
Mini-Circuits). The secondary side is connected to the
ENCODE and ENCODE pins of the converter. Since both
encode inputs are self biased, no additional components are
required.
If a logic threshold other than the nominal 1.6 V is required, the
following equations show how to use an external resistor, RX, to
raise or lower the trip point (see Figure 3; R1 = 17k, R2 = 8k).
100Ω
TTL
T1-1T
ENCODE
AD9042
ENCODE
Figure 29. TTL Source – Differential Encode
REV. A
–11–
AD9042
amplifier offset; this reference is designed to track internal circuit shifts over temperature.
If no TTL source is available, a clean sine wave may be
substituted. In the case of the sine source, the matching network is shown below. Since the matching transformer specified
is a 1:1 impedance ratio, R, the load resistor should be selected
to match the source impedance. The input impedance of the
AD9042 is negligible in most cases.
T1-1T
SINE
SOURCE
250Ω
250Ω
AIN
VOFFSET
TIED TO
VREF
THROUGH
50 OHMS
ENCODE
AD9042
R
50Ω
AD9042
+2.4V
REFERENCE
0.1µF
ENCODE
Figure 33. Analog Input Offset by +2.4 V Reference
Figure 30. Sine Source – Differential Encode
Although the AD9042 may be used in many applications, it was
specifically designed for communications systems which must
digitize wide signal bandwidths. As such, the analog input was
designed to be ac-coupled. Since most communications products
do not down-convert to dc, this should not pose a problem. One
example of a typical analog input circuit is shown below. In this
application, the analog input is coupled with a high quality chip
capacitor, the value of which can be chosen to provide a low
frequency cutoff that is consistent with the signal being
sampled; in most cases, a 0.1 µF chip capacitor will work well.
If a low jitter ECL clock is available, another option is to accouple a differential ECL signal to the encode input pins as
shown below. The capacitors shown here should be chip
capacitors but do not need to be of the low inductance variety.
0.1µF
ENCODE
ECL
GATE
AD9042
0.1µF
ENCODE
510Ω
510Ω
–VS
AD9042
0.1µF
ANALOG
SIGNAL
SOURCE
AIN
RT
Figure 31. Differential ECL for Encode
VOFFSET
50Ω
VREF
As a final alternative, the ECL gate may be replaced by an ECL
comparator. The input to the comparator could then be a logic
signal or a sine signal.
0.1µF
Figure 34. AC-Coupled Analog Input Signal
AD96687 (1/2)
0.1µF
Another option for ac-coupling is a transformer. The impedance ratio and frequency characteristics of the transformer are
determined by examining the characteristics of the input signal
source (transformer primary connection), and the AD9042 input characteristics (transformer secondary connection). “RT”
should be chosen to satisfy termination requirements of the
source, given the transformer turns ratio. A blocking capacitor
is required to prevent AD9042 dc bias currents from flowing
through the transformer.
ENCODE
AD9042
0.1µF
50Ω
ENCODE
510Ω
510Ω
–VS
Figure 32. ECL Comparator for Encode
Care should be taken not to overdrive the encode input pin
when ac coupled. Although the input circuitry is electrically
protected from over or under voltage conditions, improper
circuit operations may result from overdriving the encode input
pins.
BPF
ANALOG
SIGNAL
SOURCE
0.1µF
XFMR
AD9042
AIN
RT
VOFFSET
50Ω
LO
Driving the Analog Input
Because the AD9042 operates off of a single +5 V supply, the
analog input range is offset from ground by 2.4 volts. The
analog input, AIN, is an operational amplifier configured in an
inverting mode (ref. Equivalent Circuits: Analog Input Stage).
VOFFSET is the noninverting input which is normally tied
through a 50 ohm resistor to VREF (ref. Equivalent Circuits:
2.4 V Reference). Since the operational amplifier forces its
inputs to the same voltage, the inverting input is also at 2.4 volts.
Therefore, the analog input has a Thevenin equivalent of 250 ohms
in series with a 2.4 volt source. It is strongly recommended
that the AD9042’s internal voltage reference be used for the
VREF
0.1µF
Figure 35. Transformer-Coupled Analog Input Signal
When calculating the proper termination resistor, note that the
external load resistor is in parallel with the AD9042 analog
input resistance, 250 ohms. The external resistor value can be
calculated from the following equation:
–12–
RT =
1
1
1
−
Z 250
where Z is desired impedance.
REV. A
AD9042
A dc-coupled input configuration (shown below) is limited by
the drive amplifier performance. The AD9042’s on-chip reference is buffered using the OP279 dual, rail-to-rail operational
amplifier. The resulting voltage is combined with the analog
source using an AD9631. Pending improvements in drive
amplifiers, this dc-coupled approach is limited to ~75 dB–80 dB
of dynamic performance depending on which drive amplifier is
used. The AD9631 and OP279 run off ± 5 V.
SIGNAL
SOURCE
Layout Information
The schematic of the evaluation boards (Figures 37 and 38)
represents a typical implementation of the AD9042. The pinout
of the AD9042 facilitates ease of use and the implementation of
high frequency/high resolution design practices. All of the
digital outputs are on one side of the packages while the other
sides contain all of the inputs. It is highly recommended that
high quality ceramic chip capacitors be used to decouple each
supply pin to ground directly at the device. Depending on
the configuration used for the encode and analog inputs, one or
more capacitors are required on those input pins. The capacitors
used on the ENCODE and VREF pins must be a low inductance
chip capacitor as referenced previously in the data sheet.
AD9631
21Ω
AD9042
50Ω
200Ω
AIN
79Ω
0.1µF
0–50pF
114Ω
VOFFSET
1kΩ
Although a multilayer board is recommended, it is not required
to achieve good results. As shown in the DIP evaluation board
layout (Figures 39–42), the top layer forms a near solid ground
plane while the under side is used for routing signal. No vias
or jumpers are required to route signals in and out of the
AD9042AD. Each supply is decoupled to ground directly at the
device.
49.9Ω
571Ω
VREF
0.1µF
OP279
(1/2)
OP279
(1/2)
Figure 36. DC-Coupled Analog Input Circuit
Power Supplies
Care should be taken when selecting a power source. Linear
supplies are strongly recommended as switching supplies tend to
have radiated components that may be “received” by the
AD9042. Each of the power supply pins should be decoupled as
closely to the package as possible using 0.1 µF chip capacitors.
The AD9042 has separate digital and analog +5 V pins. The
analog supplies and the denoted AVCC digital supply pins are
denoted DVCC. Although analog and digital supplies may be
tied together, best performance is achieved when the supplies
are separate. This is because the fast digital output swings can
couple switching noise back into the analog supplies. Note that
AVCC must be held within 5% of 5 volts, however the DVCC
supply may be varied according to output digital logic family
(i.e., DVCC should be connected to the supply for the digital
circuitry).
Evaluation Boards
The evaluation board for the AD9042 is very straight forward
consisting of power, signal inputs and digital outputs. The
evaluation board includes an onboard clock oscillator for the
encode; all the user must supply is power and an analog signal.
Output Loading
Care must be taken when designing the data receivers for the
AD9042. It is recommended that the digital outputs drive a series resistor of 499 ohms followed by a CMOS gate like the
74AC574. To minimize capacitive loading, there should only
be one gate on each output pin. An example of this is shown in
the evaluation board schematics shown in Figures 37 and 38.
The digital outputs of the AD9042 have a unique constant slew
rate output stage. The output slew rate is about 1 V/ns
independent of output loading. A typical CMOS gate combined
with PCB trace and through hole will have a load of approximately 10 pF. Therefore as each bit switches, 10 mA

1V 
10 pF × 1ns  of dynamic current per bit will flow in or out of


the device. A full- scale transition can cause up to 120 mA (12
bits × 10 mA/bit) of current to flow through the digital output
stage. The series resistor will minimize the output currents that
can flow in the output stage. These switching currents are
confined between ground and the DVCC pin. Standard TTL
gates should be avoided since they can appreciably add to the
dynamic switching currents of the AD9042.
REV. A
Care should be taken when placing the digital output runs.
Because the digital outputs have such a high slew rate, the
capacitive loading on the digital outputs should be minimized.
Circuit traces for the digital outputs should be kept short and
connect directly to the receiving gate (broken only by the
insertion of the series resistor). Logic fanout for each bit should
be one CMOS gate.
Power to the analog supply pins is connected via banana jacks.
The analog supply powers the crystal oscillator and the AVCC
pins of the AD9042. The DVCC power is supplied via J3, the
digital interface. This digital supply connection also powers the
digital gates on the PCB. By maintaining separate analog and
digital power supplies, degradation in SNR and SFDR is kept
to a minimum. Total power requirement for either PCB is
approximately 140 mA. This configuration allows for easy
evaluation of different logic families (i.e., connection to a 3.3
volt logic board).
The analog input is connected via J2 and is capacitively coupled
to the AD9042 (see “Driving the Analog Input”). The onboard
termination resistor is 60.4 Ω. This resistor in parallel with
AD9042’s input resistance (250 Ω) provides a 50 Ω load to the
analog source. If a different input impedance is required,
replace R1 by using the following equation
1
R1 =
1
1
where Z is desired input impedance.
−
Z 250
The analog input range of PCB is ± 0.5 volts (i.e., signal accoupled to AD9042).
The encode signal is generated using the onboard crystal
oscillator, U1. The oscillator is socketed and may be replaced
by an external encode source via J1. If an external source is
used, it should be a high quality TTL source. A transformer
converts the single-ended TTL signal to a differential clock (see
“Encoding the AD9042”). Since the encode is coupled with a
–13–
AD9042
transformer, a sine wave could have been used; however, note
that U5 requires TTL levels to function properly.
AD9042 output data is latched using 74ACT574 (U3, U4)
latches following 499 ohm series resistors. The resistors limit
the current that would otherwise flow due to the digital output
slew rate. The resistor value was chosen to represent a time
constant of ~25% of the data rate at 40 MHz. This reduces slew
U5
74AS00
+5VA
C14
0.1µF
1
14
U3
74ACT574
U5
74AS00
4
3
9
6
8
BUFLAT
5
2
7
VCC
R2
499Ω
8
U1
K1115 OUT
VEE
7
BNC
J1
R15
100Ω
T1
T1–1T
3
4
+5V 2 DVCC
C2
0.1µF
C3
0.1µF
R5
499Ω
4
R6
499Ω
R7
499Ω
3
GND
GND
2
6Q
5D
5Q
4D
4Q
3D
3Q
2D
2Q
1D
1Q
GND 6 GND
D6 23
B10
H40DM
J3
B11
18
+5V
19
B11
OE
B10
1
B09
GND 7 GND
D5 22
B05
B04
D4 21
9 VOFFSET
D3 20
10 VREF
D2 19
GND 11 GND
D1 18
BUFLAT
B03
B02
B01
U4
74ACT574
(LSB) D0 17
GND 13 GND
NC 16
+5VA 14 AVCC
NC 15
R9
499Ω
R10
499Ω
R11
499Ω
R12
499Ω
R13
499Ω
9
8
7
6
C13
0.1µF
C15
0.1µF
C16
0.1µF
3
GND
2
8D
8Q
7D
7Q
6D
6Q
5D
5Q
4D
4Q
3D
3Q
2D
2Q
1D
1Q
CK
C9
0.1µF
17
B09
B06
GND
C8
0.1µF
16
B08
B07
+5VA
C4
10µF
15
B07
D7 24
4
+
14
B06
5 ENCODE
+5VA 12 AVCC
C12
0.1µF
6D
13
B08
+5V
C11
0.1µF
7Q
D8 25
5
C7
0.1µF
7D
12
4 ENCODE
R8
499Ω
C6
10µF
8Q
11
NC = NO CONNECT
+
8D
CK
D9 26
8 AIN
R14
49.9Ω
5
D10 27
GND 3 GND
R1
60.4Ω
R4
499Ω
(MSB) D11 28
GND 1 GND
1
6
6
R3
499Ω
U2
AD9042
2
BNC
J2
rate while not appreciably distorting the data waveform. Data is
latched in a pipeline configuration; a rising edge generates the
new AD9042 data sample, latches the previous data at the
converter output, and strobes the external data register over J3.
Power and ground must be applied to J3 to power the digital
logic section of the evaluation board.
C17
0.1µF
11
B00
GND
12
13
14
15
16
17
B00
B01
B02
GND
GND
GND
GND
1
40
2
39
3
38
4
37
5
36
6
35
7
34
8
33
9
32
10
31
11
30
12
29
13
28
14
27
15
26
16
25
17
24
18
23
19
22
20
21
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
B03
B04
B05
18
19
OE
1
Figure 37. AD9042D/PCB Schematic
Table I. AD9042D/PCB Bill of Material
Item
Quantity
Reference
Description
1
2
3
4
5
6
7
8
9
10
11
12
13
14
2
10
2
1
2
1
12
1
1
1
1
1
2
1
+5VA, GND
C2–C3, C7–C9, C11–C17
C4, C6
J3
J1, J2
R1
R2–R13
R14
R15
T1
U1
U2
U3, U4
U5
Banana Jack
Ceramic Chip Capacitor 0805, 0.1 µF
Tantalum Chip Capacitor 10 µF
40-Pin Double Row Male Header
BNC Coaxial PCB Connector
Surface Mount Resistor 1206, 60.4 ohms
Surface Mount Resistor 1206, 499 ohms
Surface Mount Resistor 1206, 49.9 ohms
Surface Mount Resistor 1206, 100 ohms
Surface Mount Transformer Mini-Circuits T1–1T
40.96 MHz Clock Oscillator
AD9042AD 12-Bit–41 MSPS ADC Converter
74ACT574 Octal Latch
74AS00 Quad Two Input NAND Gate
–14–
REV. A
AD9042
U3
74ACT574
U5
74AS00
U5
74AS00
1
3
6
5
2
+5VA
9
8
4
BUFLAT
7
R5
C14
0.1µF
6
R6
499Ω
14
GND
+5V
+5V
GND
GND
GND
D5
30
D4
7 AIN
D2
27
8 VOFFSET
D1
26
VREF
D0
GND
AVCC
AVCC
GND
GND
AVCC
NC
GND
+5VA
+5VA
GND
NC = NO CONNECT
5Q
4D
4Q
3D
3Q
2D
2Q
1D
1Q
CK
R4
499Ω
11
24 GND
R12
499Ω
23
9
R10
499Ω
8
R9
499Ω
7
6
4
+5VA
GND
GND
GND
GND
+5V
3
2
8D
C11
0.1µF
C12
0.1µF
C4
10µF
C8
0.1µF
C9
0.1µF
C17
0.1µF
C13
0.1µF
C15
0.1µF
17
B09
B10
19
1
C16
0.1µF
8Q
7Q
6D
6Q
5D
5Q
4D
4Q
3D
3Q
2D
2Q
1D
1Q
11
H40DM
J3
B11
18
OE
7D
CK
C7
0.1µF
16
B08
U4
74ACT574
R11
499Ω
5
C6
10µF
15
B07
R13
499Ω
+5VA
+
5D
14
B06
25
R8
499Ω
GND1
6Q
13
BUFLAT
16 17 18 19 20 21 22
GND
12 13 14 15
+5VA
AVCC
GND
+5VA 11
6D
12
29
28
GND
2
7Q
32
D3
10 C1
C1
0.01µF
GND
GND
DVCC
DVCC
GND
GND
4 ENCODE
9
C3
0.1µF
DVCC
31
GND
R1
60.4Ω
C18
0.01µF
D9
D6
6 GND
GND
R14
49.9Ω
DVCC
3 ENCODE
U2
AD9042
GND
8Q
7D
33
D7
GND
C2
0.1µF
D8
2 DVCC
5 GND
GND
GND
R3
499Ω
GND
1:1
BNC
J2
D10
D11
+5V
AVCC
1
6
1 DVCC
+5V
GND
2
3
R2
499Ω
44 43 42 41 40 39 38 37 36 35 34
GND
R15
100Ω
T1
T1–1T
3
4
+5VA
7
+5V
+5V
VEE
BNC
J1
4
499Ω
8
AVCC
OUT
R7
499Ω
VCC
+5VA
U1
K1115
5
8D
12
13
14
15
16
17
B00
+5V
1
40
B11
2
39
B10
3
38
B09
4
37
B08
5
36
B07
6
35
B06
7
34
B05
8
33
B04
9
32
10
31
11
30
B03 12
B02 13
29
B01 14
B00 15
27
GND 16
GND 17
25
GND 18
GND 19
23
GND 20
21
28
26
24
22
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
GND
B01
B02
B03
B04
B05
18
19
OE
1
+5VA
+
Figure 38. AD9042ST/PCB Schematic
Table II. AD9042ST/PCB Bill of Material
Item
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
REV. A
Quantity
2
10
2
1
2
1
12
1
1
1
1
1
2
1
1
Reference
+5VA, GND
C2–C3, C7–C9, C11–C17
C4, C6
J3
J1, J2
R1
R2–R13
R14
R15
T1
U1
U2
U3, U4
U5
C1, C18
Description
Banana Jack
Ceramic Chip Capacitor 0805, 0.1 µF
Tantalum Chip Capacitor 10 µF
40-Pin Double Row Male Header
BNC Coaxial PCB Connector
Surface Mount Resistor 1206, 60.4 ohms
Surface Mount Resistor 1206, 499 ohms
Surface Mount Resistor 1206, 49.9 ohms
Surface Mount Resistor 1206, 100 ohms
Surface Mount Transformer Mini-Circuits T1–1T
40.96 MHz Clock Oscillator
AD9042AST 12-Bit–41 MSPS ADC Converter
74ACT574 Octal Latch
74AS00 Quad Two Input NAND Gate
Ceramic Chip Capacitor 0805, 0.01 µF AVX05085C103MA15
–15–
AD9042
C14
M3
M4
C12
U1
U5
J1
AD9042
PIN 1
ENC
C7
U3
C13
J3
U2
GND
U4
GND
C11
R14
C3
C9
+5VA
+5VA
J2
AIN
C8
GND1
MICROPHONE
M1
TO ANTENNA™
AD9042D/PCB
EVALUATION
48391
a
1995 © USA
JB8/KU8
6 / 6 / 95
M2
Figure 39. AD9042D/PCB Top Side Silk Screen
Figure 41. AD9042D/PCB Top Side Copper (Negative)
C6
R7
R6
R5
R4
R3
R2
R8
R9
R10
R11
R15
C2 T1
R1
R12
R13
C4
Figure 40. AD9042D/PCB Bottom Side Silk Screen
Figure 42. AD9042D/PCB Bottom Side Copper (Negative)
–16–
REV. A
AD9042
C18
Figure 43. AD9042ST/PCB Top Side Silk Screen
Figure 45. AD9042ST/PCB Top Side Copper (Negative)
Figure 44. AD9042ST/PCB Bottom Side Silk Screen
Figure 46. AD9042ST/PCB Bottom Side Copper (Positive)
REV. A
–17–
AD9042
+5VA
Figure 47. AD9042ST/PCB Grounded Layer (Negative)
+5V
Figure 48. AD9042ST/PCB “Split” Power Layer (Negative)
–18–
REV. A
AD9042
DIGITAL WIDEBAND RECEIVERS
Introduction
Several key technologies are now being introduced that may
forever alter the vision of radio. Figure 49 shows the typical
dual conversion superheterodyne receiver. The signal picked up
by the antenna is mixed down to an intermediate frequency (IF)
using a mixer with a variable local oscillator (LO); the variable
LO is used to “tune-in” the desired signal. This first IF is
mixed down to a second IF using another mixer stage and a
fixed LO. Demodulation takes place at the second or third IF
using either analog or digital techniques.
LNA
RF
e.g. 900MHz
NARROWBAND
FILTER
IF1
VARIABLE
SHARED
NARROWBAND
FILTER
ADCs
I
Q
IF2
FIXED
ONE RECEIVER PER CHANNEL
Figure 49. Narrowband Digital Receiver Architecture
If demodulation takes place in the analog domain then
traditional discriminators, envelop detectors, phase locked loops
or other synchronous detectors are generally employed to strip
the modulation from the selected carrier.
However, as general purpose DSP chips such as the ADSP-2181
become more popular, they will be used in many basebandsampled applications like the one shown in Figure 49. As
shown in the figure, prior to ADC conversion, the signal must
be mixed down, filtered, and the I and Q components separated.
These functions are realizable through DSP techniques,
however several key technology breakthroughs are required:
high dynamic range ADCs such as the AD9042, new DSPs
(highly programmable with onboard memory, fast), digital tuner
& filter (with programmable frequency and BW) and wide band
mixers (high dynamic range with >12.5 MHz BW).
is used for demodulation, different routines may be used to
demodulate different standards such as AM, FM, GMSK or any
other desired standard. In addition, as new standards arise or
new software revisions are generated, they may be field installed
with standard software update channels. A radio that performs
demodulation in software as opposed to hardware is often
referred to as a soft radio because it may be changed or modified
simply through code revision.
System Description
In the wideband digital radio (Figure 50), the first down
conversion functions in much the same way as a block converter
does. An entire band is shifted in frequency to the desired
intermediate frequency. In the case of cellular base station
receivers, 5 MHz to 20 MHz of bandwidth are down-converted
simultaneously to an IF frequency suitable for digitizing with a
wideband analog-to-digital converter. Once digitized the
broadband digital data stream contains all of the in-band
signals. The remainder of the radio is constructed digitally using
special purpose and general purpose programmable DSP to
perform filtering, demodulation and signal conditioning not
unlike the analog counter parts.
In the narrowband receiver (Figure 49), the signal to be received
must be tuned. This is accomplished by using a variable local
oscillator at the first mix down stage. The first IF then uses a
narrow band filter to reject out of band signals and condition
the selected carrier for signal demodulation.
In the digital wideband receiver (Figure 50), the variable local
oscillator has been replaced with a fixed oscillator, so tuning
must be accomplished in another manner. Tuning is performed
digitally using a digital down conversion and filter chip frequently called a channelizer. The term channelizer is used
because the purpose of these chips is to select one channel out
of the many within the broadband of spectrum actually present
in the digital data stream of the ADC.
DATA
LNA
WIDEBAND
MIXER
WIDEBAND
FILTER
WIDEBAND
ADC
SHARED
CHANNEL SELECTION
Figure 50. Wideband Digital Receiver Architecture
Figure 50 shows such a wideband system. This design shows
that the front end variable local oscillator has been replaced with
a fixed oscillator (for single band radios) and the back end has
been replaced with a wide dynamic range ADC, digital tuner
and DSP. This technique offers many benefits.
First, many passive discrete components have been eliminated
that formed the tuning and filtering functions. These passive
components often require “tweaking” and special handling
during assembly and final system alignment. Digital components require no such adjustments; tuner and filter characteristics
are always exactly the same. Moreover, the tuning and filtering
characteristics can be changed through software. Since software
REV. A
I
DECIMATION
FILTER
LOW-PASS
FILTER
Q
DIGITAL
TUNER
Figure 51. Digital Channelizer
12.5MHz
(416 CHANNELS)
FIXED
LOW-PASS
FILTER
SIN
"n" CHANNELS
TO DSP
RF
e.g. 900MHz
DECIMATION
FILTER
COS
Figure 51 shows the block diagram of a typical channelizer.
Channelizers consist of a complex NCO (Numerically
Controlled Oscillator), dual multiplier (mixer), and matched
digital filters. These are the same functions that would be
required in an analog receiver, however implemented in digital
form. The digital output from the channelizer is the desired
carrier, frequently in I & Q format; all other signals have been
filtered and removed based on the filtering characteristics
desired. Since the channelizer output consists of one selected
RF channel, one tuner chip is required for each frequency
received, although only one wideband RF receiver is needed for
the entire band. Data from the channelizer may then be
processed using a digital signal processor such as the ADSP2181 or the SHARC processor, the ADSP-21062. This data
may then be processed through software to demodulate the
information from the carrier.
–19–
AD9042
+5V (A)
PRESELECT
FILTER
+5V (D)
5–15MHz
PASSBAND
LNA
ADSP-2181
D11
I&Q
DATA
12
AD9042
864MHz
CHANNELIZER
(REF. FIG 51)
499Ω
AIN
LO
DRIVE
CMOS
BUFFER
NETWORK
CONTROLLER
INTERFACE
ENCODE
M/N PLL
SYNTHESIZER
REF
IN
ENCODE
CLK
D0
40.96MHz
REFERENCE
CLOCK
Figure 52. Simplified 5 MHz Wideband “A” Carrier Receiver
System Requirements
Another option can be found through bandpass sampling. If the
analog input signal range is from dc to FS/2, then the amplifier
and filter combination must perform to the specification
required. However, if the signal is placed in the third Nyquist
zone (FS to 3 FS/2), the amplifier is no longer required to meet
the harmonic performance required by the system specifications
since all harmonics would fall outside the passband filter. For
example, the passband filter would range from FS to 3 FS/2.
The second harmonic would span from 2 FS to 3 FS, well
outside the passband filter’s range. The burden then has been
passed off to the filter design provided that the ADC meets the
basic specifications at the frequency of interest. In many
applications, this is a worthwhile tradeoff since many complex
filters can easily be realized using SAW and LCR techniques
alike at these relatively high IF frequencies. Although harmonic
performance of the drive amplifier is relaxed by this technique,
intermodulation performance cannot be sacrificed since
intermods must be assumed to fall in-band for both amplifiers
and converters.
Figure 52 shows a typical wideband receiver subsystem based
around the AD9042. This strip consists of a wideband IF filter,
amplifier, ADC, latches, channelizer and interface to a digital
signal processor. This design shows a typical clocking scheme
used in many receiver designs. All timing within the system is
referenced back to a single clock. While this is not necessary, it
does facilitate PLL design, ease of manufacturing, system test,
and calibration. Keeping in mind that the overall performance
goal is to maintain the best possible dynamic range, many
considerations must be made.
One of the biggest challenges is selecting the amplifier used to
drive the AD9042. Since this is a communications application,
the key specification for this amplifier is spurious-free dynamic
range, or SFDR. An amplifier should be selected that can
provide SFDR performance better than 80 dB into 250 ohms.
One such amplifier is the AD9631. These low spurious levels
are necessary as harmonics due to the drive amplifier and ADC
could distort the desired signals of interest.
Two other key considerations for the digital wideband receiver
are converter sample rate and IF frequency range. Since
performance of the AD9042 converter is nearly independent of
both sample rate and analog input frequency (Figures 11, 12,
and 17), the designer has greater flexibility in the selection of
these parameters. Also, since the AD9042 is a bipolar device,
power dissipation is not a function of sample rate. Thus there is
no penalty paid in power by operating at faster sample rates. All
of this is good, because by carefully selecting input frequency
range and sample rate, the drive amplifier and ADC harmonics
can actually be placed out-of-band. Thus other components
such as filters and IF amplifiers may actually end up being the
limiting factor on dynamic range.
For example, if the system has second and third harmonics that
are unacceptably high, by carefully selecting the encode rate and
signal bandwidth, these second and third harmonics can be
placed out-of-band. For the case of an encode rate equal to
40.96 MSPS and a signal bandwidth of 5.12 MHz, placing the
fundamental at 5.12 MHz places the second and third harmonics out of band as shown in the table below.
Table III.
Encode Rate
Fundamental
Second Harmonic
Third Harmonic
40.96 MSPS
5.12 MHz–10.24 MHz
10.24 MHz–20.48 MHz
15.36 MHz–10.24 MHz
Noise Floor and SNR
Oversampling is the act of sampling at a rate that is greater than
twice the bandwidth of the signal desired. Oversampling does
not have anything to do with the actual frequency of the
sampled signal, it is the bandwidth of the signal that is key.
Bandpass or “IF” sampling refers to sampling a frequency that
is higher than Nyquist and often provides additional benefits
such as down conversion using the ADC and track-and-hold as
a mixer. Oversampling leads to processing gains because the
faster the signal is digitized, the wider the distribution of noise.
Since the integrated noise must remain constant, the actual
noise floor is lowered by 3 dB each time the sample rate is
doubled. The effective noise density for an ADC may be
calculated by the equation:
V NOISE rms /
Hz =
10 − SNR /20
4 FS
For a typical SNR of 68 dB and a sample rate of 40.96 MSPS,
this is equivalent to 31 nV / Hz . This equation shows the
relationship between SNR of the converter and the sample rate
FS. This equation may be used for computational purposes to
determine overall receiver noise.
The signal-to-noise ratio (SNR) for an ADC can be predicted.
When normalized to ADC codes, the following equation
accurately predicts the SNR based on three terms. These are
jitter, average DNL error and thermal noise. Each of these
terms contributes to the noise within the converter.
–20–
REV. A
AD9042
Equation 1:
1/2
2
2

 VNOISE rms  
2  1+ ε 
SNR = –20 log  2 πFANALOG t J rms +  12  + 
 
212
2  

 


FANALOG = analog input frequency
= rms jitter of the encode (rms sum of encode source
t J rms
and internal encode circuitry)
ε
= average DNL of the ADC
VNOISE rms = V rms thermal noise referred to the analog input of
the ADC
(
)
Processing Gain
Processing gain is the improvement in signal-to-noise ratio
(SNR) gained through DSP processes. Most of this processing
gain is accomplished using the channelizer chips. These special
purpose DSP chips not only provide channel selection and
filtering but also provide a data rate reduction. Few, if any,
general purpose DSPs can accept and process data at
40.96 MSPS. The required rate reduction is accomplished
through a process called decimation. The term decimation rate
is used to indicate the ratio of input data rate to output data
rate. For example, if the input data rate is 40.96 MSPS and the
output data rate is 30 kSPS, then the decimation rate is 1365.
linearity to appear as if it were random. Then, the average
linearity over the range of dither will dominate SFDR
performance. In the AD9042, the repetitive cycle is every
15.625 mV p-p.
To insure adequate randomization, 5.3 mV rms is required;
this equates to a total dither power of –32.5 dBm. This will
randomize the DNL errors over the complete range of the
residue converter. Although lower levels of dither such as that
from previous analog stages will reduce some of the linearity
errors, the full effect will only be gained with this larger dither.
Increasing dither even more may be used to reduce some of the
global INL errors. However, signals much larger than the mVs
proposed here begin to reduce the usable dynamic range of the
converter.
Even with the 5.3 mV rms of noise suggested, SNR would be
limited to 36 dB if injected as broadband noise. To avoid this
problem, noise may be injected as an out-of-band signal. Typically,
this may be around dc but may just as well be at FS/2 or at
some other frequency not used by the receiver. The bandwidth
of the noise is several hundred kilohertz. By band-limiting and
controlling its location in frequency, large levels of dither may
be introduced into the receiver without seriously disrupting
receiver performance. The result can be a marked improvement
in the SFDR of the data converter.
Large processing gains may be achieved in the decimation and
filtering process. The purpose of the channelizer, beyond
tuning, is to provide the narrowband filtering and selectivity that
traditionally has been provided by the ceramic or crystal filters
of a narrowband receiver. This narrowband filtering is the
source of the processing gain associated with a wideband
receiver and is simply the ratio of the passband to whole band
expressed in dB. For example, if a 30 kHz AMPS signal is
being digitized with an AD9042 sampling at 40.96 MSPS, the
ratio would be 0.030 MHz/20.48 MHz. Expressed in log form,
the processing gain is –10 × log (0.030 MHz / 20.48 MHz) or
28.3 dB!
Figure 23 shows the same converter shown earlier but with this
injection of dither (ref. Figure 20). Spurious-free dynamic
range is now 94 dBFS. Figure 21 and 24 show an SFDR sweep
before and after adding dither.
To more fully appreciate the improvement that dither can have
on performance, Figures 22 and 25 show a before-and-after
dither using additional data samples in the Fourier transform.
Increasing to 128k sample points lowers the noise floor of the
FFT; this simply makes it easier to “see” the dramatic reduction
in spurious levels resulting from dither.
+15V
Additional filtering and noise reduction techniques can be
achieved through DSP techniques; many applications do use
additional process gains through proprietary noise reduction
algorithms.
16kΩ
REV. A
A
14
3
2.2kΩ
4
13
+5V
12
–5V
2kΩ
REF
5
Typically, high resolution data converters use multistage
techniques to achieve high bit resolution without large
comparator arrays that would be required if traditional “flash”
ADC techniques were employed. The multistage converter
typically provides better wafer yields meaning lower cost and
much lower power. However, since it is a multistage device,
certain portions of the circuit are used repetitively as the analog
input sweeps from one end of the converter range to the other.
Although the worst DNL error may be less than an LSB, the
repetitive nature of the transfer function can play havoc with low
level dynamic signals. Spurious signals for a full-scale input
may be –88 dBc, however 29 dB below full scale, these repetitive DNL errors may cause spurious-free dynamic range (SFDR)
to fall to 80 dBc as shown in Figure 20.
LOW CONTROL
(0–1 VOLT)
15
2
NC202
NOISE
DIODE
(NoiseCom)
Overcoming Static Nonlinearities with Dither
A common technique for randomizing and reducing the effects
of repetitive static linearity is through the use of dither. The
purpose of dither is to force the repetitive nature of static
16
1
1µF
1kΩ
11
6
A
10
7
8
0.1µF
39Ω
AD600
OP27
9
OPTIONAL HIGH
POWER DRIVE
CIRCUIT
390Ω
Figure 53. Noise Source (Dither Generator)
The simplest method for generating dither is through the use of
a noise diode (Figure 53). In this circuit, the noise diode
NC202 generates the reference noise that is gained up and
driven by the AD600 and OP27 amplifier chain. The level of
noise may be controlled by either presetting the control voltage
when the system is set up, or by using a digital-to-analog
converter (DAC) to adjust the noise level based on input signal
conditions. Once generated, the signal must be introduced to
the receiver strip. The easiest method is to inject the signal into
the drive chain after the last down conversion as shown in
Figure 54.
–21–
AD9042
present in the ADC bandwidth, then each must be placed 18 dB
below full scale to prevent ADC overdrive. In addition, 3 dB to
15 dB should be used for ADC headroom should another signal
come in-band unexpectedly. For this example, 12 dB of
headroom will be allocated. Therefore we give away 30 dB of
range and reduce the carrier-to-noise ratio (C/N)* to 54.8 dB.
FROM
RF/IF
AIN
AD9042
VOFFSET
NOISE SOURCE
LPF
Assuming that the C/N ratio must be 6 dB or better for accurate
demodulation, one of the eight signals may be reduced by 48.8 dB
before demodulation becomes unreliable. At this point, the
input signal power would be 40.6 µV rms on the ADC input or
–74.8 dBm. Referenced to the antenna, this is –104.8 dBm.
VREF
(REF. FIGURE 53)
Figure 54. Using the AD9042 with Dither
Receiver Example
To determine how the ADC performance relates to overall
receiver sensitivity, the simple receiver in Figure 55 will be
examined. This example assumes that the overall down
conversion process can be grouped into one set of specifications,
instead of individually examining all components within the
system and summing them together. Although a more detailed
analysis should be employed in a real design, this model will
provide a good approximation.
In examining a wideband digital receiver, several considerations
must be applied. Although other specifications are important,
receiver sensitivity determines the absolute limits of a radio
excluding the effects of other outside influences. Assuming that
receiver sensitivity is limited by noise and not adjacent signal
strength, several sources of noise can be identified and their
overall contribution to receiver sensitivity calculated.
GAIN = 30dB
NF = 20dB
BW =12.5MHz
SINGLE CHANNEL
BW = 30kHz
RF/IF
AD9042
REF IN
ENC
CHANNELIZER
DSP
40.96MHz
To improve sensitivity, several things can be done. First, the
noise figure of the receiver can be reduced. Since front end
noise dominates the 0.529 mV rms, each dB reduction in noise
figure translates to an additional dB of sensitivity. Second, providing broadband AGC can improve sensitivity by the range of
the AGC. However, the AGC would only provide useful improvements if all in-band signals are kept to an absolute minimal
power level so that AGC can be kept near the maximum gain.
This noise limited example does not adequately demonstrate the
true limitations in a wideband receiver. Other limitations such
as SFDR are more restrictive than SNR and noise. Assume that
the analog-to-digital converter has an SFDR specification of
–80 dBFS or –76 dBm (Full scale = +4 dBm). Also assume
that a tolerable carrier-to-interferer (C/I)** (different from C/N)
ratio is 18 dB. This means that the minimum signal level is
–62 dBFS (–80 plus 18) or –58 dBm. At the antenna, this is
–88 dBm. Therefore, as can be seen, SFDR (single or multitone) would limit receiver performance in this example.
However, as shown previously, SFDR can be greatly improved
through the use of dither (Figures 22, 25). In many cases, the
addition of the out-of-band dither can improve receiver
sensitivity nearly to that limited by thermal noise.
Multitone Performance
The first noise calculation to make is based on the signal bandwidth at the antenna. In a typical broadband cellular receiver,
the IF bandwidth is 12.5 MHz. Given that the power of noise
in a given bandwidth is defined by Pn = kTB, where B is
bandwidth, k = 1.38 × 10–23 is Boltzman’s constant and
T = 300k is absolute temperature, this gives an input noise
power of 5.18 × 10–14 watts or –102.86 dBm. If our receiver
front end has a gain of 30 dB and a noise figure of 20 dB, then
the total noise presented to the ADC input becomes –52.86 dBm
(–102.86 + 30 + 20) or 0.51 mV rms. Comparing receiver
noise to dither required for good SFDR, we see that in this
example, our receiver supplies about 10% of the dither required
for good SFDR.
The plot below shows the AD9042 in a worst case scenario of
four strong tones spaced fairly close together. In this plot no
dither was used, and the converter still maintained 85 dBFS of
spurious-free range. As illustrated previously, a modest amount
of dither introduced out-of-band could be used to lower the
nonlinear components.
0
POWER RELATIVE TO ADC FULL SCALE – dB
Figure 55. Receiver Analysis
Based on a typical ADC SNR specification of 68 dB, the
equivalent internal converter noise is 0.140 mV rms. Therefore
total broadband noise is 0.529 mV rms. Before processing gain,
this is an equivalent SNR (with respect to full scale) of 56.5 dB.
Assuming a 30 kHz AMPS signal and a sample rate of
40.96 MSPS, the SNR through processing gain is increased by
28.3 dB to 84.8 dB. However, if 8 strong and equal signals are
–20
ENCODE = 41 MSPS
–40
–60
3
6
9
7
4
2
5
8
–80
–100
–120
dc
4.1
8.2
12.3
FREQUENCY – MHz
16.4
20.5
Figure 56. Multitone Performance
**C/N is the ratio of signal to inband noise.
**C/I is the ratio of signal to inband interferer.
–22–
REV. A
AD9042
IF Sampling, Using the AD9042 as a Mix-Down Stage
Since performance of the AD9042 extends beyond the baseband
region into the second and third Nyquist zone, the converter
may find many uses as a mix down converter in both narrowband
and wideband applications. Many common IF frequencies exist
in this range of frequencies. If the ADC is used to sample these
signals, they will be aliased down to baseband during the sampling
process in much the same manner that a mixer will down-convert a
signal. For signals in various Nyquist zones, the following
equation may be used to determine the final frequency after
aliasing.
f 1NYQUISTS = f SAMPLE − f SIGNAL
f 2NYQUISTS = abs ( f SAMPLE − f SIGNAL )
f 3NYQUISTS = 2 × f SAMPLE − f SIGNAL
f 4NYQUISTS = abs (2 × f SAMPLE − f SIGNAL )
Using the converter to alias down these narrowband or wideband
signals has many potential benefits. First and foremost is the
elimination of a complete mixer stage, along with amplifiers,
filters and other devices, reducing cost and power dissipation.
RECEIVE CHAIN FOR DIGITAL BEAM-FORMING
MEDICAL ULTRASOUND USING THE AD9042
The AD9042 is an excellent digitizer for digital and analog
beam-forming medical ultrasound systems. The price/
performance ratio of the AD9042 allows ultrasound designers
the luxury of using state-of-the-art ADCs without jeopardizing
their cost budgets. ADC performance is critical for image
quality. The high dynamic range and excellent noise
performance of the AD9042 enable higher image quality
medical ultrasound systems.
Figure 58 shows the AD9042 used in one channel of the receive
chain of a medical ultrasound system. The AD604 receives its
input directly from the transducer, or from an external preamp
connected to the transducer. The AD604 contains two separate
stages. The first stage is a preamp with a fixed gain (14 dB to
20 dB) selected by a fixed resistor. The second stage is a
variable gain amplifier with the gain set by the AD7226 DAC.
The gain is increased over time to compensate for the attenuation of signal level in the body.
PRE-AMP
14 TO 20dB
One common example is the digitization of a 21.4 MHz IF
using a 10 MSPS sample clock. Using the equation above for
the fifth Nyquist zone, the resultant frequency after sampling is
1.4 MHz. Figure 57 shows performance under these conditions.
Even under these conditions, the AD9042 typically maintains
better than 80 dB SFDR.
VGA
–14 TO 34dB
TRANSDUCER/
PRE-AMP
INPUT
AD9042
AD8041
AD604
LPF
AD7226
POWER RELATIVE TO ADC FULL SCALE – dB
0
Figure 58. Using the AD9042 in Ultrasound Applications
ENCODE = 10.0 MSPS
AIN = 21.4MHz
–20
Following the AD604, a low-pass filter is used to minimize the
amount of noise presented to the ADC. The AD8041 is used to
buffer the filter from the AD9042 input. This function may not
be required depending on the filter configuration and PC board
partitioning. The digital outputs of the AD9042 are then
presented to the digital system for processing.
–40
–60
8
7
8
6
2
5
3
4
–80
–100
–120
dc
1.0
2.0
3.0
FREQUENCY – MHz
4.0
5.0
Figure 57. IF-Sampling a 21.4 MHz Input
REV. A
–23–
AD9042
AD9042AST OUTLINE DIMENSIONS
Dimensions shown in inches and (mm)
0.063 (1.60)
MAX
0.472 (12.00) BSC SQ
0.030 (0.75)
0.018 (0.45)
34
44
1
SEATING
PLANE
33
0.393
(10.0)
BSC
SQ
TOP VIEW
(PINS DOWN)
11
0.006 (0.15)
0.002 (0.05)
C2080a–10–5/96
44-Pin Thin Quad Flatpack
(ST-44)
0.039
(1.00)
REF
0.008 (0.20)
0.003 (0.09)
23
12
22
0.031 (0.80)
BSC
0.018 (0.45)
0.012 (0.30)
AD9042AD OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
28-Pin Hermetic Ceramic DIP
(DH-28)
28
15
0.595 ± 0.010
(15.11 ± 0.25)
1
PIN 1 IDENTIFIERS
0.225
(5.72)
MAX
14
0.050 ± 0.010
(1.27 ± 0.25)
1.400 ± 0.014
(35.56 ± 0.35)
0.150
(3.81)
MIN
0.100 (2.54) 0.05 (1.27)
TYP
TYP
SEATING
PLANE
0.600 (15.24)
REF
PRINTED IN U.S.A.
0.018 ± 0.002
(0.46 ± 0.05)
0.010 ± 0.002
(0.25 ± 0.05)
–24–
REV. A