Final Electrical Specifications LTC3803 Constant Frequency Current Mode Flyback DC/DC Controller in ThinSOT August 2003 U DESCRIPTION FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ The LTC®3803 is a constant frequency current mode flyback controller optimized for driving 6V-rated N-channel MOSFETs in high input voltage applications. Constant frequency operation is maintained down to very light loads, resulting in less low frequency noise generation over a wide range of load currents. Slope compensation can be programmed with an external resistor. VIN and VOUT Limited Only by External Components Adjustable Slope Compensation Internal Soft-Start Constant Frequency 200kHz Operation ±1.5% Reference Accuracy Current Mode Operation for Excellent Line and Load Transient Response No Minimum Load Requirement Low Quiescent Current: 240µA Low Profile (1mm) SOT-23 Package The LTC3803 provides ±1.5% output voltage accuracy and consumes only 240µA of quiescent current. Groundreferenced current sensing allows LTC3803-based converters to accept input supplies beyond the LTC3803’s absolute maximum VCC. A micropower hysteretic start-up feature allows efficient operation at high input voltages. For simplicity, the LTC3803 can also be powered from a high VIN through a resistor, due to its internal 9.4V shunt regulator. An internal undervoltage lockout shuts down the LTC3803 when the input voltage falls below 4.6V, guaranteeing at least 4.6V of gate drive to the external MOSFET. U APPLICATIO S ■ ■ ■ ■ Telecom Power Supplies 42V and 12V Automotive Power Supplies Auxiliary/Housekeeping Power Supplies Power Over Ethernet , LTC and LT are registered trademarks of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. The LTC3803 is available in a low profile (1mm) 6-lead SOT-23 (ThinSOTTM) package. U TYPICAL APPLICATIO 5V Output Nonisolated Telecom Housekeeping Power Supply 90 UPS840 T1 10k 10µF 10V X5R 0.0022µF • • 300µF* 6.3V X5R VCC ITH/RUN NGATE 56k 4.7µF 100V X5R VOUT 5V 2A MAX FDC2512 LTC3803 GND SENSE VFB 20k 80 VIN = 36V 70 EFFICIENCY (%) VIN 36V TO 72V Efficiency vs Load Current VIN = 48V 60 VIN = 60V 50 40 VIN = 72V 30 20 68mΩ 10 105k 3803 TA01 0 0.1 T1: COOPER CTX02-15242 *THREE 100µF UNITS IN PARALLEL 1 10 IOUT (A) 3803 TA02 3803i Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 1 LTC3803 W U U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) VCC to GND Low Impedance Source .......................... – 0.3V to 8V Current Fed ...................................... 25mA into VCC* NGATE Voltage ......................................... – 0.3V to VCC VFB, ITH/RUN Voltages ..............................– 0.3V to 3.5V SENSE Voltage ........................................... – 0.3V to 1V NGATE Peak Output Current (<10µs) ........................ 1A Operating Temperature Range (Note 2) .. – 40°C to 85°C Junction Temperature (Note 3) ............................ 150°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW LTC3803ES6 6 NGATE ITH/RUN 1 5 VCC GND 2 VFB 3 4 SENSE S6 PART MARKING S6 PACKAGE 6-LEAD PLASTIC TSOT-23 TJMAX = 150°C, θJA = 230°C/W LTACV Consult LTC Marketing for parts specified with wider operating temperature ranges. *LTC3803 internal clamp circuit self regulates VCC voltage to 9.5V. ELECTRICAL CHARACTERISTICS The ● indicates specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 8V, unless otherwise noted. (Note 2) SYMBOL VTURNON VTURNOFF VHYST VCLAMP1mA VCLAMP25mA VMARGIN ICC VITHSHDN IITHSTART VFB PARAMETER VCC Turn On Voltage VCC Turn Off Voltage VCC Hysteresis VCC Shunt Regulator Voltage VCC Shunt Regulator Voltage VCLAMP1mA – VTURNON Margin Input DC Supply Current Normal Operation Start-Up Shutdown Threshold (at ITH/RUN) Start-Up Current Source Regulated Feedback Voltage gm ∆VO(LINE) ∆VO(LOAD) Error Amplifier Transconductance Output Voltage Line Regulation Output Voltage Load Regulation IFB fOSC DCON(MIN) DCON(MAX) tRISE tFALL VIMAX ISLMAX tSFST VFB Input Current Oscillator Frequency Minimum Switch On Duty Cycle Maximum Switch On Duty Cycle Gate Drive Rise Time Gate Drive Fall Time Peak Current Sense Voltage Peak Slope Compensation Output Current Soft-Start Time CONDITIONS ● ● VTURNON – VTURNOFF ICC = 1mA, VITH/RUN = 0V ICC = 25mA, VITH/RUN = 0V ● ● ● ● (Note 4) VITH/RUN = 1.3V VCC = VTURNON – 100mV VCC = VTURNON + 100mV VITH/RUN = 0V 0°C ≤ TA␣ ≤ 85°C (Note 5) –40°C␣ ≤ TA ≤ 85°CC (Note 5) ITH/RUN Pin Load = ±5µA (Note 5) VTURNOFF < VCC < VCLAMP (Note 5) ITH/RUN Sinking 5µA (Note 5) ITH/RUN Sourcing 5µA (Note 5) (Note 5) VITH/RUN = 1.3V VITH/RUN = 1.3V, VFB = 0.8V VITH/RUN = 1.3V, VFB = 0.8V CLOAD = 3000pF CLOAD = 3000pF RSL = 0 (Note 6) (Note 7) Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3803E is guaranteed to meet specifications from 0°C to 70°C. Specifications over the – 40°C to 85°C operating temperature range MIN 7.8 4.6 1.5 8.3 8.4 0.05 ● ● ● 0.15 0.2 0.788 0.780 200 180 70 ● 90 TYP 8.7 5.7 3.0 9.4 9.5 0.6 MAX 9.2 6.8 240 40 0.28 0.3 0.800 0.800 333 0.05 3 3 10 200 6 80 40 40 100 5 1.4 350 90 0.45 0.4 0.812 0.812 500 10.3 10.5 50 240 8 90 115 UNITS V V V V V V µA µA V µA V V µA/V mV/V mV/µA mV/µA nA kHz % % ns ns mV µA ms are assured by design, characterization and correlation with statistical process controls. Note 3: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: 3803i 2 LTC3803 ELECTRICAL CHARACTERISTICS TJ = TA + (PD • 230°C/W). Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 5: The LTC3803 is tested in a feedback loop that servos VFB to the output of the error amplifier while maintaing ITH/RUN at the midpoint of the current limit range. Note 6: Peak current sense voltage is reduced dependent on duty cycle and an optional external resistor in series with the SENSE pin (RSL). For details, refer to the programmable slope compensation feature in the Applications Information section. Note 7: Guaranteed by design. U W TYPICAL PERFOR A CE CHARACTERISTICS Reference Voltage vs Supply Voltage Reference Voltage vs Temperature 801.0 VCC = 8V 799.5 799.0 798.5 798.0 802 800.4 800.2 800.0 799.8 799.6 799.0 10 30 50 70 90 6 110 6.5 TEMPERATURE (°C) 8 7.5 8.5 VCC SUPPLY VOLTAGE (V) 7 9 Oscillator Frequency vs Temperature 796 9.5 Oscillator Frequency vs Supply Voltage 210 VCC = 8V 200 195 190 185 210 TA = 25°C 110 3803 G04 10 15 ICC (mA) 20 25 TA = 25°C 208 206 204 202 200 198 196 194 192 90 5 Oscillator Frequency vs VCC Shunt Regulator Current OSCILLATOR FREQUENCY (kHz) OSCILLATOR FREQUENCY (kHz) 205 0 3803 G03 208 OSCILLATOR FREQUENCY (kHz) 799 3803 F02 3803 G01 180 –50 –30 –10 10 30 50 70 TEMPERATURE (°C) 800 797 799.2 797.0 –50 –30 –10 801 798 799.4 797.5 210 TA = 25°C 803 800.6 VFB VOLTAGE (mV) VFB VOLTAGE (mV) 800.0 804 TA = 25°C VCC ≤ VCLAMP1mA 800.8 VFB VOLTAGE (mV) 800.5 Reference Voltage vs VCC Shunt Regulator Current 206 204 202 200 198 196 194 192 190 190 6 6.5 8.5 7.5 8 7 VCC SUPPLY VOLTAGE (V) 9 3803 G05 0 5 15 10 ICC (mA) 20 25 3803 G06 3803i 3 LTC3803 U W TYPICAL PERFOR A CE CHARACTERISTICS VCC Shunt Regulator Voltage vs Temperature ICC Supply Current vs Temperature 10.0 265 9.5 9.9 260 9.8 255 9.0 VTURNON 8.5 9.7 8.0 9.6 7.5 9.4 6.5 9.3 VTURNOFF ICC = 25mA 9.5 7.0 SUPPLY CURRENT (µA) 10.0 VCC (V) VCC UNDERVOLTAGE LOCKOUT (V) VCC Undervoltage Lockout Thresholds vs Temperature ICC = 1mA 240 235 230 225 5.5 9.1 220 5.0 –50 –30 –10 10 30 50 80 TEMPERATURE (°C) 9.0 –50 –30 –10 10 30 50 70 TEMPERATURE (°C) 90 110 90 Start-Up ICC Supply Current vs Temperature ITH/RUN Shutdown Threshold vs Temperature SHUTDOWN THRESHOLD (mV) 50 40 30 20 10 600 400 350 300 250 200 150 0 –50 –30 –10 10 30 50 70 TEMPERATURE (°C) 90 110 100 –50 –30 –10 10 30 50 70 400 300 200 100 0 –50 –30 –10 10 30 50 70 TEMPERATURE (°C) 90 110 3803 G12 Soft-Start Time vs Temperature VCC = 8V 115 3.5 110 3.0 105 100 95 90 85 80 –50 –30 –10 10 30 50 70 TEMPERATURE (°C) 500 4.0 SOFT-START TIME (ms) SENSE PIN VOLTAGE (mV) 110 VCC = VTURNON + 0.1V VITH/RUN = 0V 3803 G11 Peak Current Sense Voltage vs Temperature 120 90 TEMPERATURE (°C) 3803 G10 110 ITH/RUN Start-Up Current Source vs Temperature 450 VCC = VTURNON – 0.1V 90 3803 G08 ITH/RUN PIN CURRENT SOURCE (nA) 60 215 –50 –30 –10 10 30 50 70 TEMPERATURE (°C) 110 3803 G08 3803 G07 START-UP SUPPLY CURRENT (µA) 250 245 9.2 6.0 VCC = 8V VITH/RUN = 1.3V 2.5 2.0 1.5 1.0 0.5 90 110 3803 G13 0 –50 –30 –10 10 30 50 70 TEMPERATURE (°C) 90 110 3803 G14 3803i 4 LTC3803 U U U PI FU CTIO S ITH/RUN (Pin 1): This pin performs two functions. It serves as the error amplifier compensation point as well as the run/shutdown control input. Nominal voltage range is 0.7V to 1.9V. Forcing this pin below 0.28V causes the LTC3803 to shut down. In shutdown mode, the NGATE pin is held low. SENSE (Pin 4): This pin performs two functions. It monitors switch current by reading the voltage across an external current sense resistor to ground. It also injects a current ramp that develops slope compensation voltage across an optional external programming resistor. GND (Pin 2): Ground Pin. VCC (Pin 5): Supply Pin. Must be closely decoupled to GND (Pin 2). VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output. NGATE (Pin 6): Gate Drive for the External N-Channel MOSFET. This pin swings from 0V to VCC. W BLOCK DIAGRA 5 VCC 0.3µA 0.28V 800mV REFERENCE VCC SHUNT REGULATOR + SHUTDOWN COMPARATOR VCC < VTURNON – SHUTDOWN SOFTSTART CLAMP + – ERROR AMPLIFIER CURRENT COMPARATOR 3 GND 2 VCC R + VFB UNDERVOLTAGE LOCKOUT Q S – 20mV 1.2V 200kHz OSCILLATOR SWITCHING LOGIC AND BLANKING CIRCUIT GATE DRIVER NGATE SLOPE COMP CURRENT RAMP SENSE 1 6 4 ITH/RUN 3803 BD 3803i 5 LTC3803 U OPERATIO The LTC3803 is a constant frequency current mode controller for flyback and DC/DC boost converter applications in a tiny ThinSOT package. The LTC3803 is designed so that none of its pins need to come in contact with the input or output voltages of the power supply circuit of which it is a part, allowing the conversion of voltages well beyond the LTC3803’s absolute maximum ratings. Main Control Loop Due to space limitations, the basics of current mode DC/DC conversion will not be discussed here; instead, the reader is referred to the detailed treatment in Application Note 19, or in texts such as Abraham Pressman’s Switching Power Supply Design. Please refer to the Block Diagram and the Typical Application on the front page of this data sheet. An external resistive voltage divider presents a fraction of the output voltage to the VFB pin. The divider must be designed so that when the output is at the desired voltage, the VFB pin voltage will equal the 800mV from the internal reference. If the load current increases, the output voltage will decrease slightly, causing the VFB pin voltage to fall below 800mV. The error amplifier responds by feeding current into the ITH/RUN pin. If the load current decreases, the VFB voltage will rise above 800mV and the error amplifier will sink current away from the ITH/RUN pin. The voltage at the ITH/RUN pin commands the pulse-width modulator formed by the oscillator, current comparator and RS latch. Specifically, the voltage at the ITH/RUN pin sets the current comparator’s trip threshold. The current comparator monitors the voltage across a current sense resistor in series with the source terminal of the external MOSFET. The LTC3803 turns on the external power MOSFET when the internal free-running 200kHz oscillator sets the RS latch. It turns off the MOSFET when the current comparator resets the latch or when 80% duty cycle is reached, whichever happens first. In this way, the peak current levels through the flyback transformer’s primary and secondary is controlled by the ITH/RUN voltage. Since the ITH/RUN voltage is increased by the error amplifier whenever the output voltage is below nominal, and decreased whenever output voltage exceeds nominal, the voltage regulation loop is closed. For example, whenever the load current increases, output voltage will decrease slightly, and sensing this, the error amplifier raises the ITH/RUN voltage by sourcing current into the ITH/RUN pin, raising the current comparator threshold, thus increasing the peak currents through the transformer primary and secondary. This delivers more current to the load, bringing the output voltage back up. The ITH/RUN pin serves as the compensation point for the control loop. Typically, an external series RC network is connected from ITH/RUN to ground and is chosen for optimal response to load and line transients. The impedance of this RC network converts the output current of the error amplifier to the ITH/RUN voltage which sets the current comparator threshold and commands considerable influence over the dynamics of the voltage regulation loop. Start-Up/Shutdown The LTC3803 has two shutdown mechanisms to disable and enable operation: an undervoltage lockout on the VCC supply pin voltage, and a forced shutdown whenever external circuitry drives the ITH/RUN pin low. The LTC3803 transitions into and out of shutdown according to the state diagram (Figure 1). LTC3803 SHUT DOWN VCC < VTURNOFF (NOMINALLY 5.7V) V > VITHSHDN VITH/RUN < VITHSHDN ITH/RUN AND VCC > VTURNON (NOMINALLY 0.28V) (NOMINALLY 8.7V) LTC3803 ENABLED 3803 F01 Figure 1. Start-Up/Shutdown State Diagram 3803i 6 LTC3803 U OPERATIO The undervoltage lockout (UVLO) mechanism prevents the LTC3803 from trying to drive a MOSFET with insufficient VGS. The voltage at the VCC pin must exceed VTURNON (nominally 8.7V) at least momentarily to enable LTC3803 operation. The VCC voltage is then allowed to fall to VTURNOFF (nominally 5.7V) before undervoltage lockout disables the LTC3803. This wide UVLO hysteresis range supports the use of a bias winding on the flyback transformer to power the LTC3803—see the section Powering the LTC3803. The ITH/RUN pin can be driven below VSHDN (nominally 0.28V) to force the LTC3803 into shutdown. An internal 0.3µA current source always tries to pull this pin towards VCC. When the ITH/RUN pin voltage is allowed to exceed VSHDN, and VCC exceeds VTURNON, the LTC3803 begins to operate and an internal clamp immediately pulls the ITH/RUN pin up to about 0.7V. In operation, the ITH/RUN pin voltage will vary from roughly 0.7V to 1.9V to represent current comparator thresholds from zero to maximum. Internal Soft-Start An internal soft-start feature is enabled whenever the LTC3803 comes out of shutdown. Specifically, the ITH/RUN voltage is clamped and is prevented from reaching maximum until roughly 1.4ms has passed. This allows the input and output currents of LTC3803-based power supplies to rise in a smooth and controlled manner on start-up. Powering the LTC3803 In the simplest case, the LTC3803 can be powered from a high voltage supply through a resistor. A built-in shunt regulator from the VCC pin to GND will draw as much current as needed through this resistor to regulate the VCC voltage to around 9.4V as long as the VCC pin is not forced to sink more than 25mA. This shunt regulator is always active, even when the LTC3803 is in shutdown, since it serves the vital function of protecting the VCC pin from seeing too much voltage. For higher efficiency or for wide VIN range applications, flyback controllers are typically powered through a separate bias winding on the flyback transformer. The LTC3803 has a wide UVLO hysteresis (1.5V min) and small VCC supply current draw (<90µA when VCC < VTURNON) that is needed to support such bootstrapped hysteretic start-up schemes. The VCC pin must be bypassed to ground immediately adjacent to the IC pins with a minimum of a 10µF ceramic or tantalum capacitor. Proper supply bypassing is necessary to supply the high transient currents required by the MOSFET gate driver. Adjustable Slope Compensation The LTC3803 injects a 5µA peak current ramp out through its SENSE pin which can be used for slope compensation in designs that require it. This current ramp is approximately linear and begins at zero current at 6% duty cycle, reaching peak current at 80% duty cycle. Additional details are provided in the Applications Information section. 3803i 7 LTC3803 U W U U APPLICATIO S I FOR ATIO Many LTC3803 application circuits can be derived from the topology shown in Figure 2. The LTC3803 itself imposes no limits on allowed power output, input voltage VIN or desired regulated output voltage VOUT; these are all determined by the ratings on the external power components. The key factors are: Q1’s maximum drain-source voltage (BVDSS), on-resistance (RDS(ON)) and maximum drain current, T1’s saturation flux level and winding insulation breakdown voltages, CIN and COUT’s maximum working voltage, ESR, and maximum ripple current ratings, and D1 and RSENSE’s power ratings. T1 LBIAS D2 R3 • VIN D1 VOUT • RSTART CIN LPRI LSEC COUT • 5 CVCC 1 CC 2 VCC ITH/RUN NGATE LTC3803 GND SENSE 6 4 VFB R1 3 Q1 RSL RSENSE R2 3803 F02 Figure 2. Typical LTC3803 Application Circuit SELECTING FEEDBACK RESISTOR DIVIDER VALUES The regulated output voltage is determined by the resistor divider across VOUT (R1 and R2 in Figure 2). The ratio of R2 to R1 needed to produce a desired VOUT can be calculated: R2 = VOUT – 0.8 V • R1 0.8 V Choose resistance values for R1 and R2 to be as large as possible in order to minimize any efficiency loss due to the static current drawn from VOUT, but just small enough so that when VOUT is in regulation, the error caused by the nonzero input current to the VFB pin is less than 1%. A good rule of thumb is to choose R1 to be 80k or less. TRANSFORMER DESIGN CONSIDERATIONS Transformer specification and design is perhaps the most critical part of applying the LTC3803 successfully. In addition to the usual list of caveats dealing with high frequency power transformer design, the following should prove useful. Turns Ratios Due to the use of the external feedback resistor divider ratio to set output voltage, the user has relative freedom in selecting transformer turns ratio to suit a given application. Simple ratios of small integers, e.g., 1:1, 2:1, 3:2, etc. can be employed which yield more freedom in setting total turns and mutual inductance. Simple integer turns ratios also facilitate the use of “off-the-shelf” configurable transformers such as the Coiltronics VERSA-PACTM series in applications with high input to output voltage ratios. For example, if a 6-winding VERSA-PAC is used with three windings in series on the primary and three windings in parallel on the secondary, a 3:1 turns ratio will be achieved. Turns ratio can be chosen on the basis of desired duty cycle. However, remember that the input supply voltage plus the secondary-to-primary referred version of the flyback pulse (including leakage spike) must not exceed the allowed external MOSFET breakdown rating. Leakage Inductance Transformer leakage inductance (on either the primary or secondary) causes a voltage spike to occur after the output switch (Q1) turn-off. This is increasingly prominent at higher load currents, where more stored energy must be dissipated. In some cases a “snubber” circuit will be required to avoid overvoltage breakdown at the MOSFET’s drain node. Application Note 19 is a good reference on snubber design. A bifilar or similar winding technique is a good way to minimize troublesome leakage inductances. However, remember that this will limit the primary-to-secondary breakdown voltage, so bifilar winding is not always practical. VERSA-PAC is a trademark of Coiltronics, Inc. 3803i 8 LTC3803 U W U U APPLICATIO S I FOR ATIO CURRENT SENSE RESISTOR CONSIDERATIONS The external current sense resistor (RSENSE in Figure 2) allows the user to optimize the current limit behavior for the particular application. As the current sense resistor is varied from several ohms down to tens of milliohms, peak switch current goes from a fraction of an ampere to several amperes. Care must be taken to ensure proper circuit operation, especially with small current sense resistor values. For example, a peak switch current of 5A requires a sense resistor of 0.020Ω. Note that the instantaneous peak power in the sense resistor is 0.5W and it must be rated accordingly. The LTC3803 has only a single sense line to this resistor. Therefore, any parasitic resistance in the ground side connection of the sense resistor will increase its apparent value. In the case of a 0.020Ω sense resistor, one milliohm of parasitic resistance will cause a 5% reduction in peak switch current. So the resistance of printed circuit copper traces and vias cannot necessarily be ignored. PROGRAMMABLE SLOPE COMPENSATION The LTC3803 injects a ramping current through its SENSE pin into an external slope compensation resistor (RSL in Figure 2). This current ramp starts at zero right after the NGATE pin has been high for the LTC3803’s minimum duty cycle of 6%. The current rises linearly towards a peak of 5µA at the maximum duty cycle of 80%, shutting off once the NGATE pin goes low. A series resistor (RSL) connecting the SENSE pin to the current sense resistor (RSENSE) thus develops a ramping voltage drop. From the perspective of the SENSE pin, this ramping voltage adds to the voltage across the sense resistor, effectively reducing the current comparator threshold in proportion to duty cycle. This stabilizes the control loop against subharmonic oscillation. The amount of reduction in the current comparator threshold (∆VSENSE) can be calculated using the following equation: ∆VSENSE = Duty Cycle – 6% • 5µA • RSL 74% A good starting value for RSL is 5.9k, which gives a 30mV drop in current comparator threshold at 80% duty cycle. Designs not needing slope compensation may replace RSL with a short circuit. INTERNAL WIDE HYSTERESIS UNDERVOLTAGE LOCKOUT The LTC3803 is designed to implement DC/DC converters operating from input voltages of typically 48V or more. The standard operating topology employs a third transformer winding (LBIAS in Figure 2) on the primary side that provides power for the LTC3803 via its VCC pin. However, this arrangement is not inherently self-starting. Start-up is affected by the use of an external “trickle-charge” resistor (RSTART in Figure 2) and the presence of an internal wide hysteresis undervoltage lockout circuit that monitors VCC pin voltage. Operation is as follows: “Trickle charge” resistor RSTART is connected to VIN and supplies a small current, typically on the order of 100µA, to charge CVCC. After some time, the voltage on CVCC reaches the VCC turn-on threshold. The LTC3803 then turns on abruptly and draws its normal supply current. The NGATE pin begins switching and the external MOSFET (Q1) begins to deliver power. The voltage on CVCC begins to decline as the LTC3803 draws its normal supply current, which exceeds that delivered by RSTART. After some time, typically tens of milliseconds, the output voltage approaches its desired value. By this time, the third transformer winding is providing virtually all the supply current required by the LTC3803. One potential design pitfall is undersizing the value of capacitor CVCC. In this case, the normal supply current drawn by the LTC3803 will discharge CVCC too rapidly; before the third winding drive becomes effective, the VCC turn-off threshold will be reached. The LTC3803 turns off, and the VCC node begins to charge via RSTART back up to the VCC turn-on threshold. Depending on the particular situation, this may result in either several on-off cycles before proper operation is reached or permanent relaxation oscillation at the VCC node. Note: LTC3803 enforces 6% < Duty Cycle < 80%. 3803i 9 LTC3803 U W U U APPLICATIO S I FOR ATIO Component selection is as follows: Resistor RSTART should be made small enough to yield a worst-case minimum charging current greater than the maximum rated LTC3803 start-up current, to ensure there is enough current to charge CVCC to the VCC turn-on threshold. It should be made large enough to yield a worst-case maximum charging current less than the minimum rated LTC3803 supply current, so that in operation, most of the LTC3803’s supply current is delivered through the third winding. This results in the highest possible efficiency. Capacitor CVCC should then be made large enough to avoid the relaxation oscillation behavior described above. This is complicated to determine theoretically as it depends on the particulars of the secondary circuit and load behavior. Empirical testing is recommended. The third transformer winding should be designed so that its output voltage, after accounting for the D2’s forward voltage drop, exceeds the maximum VCC turn-off threshold. Also, the third winding’s nominal output voltage should be at least 0.5V below the minimum rated VCC clamp voltage to avoid running up against the LTC3803’s VCC shunt regulator, needlessly wasting power. VCC SHUNT REGULATOR In applications including a third transformer winding, the internal VCC shunt regulator serves to protect the LTC3803 from overvoltage transients as the third winding is powering up. In applications where a third transformer winding is undesirable or unavailable, the shunt regulator allows the LTC3803 to be powered through a single dropping resistor from VIN to VCC, in conjunction with a bypass capacitor, CVCC, that closely decouples VCC to GND (see Figure 3). This simplicity comes at the expense of reduced efficiency due to the static power dissipation in the RVCC dropping resistor. The shunt regulator can draw up to 25mA through the VCC pin to GND to drop enough voltage across RVCC to regulate VCC to around 9.5V. For applications where VIN is low enough such that the static power dissipation in RVCC is acceptable, using the VCC shunt regulator is the simplest way to power the LTC3803. EXTERNAL PREREGULATOR The circuit in Figure 4 shows a third way to power the LTC3803. An external series preregulator consisting of series pass transistor Q1, Zener diode D1, and bias resistor RB brings VCC to at least 7.6V nominal, well above the maximum rated VCC turn-off threshold of 6.8V. Resistor RSTART momentarily charges the VCC node up to the VCC turn-on threshold, enabling the LTC3803. VIN VIN RVCC LTC3803 RB Q1 RSTART VCC VCC CVCC LTC3803 GND D1 8.2V 3803 F03 Figure 3. Powering the LTC3803 Via the Internal Shunt Regulator CVCC GND 3803 F04 Figure 4. Powering the LTC3803 with an External Preregulator 3803i 10 LTC3803 U PACKAGE DESCRIPTIO S6 Package 6-Lead Plastic TSOT-23 (Reference LTC DWG # 05-08-1636) 0.62 MAX 2.90 BSC (NOTE 4) 0.95 REF 1.22 REF 3.85 MAX 2.62 REF 1.4 MIN 2.80 BSC 1.50 – 1.75 (NOTE 4) PIN ONE ID RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.30 – 0.45 6 PLCS (NOTE 3) 0.95 BSC 0.80 – 0.90 0.20 BSC 0.01 – 0.10 1.00 MAX DATUM ‘A’ 0.30 – 0.50 REF 0.09 – 0.20 (NOTE 3) 1.90 BSC S6 TSOT-23 0302 NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. JEDEC PACKAGE REFERENCE IS MO-193 3803i 11 LTC3803 U TYPICAL APPLICATIO 2W Housekeeping Telecom Converter BAS516 PRIMARY SIDE 10V, 100mA OUTPUT T1 • 2.2µF 1µF VIN 36V TO 75V • 22k 806Ω 2.2µF BAS516 9.2k 1nF BAS516 1k 1 LTC3803 6 ITH/RUN NGATE 2 5 3 VCC GND VFB SENSE 220k • SECONDARY SIDE 10V, 100mA OUTPUT SECONDARY SIDE GROUND FDC2512 T1: PULSE ENGINEERING PA0648 OR TYCO TT18698 5.6k 4 1µF PRIMARY GROUND 0.1Ω 3803 TA03 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT 1425 Isolated Flyback Switching Regulator with No External Power Devices No Optoisolator or “Third Winding” Required, Up to 6W Output LT1725 General Purpose Isolated Flyback Controller No Optoisolator Required, VIN and VOUT Limited Only by External Power Components LTC1772 SOT-23 Constant Frequency Current Mode Step-Down DC/DC Controller 550kHz Switching Frequency, 2.4V to 9.8V VIN Range LTC1871 Wide Input Range, No RSENSETM Current Mode Flyback, Boost and SEPIC Controller Adjustable Switching Frequency, Programmable Undervoltage Lockout, Optional Burst Mode® Operation at Light Load LTC1872 SOT-23 Constant Frequency Current Mode Boost DC/DC Controller 550kHz Switching Frequency, 2.4V to 9.8V VIN Range LT1950 Current Mode PWM Controller Controller for Forward Converters from 30W to 300W LT3420 Photoflash Capacitor Charger with Automatic Refresh Specialized Flyback Charges High Voltage Photoflash Capacitors Quickly and Efficiently ® Burst Mode is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. 3803i 12 Linear Technology Corporation LT/TP 0803 1K • PRINTED IN THE USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2003