LT1933 600mA, 500kHz Step-Down Switching Regulator in SOT-23 U FEATURES DESCRIPTIO ■ The LT ®1933 is a current mode PWM step-down DC/DC converter with an internal 0.75A power switch, packaged in a tiny 6-lead SOT-23. The wide input range of 3.6V to 36V makes the LT1933 suitable for regulating power from a wide variety of sources, including unregulated wall transformers, 24V industrial supplies and automotive batteries. Its high operating frequency allows the use of tiny, low cost inductors and ceramic capacitors, resulting in low, predictable output ripple. ■ ■ ■ ■ ■ ■ ■ ■ ■ Wide Input Range: 3.6V to 36V 5V at 600mA from 16V to 36V Input 3.3V at 600mA from 12V to 36V Input 5V at 500mA from 6.3V to 36V Input 3.3V at 500mA from 4.5V to 36V Input Fixed Frequency 500kHz Operation Uses Tiny Capacitors and Inductors Soft-Start Internally Compensated Low Shutdown Current: <2µA Output Adjustable Down to 1.25V Low Profile (1mm) SOT-23 (ThinSOT™) Package U APPLICATIO S Cycle-by-cycle current limit provides protection against shorted outputs, and soft-start eliminates input current surge during start up. The low current (<2µA) shutdown provides output disconnect, enabling easy power management in battery-powered systems. ■ , LTC and LT are registered trademarks of Linear Technology Corporation. ThinSOT is a trademark of Linear Technology Corporation. ■ ■ ■ ■ ■ Automotive Battery Regulation Industrial Control Supplies Wall Transformer Regulation Distributed Supply Regulation Battery-Powered Equipment U ■ TYPICAL APPLICATIO 3.3V Step-Down Converter Efficiency 95 1N4148 VIN LT1933 OFF ON SHDN GND 0.1µF 90 SW VOUT 3.3V/500mA FB MBRM140 10k VOUT = 5V 22µH 16.5k 2.2µF VIN = 12V BOOST 22µF 1933 TA01a EFFICIENCY (%) VIN 4.5V TO 36V 85 VOUT = 3.3V 80 75 70 65 0 100 200 300 400 LOAD CURRENT (mA) 500 600 1933 TA01b 1933f 1 LT1933 W W W AXI U U ABSOLUTE RATI GS U U W PACKAGE/ORDER I FOR ATIO (Note 1) Input Voltage (VIN) ....................................– 0.4V to 36V BOOST Pin Voltage .................................................. 43V BOOST Pin Above SW Pin ....................................... 20V SHDN Pin ..................................................– 0.4V to 36V FB Voltage ...................................................– 0.4V to 6V Operating Temperature Range (Note 2) LT1933E ................................................. – 40°C to 85°C LT1933I ................................................ – 40°C to 125°C Maximum Junction Temperature .......................... 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW BOOST 1 6 SW GND 2 5 VIN FB 3 LT1933ES6 LT1933IS6 4 SHDN S6 PART MARKING S6 PACKAGE 6-LEAD PLASTIC TSOT-23 LTAGN LTAGP TJMAX = 125°C, θJA = 165°C/ W, θJC = 102°C/ W Consult factory for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VBOOST = 17V, unless otherwise noted. (Note 2) PARAMETER CONDITIONS MIN Undervoltage Lockout ● Feedback Voltage 1.225 ● TYP MAX UNITS 3.35 3.6 1.245 1.265 V 40 120 nA V FB Pin Bias Current VFB = Measured VREF + 10mV (Note 4) Quiescent Current Not Switching 1.6 2.5 mA Quiescent Current in Shutdown VSHDN = 0V 0.01 2 µA Reference Line Regulation VIN = 5V to 36V 0.01 Switching Frequency VFB = 1.1V 400 VFB = 0V ● Maximum Duty Cycle Switch Current Limit (Note 3) Switch VCESAT ISW = 400mA 500 %/V 600 kHz 55 kHz 88 94 % 0.75 1.05 370 Switch Leakage Current A 500 mV 2 µA Minimum Boost Voltage Above Switch ISW = 400mA 1.9 2.3 V BOOST Pin Current ISW = 400mA 18 25 mA SHDN Input Voltage High 2.3 V SHDN Input Voltage Low SHDN Bias Current VSHDN = 2.3V (Note 5) VSHDN = 0V Note 1: Absolute Maximum Ratings are those values beyond which the life of the device may be impaired. Note 2: The LT1933E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation 34 0.01 0.3 V 50 0.1 µA µA with statistical process controls. The LT1933I specifications are guaranteed over the –40°C to 125°C temperature range. Note 3: Current limit guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycle. Note 4: Current flows out of pin. Note 5: Current flows into pin. 1933f 2 LT1933 U W TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C VOUT = 5V 90 VIN = 24V 80 70 Switch Current Limit 1200 TA = 25°C VOUT = 3.3V 90 VIN = 12V EFFICIENCY (%) EFFICIENCY (%) Efficiency, VOUT = 3.3V 100 VIN = 5V VIN = 12V 80 VIN = 24V 70 D1 = MBRM140 L1 = Toko D53LCB 33µH 60 0 100 200 300 400 LOAD CURRENT (mA) 500 0 100 200 300 400 LOAD CURRENT (mA) 500 1933 G01 800 600 400 200 0 600 MINIMUM 0 20 TA = 25°C VOUT = 5V Switch Voltage Drop TA = 25°C VOUT = 3.3V L = 22µH 500 700 SWITCH VOLTAGE (mV) LOAD CURRENT (mA) LOAD CURRENT (mA) 600 100 600 500 L = 33µH 80 1933 G03 Maximum Load Current 800 700 60 40 DUTY CYCLE (%) 1933 G02 Maximum Load Current 800 1000 D1 = MBRM140 L1 = Toko D53LCB 22µH 60 600 TA = 25°C TYPICAL SWITCH CURRENT LIMIT (mA) Efficiency, VOUT = 5V 100 L = 22µH 600 L = 15µH 500 TA = 25°C 400 TA = 85°C 300 TA = –40°C 200 100 400 0 5 15 20 10 INPUT VOLTAGE (V) 25 400 30 0 5 15 20 10 INPUT VOLTAGE (V) 1933 G04 0 25 30 SWITCHING FREQUENCY (kHz) UVLO (V) FEEDBACK VOLTAGE (V) 3.6 1.250 3.4 1.240 3.2 1.235 125 1933 G07 3.0 –50 –25 0.6 600 1.255 100 0.5 Switching Frequency 3.8 0 25 50 75 TEMPERATURE (°C) 0.2 0.4 0.3 SWITCH CURRENT (A) 1933 G06 Undervoltage Lockout 1.260 1.230 –50 –25 0.1 1933 G05 Feedback Voltage 1.245 0 50 75 0 25 TEMPERATURE (°C) 100 125 1933 G08 550 500 450 400 –50 –25 50 75 0 25 TEMPERATURE (°C) 100 125 1933 G09 1933f 3 LT1933 U W TYPICAL PERFOR A CE CHARACTERISTICS Frequency Foldback Soft-Start 1.4 TA = 25°C TA = 25°C DC = 30% 1.2 500 400 300 200 100 SHDN Pin Current 200 SHDN PIN CURRENT (µA) 600 SWITCH CURRENT LIMIT (A) 1.0 0.8 0.6 0.4 0 0 0.0 0.5 1.0 FB PIN VOLTAGE (V) 150 100 50 0 0 1.5 1 2 3 SHDN PIN VOLTAGE (V) 1933 G10 4 VOUT = 3.3V TA = 25°C L = 22µH 5.5 7 INPUT VOLTAGE (V) TO START 6 TO RUN 5.0 4.5 4.0 5 TO RUN 3.5 4 10 100 LOAD CURRENT (mA) 1 16 Switch Current Limit 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 –50 –25 3.0 1 4 8 12 SHDN PIN VOLTAGE (V) 1933 G12 Typical Minimum Input Voltage 6.0 VOUT = 5V TA = 25°C L = 33µH TO START 0 1933 G11 Typical Minimum Input Voltage 8 INPUT VOLTAGE (V) TA = 25°C 0.2 SWITCH CURRENT LIMIT (A) SWITCHING FREQUENCY (kHz) 700 10 100 LOAD CURRENT (mA) 1933 G13 0 25 50 75 TEMPERATURE (°C) 1933 G14 100 125 1933 G15 Operating Waveforms, Discontinuous Mode Operating Waveforms VSW 10V/DIV VSW 10V/DIV IL 200mA/DIV IL 200mA/DIV VOUT 10mV/DIV VOUT 10mV/DIV VIN = 12V, VOUT = 3.3V, IOUT = 400mA, L = 22µH, COUT = 22µF 1933 G16 VIN = 12V, VOUT = 3.3V, IOUT = 20mA, L = 22µH, COUT = 22µF 1933 G16 1933f 4 LT1933 U U U PI FU CTIO S BOOST (Pin 1): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. SHDN (Pin 4): The SHDN pin is used to put the LT1933 in shutdown mode. Tie to ground to shut down the LT1933. Tie to 2.3V or more for normal operation. If the shutdown feature is not used, tie this pin to the VIN pin. SHDN also provides a soft-start function; see the Applications Information section. GND (Pin 2): Tie the GND pin to a local ground plane below the LT1933 and the circuit components. Return the feedback divider to this pin. VIN (Pin 5): The VIN pin supplies current to the LT1933’s internal regulator and to the internal power switch. This pin must be locally bypassed. FB (Pin 3): The LT1933 regulates its feedback pin to 1.245V. Connect the feedback resistor divider tap to this pin. Set the output voltage according to VOUT = 1.245V (1 + R1/R2). A good value for R2 is 10k. SW (Pin 6): The SW pin is the output of the internal power switch. Connect this pin to the inductor, catch diode and boost capacitor. W BLOCK DIAGRA 5 VIN VIN C2 INT REG AND UVLO ON OFF SLOPE COMP R3 4 BOOST Σ R Q S Q D2 1 SHDN C3 DRIVER C4 Q1 SW OSC L1 VOUT 6 D1 FREQUENCY FOLDBACK VC C1 gm 1.245V 2 GND 3 R2 FB R1 1933 BD 1933f 5 LT1933 U OPERATIO (Refer to Block Diagram) The LT1933 is a constant frequency, current mode step down regulator. A 500kHz oscillator enables an RS flipflop, turning on the internal 750mA power switch Q1. An amplifier and comparator monitor the current flowing between the VIN and SW pins, turning the switch off when this current reaches a level determined by the voltage at VC. An error amplifier measures the output voltage through an external resistor divider tied to the FB pin and servos the VC node. If the error amplifier’s output increases, more current is delivered to the output; if it decreases, less current is delivered. An active clamp (not shown) on the VC node provides current limit. The VC node is also clamped to the voltage on the SHDN pin; soft-start is implemented by generating a voltage ramp at the SHDN pin using an external resistor and capacitor. An internal regulator provides power to the control circuitry. This regulator includes an undervoltage lockout to prevent switching when VIN is less than ~3.35V. The SHDN pin is used to place the LT1933 in shutdown, disconnecting the output and reducing the input current to less than 2µA. The switch driver operates from either the input or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the internal bipolar NPN power switch for efficient operation. The oscillator reduces the LT1933’s operating frequency when the voltage at the FB pin is low. This frequency foldback helps to control the output current during startup and overload. U W U U APPLICATIO S I FOR ATIO FB Resistor Network The output voltage is programmed with a resistor divider between the output and the FB pin. Choose the 1% resistors according to: R1 = R2(VOUT/1.245 – 1) R2 should be 20k or less to avoid bias current errors. Reference designators refer to the Block Diagram. Input Voltage Range voltage of: VIN(MIN) = (VOUT + VD)/DCMAX – VD + VSW with DCMAX = 0.88 The maximum input voltage is determined by the absolute maximum ratings of the VIN and BOOST pins and by the minimum duty cycle DCMIN = 0.08 (corresponding to a minimum on time of 130ns): VIN(MAX) = (VOUT + VD)/DCMIN – VD + VSW The input voltage range for LT1933 applications depends on the output voltage and on the absolute maximum ratings of the VIN and BOOST pins. Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the absolute maximum ratings of the VIN and BOOST pins. The minimum input voltage is determined by either the LT1933’s minimum operating voltage of ~3.35V, or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages: Inductor Selection and Maximum Output Current DC = (VOUT + VD)/(VIN – VSW + VD) where VD is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.4V at maximum load). This leads to a minimum input A good first choice for the inductor value is: L = 5 (VOUT + VD) where VD is the voltage drop of the catch diode (~0.4V) and L is in µH. With this value the maximum load current will be above 500mA. The inductor’s RMS current rating must be greater than your maximum load current and its saturation current should be about 30% higher. For robust operation in fault conditions the saturation current should 1933f 6 LT1933 U W U U APPLICATIO S I FOR ATIO be ~1A. To keep efficiency high, the series resistance (DCR) should be less than 0.2Ω. Table 1 lists several vendors and types that are suitable. for 0.5A forward current and a maximum reverse voltage of 40V. The MBRM140 provides better efficiency, and will handle extended overload conditions. Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger value provides a slightly higher maximum load current, and will reduce the output voltage ripple. If your load is lower than 500mA, then you can decrease the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. There are several graphs in the Typical Performance Characteristics section of this data sheet that show the maximum load current as a function of input voltage and inductor value for several popular output voltages. Low inductance may result in discontinuous mode operation, which is OK, but further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN < 0.5), there is a minimum inductance required to avoid subharmonic oscillations. Choosing L greater than 3(VOUT + VD) µH prevents subharmonic oscillations at all duty cycles. Input Capacitor Catch Diode A 0.5A or 1A Schottky diode is recommended for the catch diode, D1. The diode must have a reverse voltage rating equal to or greater than the maximum input voltage. The ON Semiconductor MBR0540 is a good choice; it is rated Bypass the input of the LT1933 circuit with a 2.2µF or higher value ceramic capacitor of X7R or X5R type. Y5V types have poor performance over temperature and applied voltage, and should not be used. A 2.2µF ceramic is adequate to bypass the LT1933 and will easily handle the ripple current. However, if the input power source has high impedance, or there is significant inductance due to long wires or cables, additional bulk capacitance may be necessary. This can be provided with a low performance electrolytic capacitor. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple at the LT1933 and to force this very high frequency switching current into a tight local loop, minimizing EMI. A 2.2µF capacitor is capable of this task, but only if it is placed close to the LT1933 and the catch diode; see the PCB Layout section. A second precaution regarding the ceramic input capacitor concerns the maximum input voltage rating of the LT1933. A ceramic input capacitor combined with trace or cable inductance forms a high quality (under damped) tank circuit. If the LT1933 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding the LT1933’s Table 1. Inductor Vendors Vendor URL Part Series Inductance Range (µH) Size (mm) Coilcraft www.coi1craft.com DO1608C 10 to 22 2.9 × 4.5 × 6.6 MSS5131 10 to 22 3.1 × 5.1 × 5.1 MSS6122 10 to 33 2.2 × 6.1 × 6.1 CR43 10 to 22 3.5 × 4.3 × 4.8 CDRH4D28 10 to 33 3.0 × 5.0 × 5.0 CDRH5D28 22 to 47 3.0 × 5.7 × 5.7 D52LC 10 to 22 2.0 × 5.0 × 5.0 D53LC 22 to 47 3.0 × 5.0 × 5.0 WE-TPC MH 10 to 22 2.8 × 4.8 × 4.8 WE-PD4 S 10 to 22 2.9 × 4.5 × 6.6 WE-PD2 S 10 to 47 3.2 × 4.0 × 4.5 Sumida www.sumida.com Toko www.toko.com Würth Elektronik www.we-online.com 1933f 7 LT1933 U W U U APPLICATIO S I FOR ATIO voltage rating. This situation is easily avoided; see the Hot Plugging Safely section. may be required to get the full benefit (see the Compensation section). Output Capacitor High performance electrolytic capacitors can be used for the output capacitor. Low ESR is important, so choose one that is intended for use in switching regulators. The ESR should be specified by the supplier, and should be 0.1Ω or less. Such a capacitor will be larger than a ceramic capacitor and will have a larger capacitance, because the capacitor must be large to achieve low ESR. Table 2 lists several capacitor vendors. The output capacitor has two essential functions. Along with the inductor, it filters the square wave generated by the LT1933 to produce the DC output. In this role it determines the output ripple, and low impedance at the switching frequency is important. The second function is to store energy in order to satisfy transient loads and stabilize the LT1933’s control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good value is COUT = 60/VOUT where COUT is in µF. Use X5R or X7R types, and keep in mind that a ceramic capacitor biased with VOUT will have less than its nominal capacitance. This choice will provide low output ripple and good transient response. Transient performance can be improved with a high value capacitor, but a phase lead capacitor across the feedback resistor R1 Figure 1 shows the transient response of the LT1933 with several output capacitor choices. The output is 3.3V. The load current is stepped from 100mA to 400mA and back to 100mA, and the oscilloscope traces show the output voltage. The upper photo shows the recommended value. The second photo shows the improved response (less voltage drop) resulting from a larger output capacitor and a phase lead capacitor. The last photo shows the response to a high performance electrolytic capacitor. Transient performance is improved due to the large output capacitance, but output ripple (as shown by the broad trace) has increased because of the higher ESR of this capacitor. Table 2. Capacitor Vendors Vendor Phone URL Part Series Comments Panasonic (714) 373-7366 www.panasonic.com Ceramic, Polymer, Tantalum EEF Series Kemet Sanyo Murata (864) 963-6300 (408) 749-9714 (404) 436-1300 AVX Taiyo Yuden (864) 963-6300 www.kemet.com www.sanyovideo.com Ceramic, Tantalum Ceramic, Polymer, Tantalum www.murata.com Ceramic www.avxcorp.com Ceramic, Tantalum www.taiyo-yuden.com T494, T495 POSCAP TPS Series Ceramic 1933f 8 LT1933 U U W U APPLICATIO S I FOR ATIO VOUT VOUT 50mV/DIV 16.5k FB 22µF 10k IOUT 200mA/DIV 1933 F01a VOUT VOUT 50mV/DIV 470pF 16.5k FB 22µF 2x 10k IOUT 200mA/DIV 1933 F01b VOUT VOUT 50mV/DIV 16.5k + FB 10k 100µF SANYO 4TPB100M IOUT 200mA/DIV 1933 F01c Figure 1. Transient Load Response of the LT1933 with Different Output Capacitors as the Load Current is Stepped from 100mA to 400mA. VIN = 12V, VOUT = 3.3V, L = 22µH. 1933f 9 LT1933 U W U U APPLICATIO S I FOR ATIO Capacitor C3 and diode D2 are used to generate a boost voltage that is higher than the input voltage. In most cases a 0.1µF capacitor and fast switching diode (such as the 1N4148 or 1N914) will work well. Figure 2 shows two ways to arrange the boost circuit. The BOOST pin must be at least 2.3V above the SW pin for best efficiency. For outputs of 3V and above, the standard circuit (Figure 2a) is best. For outputs between 2.5V and 3V, use a 0.47µF capacitor and a small Schottky diode (such as the BAT54). For lower output voltages the boost diode can be tied to the input (Figure 2b). The circuit in Figure 2a is more efficient because the BOOST pin current comes from a lower voltage source. You must also be sure that the maximum voltage rating of the BOOST pin is not exceeded. The minimum operating voltage of an LT1933 application is limited by the undervoltage lockout (~3.35V) and by the maximum duty cycle as outlined above. For proper startup, the minimum input voltage is also limited by the boost circuit. If the input voltage is ramped slowly, or the LT1933 is turned on with its SHDN pin when the output is already in regulation, then the boost capacitor may not be fully charged. Because the boost capacitor is charged with the energy stored in the inductor, the circuit will rely on some minimum load current to get the boost circuit running properly. This minimum load will depend on input and output voltages, and on the arrangement of the boost circuit. The minimum load generally goes to zero once the circuit has started. Figure 3 shows a plot of minimum load to start and to run as a function of input voltage. In many cases the discharged output capacitor will present a load to the switcher which will allow it to start. The plots show the worst-case situation where VIN is ramping very slowly. For lower start-up voltage, the boost diode can be tied to VIN; however, this restricts the input range to one-half of the absolute maximum rating of the BOOST pin. Minimum Input Voltage VOUT = 3.3V 6.0 VOUT = 3.3V TA = 25°C L = 22µH 5.5 TO START INPUT VOLTAGE (V) BOOST Pin Considerations 5.0 4.5 4.0 TO RUN 3.5 3.0 D2 1 1933 F03a C3 BOOST LT1933 VIN VIN 10 100 LOAD CURRENT (mA) Minimum Input Voltage VOUT = 5V VOUT SW 8 VOUT = 5V TA = 25°C L = 33µH GND 1933 F02a 7 INPUT VOLTAGE (V) VBOOST – VSW ≅ VOUT MAX VBOOST ≅ VIN + VOUT (2a) D2 C3 BOOST LT1933 VIN VIN TO START 6 TO RUN 5 SW VOUT GND 4 1933 F02b VBOOST – VSW ≅ VIN MAX VBOOST ≅ 2VIN 1 10 100 LOAD CURRENT (mA) 1933 F03b (2b) Figure 2. Two Circuits for Generating the Boost Voltage Figure 3. The Minimum Input Voltage Depends on Output Voltage, Load Current and Boost Circuit 1933f 10 LT1933 U U W U APPLICATIO S I FOR ATIO At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This reduces the minimum input voltage to approximately 300mV above VOUT. At higher load currents, the inductor current is continuous and the duty cycle is limited by the maximum duty cycle of the LT1933, requiring a higher input voltage to maintain regulation. Soft-Start The SHDN pin can be used to soft-start the LT1933, reducing the maximum input current during start up. The SHDN pin is driven through an external RC filter to create a voltage ramp at this pin. Figure 4 shows the start up waveforms with and without the soft-start circuit. By choosing a large RC time constant, the peak start up current can be reduced to the current that is required to regulate the output, with no overshoot. Choose the value of the resistor so that it can supply 60µA when the SHDN pin reaches 2.3V. Shorted and Reversed Input Protection If the inductor is chosen so that it won’t saturate excessively, an LT1933 buck regulator will tolerate a shorted output. There is another situation to consider in systems where the output will be held high when the input to the LT1933 is absent. This may occur in battery charging applications or in battery backup systems where a battery or some other supply is diode OR-ed with the LT1933’s output. If the VIN pin is allowed to float and the SHDN pin is held high (either by a logic signal or because it is tied to VIN), then the LT1933’s internal circuitry will pull its quiescent current through its SW pin. This is fine if your system can tolerate a few mA in this state. If you ground the SHDN pin, the SW pin current will drop to essentially zero. However, if the VIN pin is grounded while the output is held high, then parasitic diodes inside the LT1933 can RUN 5V/DIV RUN SHDN GND 1933 F04a IIN 100mA/DIV VOUT 5V/DIV 50µs/DIV RUN 5V/DIV RUN 15k SHDN 0.1µF GND 1933 F04b IIN 100mA/DIV VOUT 5V/DIV 0.5ms/DIV Figure 4. To Soft-Start the LT1933, Add a Resistor and Capacitor to the SHDN Pin. VIN = 12V, VOUT = 3.3V, COUT = 22µF, RLOAD = 10Ω. 1933f 11 LT1933 U W U U APPLICATIO S I FOR ATIO pull large currents from the output through the SW pin and the VIN pin. Figure 5 shows a circuit that will run only when the input voltage is present and that protects against a shorted or reversed input. D4 5 VIN VIN 1 BOOST LT1933 4 SHDN GND 2 6 SW VOUT FB 3 BACKUP D4: MBR0540 1933 F05 Figure 5. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied to the Output; It Also Protects the Circuit from a Reversed Input. The LT1933 Runs Only When the Input is Present Hot Plugging Safely The small size, robustness and low impedance of ceramic capacitors make them an attractive option for the input bypass capacitor of LT1933 circuits. However, these capacitors can cause problems if the LT1933 is plugged into a live supply (see Linear Technology Application Note 88 for a complete discussion). The low loss ceramic capacitor combined with stray inductance in series with the power source forms an under damped tank circuit, and the voltage at the VIN pin of the LT1933 can ring to twice the nominal input voltage, possibly exceeding the LT1933’s rating and damaging the part. If the input supply is poorly controlled or the user will be plugging the LT1933 into an energized supply, the input network should be designed to prevent this overshoot. Figure 6 shows the waveforms that result when an LT1933 circuit is connected to a 24V supply through six feet of 24gauge twisted pair. The first plot is the response with a CLOSING SWITCH SIMULATES HOT PLUG IIN VIN DANGER! LT1933 + VIN 20V/DIV 2.2µF LOW IMPEDANCE ENERGIZED 24V SUPPLY IIN 5A/DIV STRAY INDUCTANCE DUE TO 6 FEET (2 METERS) OF TWISTED PAIR 20µs/DIV (6a) LT1933 + 10µF 35V AI.EI. RINGING VIN MAY EXCEED ABSOLUTE MAXIMUM RATING OF THE LT1933 + VIN 20V/DIV 2.2µF IIN 5A/DIV (6b) 20µs/DIV 1Ω LT1933 + 0.1µF VIN 20V/DIV 2.2µF IIN 5A/DIV (6c) 20µs/DIV 1933 F06 Figure 6. A Well Chosen Input Network Prevents Input Voltage Overshoot and Ensures Reliable Operation When the LT1933 is Connected to a Live Supply 1933f 12 LT1933 U U W U APPLICATIO S I FOR ATIO 2.2µF ceramic capacitor at the input. The input voltage rings as high as 35V and the input current peaks at 20A. One method of damping the tank circuit is to add another capacitor with a series resistor to the circuit. In Figure 6b an aluminum electrolytic capacitor has been added. This capacitor’s high equivalent series resistance damps the circuit and eliminates the voltage overshoot. The extra capacitor improves low frequency ripple filtering and can slightly improve the efficiency of the circuit, though it is likely to be the largest component in the circuit. An alternative solution is shown in Figure 6c. A 1Ω resistor is added in series with the input to eliminate the voltage overshoot (it also reduces the peak input current). A 0.1µF capacitor improves high frequency filtering. This solution is smaller and less expensive than the electrolytic capacitor. For high input voltages its impact on efficiency is minor, reducing efficiency less than one half percent for a 5V output at full load operating from 24V. Frequency Compensation The LT1933 uses current mode control to regulate the output. This simplifies loop compensation. In particular, the LT1933 does not require the ESR of the output capacitor for stability allowing the use of ceramic capacitors to achieve low output ripple and small circuit size. Figure 7 shows an equivalent circuit for the LT1933 control loop. The error amp is a transconductance amplifier with finite output impedance. The power section, consisting of the modulator, power switch and inductor, is modeled as a transconductance amplifier generating an output current proportional to the voltage at the VC node. Note that the output capacitor integrates this current, and LT1933 – 0.7V + CURRENT MODE POWER STAGE SW gm GND ERROR AMPLIFIER 500k PCB Layout For proper operation and minimum EMI, care must be taken during printed circuit board layout. Figure 8 shows the recommended component placement with trace, ground plane and via locations. Note that large, switched currents flow in the LT1933’s VIN and SW pins, the catch diode (D1) and the input capacitor (C2). The loop formed by these components should be as small as possible and tied to system ground in only one place. These components, along with the inductor and output capacitor, CPL SHUTDOWN FB gm = 150µmhos + CC 80pF – VC RC 100k If the output capacitor is different than the recommended capacitor, stability should be checked across all operating conditions, including load current, input voltage and temperature. The LT1375 data sheet contains a more thorough discussion of loop compensation and describes how to test the stability using a transient load. OUT R1 1.1mho that the capacitor on the VC node (CC) integrates the error amplifier output current, resulting in two poles in the loop. RC provides a zero. With the recommended output capacitor, the loop crossover occurs above the RCCC zero. This simple model works well as long as the value of the inductor is not too high and the loop crossover frequency is much lower than the switching frequency. With a larger ceramic capacitor (very low ESR), crossover may be lower and a phase lead capacitor (CPL) across the feedback divider may improve the phase margin and transient response. Large electrolytic capacitors may have an ESR large enough to create an additional zero, and the phase lead may not be necessary. VIN ESR 1.245V + C1 VOUT C1 C2 C1 SYSTEM GROUND D1 R2 VIAS TO LOCAL GROUND PLANE OUTLINE OF LOCAL GROUND PLANE 1933 F08 1933 F07 Figure 7. Model for Loop Response Figure 8. A Good PCB Layout Ensures Proper, Low EMI Operation 1933f 13 LT1933 should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components, and tie this ground plane to system ground at one location, ideally at the ground terminal of the output capacitor C1. The SW and BOOST nodes should be as small as possible. Finally, keep the FB node small so that the ground pin and ground traces will shield it from the SW and BOOST nodes. Include two vias near the GND pin of the LT1933 to help remove heat from the LT1933 to the ground plane. High Temperature Considerations The die temperature of the LT1933 must be lower than the maximum rating of 125°C. This is generally not a concern unless the ambient temperature is above 85°C. For higher temperatures, care should be taken in the layout of the circuit to ensure good heat sinking of the LT1933. The maximum load current should be derated as the ambient temperature approaches 125°C. The die temperature is calculated by multiplying the LT1933 power dissipation by the thermal resistance from junction to ambient. Power dissipation within the LT1933 can be estimated by calculating the total power loss from an efficiency measurement and subtracting the catch diode loss. The resulting temperature rise at full load is nearly independent of input voltage. Thermal resistance depends on the layout of the circuit board, but a value of 125°C/W is typical. Die temperature rise was measured on a two-layer, five by five cm circuit board in still air. The LT1933 producing 5V at 500mA showed a temperature rise of 28°C, allowing it to deliver full load to 97°C ambient. Above this temperature the load current should be reduced. For 3.3V at 500mA the temperature rise is 24°C. Other Linear Technology Publications Application notes AN19, AN35 and AN44 contain more detailed descriptions and design information for Buck regulators and other switching regulators. The LT1376 data sheet has a more extensive discussion of output ripple, loop compensation and stability testing. Design Note DN100 shows how to generate a bipolar output supply using a Buck regulator. U TYPICAL APPLICATIO S 1.8V Step-Down Converter 5V Step-Down Converter D2 VIN 6.3V TO 36V 5 VIN BOOST LT1933 VIN 3.6V TO 20V 5 VIN BOOST 1 LT1933 OFF ON 4 SHDN SW GND FB 2 3 C2 2.2µF 6 R1 4.42k OFF ON C3 0.1µF 4 SHDN L1 10µH GND FB 2 VOUT 1.8V/500mA D1 SW 3 C2 2.2µF D2 1 6 R1 30.1k C3 0.1µF L1 33µH VOUT 5V/500mA D1 C1 22µF 6.3V R2 10k C1 22µF 2x R2 10k 1933 TA02c 1933 TA02a 3.3V Step-Down Converter VIN 4.5V TO 36V 5 VIN BOOST 1 LT1933 OFF ON 4 SHDN GND 2 C2 2.2µF SW FB 3 R2 10k 6 R1 16.5k 12V Step-Down Converter D2 C3 0.1µF D1 VIN 14.5V TO 36V L1 22µH 5 VIN BOOST 1 LT1933 VOUT 3.3V/500mA OFF ON 4 SHDN GND 2 C1 22µF 6.3V 1933 TA02b C2 2.2µF SW FB 3 R2 10k 6 R1 86.6k D3, 6V C3 0.1µF D1 D2 L1 47µH VOUT 12V/450mA C1 10µF 1933 TA02d 1933f 14 LT1933 U PACKAGE DESCRIPTION S6 Package 6-Lead Plastic SOT-23 (Reference LTC DWG # 05-08-1634) 0.62 MAX 2.80 – 3.10 (NOTE 4) 0.95 REF 1.22 REF 1.4 MIN 3.85 MAX 2.62 REF 2.60 – 3.00 1.50 – 1.75 (NOTE 4) PIN ONE ID RECOMMENDED SOLDER PAD LAYOUT PER IPC CALCULATOR 0.25 – 0.50 TYP 6 PLCS NOTE 3 0.95 BSC 0.90 – 1.30 0.20 BSC 0.90 – 1.45 DATUM ‘A’ 0.35 – 0.55 REF 0.09 – 0.20 (NOTE 3) NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. DRAWING NOT TO SCALE 3. DIMENSIONS ARE INCLUSIVE OF PLATING 4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR 5. MOLD FLASH SHALL NOT EXCEED 0.254mm 6. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ) 1.90 BSC 0.09 – 0.15 NOTE 3 S6 SOT-23 0502 ATTENTION: ORIGINAL SOT23-6L PACKAGE. MOST SOT23-6L PRODUCTS CONVERTED TO THIN SOT23 PACKAGE, DRAWING # 05-08-1636 AFTER APPROXIMATELY APRIL 2001 SHIP DATE 1933f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1933 U TYPICAL APPLICATIO S 2.5V Step-Down Converter D2 5 VIN 3.6V TO 36V VIN BOOST 1 LT1933 4 OFF ON SHDN GND SW FB 2 C2 2.2µF 3 6 R1 10.5k C3 0.47µF L1 15µH D1 VOUT 2.5V/500mA C1 22µF R2 10k 1933 TA03 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1074/LT1074HV 4.4A IOUT, 100kHz, High Efficiency Step-Down DC/DC Converter VIN: 7.3V to 45V/64V, VOUT(MIN) = 2.21V, IQ = 8.5mA, ISD = 10µA, DD-5/DD-7, TO220-5/ TO220-7 Packages LT1076/LT1076HV 1.6A IOUT, 100kHz, High Efficiency Step-Down DC/DC Converter VIN: 7.3V to 45V/64V, VOUT(MIN) = 2.21V, IQ = 8.5mA, ISD = 10µA, DD-5/DD-7, TO220-5/ TO220-7 Packages LT1676 60V, 440mA IOUT, 100kHz, High Efficiency Step-Down DC/DC Converter VIN: 7.4V to 60V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD = 2.5µA, S8 Package LT1765 25V, 2.75A IOUT, 1.25MHz, High Efficiency Step-Down DC/DC Converter VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 1mA, ISD = 15µA, S8, TSSOP16E Packages LT1766 60V, 1.2A IOUT, 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, ISD = 25µA, TSSOP16/TSSOP16E Packages LT1767 25V, 1.2A IOUT, 1.25MHz, High Efficiency Step-Down DC/DC Converter VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 1mA, ISD = 6µA, MS8/MS8E Packages LT1776 40V, 550mA IOUT, 200kHz, High Efficiency Step-Down DC/DC Converter VIN: 7.4V to 40V, VOUT(MIN) = 1.24V, IQ = 3.2mA, ISD = 30µA, N8, S8 Packages LT1940 25V, Dual 1.4A IOUT, 1.1MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 25V, VOUT(MIN) = 1.25V, IQ = 3.8mA, ISD = <30µA, TSSOP16E Package LT1956 60V, 1.2A IOUT, 500kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, ISD = 25µA, TSSOP16/TSSOP16E Packages LT1976 60V, 1.2A IOUT, 200kHz, High Efficiency Step-Down DC/DC Converter with Burst-Mode® VIN: 3.3V to 60V, VOUT(MIN) = 1.2V, IQ = 100µA, ISD = <1µA, TSSOP16E Package LT3010 80V, 50mA, Low Noise Linear Regulator VIN: 1.5V to 80V, VOUT(MIN) = 1.28V, IQ = 30µA, ISD = <1µA, MS8E Package LT3407 Dual 600mA IOUT, 1.5MHz, Synchronous Step-Down DC/DC Converter VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, ISD = <1µA, MS10E Package LT3412 2.5A IOUT, 4MHz, Synchronous Step-Down DC/DC Converter VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA, ISD = <1µA, TSSOP16E Package LTC3414 4A IOUT, 4MHz, Synchronous Step-Down DC/DC Converter VIN: 2.3V to 5.5V, VOUT(MIN) = 0.8V, IQ = 64µA, ISD = <1µA, TSSOP20E Package LT3430/LT3431 60V, 2.75A IOUT, 200kHz/500kHz, High Efficiency Step-Down DC/DC Converter VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, ISD = 30µA, TSSOP16E Package Burst Mode is a registered trademark of Linear Technology Corporation. 1933f 16 Linear Technology Corporation LT/TP 0704 1K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2004