MAXIM MAX1636EAP

19-1268; Rev 0; 8/97
KIT
ATION
EVALU
E
L
B
A
AVAIL
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
____________________________Features
The MAX1636 is a synchronous, buck, switch-mode,
power-supply controller that generates the CPU supply
voltage in battery-powered systems. It achieves ±1%
output voltage accuracy and offers the excellent loadtransient response needed by upcoming generations of
dynamic-clock CPUs.
Up to 95% efficiency is achieved through synchronous
rectification and Maxim’s proprietary Idle Mode™ control scheme. Efficiency is greater than 80% over a
1000:1 load-current range, extending battery life in system-suspend or standby modes. Excellent dynamic
response corrects output load transients caused by the
latest dynamic-clock CPUs within five 300kHz clock
cycles. Strong, 1A, on-board gate drivers ensure fast,
external N-channel MOSFET switching.
♦ ±1% DC Accuracy (Adjustable Mode)
The MAX1636 features a logic-controlled and synchronizable, fixed-frequency, pulse-width-modulation (PWM)
operating mode that reduces noise and RF interference
in sensitive mobile communications and pen-entry applications. Holding SKIP high forces fixed-frequency mode
for lowest noise under all load conditions.
For a low-cost version that omits the +5V VL linearregulator block and comes in a smaller 16-pin QSOP
package, refer to the MAX1637 data sheet.
♦ 3µA (typ) Shutdown Current
♦ Output Overvoltage Crowbar Protection
♦ Output Undervoltage Shutdown
♦ Adjustable Switching Frequency to 340kHz
♦ Low-Dropout Operation
♦ Idle Mode Pulse-Skipping Operation
♦ 1.10V to 5.5V Adjustable Output Voltage
♦ 2.5V/3.3V Dual-Mode Fixed-Output Settings
♦ Internal Digital Soft-Start
♦ 1.1V ±1% Reference Output
RESET)
♦ Open-Drain Power-Good Output (R
♦ 20-Pin SSOP Package
______________Ordering Information
PART
TEMP. RANGE
PIN-PACKAGE
MAX1636EAP
-40°C to +85°C
20 SSOP
________________________Applications
__________Typical Operating Circuit
Notebook Computers
Subnotebook Computers
VIN
Desktop Computers
Bus-Termination Supplies
V+
OVP
VL
VCC
SHDN
__________________Pin Configuration
TOP VIEW
CSH 1
20 SKIP
CSL 2
19 LX
RESET 3
18 DH
SHDN 4
OVP 5
CC
LX
17 BST
MAX1636
CC 6
DH
BST
MAX1636
16 PGND
15 DL
REF 7
14 VL
SYNC 8
13 V+
GND 9
12 VCC
GND 10
11 FB
SSOP
SKIP
DL
PGND
SYNC
CSH
GND
REF
GND
CSL
FB
RESET
TO µP
Idle Mode is a trademark of Maxim Integrated Products.
________________________________________________________________ Maxim Integrated Products
1
For free samples & the latest literature: http://www.maxim-ic.com, or phone 1-800-998-8800
For small orders, phone 408-737-7600 ext. 3468.
MAX1636
_______________General Description
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
ABSOLUTE MAXIMUM RATINGS
V+ to GND ...............................................................-0.3V to 36V
GND to PGND........................................................................±2V
SHDN to GND. ......................................................... -0.3V to 36V
LX, BST to GND. ...................................................... -0.3V to 36V
DH, BST to LX .............................................................-0.3V to 6V
VL, VCC, CSL, CSH, FB, SKIP to GND ...................... -0.3V to 6V
DL to GND.. ..................................................-0.3V to (VL + 0.3V)
REF, RESET, SYNC, CC, OVP to GND. ..... -0.3V to (VCC + 0.3V)
VL Output Current... ............................................................50mA
VL Short Circuit to GND..............................................Momentary
REF Output Current ............................................................20mA
REF Short Circuit to GND ....... ......................................Indefinite
Continuous Power Dissipation (TA = +70°C)
SSOP (derate 8.00mW/°C above +70°C) .....................640mW
Operating Temperature Range
MAX1636EAP. ..................................................-40°C to +85°C
Storage Temperature Range .............................-65°C to +160°C
Junction Temperature ......................................................+150°C
Lead Temperature (soldering, 10sec) .............................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, V+ = 15V, SYNC = VL = VCC, IVL = 0mA, IREF = 0mA, TA = 0°C to +85°C, unless otherwise noted. Typical values
are at TA = +25°C.)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
SMPS CONTROLLER
Input Voltage Range, V+
Input source for VL regulator
4.5
30
V
Input Voltage Range, VL
Gate-driver supply rail
4.2
5.5
V
Input Voltage Range, VCC
Internal chip supply rail
3.15
5.5
V
Output Voltage, Adj Mode
FB tied to VOUT, 0mV < (CSH - CSL) < 80mV, 4.5V
< V+ < 30V (includes line and load regulation)
1.090
1.100
1.110
V
Output Voltage, Fixed 2.5V Mode
FB tied to VCC, 0mV < (CSH - CSL) < 80mV, 4.5V
< V+ < 30V (includes line and load regulation)
2.486
2.55
2.614
V
Output Voltage, Fixed 3.3V Mode
FB tied to GND, 0mV < (CSH - CSL) < 80mV, 4.5V
< V+ < 30V (includes line and load regulation)
3.282
3.366
3.450
V
VCC = VL = 5V
VREF
5.5
VCC = 3.3V, VL = 5V
VREF
3.6
Output Adjustment Range
Current-Limit Threshold
Power Consumption
Positive direction
80
100
120
Negative direction
-145
-100
-55
VCC = 5V, output not switching
2.0
VCC = 3.3V, output not switching
1.5
Shutdown Supply Current (V+)
SHDN = GND, OVP = GND
FB Input Current
FB forced to REF
Soft-Start Ramp Time
SHDN to full current limit, five levels
Idle Mode Switchover Threshold
CSH - CSL
2
3
-50
10
50
512
20
30
_______________________________________________________________________________________
V
mV
mV
mW
µA
nA
clks
40
mV
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
(Circuit of Figure 1, V+ = 15V, SYNC = VL = VCC, IVL = 0mA, IREF = 0mA, TA = 0°C to +85°C, unless otherwise noted. Typical values
are at TA = +25°C.)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
INTERNAL VL REGULATOR AND REFERENCE
VCC = 5V, I(VL) = 0
60
Regulator Supply Current (V+)
VCC = 5V, I(VL) = 0, V+ = 4.5V
(includes PNP base current)
500
Standby Supply Current (V+)
SHDN = GND, OVP = VCC
60
I(VL) = 0 to 25mA, 5V < V+ < 30V
4.5
5.0
5.3
I(VL) = 0 to 25mA, 6V < V+ < 30V
4.7
5.0
5.3
VL Undervoltage Lockout Threshold
Rising edge, hysteresis = 25mV
3.45
3.60
3.75
VL/ VCC Switchover Threshold
Rising edge, hysteresis = 25mV
REF Output Voltage
No REF load
1.090
1.100
REF Load Regulation
REF Line Regulation
VL Output Voltage
3.15
µA
µA
V
V
V
1.110
V
REF load = 0 to 50µA
10
mV
VCC = 3.3V to 5.5V
3
mV
OSCILLATOR
Oscillator Frequency
Maximum Duty Factor
Maximum Duty Factor, Dropout Mode
SYNC = VCC
270
300
330
SYNC = GND
170
200
230
SYNC = VCC
91
94
SYNC = GND
93
96
98
99
SYNC = GND
%
ns
200
SYNC Input Pulse Width Low
SYNC Input Rise/Fall Time
%
200
SYNC Input Pulse Width High
ns
Guaranteed by design
240
SYNC Input Frequency Range
kHz
200
ns
340
kHz
OVERVOLTAGE PROTECTION
4
7
10
Overvoltage Trip Threshold
FB, with respect to regulation point
Overvoltage Fault Propagation Delay
FB to DL delay, 22mV overdrive, CGATE = 2000pF
1.25
µs
Thermal Shutdown Threshold
Hysteresis = 10°C
150
°C
Catastrophic Output Undervoltage
Lockout Threshold
% of nominal output
Catastrophic Output Undervoltage
Lockout Delay
From shutdown or power-on-reset state
RESET Trip Threshold
Falling edge (hysteresis = 1%)
60
70
80
6144
-6
RESET Delay Time
%
%
clks
-3
32768
%
clks
INPUTS AND OUTPUTS
Logic Input Voltage High
SHDN, SKIP, OVP, SYNC
Logic Input Voltage Low
SHDN, SKIP, OVP, SYNC
Logic Input Bias Current
Pin at GND or VCC; SKIP, OVP, SYNC
SHDN Input Bias Current
SHDN = GND or V+
RESET Output Voltage Low
ISINK = 4mA
RESET Output Leakage Current
+5.5V stress voltage applied
2.4
V
0.8
V
-1
1
µA
-3
3
µA
0.4
V
1
µA
_______________________________________________________________________________________
3
MAX1636
ELECTRICAL CHARACTERISTICS (continued)
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, V+ = 15V, SYNC = VL = VCC, IVL = 0mA, IREF = 0mA, TA = 0°C to +85°C, unless otherwise noted. Typical values
are at TA = +25°C.)
PARAMETER
CONDITIONS
Current-Sense Input
Leakage Current
CSH = CSL = 5V, V+ = VL = VCC =
GND, either CSH or CSL input
Gate-Driver Sink/Source Current
DH or DL forced to 2V
Gate-Driver On-Resistance
High or low, DH or DL
MIN
TYP
MAX
UNITS
10
µA
7
Ω
1
A
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1, V+ = 15V, SYNC = VL = VCC, IVL = 0mA, IREF = 0mA, TA =-40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
SMPS CONTROLLER
Input Voltage Range, V+
Input source for VL regulator
4.5
30
V
Input Voltage Range, VL
Gate-driver supply rail
4.2
5.5
V
Input Voltage Range, VCC
Internal chip supply rail
3.15
5.5
V
Output Voltage, Adj Mode
FB tied to VOUT, 0mV < (CSH - CSL) < 80mV, 4.5V
< V+ < 30V (includes line and load regulation)
1.086
1.114
V
Output Voltage, Fixed 2.5V Mode
FB tied to VCC, 0mV < (CSH - CSL) < 80mV, 4.5V
< V+ < 30V (includes line and load regulation)
2.432
2.635
V
Output Voltage, Fixed 3.3V Mode
FB tied to GND, 0mV < (CSH - CSL) < 80mV, 4.5V
< V+ < 30V (includes line and load regulation)
3.195
3.497
V
VCC = VL = 5V
VREF
5.5
VCC = 3.3V, VL = 5V
VREF
3.6
70
130
Output Adjustment Range
Current-Limit Threshold
Power Consumption
Positive direction
VCC = 5V, output not switching
2.0
VCC = 3.3V, output not switching
1.5
V
mV
mW
INTERNAL VL REGULATOR AND REFERENCE
VCC = 5V, I(VL) = 0
60
Regulator Supply Current (V+)
VCC = 5V, I(VL) = 0, V+ = 4.5V
(includes PNP base current)
500
Standby Supply Current (V+)
SHDN = GND, OVP = VCC
VL Output Voltage
VL Undervoltage Lockout
Threshold
4
60
I(VL) = 0 to 25mA, 5V < V+ < 30V
4.5
5.3
I(VL) = 0 to 25mA, 6V < V+ < 30V
4.7
5.3
Rising edge, hysteresis = 25mV
3.45
3.91
_______________________________________________________________________________________
µA
µA
V
V
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
MAX1636
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1, V+ = 15V, SYNC = VL = VCC, IVL = 0mA, IREF = 0mA, TA = -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
OSCILLATOR
Oscillator Frequency
SYNC = VCC
270
330
SYNC = GND
170
230
kHz
200
SYNC Input Pulse Width High
ns
200
SYNC Input Pulse Width Low
SYNC Input Rise/Fall Time
ns
200
ns
240
340
kHz
Guaranteed by design
SYNC Input Frequency Range
OVERVOLTAGE PROTECTION
Overvoltage Trip Threshold
FB, with respect to regulation point
3.5
10
%
Catastrophic Output Undervoltage
Lockout Threshold
% of nominal output
60
80
%
RESET Trip Threshold
Falling edge (hysteresis = 1%)
-7
-1.5
%
Logic Input Voltage High
SHDN, SKIP, OVP, SYNC
2.4
Logic Input Voltage Low
SHDN, SKIP, OVP, SYNC
0.8
V
RESET Output Voltage Low
ISINK = 4mA
0.4
V
Gate-Driver On-Resistance
High or low, DH or DL
7
Ω
INPUTS AND OUTPUTS
V
Note 1: Specifications to -40°C are guaranteed by design and not production tested.
__________________________________________Typical Operating Characteristics
(Circuit of Figure 1, VIN = 7V, TA = +25°C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT
(5V/3A CIRCUIT)
EFFICIENCY (%)
80
VIN = 30V
VIN = 22V
70
VIN = 15V
80
VIN = 22V
VIN = 30V
70
MAX1636 TOC03
90
100
MAX1636 TOC02
VIN = 15V
VIN = 7V
EFFICIENCY (%)
VIN = 7V
90
100
MAX1636 TOC01
100
EFFICIENCY vs. LOAD CURRENT
(1.8V/1A CIRCUIT)
90
EFFICIENCY (%)
EFFICIENCY vs. LOAD CURRENT
(3.3V/3A CIRCUIT)
VIN = 7V
80
VIN = 15V
70
VIN = 22V
60
60
50
60
50
1m
10m
100m
LOAD CURRENT (A)
1
10
50
1m
10m
100m
LOAD CURRENT (A)
1
10
1m
10m
100m
1
10
LOAD CURRENT (A)
_______________________________________________________________________________________
5
_____________________________Typical Operating Characteristics (continued)
(Circuit of Figure 1, VIN = 7V, TA = +25°C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT
(1.8V/4A CIRCUIT)
VIN = 15V
VIN = 22V
60
80
VIN = 15V
70
60
MAX1636 TOC06
MAX1636 TOC05
VIN = 7V
1000
QUIESCENT SUPPLY CURRENT (mA)
80
70
90
EFFICIENCY (%)
EFFICIENCY (%)
100
MAX1636 TOC04
VIN = 7V
90
QUIESCENT SUPPLY CURRENT
vs. INPUT VOLTAGE
EFFICIENCY vs. LOAD CURRENT
(1.8V/7A CIRCUIT)
100
100
10
1
0.10
VIN = 22V
1
10
1m
10m
2
1
0
-1
-2
30
25
20
15
10
10m
100m
1
0
10
5
10 15 20 25 30 35 40 45 50
LOAD CURRENT (A)
VL LOAD CURRENT (mA)
DROPOUT VOLTAGE
vs. LOAD CURRENT
RESET TIME DELAY
vs. OSC FREQUENCY
300
250
200
150
100
0.5
0.4
0.3
0.2
0
0
10 20 30 40 50 60 70 80 90 100
LOAD-TRANSIENT RESPONSE
(3.3V/3A, PWM MODE)
225
RESET TIME DELAY (ms)
350
200
150
4A
125
LOAD
2A CURRENT
100
0A
75
50
0
0.5
1.0
1.5
2.0
2.5
LOAD CURRENT (A)
6
3.0
3.5
VOUT
50mV/div
175
VOUT FORCED TO 4.95V
50
30
REF LOAD CURRENT (µA)
250
MAX1636 TOC10
400
25
0.6
MAX1636 TOC11
1m
20
0.1
0
-5
15
REF LOAD-REGULATION ERROR
vs. REF LOAD CURRENT
5
-4
10
VL LOAD-REGULATION ERROR
vs. VL LOAD CURRENT
35
-3
5
INPUT VOLTAGE (V)
MAX1636 TOC08
3
0
10
40
LOAD REGULATION ∆V (mV)
PWM MODE
VOUT = 5V
4
LOAD REGULATION ∆ VOUT (mV)
MAX1636 TOC07
5
1
LOAD CURRENT (A)
LOAD CURRENT (A)
LOAD REGULATION
vs. LOAD CURRENT
100m
MAX1636 TOC09
100m
LOAD REGULATION ∆V (mV)
10m
VOUT = 3.3V
0.01
50
1m
MAX1636 TOC12
50
VIN - VOUT (mV)
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
150
200
250
300
350
FOSC (kHz)
400
450
500
100µs/div)
_______________________________________________________________________________________
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
(Circuit of Figure 1, VIN = 7V, TA = +25°C, unless otherwise noted.)
LOAD-TRANSIENT RESPONSE
(1.8V, PWM MODE)
MAX1636 TOC14
MAX1636 TOC13
SWITCHING WAVEFORMS
(PWM MODE)
VOUT
50mV/div
VOUT
20mV/div
5V
VLX
0V
5A LOAD CURRENT
1A
0A
0A
1µs/div
SWITCHING WAVEFORMS
(PFM MODE)
SWITCHING WAVEFORMS
(DROPOUT OPERATION)
VOUT = 1.8V
MAX1636 TOC15
100µs/div
VOUT = 5V
VOUT
50mV/div
MAX1636 TOC16
10A
VLX
VLX
0
0V
0A
VOUT
20mV/div
5V
5V
1A
INDUCTOR
CURRENT
4A
INDUCTOR
CURRENT
INDUCTOR
CURRENT
2A
0A
STANDBY AND STARTUP RESPONSE
(VOUT = 1.8V, NO LOAD)
MAX1636 TOC18
5µs/div
OVERVOLTAGE-PROTECTION WAVEFORMS
(VIN SHORTED TO VOUT
THROUGH a 0.5Ω RESISTOR)
MAX1636 TOC17
20µs/div
VOUT
1V/div
VOUT
100mV/div
5V
VDL
0
0
VSHDN
5V/div
-5A
INDUCTOR
CURRENT
-10A
1ms/div
10µs/div
_______________________________________________________________________________________
7
MAX1636
_____________________________Typical Operating Characteristics (continued)
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
______________________________________________________________Pin Description
PIN
NAME
FUNCTION
1
CSH
Current-Sense Input, High Side
2
CSL
Current-Sense Input, Low Side. Also serves as a feedback input in fixed output modes.
3
RESET
Timed Reset Output. Low for at least 100ms after output voltage is valid, then goes high impedance
(open drain).
4
SHDN
Shutdown Control Input. Puts chip in shutdown or standby mode, depending on OVP (Table 5).
5
OVP
Overvoltage Protection Enable/Disable. Tie to GND to disable OVP; tie to VCC to enable OVP.
6
CC
Compensation pin. Connect a small capacitor to GND to set the integration time constant.
7
REF
1.100V Reference Output. Capable of sourcing 50µA for external loads; bypass with a 0.22µF
(min) capacitor.
8
SYNC
Oscillator Frequency Select and Synchronization Input. Tie to VCC for 300kHz operation; tie to GND for
200kHz operation.
9, 10
GND
Analog Ground
11
FB
12
VCC
Main Supply Voltage Input. Powers the PWM controller, logic, and reference. Input range is +3.15V to
+5.5V.
13
V+
5V VL Linear-Regulator Input. The VL linear regulator automatically shuts off if V+ is left open or shorted to
VL. Bypass V+ to GND with a 0.1µF capacitor close to the IC.
14
VL
5V Linear-Regulator Output. Powers the DL low-side gate driver. Bypass with a 2.2µF (min) capacitor.
15
DL
Low-Side Gate-Driver Output
16
PGND
17
BST
Boost-Capacitor Connection
18
DH
High-Side Gate-Driver Output
19
LX
Inductor Connection
20
SKIP
Feedback Input. Tie to GND for fixed 3.3V output; tie to VCC for fixed 2.5V output; tie to resistor divider for
adjustable mode.
Power Ground
Low-Noise Mode Control. Forces fixed-frequency PWM operation when high.
______Standard Application Circuit
The basic MAX1636 buck converter (Figure 1) is easily
adapted to meet a wide range of applications with
inputs up to 30V by substituting components from
Table 1. These circuits represent a good set of tradeoffs between cost, size, and efficiency, while staying
within the worst-case specification limits for stressrelated parameters, such as capacitor ripple current.
Do not change the circuits’ switching frequency without
8
first recalculating component values (particularly inductance value at maximum battery voltage). Adding a
Schottky rectifier across the synchronous rectifier
improves circuit efficiency by approximately 1%. This
rectifier is otherwise not needed because the MOSFET
required typically incorporates a high-speed silicon
diode from drain to source. Use a Schottky rectifier
rated at a DC current equal to at least one-third of the
load current.
_______________________________________________________________________________________
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
MAX1636
Table 1. Component Selection for Standard Applications
LOAD CURRENT
COMPONENT
1A
4A
7A (EV KIT)
3A
3A
Input Voltage
Range
7V to 22V
7V to 22V
7V to 22V
4.75V to 30V
6V to 30V
Output Voltage
Range
1.8V
1.8V
1.25V to 2V
3.3V
5V
Application
CPU I/O
CPU Core
CPU Core
Frequency
300kHz
300kHz
300kHz
300kHz
300kHz
Q1 High-Side
MOSFET
1/2 Si4902DY or
1/2 MMDF3NO3HD
International Rectifier
International Rectifier
IRF7413,
IRF7403 or
Fairchild NDS8410A,
Siliconix Si9804DY
or Siliconix Si4410DY
International Rectifier
IRF7413,
Fairchild NDS8410A,
or Siliconix Si4410DY
International Rectifier
IRF7413,
Fairchild NDS8410A,
or Siliconix Si4410DY
Q2 Low-Side
MOSFET
1/2 Si4902DY or
1/2 MMDF3NO3HD
International Rectifier
IRF7413,
Fairchild FDS6680 or
Fairchild NDS8410A, Siliconix Si4420DY
or Siliconix Si4410DY
International Rectifier
IRF7413,
Fairchild NDS8410A,
or Siliconix Si4410DY
International Rectifier
IRF7413,
Fairchild NDS8410A,
or Siliconix Si4410DY
C1 Input
Capacitor
4.7µF, 25V ceramic
Tokin C34Y5U1E475Z
or Marcon/United
Chemicon
THCR40E1E475Z
2 x 10µF, 25V ceramic
Tokin C34Y5U1E106Z
or Marcon/United
Chemicon
THCR50E1E106ZT
4 x 10µF, 25V ceramic
Tokin C34Y5U1E106Z
or Marcon/United
Chemicon
THCR50E1E106ZT
2 x 22µF, 35V
AVX
TPSE226M035R0300
or
Sprague
593D226X0035E2W
2 x 22µF, 35V
AVX
TPSE226M035R0300
or
Sprague
593D226X0035E2W
C2 Output
Capacitor
220µF, 6.3V tantalum
Sprague
595D227X96R3C2
2 x 470µF,
4V low-ESR
Sprague
594D477X0004R2T
4 x 390µF, 6.3V lowESR, Sprague
594D397X06R3R2T,
or 4 x 470µF, 4V
Sprague
594D477X0004R2T
2x 220µF
Sprague 594D
594D227X0010D2T
R1 Resistor
0.070Ω, 1% (1206)
Dale
WSL-1206-R070F
0.015Ω, 1% (2512)
Dale
WSL-2512-R015F
0.010Ω, 1% (2512)
Dale
WSL-2512-R010F
0.020Ω, 1% (2010)
Dale
WSL-2010-R020F
0.020Ω, 1% (2010)
Dale
WSL-2010-R020F
15µH
Sumida CD54-150
4.6µH
Panasonic
ETQP1F4R6H,
Sumida
CDRH127-4R7,
Coiltronics
UP2-4R7, or
Coilcraft
DO3316P-472
2.2µH
Panasonic P1F2R0HL,
Sumida
CDRH127-2R4,
Coiltronics
UP4-2R2, or
Coilcraft
DO5022P-222HC
10µH
Sumida
CDRH125-100,
Coiltronics
UP2-100, or
Coilcraft
DO3316-103
10µH
Sumida
CDRH125-100,
Coiltronics
UP2-100, or
Coilcraft
DO3316-103
L1 Inductor
2x 220µF
Sprague 594D
594D227X0010D2T
_______________________________________________________________________________________
9
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
Table 2. Component Suppliers
POWER INPUT
COMPANY
0.1µF
SHDN
V+
OVP
SYNC
VL
VCC
4.7µF
C1
1nF
CC
MAX1636
BST
CMPSH-3
DH
REF
Q1
0.1µF
1µF
L1
R1
OUTPUT
LX
GND
DL
GND
Q2
C2
*
PGND
CSH
VCC
CSL
R2
10k
RESET
FB
SKIP
R3
*SEE RECTIFIER CLAMP DIODE SECTION
FACTORY FAX
(COUNTRY CODE)
USA PHONE
AVX
(1) 803-626-3123
(803) 946-0690
Central
Semiconductor
(1) 516-435-1824
(516) 435-1110
Coilcraft
(1) 847-639-1469
(847) 639-6400
Coiltronics
(1) 561-241-9339
(561) 241-7876
Dale
(1) 605-665-1627
(605) 668-4131
Fairchild
(1) 408-721-1635
(408) 721-2181
International
Rectifier (IR)
(1) 310-322-3332
(310) 322-3331
IRC
(1) 512-992-3377
(512) 992-7900
Marcon/United
Chemi-Con
IRC
(1) 847-696-9278
(1) 512-992-3377
(847) 696-2000
(512) 992-7900
Matsuo
Motorola
(1) 714-960-6492
(1) 602-994-6430
(714) 969-2491
(602) 303-5454
Panasonic
(1) 714-373-7183
(714) 373-7939
Sanyo
(81) 7-2070-1174
(619) 661-6835
Siliconix
(1) 408-970-3950
(408) 988-8000
Sprague
(1) 603-224-1430
(603) 224-1961
Sumida
(81) 3-3607-5144
(847) 956-0666
TDK
(1) 847-390-4428
(847) 390-4373
Tokin
(1) 408-434-0375
(408) 432-8020
Figure 1. Standard Application Circuit
_______________Detailed Description
The MAX1636 is a BiCMOS, switch-mode, power-supply controller designed primarily for buck-topology regulators in battery-powered applications where high
efficiency and low quiescent supply current are critical.
Light-load efficiency is enhanced by automatic Idle
Mode operation, a variable-frequency, pulse-skipping
mode that reduces transition and gate-charge losses.
The step-down, power-switching circuit consists of two
N-channel MOSFETs, a rectifier, and an LC output filter.
The output voltage is the average AC voltage at the
switching node, which is regulated by changing the
duty cycle of the MOSFET switches. The gate-drive signal to the N-channel high-side MOSFET, which must
exceed the battery voltage, is provided by a flyingcapacitor boost circuit that uses a 100nF capacitor
between BST and LX. The MAX1636 contains 10 major
circuit blocks (Figure 2).
The pulse-width-modulation (PWM) controller consists
of a Dual Mode™ feedback network and multiplexer, a
multi-input PWM comparator, high-side and low-side
gate drivers, and logic. The MAX1636 contains faultprotection circuits that monitor the PWM output for
undervoltage and overvoltage. Bias generator blocks
include the 5V (VL) linear regulator and the 1.1V precision reference. The PWM uses a 200kHz/300kHz synchronizable oscillator. The circuit blocks are powered
from an internal IC power rail that receives power from
either VL or VCC. The synchronous-switch gate driver is
powered directly from VL, while the high-side-switch
gate driver is powered indirectly from VL via an external
diode-capacitor boost circuit.
Dual Mode is a trademark of Maxim Integrated Products.
10
______________________________________________________________________________________
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
MAX1636
INPUT
V+
SYNC
TO VL
MAX1636
VCC
POWER
SWITCHOVER
IC
POWER
BST
200kHz
TO
300kHz
OSC
5V
LINEAR
REG.
VL
SKIP
DH
LX
PWM
LOGIC
VL
DL
+
PGND
1.1V
REF.
REF
SHDN
SHUTDOWN
CONTROL
OFF
+
-
REF
UNDERVOLTAGE
FAULT
CC
REF
FB
+
VREF +7%
VREF -5%
+
60kHz
LP FILTER
-
RESET
gm
-
OVERVOLTAGE
FAULT
VREF -30%
ERROR
INTEGRATOR
+
+
OVP
-
SLOPE
COMPENSATION
CSH
CSL
TIMER
POWER
GOOD
+
0.2V
GND
Figure 2. Functional Diagram
______________________________________________________________________________________
11
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
PWM Controller
Table 3. SKIP PWM Table
SKIP
LOAD
CURRENT
MODE
Low
Light
Idle
Low
Heavy
PWM
Constant-frequency PWM,
continuous inductor current
High
Light
PWM
Constant-frequency PWM,
continuous inductor current
High
Heavy
PWM
Constant-frequency PWM,
continuous inductor current
The heart of the current-mode PWM controller is a
multi-input, open-loop comparator that sums four signals: the output voltage error signal with respect to the
reference voltage, the current-sense signal, the integrated voltage-feedback signal, and the slopecompensation ramp (Figure 3).
The PWM controller is a direct-summing type, lacking a
traditional error amplifier and the phase shift associated with it. This direct-summing configuration approaches ideal cycle-by-cycle control over the output voltage
(Figure 4).
When SKIP = low, Idle Mode circuitry automatically
optimizes efficiency throughout the load-current range.
Idle Mode dramatically improves light-load efficiency
DESCRIPTION
Pulse-skipping,
discontinuous inductor
current
CSH
1X
CSL
FB
2X
REF
CC
gm
BST
R
S
LEVEL
SHIFT
Q
DH
LX
OSC
SLOPE
COMPENSATION
30mV
SKIP
CURRENT
LIMIT
DAC
SHDN
COUNTER
SHOOTTHROUGH
CONTROL
CK
SOFT-START
SYNCHRONOUS
RECTIFIER CONTROL
R
-100mV
S
VL
Q
LEVEL
SHIFT
DL
PGND
Figure 3. PWM Controller Functional Diagram
12
______________________________________________________________________________________
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
MAX1636
VCC
R1
R2
TO PWM
LOGIC
UNCOMPENSATED
HIGH-SPEED
LEVEL TRANSLATOR
AND BUFFER
OUTPUT DRIVER
FB
CC
I1
I2
I3
I4
VBIAS
REF
CSH
CSL
SLOPE COMPENSATION
Figure 4. Main PWM Comparator Functional Diagram
by reducing the effective frequency, subsequently
reducing switching losses. It forces the peak inductor
current to ramp to 30% of the full current limit, delivering extra energy to the output and allowing subsequent
cycles to be skipped. Idle Mode transitions seamlessly
to fixed-frequency PWM operation as load current
increases.
With SKIP = high, the controller always operates in
fixed-frequency PWM mode for lowest noise. Each
pulse from the oscillator sets the main PWM latch that
turns on the high-side switch for a period determined
by the duty factor (approximately VOUT / VIN). As the
high-side switch turns off, the synchronous rectifier
latch sets; 60ns later, the low-side switch turns on. The
low-side switch stays on until the beginning of the next
clock cycle.
In PWM mode, the controller operates as a fixed-frequency, current-mode controller in which the duty factor is set by the input/output voltage ratio. The
current-mode feedback system regulates the peak
inductor current value as a function of the output voltage error signal. In continuous-conduction mode, the
average inductor current is nearly the same as the
peak current, so the circuit acts as a switch-mode
transconductance amplifier. This pushes the second
output LC filter pole, normally found in a duty-factorcontrolled (voltage-mode) PWM, to a higher frequency.
To preserve inner-loop stability and eliminate regenerative inductor current “staircasing,” a slope-compensation ramp is summed into the main PWM comparator to
make the apparent duty factor less than 50%.
The relative gains of the voltage-sense and currentsense inputs are weighted by the values of current
sources that bias four differential input stages in the
main PWM comparator (Figure 4). The voltage sense
into the PWM has been conditioned by an integrated
component of the feedback voltage, yielding excellent
DC output voltage accuracy. See the Output Voltage
Accuracy section for more information.
Synchronous Rectifier Driver (DL)
Synchronous rectification reduces conduction losses in
the rectifier by shunting the normal Schottky catch
diode with a low-resistance MOSFET switch. Also, the
synchronous rectifier ensures proper start-up of the
boost gate-driver circuit. If the synchronous power
MOSFET is omitted for cost or other reasons, replace it
with a small-signal MOSFET, such as a 2N7002.
If the circuit is operating in continuous-conduction
mode, the DL drive waveform is simply the complement of the DH high-side-drive waveform (with controlled dead time to prevent cross-conduction or
“shoot-through”). In discontinuous (light-load) mode,
the synchronous switch is turned off as the inductor
current falls through zero.
______________________________________________________________________________________
13
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
Table 4. Powering the MAX1636
AVAILABLE
POWER SOURCES
Battery, 3.3V, and 5V
Battery and 5V
Battery and 3.3V
Battery only
VCC
CONNECTS
TO
V+
CONNECTS
TO
VL
CONNECTS
TO
COMMENT
3.3V
5V
5V
Most efficient
5V
5V
5V
3.3V
Battery
Bypass capacitor only
VL
Battery
Bypass capacitor only
REF and VL Supplies and VCC Input
The 1.1V reference (REF) is accurate to ±1% over temperature, making REF useful as a precision system reference. Bypass REF to GND with a 0.22µF (min)
capacitor. REF can supply up to 50µA for external
loads. Loading REF reduces the main output voltage
slightly because of the reference load-regulation error.
The 5V VL linear-regulator output can be tied to the
system +5V supply in order to obtain gate-drive power
from an efficient source. The two supply pins (VCC and
VL) are independent of each other (no protection
diodes or sequencing requirements), allowing you to
choose the most efficient sources for chip biasing from
among existing system supply voltages without having
to worry about sequencing or latch-up problems
(Table 4).
The V CC input runs the chip if the V CC voltage is
greater than 3.15V. Otherwise, the chip supply is powered from VL via the internal VCC-VL switchover circuit.
If a system supply between 3.3V and 5V is not available, tie VCC directly to VL.
In shutdown mode, the VL regulator and reference are
completely turned off. In standby mode, the VL regulator and DL stay alive so that the overvoltage-protection
circuit can operate (Table 5).
Important: Ensure that VL and VCC do not exceed 6V.
Measure VL with the main output fully loaded. If it is
pumped above 5.5V, either excessive boost-diode
capacitance or excessive ripple at V+ is the probable
cause. Use only small-signal diodes for the boost circuit (10mA to 100mA Schottky or 1N4148 are preferred) and bypass VL to PGND with a 4.7µF capacitor
directly at the package pins.
Shutdown and Standby Modes
Holding SHDN low puts the IC into its 3µA shutdown
mode. SHDN is a logic input with a threshold of about
1V (the VTH of an internal N-channel MOSFET). For
automatic start-up, tie SHDN to V+.
14
Least efficient
Standby operation is entered when SHDN = low and
OVP = high (Table 5). In standby mode, the VL regulator stays active, and the DL output is forced high to
provide overvoltage protection by clamping the output
to GND. However, DL is not forced high until the output
sags below VREF, so that the output can be held high
by external keep-alive supplies.
RESET Power-Good Voltage Monitor
The power-good monitor generates a system-reset signal. The RESET output is an open drain that needs to
be pulled up to the appropriate logic supply. At first
power-up, RESET is held low until output is in regulation. At this point, an internal timer begins counting
oscillator pulses, and RESET continues to be held low
until 32,000 cycles have elapsed. After this timeout
period (107ms at 300kHz or 160ms at 200kHz), the
RESET output is released.
Output Undervoltage Lockout
The output undervoltage-lockout circuit is similar to
foldback current limiting but employs a timer rather
than a variable current limit. The SMPS has an undervoltage-protection circuit that is activated 6144 clock
cycles after the SMPS is enabled. If the SMPS output is
under 70% of the nominal value, output is latched off
and does not restart until SHDN is toggled or until V+
power is cycled below 1V. Note that undervoltage protection can make prototype troubleshooting difficult,
since only 20ms or 30ms elapse before the SMPS is
latched off.
The output undervoltage lockout circuit protects
against heavy overloads and shorts to the main SMPS
output. The circuit trips if the output is less than 70% of
the nominal output value any time after the timeout has
expired upon start-up. When the comparator trips, the
output is turned off (the SMPS stops switching). This
state is similar to thermal shutdown and can be exited
by a power-on reset or by a rising edge on SHDN. The
overvoltage crowbar is disabled in output undervoltage
or thermal shutdown modes.
______________________________________________________________________________________
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
MAX1636
Table 5. Operating Modes
MODE
SHDN
OVP
Run
High
High
All circuit blocks
active
Normal operation
DL = high to enforce overvoltage
protection
HOW ENTERED
STATUS
NOTES
Standby
Low
High
VL = on
REF = off
DL = high
RESET = high-Z
(high state)
Shutdown
Low
Low
All circuit blocks
inactive
Lowest possible quiescent consumption
High
VOUT > 7% too high
VL = on
REF = off
DL = high
RESET = low
Cycling SHDN or a power-on reset exits
crowbar.
VOUT < 70% of
nominal after
20–30ms timeout
expires
VL = on
REF = off
DL = low
RESET = low
Cycling SHDN or a power-on reset exits
output UVLO.
Cycling SHDN or a power-on reset exits
thermal shutdown.
Cycling SHDN or a power-on reset exits
thermal shutdown.
Overvoltage
(crowbar)
Output
UVLO
High
High
Don’t care
Thermal
Shutdown
High
High
TJ > +150°C
VL = on
REF = off
DL = high
RESET = low
Thermal
Shutdown
High
Low
TJ > +150°C
All circuit blocks
inactive
Output Overvoltage Protection (OVP)
The overvoltage crowbar protection circuit is intended
to blow a fuse in series with the battery if the main
SMPS output rises significantly higher than its preset
level. In normal operation, the output is compared to
the internal precision reference voltage. If the output
goes 7% above nominal, the synchronous rectifier
MOSFET turns on 100% (the high-side MOSFET is
simultaneously forced off) in order to draw massive
amounts of battery current to blow the fuse. This safety
feature does not protect the system against a failure of
the controller IC itself but is intended primarily to guard
against a short across the high-side MOSFET. A crowbar event is latched and can only be reset by a rising
edge on SHDN (or by removal of the V+ supply voltage). The overvoltage-detection decision is made relative to the regulation point.
The overvoltage comparators are kept inactive in
standby mode. Instead, the DL driver is simply left in
the high state. However, DL does not turn on until the
output has decayed to less than 1V. This prevents con-
flicts in systems where the output is held up by an
external source in suspend or backup mode. The OVP
pin has an internal pulldown resistor that is only turned
on during the reset phase. The OVP pin’s state is then
sampled and stored internally. A floating OVP pin
implies no overvoltage protection.
Boost High-Side Gate-Drive Supply (BST)
Gate-drive voltage for the high-side N-channel switch is
generated by a flying-capacitor boost circuit (Figure 2).
The capacitor between BST and LX is alternately
charged from the VL supply and placed parallel to the
high-side MOSFET's gate-source terminals.
On start-up, the synchronous rectifier (low-side MOSFET) forces LX to 0V and charges the boost capacitor
to 5V. On the second half-cycle, the SMPS turns on the
high-side MOSFET by closing an internal switch
between BST and DH. This provides the necessary
enhancement voltage to turn on the high-side switch,
an action that boosts the 5V gate-drive signal above
the battery voltage.
______________________________________________________________________________________
15
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
Ringing at the high-side MOSFET gate (DH) in discontinuous-conduction mode (light loads) is a natural operating condition. It is caused by residual energy in the
tank circuit, formed by the inductor and stray capacitance at the switching node, LX. The gate-drive negative rail is referred to LX, so any ringing there is directly
coupled to the gate-drive output.
Current-Limiting and Current-Sense Inputs
(CSH and CSL)
The current-limit circuit resets the main PWM latch and
turns off the high-side MOSFET switch whenever the
voltage difference between CSH and CSL exceeds
100mV. This limiting is effective for both current flow
directions, putting the threshold limit at ±100mV. The
tolerance on the positive current limit is ±20%, so the
external low-value sense resistor (R1) must be sized for
80mV/IPEAK, where IPEAK is the required peak inductor
current to support the full load current. Components
must be designed to withstand continuous current
stresses of 120mV/R1.
For breadboarding or for very high current applications,
it may be useful to wire the current-sense inputs with a
twisted pair rather than PC traces (two pieces of
wrapped wire twisted together are sufficient.) This
reduces the noise picked up at CSH and CSL, which
can cause unstable switching and reduced output current.
Oscillator Frequency and Synchronization
(SYNC)
The SYNC input controls the oscillator frequency. Low
selects 200kHz; high selects 300kHz. SYNC can also
be used to synchronize with an external 5V CMOS or
TTL clock generator. SYNC has a guaranteed 240kHz
to 340kHz capture range. A high-to-low transition on
SYNC initiates a new cycle.
Operation at 300kHz optimizes the application circuit
for component size and cost. Operation at 200kHz provides increased efficiency, lower dropout, and
improved load-transient response at low input-output
voltage differences (see the Low-Voltage Operation
section).
Output Voltage Accuracy (GND, CC)
Output voltage error is guaranteed to be within ±1%
over all conditions of line, load, and temperature. The
DC load regulation is typically better than 0.1% due to
the integrator amplifier. Transient response is optimized
by providing a feedback signal that has a direct path
from the output to the main summing PWM comparator.
The integrated feedback signal is also summed into the
16
PWM comparator, with the gain weighted so that the
integrated signal has only enough gain to correct the
DC inaccuracies. The integrator’s response time is
determined by the time constant set by the capacitor
placed on the CC pin. The time constant should not be
so fast that the integrator responds to the normal VOUT
ripple or too slow to negate the integrator’s effect. A
470pF to 1500pF CC capacitor is sufficient for 200kHz
to 300kHz frequencies.
Figure 5 shows the output voltage response to a 0A to
3A load transient with and without the integrator. With
the integrator, the output voltage returns to within 0.1%
of its no-load value with only a small AC excursion.
Without the integrator, the typical load-transient
response with the AC and DC output voltage changes.
Asymmetrical clamping at the integrator output prevents worsening of load transients during pulseskipping mode.
Internal Digital Soft-Start Circuit
Soft-start allows a gradual increase of the internal current-limit level at start-up to reduce input surge currents. The SMPS contains an internal digital soft-start
circuit controlled by a counter, a digital-to-analog converter (DAC), and a current-limit comparator. In shutdown or standby mode, the soft-start counter is reset to
zero. When the SMPS is enabled, its counter starts
counting oscillator pulses, and the DAC begins incrementing the comparison voltage applied to the currentlimit comparator. The DAC output increases from 0mV
to 100mV in five equal steps as the count increases to
512 clocks. As a result, the main output capacitor
charges up relatively slowly. The exact time of the output rise depends on output capacitance and load current, but it is typically 1ms with a 300kHz oscillator.
Overload and Dropout Operation
Dropout (low input-output differential) operation is
enhanced by stretching the clock pulse width to
increase the maximum duty factor. The algorithm follows: If the output voltage (VOUT) drops out of regulation without the current limit having been reached, the
SMPS skips an off-time period (extending the on-time).
At the end of the cycle, if the output is still out of regulation, the SMPS skips another off-time period. This
action can continue until three off-time periods are
skipped, effectively dividing the clock frequency by as
much as four. This behavior also slightly improves loadtransient response. Dividing the clock frequency by
four raises the maximum duty factor to above 98%. The
typical PWM minimum off-time is 300ns, regardless of
the operating frequency.
______________________________________________________________________________________
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
50
INTEGRATOR
DEFEATED
CC = REF
VOUT = 3.3V
VOUT
(mV)
VOUT
(mV)
-50
-50
4
4
IOUT
(A)
MAX1636
50
INTEGRATOR
ACTIVE
CC = 470pF
VOUT = 3.3V
IOUT
(A)
2
2
0
0
(100µs/div)
Figure 5a. Load-Transient Response with Integrator Active
Adjustable-Output Feedback
(Dual-Mode FB)
A fixed, preset output voltage of 2.5V and 3.3V is
selected when FB is connected to VCC or ground. In
this mode, internal resistors monitor the voltage on
CSL. For voltages other than the fixed-output options,
adjust the output voltage through a resistor divider connected to FB (Figure 2). Calculate the output voltage
with the following formula:
VOUT = VREF (1 + R1 / R2)
where VREF = 1.1V nominal. Recommended normal values for R2 range from 5kΩ to 100kΩ. To achieve a 1.1V
nominal output, simply connect FB directly to CSL.
Remote output voltage sensing is not possible in fixed
output mode due to the combined nature of the voltage-sense and current-sense inputs (CSL). It is, however, easy to do in adjustable mode by using the top of
the external resistor divider as the remote sense point.
Low-Noise Operation (PWM Mode)
PWM mode (SKIP = high) minimizes RF and audio
interference in noise-sensitive applications such as hi-fi
multimedia-equipped systems, cellular phones, RF
communicating computers, and electromagnetic penentry systems. See the summary of operating modes in
Table 5. SKIP can be driven from an external logic signal.
(100µs/div)
Figure 5b. Load-Transient Response with Integrator Defeated
PWM mode forces a constant switching frequency,
reducing interference due to switching noise by concentrating the emissions at a known frequency outside
the system audio or IF bands. Choose an oscillator frequency for which switching frequency harmonics do
not overlap a sensitive frequency band. If necessary,
synchronize the oscillator to a tight-tolerance external
clock generator. To extend the output voltage-regulation range, constant operating frequency is not maintained under overload or dropout conditions (see the
Overload and Dropout Operation section).
PWM mode (SKIP = high) forces two changes on the
PWM controller. First, it disables the minimum-current
comparator, ensuring fixed-frequency operation.
Second, it changes the detection threshold for reversecurrent limit from 0mV to -100mV, allowing the inductor
current to reverse at light loads. This results in fixed-frequency operation and continuous inductor-current flow.
PWM mode eliminates discontinuous-mode inductor
ringing and improves cross-regulation of transformercoupled, multiple-output supplies.
In most applications, tie SKIP to GND to minimize quiescent supply current. VL supply current with SKIP high
is typically 20mA, depending on external MOSFET gate
capacitance and switching losses.
______________________________________________________________________________________
17
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
__________________Design Procedure
L = VOUT(VIN(MAX) - VOUT) / (VIN(MIN) x f x IOUT x LIR)
The five predesigned standard application circuits
(Figure 1 and Table 1) contain ready-to-use solutions
for common application needs. Use the following
design procedure to optimize these basic schematics
for different voltage or current requirements. But before
beginning a design, firmly establish the following:
• Maximum input (battery) voltage, V IN(MAX) . This
value should include the worst-case conditions, such
as no-load operation when a battery charger or AC
adapter is connected but no battery is installed.
VIN(MAX) must not exceed 30V.
• Minimum input (battery) voltage, VIN(MIN). This should
be taken at full load under the lowest battery conditions. If VIN(MIN) is less than 4.5V, use an external circuit to externally hold VL above the VL undervoltage
lockout threshold. If the minimum input-output difference is less than 1.5V, the filter capacitance required
to maintain good AC load regulation increases (see
Low-Voltage Operation section).
where f = switching frequency, normally 200kHz or
300kHz, and IOUT = maximum DC load current. The
peak current can be calculated by:
IPEAK = ILOAD + [VOUT(VIN(MAX) - VOUT) / (2 x f x L x VIN(MAX))]
The inductor's DC resistance should be low enough
that RDC x IPEAK < 100mV, as it is a key parameter for
efficiency performance. If a standard, off-the-shelf
inductor is not available, choose a core with an LI2 rating greater than L x IPEAK2 and wind it with the largest
diameter wire that fits the winding area. For 300kHz
applications, ferrite-core material is strongly preferred;
for 200kHz applications, Kool-Mu® (aluminum alloy) or
even powdered iron is acceptable. If light-load efficiency is unimportant (in desktop PC applications, for
example), then low-permeability iron-powder cores may
be acceptable, even at 300kHz. For high-current applications, shielded-core geometries, such as toroidal or
pot core, help keep noise, EMI, and switching-waveform jitter low.
Inductor Value
Current-Sense Resistor Value
The exact inductor value is not critical and can be
freely adjusted to make trade-offs between size, cost,
and efficiency. Lower inductor values minimize size
and cost but reduce efficiency due to higher peak-current levels. The smallest inductor is achieved by lowering the inductance until the circuit operates at the
border between continuous and discontinuous mode.
Further reducing the inductor value below this
crossover point results in discontinuous-conduction
operation even at full load. This helps lower output filter
capacitance requirements, but efficiency suffers due to
high I2R losses. On the other hand, higher inductor values mean greater efficiency, but resistive losses due to
extra wire turns eventually exceed the benefit gained
from lower peak-current levels. Also, high inductor values can affect load-transient response (see the VSAG
equation in the Low-Voltage Operation section). The
equations in this section are for continuous-conduction
operation.
Three key inductor parameters must be specified:
inductance value (L), peak current (IPEAK), and DC
resistance (RDC). The following equation includes a
constant, LIR, which is the ratio of inductor peak-topeak AC current to DC load current. A higher LIR value
allows smaller inductance but results in higher losses
and higher ripple. A good compromise between size
and losses is a 30% ripple-current to load-current ratio
(LIR = 0.3), which corresponds to a peak inductor current 1.15 times higher than the DC load current.
The current-sense resistor value is calculated according to the worst-case, low-current-limit threshold voltage (from the Electrical Characteristics table) and the
peak inductor current:
RSENSE = 80mV / IPEAK
Use IPEAK from the second equation in the Inductor
Value section. Use the calculated value of RSENSE to
size the MOSFET switches and specify inductor saturation-current ratings according to the worst-case highcurrent-limit threshold voltage:
IPEAK = 120mV / RSENSE
Low-inductance resistors, such as surface-mount metal
film, are recommended.
Input Capacitor Value
Connect low-ESR bulk capacitors directly to the drain
on the high-side MOSFET. The bulk input filter capacitor is usually selected according to input ripple current
requirements and voltage rating, rather than capacitor
value. Electrolytic capacitors with low enough equivalent series resistance (ESR) to meet the ripple-current
requirement invariably have sufficient capacitance values. Aluminum electrolytic capacitors, such as Sanyo
OS-CON or Nichicon PL, are superior to tantalum
types, which risk power-up surge-current failure, especially when connecting to robust AC adapters or lowimpedance batteries. RMS input ripple current (IRMS) is
Kool-Mu is a registered trademark of Magnetics, Inc.
18
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Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
IRMS = ILOAD ×
(
)
VOUT VIN − VOUT / VIN
Therefore, when VIN is 2 x VOUT:
IRMS = ILOAD / 2
Output Filter Capacitor Value
The output filter capacitor values are generally determined by the ESR and voltage-rating requirements
rather than actual capacitance requirements for loop
stability. In other words, the low-ESR electrolytic capacitor that meets the ESR requirement usually has more
output capacitance than is required for AC stability.
Use only specialized low-ESR capacitors intended for
switching-regulator applications, such as AVX TPS,
Sprague 595D, Sanyo OS-CON, or Nichicon PL series.
To ensure stability, the capacitor must meet both minimum capacitance and maximum ESR values as given
in the following equations:
COUT > VREF(1 + VOUT / VIN(MIN)) / VOUT x RSENSE x f
RESR < RSENSE x VOUT / VREF
where RESR can be multiplied by 1.5, as discussed
below.
These equations are worst case, with 45 degrees of
phase margin to ensure jitter-free, fixed-frequency
operation, and provide a nicely damped output
response for zero to full-load step changes. Some costconscious designers may wish to bend these rules with
less-expensive capacitors, particularly if the load lacks
large step changes. This practice is tolerable if some
bench testing over temperature is done to verify
acceptable noise and transient response.
No well-defined boundary exists between stable and
unstable operation. As phase margin is reduced, the
first symptom is timing jitter, which shows up as blurred
edges in the switching waveforms where the scope
does not quite sync up. Technically speaking, this jitter
(usually harmless) is unstable operation, since the duty
factor varies slightly. As capacitors with higher ESRs
are used, the jitter becomes more pronounced, and the
load-transient output voltage waveform starts looking
ragged at the edges. Eventually, the load-transient
waveform has enough ringing on it that the peak noise
levels exceed the allowable output voltage tolerance.
Note that even with zero phase margin and gross instability, the output voltage noise never gets much worse
than IPEAK x RESR (under constant loads).
Designers of RF communicators or other noise-sensitive analog equipment should be conservative and stay
within the guidelines. Designers of notebook computers
and similar commercial-temperature-range digital systems can multiply the RESR value by a factor of 1.5
without hurting stability or transient response.
The output voltage ripple, which is usually dominated
by the filter capacitor’s ESR, can be approximated as
IRIPPLE x RESR. There is also a capacitive term, so the
full equation for ripple in continuous-conduction mode
is V NOISE(p-p) = I RIPPLE x [R ESR + 1 / (2 x p x f x
COUT)]. In Idle Mode, the inductor current becomes
discontinuous, with high peaks and widely spaced
pulses, so the noise can actually be higher at light load
(compared to full load). In Idle Mode, calculate the output ripple as follows:
VNOISE(p−p) =
[
0.02 x RESR
+
RSENSE
(
0.0003 x L x 1/VOUT + 1/ VIN − VOUT
(RSENSE )
2
)]
x CF
Selecting Other Components
MOSFET Switches
The high-current N-channel MOSFETs must be logiclevel types with guaranteed on-resistance specifications at VGS = 4.5V. Lower gate-threshold
specifications are better (i.e., 2V max rather than 3V
max). Drain-source breakdown voltage ratings must at
least equal the maximum input voltage, preferably with
a 20% derating factor. The best MOSFETs have the
lowest on-resistance per nanocoulomb of gate charge.
Multiplying RDS(ON) by Qg provides a good figure of
merit for comparing various MOSFETs. Newer MOSFET
process technologies with dense cell structures generally perform best. The internal gate drivers tolerate
>100nC total gate charge, but 70nC is a more practical
upper limit to maintain best switching times.
In high-current applications, MOSFET package power
dissipation often becomes a dominant design factor.
I2R power losses are the greatest heat contributor for
both high-side and low-side MOSFETs. I2R losses are
distributed between Q1 and Q2 according to duty factor as shown in the equations below. Generally, switching losses affect only the upper MOSFET, since the
Schottky rectifier usually clamps the switching node
before the synchronous rectifier turns on. Gate-charge
losses are dissipated by the driver and do not heat the
MOSFET. Calculate the temperature rise according to
package thermal-resistance specifications to ensure
______________________________________________________________________________________
19
MAX1636
determined by the input voltage and load current, with
the worst case occurring at VIN = 2 x VOUT:
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
that both MOSFETs are within their maximum junction
temperature at high ambient temperature. The worstcase dissipation for the high-side MOSFET occurs at
both extremes of input voltage, and the worst-case dissipation for the low-side MOSFET occurs at maximum
input voltage.
Duty = (VOUT + VQ2) / (VIN - VQ1)
PD (upper FET) = ILOAD2 x RDS(ON) x Duty + VIN x
ILOAD x f x [(VIN x CRSS) / IGATE + 20ns]
PD (lower FET) = ILOAD2 x RDS(ON) x (1 - Duty)
where on-state voltage drop V Q = ILOAD x RDS(ON),
CRSS = MOSFET reverse transfer capacitance, IGATE =
DH driver peak output current capability (1A typ), and
20ns = DH driver inherent rise/fall time. The MAX1636’s
output undervoltage shutdown protects the synchronous rectifier under output short-circuit conditions. To
reduce EMI, add a 0.1µF ceramic capacitor from the
high-side switch drain to the low-side switch source.
Rectifier Clamp Diode
The rectifier is a clamp across the low-side MOSFET
that catches the negative inductor swing during the
60ns dead time between turning one MOSFET off and
each low-side MOSFET on. The latest generations of
MOSFETs incorporate a high-speed silicon body
diode, which serves as an adequate clamp diode if
efficiency is not of primary importance. A Schottky
diode can be placed in parallel with the body diode to
reduce the forward voltage drop, typically improving
efficiency 1% to 2%. Use a diode with a DC current rating equal to one-third of the load current; for example,
use an MBR0530 (500mA-rated) type for loads up to
1.5A, a 1N5819 type for loads up to 3A, or a 1N5822
type for loads up to 10A. The rectifier’s rated reversebreakdown voltage must be at least equal to the maximum input voltage, preferably with a 20% derating
factor.
Boost-Supply Diode
A signal diode such as a 1N4148 works well in most
applications. If the input voltage can go below +6V,
use a small (20mA) Schottky diode for slightly
improved efficiency and dropout characteristics. Do
not use large power diodes, such as 1N5817 or
1N4001, since high junction capacitance can pump up
VL to excessive voltages.
20
Low-Voltage Operation
Low input voltages and low input-output differential
voltages each require extra care in their design. Low
absolute input voltages can cause the VL linear regulator to enter dropout and eventually shut itself off. Low
VIN - VOUT differentials can cause the output voltage to
sag when the load current changes abruptly. The sag’s
amplitude is a function of inductor value and maximum
duty factor (D MAX , an Electrical Characteristics
parameter, 98% guaranteed over temperature at f =
200kHz) as follows:
VSAG =
(ISTEP )2 x L
2 x CF × (VIN(MIN) x DMAX − VOUT )
Table 6 is a low-voltage troubleshooting guide. The
cure for low-voltage sag is to increase the output
capacitor’s value. For example, at VIN = +5.5V, VOUT =
5V, L = 10µH, f = 200kHz, and ISTEP = 3A, a total
capacitance of 660µF keeps the sag less than 200mV.
Note that only the capacitance requirement increases;
the ESR requirements do not change. Therefore, the
added capacitance can be supplied by a low-cost bulk
capacitor in parallel with the normal low-ESR capacitor.
__________Applications Information
Heavy-Load Efficiency Considerations
The major efficiency-loss mechanisms under loads are
as follows, in the usual order of importance:
• P(I2R) = I2R losses
• P(tran) = transition losses
• P(gate) = gate-charge losses
• P(diode) = diode-conduction losses
• P(cap) = capacitor ESR losses
• P(IC) = losses due to the IC’s operating supply
current
Inductor core losses are fairly low at heavy loads
because the inductor’s AC current component is small.
Therefore, they are not accounted for in this analysis.
Ferrite cores are preferred, especially at 300kHz, but
powdered cores, such as Kool-Mu, can also work well.
Efficiency = POUT / PIN x 100%
= POUT / (POUT + PTOTAL) x 100%
PTOTAL = P(I2R) + P(tran) + P(gate) + P(diode)
+ P(cap) + P(IC)
P = (I2R) = (ILOAD)2 x (RDC + RDS(ON) +RSENSE)
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Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
SYMPTOM
CONDITION
ROOT CAUSE
SOLUTION
Sag or droop in VOUT under
step-load change
Low VIN-VOUT
differential, <1.5V
Limited inductor-current
slew rate per cycle.
Increase bulk output capacitance
per formula (see Low-Voltage
Operation section). Reduce inductor
value.
Dropout voltage is too high
(VOUT follows VIN as VIN
decreases)
Low VIN-VOUT
differential, <1V
Maximum duty-cycle limits
exceeded.
Reduce operation to 200kHz.
Reduce MOSFET on-resistance and
coil DCR.
Unstable—jitters between
different duty factors and
frequencies
Low VIN-VOUT
differential, <0.5V
Normal function of internal
low-dropout circuitry.
Increase the minimum input voltage
or ignore.
Poor efficiency
Low input voltage, <5V
VL linear regulator is going
into dropout and isn’t providing good gate-drive levels.
Use a small 20mA Schottky diode
for boost diode. Supply VL from an
external source.
Won’t start under load or
quits before battery is
completely dead
Low input voltage, <4.5V
VL output is so low that it
hits the VL UVLO threshold.
Supply VL from an external source
other than VIN, such as the system
+5V supply.
where RDC is the DC resistance of the coil, RDS(ON) is
the MOSFET on-resistance, and RSENSE is the currentsense resistor value. The RDS(ON) term assumes identical MOSFETs for the high-side and low-side switches
because they time-share the inductor current. If the
MOSFETs are not identical, their losses can be estimated by averaging the losses according to duty factor.
PD(tran) = transition loss = VIN x ILOAD x f x 3/2 x
[(VIN CRSS / IGATE ) + 20ns]
where CRSS is the reverse transfer capacitance of the
high-side MOSFET (a data-sheet parameter), IGATE is
the DH gate-driver peak output current (1.5A typ), and
20ns is the rise/fall time of the DH driver (20ns typ).
P(gate) = Qg x f x VL
where VL is the internal logic-supply voltage (+5V), and
Qg is the sum of the gate-charge values for low-side
and high-side switches. For matched MOSFETs, Qg is
twice the data-sheet value of an individual MOSFET. If
VOUT is set to less than 4.5V, replace VL in this equation with V BATT . In this case, efficiency can be
improved by connecting VL to an efficient 5V source,
such as the system +5V supply.
P(diode) = diode conduction losses =
ILOAD x VFWD x tD x f
where tD is the diode-conduction time (120ns typ), and
VFWD is the forward voltage of the diode. This power is
dissipated in the MOSFET body diode if no external
Schottky diode is used.
P(cap) = input capacitor ESR loss = IRMS2 x RESR
where IRMS is the input ripple current as calculated in
the Input Capacitor Value section.
Light-Load Efficiency Considerations
Under light loads, the PWM operates in discontinuous
mode, where the inductor current discharges to zero at
some point during the switching cycle. This makes the
inductor current’s AC component high compared to the
load current, which increases core losses and I2R losses in the output filter capacitors. For best light-load efficiency, use MOSFETs with moderate gate-charge
levels and use ferrite, MPP, or other low-loss core material. Avoid powdered-iron cores; even Kool-Mu
(aluminum alloy) is not as good as ferrite.
PC Board Layout Considerations
Good PC board layout is required in order to achieve
specified noise, efficiency, and stable performance.
The PC board layout artist must be given explicit
instructions, preferably a pencil sketch showing the
placement of power-switching components and highcurrent routing. See the PC board layout in the
MAX1636 evaluation kit manual for examples. A ground
plane is essential for optimum performance. In most
applications, the circuit will be located on a multi-layer
board, and full use of the four or more copper layers
is recommended. Use the top layer for high-current
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21
MAX1636
Table 6. Low-Voltage Troubleshooting Chart
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
HIGH-CURRENT PATH
VCC
SENSE RESISTOR
MAX1636
VL
GND
BST
LX
MAX1636
Figure 6. Kelvin Connections for the Current-Sense Resistors
Figure 7. Capacitor Placement
connections, the bottom layer for quiet connections
(REF, CC, GND), and the inner layers for an uninterrupted ground plane. Use the following step-by-step
guide:
terminals, which ensures that the IC’s analog ground is
sensing at the supply’s output terminals without interference from IR drops and ground noise. Other high-current paths should also be minimized, but focusing
primarily on short ground and current-sense connections eliminates about 90% of all PC board layout problems (see the PC board layouts in the MAX1636
evaluation kit manual for examples).
2) Place the IC and signal components. Keep the main
switching nodes (LX nodes) away from sensitive
analog components (current-sense traces and REF
capacitor). Place the IC and analog components on
the opposite side of the board from the powerswitching node. Important: The IC must be no further than 10mm from the current-sense resistors.
Keep the gate-drive traces (DH, DL, and BST) shorter than 20mm and route them away from CSH, CSL,
and REF. Place ceramic bypass capacitors close to
the IC. The bulk capacitors can be placed further
away. If using VL to power VCC, minimize noise by
placing a 0.1µF capacitor close to the VCC pin and
placing the 4.7µF capacitor further away, but closer
than the boost diode (Figure 7).
3) Use a single-point star ground where the input
ground trace, power ground (subground plane), and
normal ground plane meet at the supply's output
ground terminal. Connect both IC ground pins and
all IC bypass capacitors to the normal ground plane.
1) Place the high-power components (C1, C2, Q1, Q2,
D1, L1, and R1) first, with their grounds adjacent.
• Minimize current-sense resistor trace lengths and
ensure accurate current sensing with Kelvin connections (Figure 6).
• Minimize ground trace lengths in the high-current
paths.
• Minimize other trace lengths in the high-current
paths.
— Use >5mm-wide traces.
— CIN to high-side MOSFET drain: 10mm
max length
— Rectifier diode cathode to low side
— MOSFET: 5mm max length
— LX node (MOSFETs, rectifier cathode, inductor): 15mm max length
Ideally, surface-mount power components are butted
up to one another with their ground terminals almost
touching. These high-current grounds are then connected to each other with a wide, filled zone of
top-layer copper so they do not go through vias. The
resulting top-layer subground plane is connected to the
normal inner-layer ground plane at the output ground
___________________Chip Information
TRANSISTOR COUNT: 3472
22
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Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
SSOP.EPS
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23
MAX1636
________________________________________________________Package Information
MAX1636
Low-Voltage, Precision Step-Down
Controller for Portable CPU Power
24
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